UNIVERSITY OF CALIFORNIA Santa Barbara The Integration of ...

UNIVERSITY OF CALIFORNIA
Santa Barbara
The Integration of Mach-Zehnder Modulators with Sampled Grating DBR
Committee in charge:
Lasers
A Dissertation submitted in partial satisfaction of the
requirements for the degree Doctor of Philosophy
in Materials
by
Jonathon Scott Barton
Professor Larry A. Coldren, Chair
Professor Daniel J. Blumenthal
Professor Nadir Dagli
Professor Steven DenBaars
Professor Evelyn L. Hu
September 2004

The dissertation of Jonathon Scott Barton is approved.
________________________________________
Evelyn L. Hu
________________________________________
Daniel J. Blumenthal
________________________________________
Steven DenBaars
________________________________________
Nadir Dagli
________________________________________
Larry A. Coldren, Committee Chair
September 2004

The Integration of Mach-Zehnder Modulators with Sampled Grating DBR lasers
Copyright © 2004
By
Jonathon Scott Barton
iii

TABLE OF CONTENTS
Abstract ................................................................................................ vii
Vita......................................................................................................... ix
Symbols and acronyms ....................................................................... xv
Acknowledgements.............................................................................. xvi
INTRODUCTION .................................................................................................1
0.1 Direct Modulation.......................................................................7
0.2 External Modulation .................................................................10
0.3 Electro-absorption Modulators.................................................14
0.4 Mach-Zehnder Modulators ......................................................15
0.5 Mach-Zehnder Bias Approaches.............................................17
0.6 Traveling Wave Devices..........................................................20
References..............................................................................21
CHAPTER I: DEVICE INTEGRATION .............................................. 25
1.1 Widely-tunable laser design ...................................................27
1.2 Semiconductor Optical Amplifier(SOA) Integration ................33
1.3 Dual SOAs ..............................................................................36
1.4 Optical Feedback and Reflection............................................40
1.5 Linewidth Measurements........................................................43
1.6 Relative Intensity Noise ..........................................................47
References..............................................................................50
CHAPTER II: MOCVD GROWTH & FABRICATION............. 54
2.1 Semiconductor Epitaxial Structure ..........................................56
2.2 Quantum Well Design..............................................................57
2.3 Conducting Substrate Base structure......................................58
2.4 Growth Characterization..........................................................60
2.5 Semi-insulating substrate growth ............................................63
2.6 Regrowth..................................................................................64
2.7 Zn Doping of InP and InGaAsP ...............................................66
2.8 Transmitter Fabrication............................................................70
References...............................................................................73
CHAPTER III: LUMPED MODULATOR DESIGNS.................. 76
3.1 Device Efficiency ....................................................................77
3.2 DC Extinction Curves..............................................................79
3.3 Franz-Keldysh Absorption .....................................................81
iv

3.4 Electric Field Effects ...............................................................84
Linear Electro-optic effect ...............................................84
Kerr Effect .......................................................................86
3.5 Carrier Based Effects..............................................................88
Plasma Effect..................................................................88
Bandfilling Effect .............................................................91
Carrier induced bandgap shrinkage ...............................92
3.6 Temperature induced bandgap shrinkage..............................93
3.7 Accumulation of Effects .........................................................96
3.8 High Speed Design................................................................99
3.5 Junction Capacitance Minimization .....................................101
3.6 Parasitic Capacitance Minimization.....................................103
3.7 Fringing Capacitance ..........................................................107
3.8 Multimode Interference (MMI) Design .................................109
3.9 Phase Shifter .......................................................................112
3.10 1 st Generation Designs ........................................................114
3.11 2 nd Generation Designs .......................................................116
References...........................................................................119
CHAPTER IV: SERIES PUSH-PULL DESIGNS ......................125
4.1 Lumped Series push-pull bandwidth ..................................126
4.2 Dual RF series push-pull devices........................................129
4.3 Traveling wave modulators ...............................................131
4.4 Traveling wave matching ....................................................133
4.5 Transmission line model......................................................137
4.6 Traveling wave bandwidth ...................................................140
4.7 RF loss.................................................................................144
4.8 CPS T-Electrode devices ....................................................150
4.9 Characteristic Impedance Comparison ...............................153
4.10 Measured Bandwidth...........................................................154
References...........................................................................158
CHAPTER V: DEVICE COMPARISONS.......................................165
5.1 DC Modulation Efficiency......................................................170
5.2 RF Extinction of devices........................................................174
5.3 Bandwidth Comparison.........................................................178
5.4 Chirp Measurements.............................................................179
5.5 Chirp Measurement Techniques...........................................179
5.6 Linearization of Modulators...................................................184
References............................................................................188
v

CHAPTER VI: CONCLUSIONS AND FUTURE WORK .....193
6.1 Wavelength Converters .......................................................196
References...........................................................................199
Appendix A: Relevant Material Constants..........................................204
Appendix B: RF Spectrum Analyzer Deimbedding ............................208
Appendix C: Process .........................................................................210
vi

ABSTRACT
JONATHON S. BARTON
THE INTEGRATION OF MACH-ZEHNDER MODULATORS WITH
SAMPLED GRATING DBR LASERS
Some of the latest results of InP based widely-tunable optical transmitters will
be presented. Widely-tunable transmitters are seen as a crucial component
in Dense Wavelength Division Multiplexing (DWDM) communication systems.
Integration of different optical components reduces the costs, insertion losses,
and footprint of the device. This work outlines the material design and
fabrication aspects to produce high bandwidth, low drive voltage modulation
without degradation of the laser and modulator device characteristics with
optical/electrical crosstalk. Photonic integrated circuits are particularly
susceptible to optical reflections - which can cause excessive chirp, gain
ripple, lasing of the Semiconductor Optical Amplifier, higher noise figure, and
inter-modulation distortion. Careful design is required to minimize refractive
index discontinuities in active/passive interfaces, Multi-mode interference
devices, and through angling and flaring at the output waveguide.
With the use of a Mach-Zehnder modulator, the chirp parameter can be tailored
to maximize the transmission distance through fiber - particularly important at
high data rates (10Gbit/s). Using traveling wave electrodes in a series push-pull
electrode structure, we are able to demonstrate some of the highest speed
widely tunable lasers to date with 40GHz bandwidth. DC extinction Vpi as low
vii

as 0.6V is demonstrated with high saturation power dual SOA structures. This
is all achieved using low k dielectrics (BCB) and highly efficient PN junctions in
which compromises must be made to insure high performance in each
integrated device region.
viii

1975 Born in Sacramento, California
VITA
JONATHON S. BARTON
1993 Bachelor degree in Electrical Engineering and Material Science -
University of California, Davis
2003 Intel Fellow
2004 PH.D in Electronic Materials - University of California Santa Barbara
LIST OF PUBLICATIONS
[1] Mason B, Barton J, Fish GA, Coldren LA, Denbaars SP. Design of sampled
grating DBR lasers with integrated semiconductor optical amplifiers. IEEE
Photonics Technology Letters, vol.12, no.7, July 2000, pp.762-4.
[2] Blumenthal DJ, Olsson B-E, Rossi G, Dimmick TE, Rau L, Masanovic M,
Lavrova O, Doshi R, Jerphagnon O, Bowers JE, Kaman V, Coldren LA, Barton
J. “All-optical label swapping networks and technologies.” Journal of Lightwave
Technology, vol.18, no.12, Dec. 2000, pp.2058-75.
[3] Hanxing Shi, Cohen D, Barton J, Majewski M, Coldren LA, Larson MC, Fish
GA. “Relative intensity noise measurements of a widely tunable sampledgrating
DBR laser.” IEEE Photonics Technology Letters, vol.14, no.6, June
2002, pp.759-61.
[4] Majewski ML, Barton J.S., Coldren LA, Akulova Y, Larson MC. “Direct
intensity modulation in sampled-grating DBR lasers.” IEEE Photonics
Technology Letters, vol.14, no.6, June 2002, pp.747-9.
[5] Shi HX, Cohen DA, Barton J, Majewski M, Coldren LA, Larson MC, Fish
GA. “Dynamic range of widely tunable sampled grating DBR lasers.”
Electronics Letters, vol.38, no.4, 14 Feb. 2002, pp.180-1.
ix

[6] Skogen EJ, Barton JS, DenBaars SP, Coldren LA. “Tunable sampledgrating
DBR lasers using quantum-well intermixing.” IEEE Photonics
Technology Letters, vol.14, no.9, Sept. 2002, pp.1243-5.
[7] Skogen EJ, Barton JS, Denbaars SP, Coldren LA. “A quantum-wellintermixing
process for wavelength-agile photonic integrated circuits.” IEEE
Journal of Selected Topics in Quantum Electronics, vol.8, no.4, July-Aug. 2002,
pp.863-9.
[8] Raring J.W., E. J. Skogen, L. A. Johansson, M. N. Sysak, J. S. Barton, M.
L. Mašanović, L. A. Coldren Demonstration of Widely-Tunable Single-Chip 10
Gb/s Laser-Modulators Using Multiple-Bandgap InGaAsP Quantum-Well
Intermixing, Photonics Technology Letts July 2003.
[9] Barton J.S., Skogen E.J., Mašanović M.L., DenBaars S.P., and Coldren
L.A., “Widely-tunable high-speed transmitters using integrated SGDBRs and
Mach-Zehnder modulators.” IEEE Journal of selected topics in quantum
electronics, Vol. 9, NO. 5, pp.1113-17. September /October 2003.
[10] Coldren LA, Fish GA, Akulova Y, Barton JS, Johansson L, Coldren CW.
“Tunable semiconductor lasers: a tutorial.” Journal of Lightwave Technology,
vol.22, no.1, Jan. 2004, pp.193-202.
[11] Masanovic M.L., V. Lal, J. A. Summers, J. S. Barton, E. J. Skogen, L. A.
Coldren, and D. J. Blumenthal, "Design and Performance of a Monolithically-
Integrated Widely-Tunable All-Optical Wavelength Converter with Independent
Phase Control," accepted for publication in IEEE Photonics Technology
Letters, 2004.
[12] Masanovic M.L., V. Lal, J. S. Barton, E. J. Skogen, J. A. Summers, L. Rau,
L. A. Coldren, and D. J. Blumenthal, "Widely-Tunable Monolithically-Integrated
All-Optical Wavelength Converters in InP," to be published in IEEE Journal of
Lightwave Technology, 2004.
[13] Hutchinson J.M., J. F. Zheng, J. S. Barton, M. L. Masanovic, M. N. Sysak,
J. A. Henness, L. A. Johansson, D. J. Blumenthal, L. A. Coldren, H. V. Demir,
V. A. Sabnis, O. Fidaner, J. S. Harris, and D. A. B. Miller, " Indium Phosphide
based Wavelength Conversion for High Speed Optical Networks," Intel
Technology Journal, 2004.
[14] Barton JS, Masanovic ML, Sysak MN, Hutchinson JM, Skogen EJ,
Blumenthal DJ, Coldren LA. “2.5-Gb/s error-free wavelength conversion using
a monolithically integrated widely tunable SGDBR-SOA-MZ transmitter and
x

integrated photodetector.” IEEE Photonics Technology Letters, vol.16, no.6,
June 2004, pp.1531-3.
[15] Masanovic M.L., V. Lal, J. S. Barton, E. J. Skogen, L. A. Coldren, and D.
J. Blumenthal, "Monolithically integrated Mach-Zehnder interferometer
wavelength converter and widely tunable laser in InP," IEEE Photonics
Technology Letters, vol. 15, pp. 1117-19, 2003.
[16] Masanovic M.L., E. J. Skogen, J. S. Barton, J. M. Sullivan, D. J.
Blumenthal, and L. A. Coldren, "Multimode interference-based two-stage 1 * 2
light splitter for compact photonic integrated circuits," IEEE Photonics
Technology Letters, vol. 15, pp. 706-8, 2003.
[17] Skogen EJ, Raring JW, Barton JS, DenBaars SP, Coldren LA. “Postgrowth
control of the quantum-well band edge for the monolithic integration of widely
tunable lasers and electroabsorption modulators.” IEEE Journal of Selected
Topics in Quantum Electronics, vol.9, no.5, Sept.-Oct. 2003, pp.1183-90.
[18] Sysak, M.N., J. S. Barton, L. A. Johansson, J. W. Raring, E. J. Skogen, M.
L. Mašanović, D. Blumenthal, and L. A. Coldren, Single Chip Wavelength
Conversion using a Photocurrent Driven (PD) EA Modulator integrated with a
Widely Tunable Sampled Grating DBR (SGDBR) Laser. Submitted to
Photonics Tech. Letts.
CONFERENCE TALKS
[19] Mason B, Fish GA, Barton J, Coldren LA, DenBaars SP. Characteristics of
sampled grating DBR lasers with integrated semiconductor optical amplifiers.
Optical Fiber Communication Conference. Technical Digest Postconference
Edition. Trends in Optics and Photonics Vol.37 (IEEE Cat. No. 00CH37079).
Opt. Soc. America. Part vol.1, 2000, pp.193-5 vol.1.
[20] Majewski ML, Barton J, Coldren LA, Akulova Y, Larson MC. “Widely
tunable directly modulated sampled-grating DBR lasers.” Optical Fiber
Communications Conference. (OFC). Postconference Technical Digest (IEEE
Cat. No.02CH37339). Opt Soc. America. Part vol.1, 2002, pp.537-8 vol.1.
[21] Barton J, Coldren L, Fish GA. Tunable lasers using sampled grating DBRs.
2001 Digest of LEOS Summer Topical Meetings: Advanced Semiconductor
Lasers and Applications/Ultraviolet and Blue Lasers and Their
Applications/Ultralong Haul DWDM Transmission and Networking/WDM
Components (IEEE Cat. No.01TH8572). IEEE. 2001, pp.2 pp. Invited
xi

[22] Skogen EJ, Barton J, DenBaars SP, Coldren LA. “Tunable buried ridge
stripe sampled grating distributed Bragg reflector lasers utilizing quantum well
intermixing.” LEOS 2001. 14th Annual Meeting of the IEEE Lasers and Electro-
Optics Society (Cat. No.01CH37242). IEEE. Part vol.1, 2001, pp.169-70 vol.1.
[23] Barton JS., Skogen, E.J. , Masanovic M., S. Denbaars, L. A. Coldren,
“Integration of a Mach-Zehnder Modulator with Sampled Grating Distributed
Bragg Reflector Laser,” Proc. Integrated Photonics Research Conference,
paper no. 1FC3-1, Vancouver, Canada, July 17-19 2002.
[24] Skogen E.J., Barton J.S., DenBaars S.P., Coldren, L.A., “On Tuning
Efficiency of Sampled Grating DBR Lasers using Quantum Well Intermixing”,
Proc.Integrated Photonics Research Conference, paper no. IFC2, Vancouver,
Canada July 17-19 2002.
[25] Mašanović M., Skogen E.J., Barton J.S., Sullivan J., Blumenthal D.J.,
Coldren L.A., “Cascaded Multimode Interference-Based 1x2 Light Splitter for
Photonic Integrated Circuits” Integrated Photonics Research Conference,
Vancouver, Canada, July 2002.
[26] Majewski ML, Barton J, Coldren LA, Akulova Y, Fish G. “Wavelength
monitoring in widely tunable sampled-grating DBR lasers integrated with
semiconductor optical amplifiers.” Technical Digest. Summaries of papers
presented at the Conference on Lasers and Electro-Optics. Conference Edition
(IEEE Cat. No.02CH37337). Opt. Soc. America. Part vol.1, 2002, pp.414-16
vol.1.
[27] Barton JS, Skogen EJ, Masanovic ML, DenBaars SP, Coldren LA.
“Tailorable chirp using integrated Mach-Zehnder modulators with tunable
sampled grating distributed Bragg reflector lasers.” 2002 IEEE 18th
International Semiconductor Laser Conference. Conference Digest (Cat.
No.02CH37390). paper no. TuB3, Garmisch, Germany (Sept. 29- Oct. 3) IEEE.
2002, pp.49-50.
[28] Skogen EJ, Barton JS, Masanovic ML, Getty JT, DenBaars SP, Coldren
LA. “Use of post-growth control of the quantum-well band edge for optimized
widely-tunable laser-x devices.” 2002 IEEE 18th International Semiconductor
Laser Conference. Conference Digest (Cat. No.02CH37390). IEEE. 2002,
pp.53-4.
[29] Barton JS, Skogen EJ, Masanovic ML, Raring J, Sysak MN, Johansson L,
DenBaars SP, Coldren LA. “Photonic integrated circuits based on sampledgrating
distributed-Bragg-reflector lasers.” SPIE-Int. Soc. Opt. Eng.
xii

Proceedings of Spie - the International Society for Optical Engineering,
vol.4998, 2003, pp.43-54. Invited
[30] Mašanović M., Skogen EJ, Barton JS, Lal V, Blumenthal DJ, Coldren LA.
“Demonstration of monolithically-integrated InP widely-tunable laser and SOA-
MZI wavelength converter.” 2003 International Conference Indium Phosphide
and Related Materials. Conference Proceedings (Cat. No.03CH37413). IEEE.
2003, pp.289-91. Santa Barbara, California (May 12-16, 2003)
[31] Skogen E.J., J.S. Barton, J.W. Raring, L.A. Coldren, S.P. DenBaars,
"High Contrast InP/InGaAsP Grating MOCVD Regrowth Using TBA and
TBP", Conference Proceedings from ICMOVPE conference, Hawaii. 2004.
[32] Skogen EJ, Barton JS, DenBaars SP, Coldren LA. “Wavelength agile
photonic integrated circuits using a novel quantum well intermixing process.”
Optical Fiber Communications Conference. (OFC). Postconference Technical
Digest. Postdeadline Papers (IEEE Cat. No.02CH37339). Opt Soc. America.
Part vol.2, 2002, pp.FB8-1-3 vol.2.
[33] Johansson LA, Barton JS, Coldren L. “High-performance EAM-integrated
SGDBR laser for WDM microwave photonic applications.” 2002 International
Topical Meeting on Microwave Photonics. Technical Digest (IEEE Cat.
No.02EX638). IEICE. 2002, pp.61-4. Tokyo, Japan.
[34] Barton J. S., Milan L. Mašanović, Matthew N. Sysak, Erik J. Skogen, John
Hutchinson, Daniel J. Blumenthal, Larry A. Coldren, “ A Novel Monolithically-
Integrated Widely-Tunable Wavelength Converter Based on a SGDBR-SOA-
MZ Transmitter and Integrated Photo-Detector,” Proc. Photonics in Switching
2003, paper no. PS.Mo.A9, pp. 34-36, Versailles, France (September 2003)
[35] Mašanović M.L., Roopesh R. Doshi, Vikrant Lal, Jonathon. S. Barton, Larry
A. Coldren, Daniel J. Blumenthal, “First Demonstration of both Analog and
Digital Wavelength Conversion using a Monolithically-Integrated InP Widely
Tunable All-Optical Wavelength Converter (TAO-WC),” Proc. Photonics in
Switching 2003, paper no. PS.Mo.A10, pp. 37-39, Versailles, France
(September 2003)
[36] Mašanović M. L., Vikrant Lal, Jonathon S. Barton, Larry A. Coldren and
Daniel J. Blumenthal, “Wavelength Conversion Over a 50nm Input and 21nm
Output Wavelength Range Using a Monolithically Integrated Tunable All-
Optical MMI-MZI (TAOMI) Wavelength Converter,” Proc. ECOC-IOOC 2003,
paper no. Th1.6.5, pp.930-931, Rimini, Italy (September 21-25, 2003)
xiii

[37] Johansson L.A., J.S. Barton, M.L. Masanovic, J.M. Hutchinson, J.A.
Henness, Y.A. Akulova, G.A. Fish and L.A. Coldren, “Integrated Optical
Components for WDM Optical/Wireless Applications,” Proc. Microwave
Photonics, pp. 161-164, Budapest, Hungary (September 2003)
[38] Johansson L.A., J.S. Barton and L.A. Coldren, G.A. Fish, “Generation of
High-Speed Optical Frequency Modulation Using a Phase Modulator” OFC
2004.
[39] Hutchinson J.M., Jonathon S. Barton, Milan L. Mašanović, Matthew N.
Sysak, Jeffrey A. Henness, Leif A. Johansson, Larry A. Coldren, “Monolithically
integrated InP-based tunable wavelength conversion”, Presented at Photonics
West, San Jose, 2004.
[40] Mašanović M.L., Vikrant Lal, Leif A. Johansson, Jonathon S. Barton, Larry
A. Coldren, Daniel J. Blumenthal, “Characterization of the Chirp Properties of a
Monolithically-Integrated Widely-Tunable All-Optical Wavelength Converter
(TAO-WC)” Proc. LEOS 2003, paper no. TuCC3, pp. 433-434, Tucson,
Arizona (October 2003)
[41] Hutchinson J.M., Jeffery A. Henness, Leif A. Johansson, Jonathon S.
Barton, Milan L. Mašanović, Larry A. Coldren, “2.5 Gb/sec Wavelength
Conversion Using Monolithically-Integrated Photodetector and Directly
Modulated Widely-Tunable SGDBR Laser,” Proc. LEOS 2003 , paper no.
WU4, pp. 650-651, Tucson, Arizona (October 2003)
[42] Raring J.W., E. J. Skogen, L. A. Johansson, M. N. Sysak, J. S. Barton, M.
L. Masanovic, and L. A. Coldren, "Quantum Well Intermixing for Monolithic
Integration: A Demonstration of Novel Widely-Tunable 10Gb/s Transmitters
and Wavelength Converters," presented at Integrated Photonics Research
Conference, San Francisco, California, USA, 2004.
[43] Masanovic M.L., V. Lal, J. A. Summers, J. S. Barton, E. J. Skogen, L. A.
Coldren, and D. J. Blumenthal, "10 Gbps and 2.5 Gbps error-free operation of
a monolithically integrated widely-tunable all-optical wavelength converter with
independent phase control and output 35nm tuning range," presented at
Optical Fiber Communications Conference, OFC, Los Angeles, California,
USA, 2004.
[44] Choquette KD, Barton JS, Geib KM, Allerman AA, Hindi JJ. Short
wavelength bottom-emitting VCSELs. SPIE-Int. Soc. Opt. Eng. Proceedings of
Spie - the International Society for Optical Engineering, vol.3627, 1999, pp.56-
9.
xiv

SYMBOLS AND ACRONYMS
AFM Atomic Force Microscope
BCB Benzocyclobutene
BER Bit Error Rate
CPW Coplanar Waveguide.
CPS Coplanar Stripline
DBR Distributed Bragg Reflector
EAM Electroabsorption Modulator
EO Electo-optic
FESEM Field Emission Scanning Electron Microscope
FKE Franz-Keldysh effect
FWHM Full-width Half Maximum
LEO Linear Electrooptic effect
MOCVD Metal-organic chemical vapor deposition
MQW Multiple Quantum Well
MZ Mach-Zehnder
MZM Mach-Zehnder Modulator
PIC Photonic integrated circuit
RIE Reactive Ion Etch
SOA Semiconductor Optical Amplifier
SEM Scanning Electron Microscope
SIMS Secondary Ion Mass Spectroscopy
SGDBR Sampled Grating Distributed Bragg Reflector (laser)
SMSR Side Mode Suppression Ratio
TW Traveling wave
QCSE Quantum Confined start effect
QWI Quantum Well Intermixing
WDM Wavelength Division Multiplexing
WG Waveguide
SYMBOLS
A. amplitude factor,
� damping factor,
�r . Angular relaxation resonance frequency.
go differential gain
�i internal quantum efficiency
v cavity volume
xv

ACKNOWLEDGEMENTS
I have been lucky to have the opportunity to work with a great set of people in a
first-rate facility that truly allows the full process of events to take place – from
MOCVD growth, to processing, to complicated RF testing of the devices. This
work would not be possible without the help of my committee members which
span a wide knowledge base – giving me the opportunity to grow in the
MOCVD lab under Steve DenBaars, processing expertise from Evelyn Hu, RF
experience from Nadir Dagli, Opto-electronic systems work from Dan
Blumenthal, and optoelectronic design expertise from Larry Coldren. Everyone
in the Coldren group has made an impact in one way or another. Special
thanks to Erik Skogen for showing me how to grow in MOCVD and process InP
based materials, Beck Mason and Greg Fish for helping me in my early career
in SGDBR design and testing, Dan Lofgreen for showing me the intricacies of
MATLAB. Milan Mašanović and Leif Johansson for helping me out with RF
testing. The folks at Agility Communications who helped with some of the
regrowths and AR coating runs. Also, thanks to Intel for providing me with the
2003 Intel fellowship. Last but not least, special thanks to Jennifer Hale, who
has endeared the long hours and stress.
xvi

INTRODUCTION
Photonic integrated circuits[11] are being pursued to keep up with the ever-
increasing desire for high optical bandwidth with a small footprint, high power,
cost effective packaging and high degree of functionality. With the recent
downturn in the telecom market, optical components that will provide value to
optical networks at low cost are particularly desired. The cost primarily comes
from packaging and the yield improvement of chips in tunable laser fabrication.
Because of this, tunable lasers are seen as able to provide value by the
integration of lasers with optical amplifiers and modulators on one chip
reducing fiber alignments from as high as 5 in the discrete case to 1 in the
integrated case. Each fiber alignment adds 3-5dB of insertion loss which
demonstrates the obvious desire to integrate. Also integrated devices enable
new small form factor 10 Gbit/s transponders – in which packaging of
conventional bulky discrete parts simply is not possible due to space
limitations.
1

Mach-Zehnder SOA SGDBR
Fig. 1. Integrated SGDBR-SOA-MZ
This dissertation explores the integration of interferometric Mach-Zehnder (MZ)
modulators, semiconductor optical amplifiers (SOA) and sampled grating
(SGDBR) tunable lasers as illustrated in fig. 1. This introduction will attempt to
outline the choices of opto-electronic building blocks available to achieve high-
performance optical transmission. As will be outlined in the following chapters,
these devices are capable of achieving 10 Gbit/s operation with tailorable chirp.
Monolithic integration of these devices presents a number of challenges.
Optimization of the laser and modulator structures needs to take into account
often competing design specifications. Additionally, careful design is
necessary to minimize optical reflections[1,227,224] and retain single mode
operation. Electrical crosstalk must be reduced to control unwanted chirp and
improve the tuning mechanism. Also, thermal crosstalk plays a role in the
2

integrated device performance[25]. An in-depth look at these integration
challenges is presented in Chapter 1.
Desire for increased bandwidth has prompted the change of SONNET
standards as shown in table 1. Suppliers are gearing up to provide data rates
greater than 40 Gb/s to meet perceived bandwidth constraints as fiber is being
implemented in not only long-haul applications, but increasingly, metro
networks as well.
Table 1. Sonnet Data Rate
Rates
OC-12 622 Mb/s
OC-48 2.5 Gb/s
OC-192 10 Gb/s
OC-768 40 Gb/s
Cost has also driven the push for devices that enable CWDM and DWDM
systems as new fiber tends to be expensive to deploy – however replacement
of transmitters and receivers considerably less expensive. System designers
generally would like single devices that cover the C- Band (1530-1562) and/or
L-Band (1565-1610) and S-Band(1485-1520).
Another major concern for communication applications– particularly with higher
bit rates (10 and 40 Gbit/s) is the control of a parameter called chirp. Chirp (�)
is defined as the ratio between the change in the real part to the imaginary part
of the refractive index.
3

�n
real
� �
[1]
�n
This parameter strongly affects the maximum transmission distance possible
before signal regeneration. Figure 2a&b shows the influence of the chirp
parameter [�] on the transmission distance for standard non dispersion shifted
fiber.
Fig 2.A &2B Dispersion Penalty vs transmission distance for OC-192 signal 9.95 Gbit/s for various alpha parameters for
standard non-dispersion shifted Corning SMF-28 fiber. Maximum transmission distance for 10 and 40 Gbit/s signal
The dispersion penalty is given by[14] :
imag
2 2
2 2
� � 5log
[( 1�
8��B
L)
� ( 8�B
L)
]
[2]
c
10
Where � is chirp parameter, � is group velocity dispersion parameter, B is the bit rate, L is the
fiber length. standard Corning SMF-28 fiber � =16.45 ps/nm/km at 1555nm
This chirp is often split into three major components[521] including:
1.Transient chirp due to the sudden changes of the current in the device
2. Adiabatic chirp due to the induced change in refractive index
4

3. Thermal chirp due to the large resistance change – and resulting
heating/cooling of the active region.
The power penalty is also influenced by degradation of the extinction ratio as
shown in fig. 3, the signal to noise ratio, and jitter [24].
Fig. 3. Power penalty due to extinction ratio degradation
The total transmission distance is often described at the point at which the total
power penalty reaches 2dB. With proper chirp management, for a 2dB power
penalty at 10Gbit/s – one can transmit over 125km in between repeaters. At
40Gbit/s, this distance is dramatically lower – just above 7km. This means that
chirp management is crucial at high bit rates – and for 40 Gbit/s this is more
important than high output power as the signal needs to be regenerated
frequently without dispersion compensation.
5

First we should explore the modulation possibilities of tunable lasers. In
principle tunable lasers may be either directly modulated 1 or externally
modulated 2 with either a separate modulator or as an integrated device. Next
we will compare the performance of different external modulator
designs/materials which typically use LiNbO3, AlGaAs/GaAs, Polymers, or InP
in interferometric Mach-Zehnder modulators or in Electro-absorption based
modulators InP devices. Chapter 1 explores the integration platform and the
influence of design and materials on laser, Semiconductor Optical Amplifier
(SOA) and modulator properties and an analysis of the reflections in the
device. Others [306,322] have attempted integration of MZ modulators with
lasers, however without very careful design the performance can suffer from
optical reflections. In fact in some cases these problems have lead people to
believe that integration yields diminishing returns – and lead to exploration of
the copackaging of discrete components[26]. In chapter 2 we will examine the
material properties with respect to the growth structure and doping of the
device.
1 As discussed in Section 2
2 As discussed in Section 3
6

0.1 DIRECT MODULATION
The direct modulation of widely-tunable lasers is desirable[15] – due to its
simplicity and reduced optical absorption path. Fairly large optical bandwidths
have been demonstrated for the direct modulation of DFB lasers, narrowly
tunable DBR lasers 3 , and SGDBR lasers 4 . A demonstration of the small-signal
modulation response S21 for an SGDBR is shown in fig. 4. As the bias to the
laser is increased the response exhibits less dampening.
Intensity Modulation Magnitude (dB)
6
4
2
0
-2
-4
-6
-8
-10
50mA
60mA
70mA
80mA
100mA
120mA
150mA
0 1 2 3 4 5 6 7 8
Frequency (GHz)
P[RF]= -20dBm
Fig. 4 Directly Modulated SGDBR – laser provided by Agility Comm. Small-signal intensity
modulation responses of the SGDBR laser for different gain section currents. λ = 1552.72 nm.
A maximum in the modulation bandwidth occurs for short cavity lengths –
taking into account the carrier-density dependent differential gain and the
3 �31GHz [27]
4 6-8 GHz [28-30]
7

photon lifetime. However, in order to form a SGDBR the gain section length
generally needs to be fairly long 5 to achieve enough gain for operation. At
these lengths one can expect a maximum bandwidth on the order of 15-20
GHz ignoring parasitics[312]. A higher internal loss leads to a reduced photon
lifetime – and improved modulation bandwidth.
High-speed direct modulation of SGDBR lasers requires high differential gain,
low nonlinear gain saturation, high optical confinement, and short and narrow
cavities. The small signal frequency response is given by[312]:
� 1 ��
1 � A
H ( w)
� � ��
�
[3]
2 2
�1�
j��
s ��1
� j��
RC � �r
� � � j��
where A is an amplitude factor, � is the damping factor, �r is the angular relaxation resonance
frequency.
The first two terms in (1) correspond to the low-frequency limitations imposed
on the modulation response. The first term depends on the carrier transport
time (�s) over the separate carrier confinement region, the second term is
determined by the time constant �RC of the RC parasitic elements of the chip.
These two parameters are often related through the so called K-factor. The
K-factor calculated for standard SGDBR lasers is close to 0.6-ns, therefore the
maximum achievable 3-dB modulation bandwidth will be 14.8-GHz. The K-
Factor is related to the maximum achievable bandwidth by:
5 >450�m
8

The K factor is given by
f
K �
max
�
�2�
2
�
� K
�
�
�
�
� ( � p �
v g
4 2
where go is the differential gain, vg is group velocity, � is dielectric constant. Published K-factors
are usually in the range 0.13 to 2.4ns.
The maximum modulation bandwidth is limited by RC parasitics, device
heating, and maximum power handling capabilities [312].
Unfortunately, modulating the gain section will modulate the phase and front
mirror sections due to current leakage. So without adequate isolation between
the sections, this can cause the wavelength to change – broadening the
linewidth of the laser known as chirp. When adjacent electrodes are not biased
– the impedance is high leading to little modulation [314] however the isolation
resistance decreases with frequency. Fortunately, since the SGDBR consists of
multiple sections, modulating the gain section alone will only affect part of the
phase within the cavity[330]. Chirp is fairly large and positive for direct
modulation of widely tunable lasers as typical linewidth enhancement factors
range from 2-9[28] depending on the tuning range of the laser and the
placement of the grating with respect to gain spectra[314]. This means the
maximum transmission distance before a repeater is necessary is fairly short –
9
g
o
)
[4]
[5]

on the order of 10’s of km as shown in fig 1. For small-signal modulation the
chirp during direct modulation can be written as:
�v
��P
� f
� 1�
�
f m 2P
� f
where fg is the characteristic frequency of the chirp and � is the chirp parameter, also termed
the linewidth enhancement factor.
A number of approaches have been employed to minimize the chirp of direct
modulation. Chirp has been shown to be improved with tensile-strained MQW
material – giving a smaller linewidth enhancement factor (�). Also,
‘prechirping’ has been employed which involves the simultaneous modulation
of the laser and an external modulator at the same time[532].
0.2 EXTERNAL MODULATION
External modulators refer to modulators that operate external to the cavity of
the laser. The most common materials for use in external modulators are
LiNbO3, electro-optic polymers and III-V compound semiconductors. LiNbO3
has been the material of choice due to its high linear electro-optic coefficient
g
m
�
�
�
and low optical loss(>5dB) as shown in table 2.
10
2
[6]

Table 2
Chemical Formula LiNbO3
Crystal Structure Hexagonal
Space Group: R3c
Point Group: 3m
Lattice Parameters, Е a = 5.148
c = 13.863
Density, g/cm 3 4.64
Thermal Expansion Coefficients, °
C -1
aa = 16.7 . 10 -6
ac = 4.0 . 10 -6
Transparency Range, mm 0.4 - 5.0
Propagation loss 0.2dB/cm [4]
Index of refraction (no) 2.15 (typically) [4]
Electro-Optical Coefficients, pmV -1 r33 = 30.8; r31 = 8.6; r22 = 3.4;r51 = 28
Where the index shift is given by:
3
no
2
n � �r33E z � s33E
� � [7]
2
r33 is the linear electro-optic coefficient and s33 is the quadratic coefficient – which is negligible
with LiNbO3.
Also, with traveling wave electrodes, and Ti-diffused ridge waveguide optical
structures, LiNbO3 modulators have been demonstrated with bandwidths
greater than 100GHz[28] or with drive voltages

Table 3. Comparison of LiNbO3 and InP Modulators
Material
System
Ref Vpi
3dB Frequency Extinction Figure of
Response
(GHz)
Ratio (dB) Merit
(GHz/V)
Comments
LiNbO3 [32] Noguchi 1998. 5.1 105 20 20.6 Highest speed to date
[21] Sugiyama M et. al.
OFC 2002’
0.9 24.4 22 Lowest Drive voltage LiNbO3
Mitomi, 1998. 3 40 13.3
– suitable for SiGe driver
[23] Dolfi D. W. 1992. 12.3 50 3.58

A more desirable and less costly approach with respect to packaging would be
to integrate the modulator with the laser chip. Common modulator designs
employ either the Franz-Keldysh effect (FKE)[319] or the Multi-Quantum Well
(MQW) Quantum Confined Stark effect (QCSE)[339] with either electro-
absorption (EA) modulators or Mach-Zehnder phase modulators. Additionally,
electro-absorption modulators can be either designed as lumped circuit
components or in more sophisticated traveling wave designs to reach
bandwidths exceeding 50GHz[16,17]. Of course, each design has tradeoffs.
Key parameters to consider are drive voltage, bandwidth, optical power-
handling capability, bias stability, wavelength sensitivity and insertion loss. Due
to the decoupling of the laser functions from the modulation, external
modulation leads to a simpler tuning mechanism of the SGDBR, with a simpler
layout of the control circuits, and the prevention of chirping and mode-hopping.
Additionally, the extinction ratio can be far more desirable for external
modulation[214]. External modulators, however introduce loss to the
transmitted power due to their coupling and insertion losses. To compensate
for these losses optical amplifiers are typically added to the transmitter or the
laser must be very high power (>10dBm). This approach leads to increased
costs and complexity of the transmitter. Another inconvenience associated with
the use of semiconductor modulators is the wavelength sensitivity of their
extinction ratio which is present in both Franz-Keldysh(FKE) and Quantum
13

Confined Stark-effect(QCSE) based EAMs and Mach Zehnders. The different
types of modulators will be outlined in the following sections.
0.3 ELECTRO-ABSORPTION MODULATORS
EAMs are attractive due to their short length (75-400µm), ability to integrate
with lasers[31], and low drive voltages. Most discrete devices are based on the
QCSE – that offers superior attenuation to FKE devices. However, QCSE
devices although well suited to single wavelength lasers such as DFBs, can be
highly wavelength dependent without careful design – and not as desirable for
widely-tunable laser integration. FKE devices exhibit a lowering of the chirp
parameter with higher reverse biases - however achieving negative chirp is
difficult as demonstrated in fig. 5 without very high biases and large insertion
losses.
Chirp factor, alpha
5
4
3
2
1
1530 nm
1545 nm
1560 nm
0
0 0.5 1 1.5 2 2.5 3
Reverse bias (V)
Fig 5. Alpha parameter vs wavelength for a FKE EA-Modulator. Printed with permission from L.
A. Johansson
14

EAMs, particularly with high power integrated lasers, will dissipate large
amounts of power due to the photocurrent generated in the EAM. The thermal
management of integrated devices require careful material design, heatsinking,
and thick metal electrodes to dissipate heat[25].
Having said this, quantum well intermixed shallow well EAMs have shown
exceptional promise [228] in providing wideband operation with high bandwidth,
low drive voltage and negative chirp[34]. EAMs based on QCSE have shown
that the chirp parameter may be set negative/positive by adjusting the bias
voltage – of course increases in the bias results in more optical loss[34].
0.4 MACH-ZEHNDER MODULATORS
Mach-Zehnder modulators are a class of interferometric based modulators that
rely on the relative phase shift of one branch with respect to the other to
achieve partial – to full canceling of the signal at the output as shown in fig. 6.
15

Intensity
1
0.8
0.6
0.4
0.2
0
Voltage (V)
Fig. 6 Mach-Zehnder configuration and idealized light intensity output at the output.
In practice, particularly with the use of InGaAsP waveguides with compositions
corresponding to the emission wavelengths close to the operating wavelength,
this index change is accompanied by an absorption change as well – due to the
Kramers-Kronig relations.
The argument for using Mach-Zehnder modulators consists of a few reasons.
First of all, MZMs offer better power handling than EAMs since less
photocurrent is generated in most designs and the optical power is split into
two. High optical bandwidths (>10 GHz) are achievable, with fairly low drive
voltages Vpi

the device is ‘off’ without bias. The other advantage of these devices is their
higher wavelength independence which is beneficial for WDM systems.
Despite the above advantages, Mach Zehnder devices must be fairly long in
comparison with EAMs. The length of the modulator is an optimization
between the insertion loss and efficiency of the modulator. Most Mach-
Zehnder devices provide phase-modulation due to the linear electro-optic
effect. The devices shown in this dissertation make use of carrier based
effects and electric field based effects – that are fairly efficient by designing the
waveguide bandgap close to operating wavelength and doping the waveguide.
Doing this makes the device more efficient and compact at the expense of bias
dependent absorption loss and wavelength dependence as the device is a
cross between an EAM and typical MZ modulator. These losses can be
mitigated by integration of a laser. Unlike LiNbO3 modulators, high
performance InP based Mach-Zehnder modulators induce considerable loss
with reverse bias due to the Franz-Keldysh effect.
0.5 MACH-ZEHDNER BIASING APPROACHES
There are a number of different approaches to biasing MZ modulators as
shown in table 4. The simplest approach uses a single RF input on one
branch. The drive voltage requirements may be reduced by using a push-pull
bias scheme. By using both the normal data output and inverting output from a
17

modulator driver, the drive voltage requirement for each is cut in half with a
parallel push-pull bias scheme. In Chapter 3, lumped modulators are
presented using the above modulation approaches and the performance
explored with respect to efficiency and speed – and the relations to material
properties of the passivation dielectric and semiconductor materials.
Alternatively, by using a series push-pull configuration on the two modulator
sections, the bandwidth can be roughly doubled with the same drive voltage as
the single-sided case (see Chap. 4). The ultimate in figure-of-merit is achieved
with the dual RF push-pull modulation approach. This uses both RF inputs with
4 electrodes to produce two sets of series push-pull electrodes. This not only
doubles the bandwidth with respect to the single sided case – but additionally
requires half the drive voltage. For an integrated modulator with a high power
tunable laser – the insertion loss is not as much of a problem as the drive
voltage and bandwidth. These devices are also shown in chapter 4.
18

Table 4 MZ Bias Techniques
Single-
Sided
Modulation
Doublesided
Parallel
push-pull
Doublesided
Series
Push-pull
Dual RF
Series
Push-pull
Bandwidth Voltage Device
Length
Advantages Disadvantages
B Vpi L Simple
Configuration Easy
Not highest
figure of merit.
to get negative
B Vpi/2 L
chirp – harder for 0
chirp
Utilize both
inverting and non-
2 RF
more
inputs
inverting
from driver
outputs complexity
2B Vpi L Higher
better
speed,
wavelength
Requires
decoupling
sensitivity. Easy to and bias
get 0 chirp – more circuitry.
difficult to get Semi-
negative chirp insulating
substrate
necessary.
is
2B Vpi/2 2L Highest
merit
figure
–
of
for
4 electrodes.
Highest
integrated devices. complexity –
good isolation
is necessary
Chapter 4 examines the higher speed series push-pull devices. Although
lumped electrode devices operate at sufficient bandwidths to enable 10 Gbit/s
operation, even higher speeds are possible by taking advantage of traveling
wave effects[5,6,16,19,23]. These devices exhibit higher optical bandwidth,
and with improved transmission line characteristics, lower return losses (S11).
Finally Chapter 5 gives some comparison of performance – with respect to the
chirp and bandwidth of different designs. Additionally, a conclusion and future
work session explores the gains that could be made to the device with
improved doping, bias schemes, etc. as well as more complex PICs that could
be made such as photocurrent-driven wavelength converters.
19

0.6 TRAVELING WAVE DEVICES
Traveling wave modulators have electrodes that are transmission lines to
distribute the capacitance over the whole device length. For lumped electrode
devices, low drive voltages require long devices, however large bandwidths
require short devices.
Fig. 6 Scanning Electron Microscope image of traveling wave electrode device
Traveling wave devices such as shown in Fig. 6 are not limited by the RC time
constant so they may be made longer to achieve superior extinction and lower
drive voltages. The maximum length of the electrodes is limited primarily by
the optical and electrical propagation losses. Optical losses can be high if
doped waveguides are used 6 , and microwave loss can be high if highly doped
layers are underneath the contacts. TW structures benefit greatly by
decreasing the capacitance per unit length and lowering the microwave losses.
6 where 10cm -1 is typical for a SGDBR structure with 3�m wide ridge
20

This increases the modulator impedance and the microwave phase velocity. A
more extensive look at these designs is presented in Chap. 4.
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23

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24

Chapter 1
Device Integration
This dissertation demonstrates an integration platform to enable widely-tunable
laser functionality, semiconductor optical amplifier (SOA) technology, as well as
efficient and high-speed modulation all on one chip. In this chapter we explore
the laser, SOA, and modulator integration considerations. First we look at the
SGDBR laser design and its suitability for integration. Next the SOA design is
examined and finally the susceptibility of reflections is explored.
Fig 1-1 Integration Platforms for 1.55�m based photonic integrated circuits [128]
Printed with permission from Dr. Erik Skogen
25

Modulator-Laser integration is usually performed using the following
approaches:
1) Butt-coupling approach – allows for independent optimization of
device structures – however there is a need for selective area
growth. Also, more optical loss at the interface. Relatively
complicated [125].
2) QW intermixing – low interfacial mode mismatch. Allows for multiple
bandgaps due to differing amount of intermixing in the modulator and
passive regions. This is a maturing technology for the fabrication of
SGDBRs as developed by Skogen et al.[128]
3) Offset QW structure – where QWs are removed in passive regions
and Franz-Keldysh effect (Bulk effect) utilized in modulator sections.
Devices in this dissertation use this approach. See Appendix D.
26

1.1 WIDELY-TUNABLE LASER DESIGN
Over the last 20 years a few different implementations of monolithic-tunable
lasers have been fabricated including Y—cavity lasers, grating assisted
Table 1.1 Tunable Laser Technologies
Technology Tuning
range
(nm)
Fiber
Coupled
power
(dBm)
Companie
s
SMSR RIN Tuning Speed Integrated Amplifiers
or modulators
DBR [208] 17 Agere Ns-ms Easy
SGDBR 40-72 4/11 Agility/
Marconi
35-55dB -150dB/Hz Ns-ms Easy
SSGDBR 100 NEL
DS-DBR Bookham Easy - -OFC 2003
Y-Branch 46 13-14 Syntune 40 Ns-ms Not possible with
GCSR 40 3 ADC/
Altitune
Ns – ms
current implementation
OFC 2003.
Not possible
MEMS 20 .3 Bandwidth
Possible but very
VCSEL
electrically
pumped
–
9
difficult
MEMS >50nm >7mW Nortel,
1-10ms
VCSEL tuning achieva Cortek,
Possible not easy
Optically
ble with Cielo,
pumped
120mW
pump
Picolight,
Wavelength
selectable
DFB array
with MMI
DFB array
with MEMs
Tunable
External
Cavity
Diode lasers
(ECDL)
40 15 NEC/
Fujitsu
20 Santur
100 7 Iolon,
Intel
Princeton
Optronics,
Blue sky
Research
55 -150dB/Hz
27
Temp Tune Possible
Not possible

coupled cavity (GACC), grating coupled Sampled Reflector (GCSR) lasers,
Superstructure Sampled Grating Distributed Bragg reflector
(SSGDBR)[212,219], DS-DBR, and Sampled Grating Distributed Bragg
Reflector (SGDBR) lasers[207,216,217,218], and narrowly tunable DBR
lasers[208,209,211] as shown in table 1-1. For the work described here, a
SGDBR laser was employed as it has many advantages over other tunable
lasers. The simple holographic grating definition gives a wide tuning range
without requiring facets for operation leading to highly integrated functionality.
Also, there are no moving parts, wafer level testing is possible, and the
technology has been demonstrated with high volume manufacturing as only
one regrowth is required. Recently devices have been demonstrated
commercially demonstrating high output power [129] with fast tuning(35dB). Fig. 1. shows a SEM of an SGDBR device. While large tuning
ranges have been demonstrated [as high as 72nm] by using a large mirror
comb spacing difference – this is a tradeoff with power output and flatness over
the tuning range.
28

Figure 1-2 Scanning Electron Micrograph of SGDBR Laser
Detector
Rear Mirror
Phase
Gain
Front Mirror
All tunable transmitters in this work use SGDBR devices with the same design
parameters as shown in table 2-1. These parameters are fairly conservative –
as one may obtain higher power with less mirror periods on the front mirror.
Table 2-1 SGDBR Device Parameters
Front Mirror Burst width 4�m (x5) Typical threshold current
(mA)
27.5 mA
Front Burst Period 68.5�m Typical threshold Voltage 1.1V
Rear Mirror Burst width 6�m (x12) Max Power 20mW
Rear Burst Period 61.5�m
Gain Length 550�m
Ridge Width 3�m
29
.

The tunable laser structure includes two sampled grating mirror sections with
parameters as shown in table 2-1. The basic operation is based on the fact
that the SGDBR takes advantage of a tuning enhancement by using two
mirrors with slightly different mirror periods – which results in slightly differing
reflection spectra from the Vernier effect [207,220]. The two reflectivity spectra
are shown in Fig 1-3.
Fig. 1-3 Reflectivity spectra for front and rear mirrors.
The device relies on refractive index tuning with current injection into the front
and rear mirror sections – in which the wavelength can be tuned approximately
38nm with the design above as shown in Fig 1-4.
30

Fig 1-4 Wavelength as a function of biases on each mirror section
Typical IV and LI output plots are shown in Fig 1-5 for a untuned device. The
laser threshold for devices ranges from approximately 25-35mA for different
tuning biases. As current is injected into the mirrors – this adds optical loss in
the laser and increases the threshold current.
31

L [mW] CW
25
20
15
10
5
V [V]
L [mW]
S020714B_DIE#8_Device#2
0
0.5
0 20 40 60 80 100 120
I [mA]
Fig. 1-5. Typical LIV plot for a ridge SGDBR device
From Fig 1-6. one can see the ‘supermode’ boundaries as the mirrors are
tuned differentially.
Fig 1-6a. Output Power (mW) vs tuning on front and rear mirrors
Fig 1-6b corresponding Gain Voltage. Gain section = 100mA SOA = 120mA
32
3
2.5
2
1.5
1
V [V]

There are optimum mirror alignments in the center of each hexagon region for
a given phase section bias (in this case 0mA). The more drastic changes in
power correspond to wrap-around points where the device tunes from one side
of the tuning range to the other. These devices also demonstrate a low level of
spurious reflections in the longitudinal cavity as the devices tune normally. Fig.
1-6b shows the corresponding gain section voltage as the device is tuned.
This plot closely relates to the output power plot – and can be used in
electronic control circuitry to determine the best alignment of the mirrors for a
given WDM channel wavelength.
1.2 SEMICONDUCTOR OPTICAL AMPLIFIER
INTEGRATION
As seen in the previous section, the SGDBR lasers can output as high as
20mW. However, due to the fairly thin waveguide (0.35�m) and curved flared
output – the output mode does not couple well with lensed fiber. A mode-
converter would be desirable to increase the output power without introducing
more optical noise sources, however the integration adds complexity to the
processing and was not pursued in this work. Coupling losses can be as high
as �10dB for discrete SOA devices without mode converters[101]. In this
device, with only one output, the typical coupling losses range from 4-5dB.
33

Additionally zero bias insertion losses in the modulator which range from 4-7
(see Fig. 1-7) as well as bias induced losses make SOA integration desirable.
Fig 1-7 Unbiased insertion loss for a MZ Modulator for different wavelengths. Total length
1100�m (10.5cm -1 at 1550)
Semiconductor optical amplifiers were integrated[114] to improve the overall
power output and to even out the wavelength dependent power as lower input
optical power results in higher gain. The SOA was chosen to be before the
modulator so that the extinction ratio would not be degraded. SOA integration
provides for added functionality of the device. Not only higher output power is
now possible – but it can be used as a variable attenuator since it has high
dynamic range. Also it can be readily applied for power leveling the channels
across the wavelength range since it is an independent power control.
34

Additionally, SOAs have been used successfully as modulators, gates,
frequency converters, or detectors[200]. SGDBRs require a sufficient AR
coating [>10 -3 ][100] for consistent tuning characteristics. Integration of an SOA
and modulator requires an improvement in the reflectivity to 10 -4 -10 -5 to prevent
device degradation as described in the following section.
The most important measures of performance for linear applications are the
maximum gain, saturation power and maximum noise figure[106]. In general,
SOAs are not competitive with EDFAs because the noise figure is so high.
Typical relative noise figures of EDFAs to SOAs at 1.55um (3.1 vs 5.2). [101]
In the integrated device the SGDBR laser is a CW source – in which the SOAs
are biased to saturation – so the noise characteristics do not suffer much.
Devices 4-9 have 400µm and 500µm long SOAs before the MZ modulator.
The gain from these devices is shown in Fig. 1-8.
Fig 1-7 Gain at 1554um for a 400�m and 500�m long, 3�m wide SOA at T = 16C
35

For high power from the SGDBR (>10dBm), this means the SOA only
contributes approximately 6-7dB of gain at high SOA biases. This effectively
cancels out the insertion losses of the modulator.
1.3 DUAL SOAS
A number of different approaches have been explored to improve the
saturation power and gain of SOA devices. Gradual tapering of the waveguide
has been explored[103] as well as using multiple waveguides coupled using
MMIs to improve the gain saturation[105].
Fig. 1-8 DUAL SOA Design Layout
Saturation power is generally limited by the small cross sectional active area
required for single-mode operation. By using N waveguides in parallel the
saturation power is improved by 10log (N) dB with respect to a single
waveguide device without degrading the noise figure. The devices presented
36

in this dissertation have Dual SOAs ranging from 350�m each to 575�m each.
The SOA region was tapered out to 3.5�m wide ridges in the SOA region from
2.5�m wide ridges in the modulator region.
The gain of the Dual SOAs was measured with one SOA reverse biased – then
compared the gain with one biased to 150mA and the other SOA bias varied.
Clearly thermal crosstalk affects the SOAs so that the gain is reduced 1-2dB
due to heating from the adjacent SOA that is 16�m away. Overall the
saturation power is improved – and the Dual SOA approach is beneficial.
These heating effects are beneficial to the efficiency of the modulator – as
shown in more detail in chapter 5.
37

Fig 1-9a&b Output power for Dual SOA with 575µm electrodes 13.5um transparency current –
second SOA reverse biased
38

Fig 1-10a&b Output power and gain with SOA #2 biased to 150mA [575µm SOA]
39

1.4 OPTICAL FEEDBACK AND REFLECTION
Highly integrated devices require the minimization of optical feedback in order
to reduce unwanted chirp, lasing of the SOA, gain ripple, higher noise figure,
and inter-modulation distortion. Reflections lower the gain bandwidth and
output saturation power. The design needs to consider reflections due to the
facets and active/passive interfaces[127], waveguide design, tapers, and multi-
mode interference (MMI) devices within the Mach-Zehnder modulator.
Reflections at the facets are minimized by flaring and angling the waveguide at
the outputs and backside absorber facets. This reduces the requirements of
the AR coating by an order of magnitude – often providing continuous tuning
even without an AR coating. In order to operate properly, SOA based devices
require sufficiently low Anti reflection (AR) coating reflectivity - generally
accepted as < 10 -4 . A multi-layer AR coating is used to achieve greater than
10 -4 reflectivity back into the optical cavity.
Active/Passive interfaces are angled to minimize reflections due to the index
discontinuity. Additionally, the MMI lengths are optimized for minimum
reflections and are tapered so that reflections are not coupled back into the
laser cavity – mostly important in the ‘off’ state. Due to the nature of the
multimode interference splitter, changes in the widths, growth thicknesses etc.
will influence the light propagation. Biasing of sections on top of the MMIs will
allow for tuning the power splitting ratio[608] – important in MZ design for the
40

chirp and extinction ratio tuning. Note however, that the imaging length is
modified potentially leading to large reflections when not optimally biased. A
safer approach is to adjust the different branch gain or loss in the two
branches. Finally, the waveguide design is weakly-guided throughout the
structure. Also, as demonstrated in [124] shallow ridge technology in the MZ
lowers the reflections in the device considerably in comparison with previous
deep ridge etched MZs[336, 337,344]. It has been shown that when parasitic
reflections are generated in the MZ – the extinction ratio is degraded when the
chirp is best[306].
The optical feedback in the device can be explored using a few different
approaches. Optical low coherence reflectometry (OLCR) is a good approach
to measure the reflections at each interface in the structure[124]. Also, one
can look at the tuning modemap as a function of biases on the rear detector
and front detector biases. An example of this is shown for a device with two
575µm long SOAs in the branches of the Mach-Zehnder in Fig. 1-11a with poor
AR coating.
41

Fig 1-11a Device #10 Comparison of Dual SOA with significant optical feedback with SOAs
biased to 80mA. Fig 1-11b demonstrates reflection reduction by reverse biasing front and rear
detectors -4V
Reverse biasing the active rear absorber and the passive front detector 7
reduces the feedback from the facets – thereby improving the tuning map
profile as shown in fig 1-11b.
7 As shown in Fig 1-12
Fig 1-12 Front detector reverse biased to absorb off-state light
42

Another indication of the optical feedback in the device is the degradation of
the linewidth.
1.5 LINEWIDTH MEASUREMENTS
Linewidth arises from coupling between variations of phase and intensity. The
linewidth for a conventional Fabry-perot, DFB, or SGDBR is given by:
where the threshold gain is given by:
and the mirror loss:
2
2
vg hvgthnsp�
m ( 1��
)
� � �
[1.1]
8�P
gth � �i ��
�
1
m
[1.2]
1 1
� m � ln
[1.3]
2L
R ( �) R ( �)
1
P is optical power, � is the linewidth enhancement factor, L is the cavity length, and R1 and R2
are the reflection coefficients for the mirrors. vg is the group velocity, nsp spontaneous factor, hv
is the optical energy
In general, linewidth decreases with increased laser power and increases with
tuning of the mirror and phase sections – due to misalignment of the
mirrors[29].
43
2

Accurate measurement of linewidth in multi-section devices requires careful
control. Battery powered sources are preferable to source less noise – and
also biasing that shorts high frequency noise to GND are required – particularly
when biasing sections highly sensitive to noise such as the phase section.
Also, the setup requires sufficient optical isolation(>50dB). In order to explore
the susceptibility of the device to optical reflections, a linewidth measurement
setup that was used as shown in Fig 1-13.
SGDBR-MZ
LiNbO 3
2GHz
AMP
Fig. 1-13 Linewidth measurement setup Delayed Self-Heterodyne Method
The output light is inserted in a LiNbO3 MZ modulator in order to frequency shift
the linewidth away from 0Hz reducing the noise in the measurement. This
shifted Full-Width Half Max (FWHM) linewidth is double the true linewidth of the
laser assuming that the lineshape is near Lorentzian in shape due to the
heterodyning of the two signals. Then the signal goes into another
interferometer – in which one arm has a large delay (1km) which must be
longer than the reciprocal of the linewidth in order to have a small
measurement error. This means a system with 3.5�s delay will be able to make
44

linewidth measurements as low as 225kHz. With a 25�s delay one can
measure to 30kHz.
The linewidth of a commercial SGDBR-SOA was compared with a device with
a 400�m SOA before the MZ (Device #8) and a device with Dual 350�m long
SOAs in Fig 1-14. Each was compared at the same bias point of 150mA on
the Gain section and SOAs.
Fig 1-14 Linewidth measurement comparing Commercial SGDBR-SOA device to Device 8
(single 400µm SOA) and Device 1 (Dual SOA device with 350�m long SOAs)
As can be seen, there was very little difference between the linewidth of the
three devices. This actually is a coincidence as the linewidth varied
45

approximately from 4-9 for different Gain and SOA biases. However, this
demonstrates that the external cavity (MZ) does not adversely affect the laser’s
linewidth with optical feedback.
Although this demonstrates the lack of optical reflections in the device, in order
to look at the quality of signal under transmission, it is helpful to look at the
linewidth as a function of frequency. The noise is not constant with frequency
for the laser 8 and consists of non-white and white components. A relaxation
oscillation induced noise resonance is found between 1-6GHz depending on
the output power. Also, phase noise below 500MHz is fairly large due to carrier
fluctuations in the tuning sections from noisy current sources. This low
frequency non-white phase noise has been shown to not affect the
transmission performance of SGDBR lasers after fiber[126]. As these devices
are most likely to be used in a 10Gbit/s system where the phase noise to
intensity noise conversion efficiency is greatest at 5GHz, the noise at higher
frequencies is more important than at low frequencies[506]. The integration
time affects the measurement – and ideally the linewidth is measured as a
function of frequency as shown in [126]. The instantaneous linewidth is much
narrower than shown in this measurement – as the line tends to drift over time.
This is why the seemingly large linewidths shown here are not detrimental to
practical applications – as the white noise at higher frequencies are the most
important to the transmission properties of the laser. Nonetheless, for a typical
8 See RIN section 1.6
46

SGDBR measured using the coherent optical frequency discriminator
technique[131], linewidths are usually measured between 1-2 MHz[126].
1.6 RELATIVE INTENSITY NOISE
The relative intensity noise (RIN) is a measure of the quality of laser diodes for
broadband digital or analog systems. It is defined as
2
� �P
�
RIN � dB / Hz . [1-4]
2
P
The numerator is the mean square optical intensity fluctuation at a specified
frequency and P is the average output power. Since the ratio of the optical
powers squared is equivalent to the ratio of the detected electrical powers, this
equation can be written as
N electrical
RIN � dB / Hz
[1-5]
P ( electrical)
avg
Nelectrical is the power-spectral density of the photocurrent at a specified
frequency and Pavg is the average power of the photocurrent. It can also be
expressed as:
2
N
4
�R
2
1/
� � � � 2 2h�
2q
RIN � 16�
( ��)
ST
H ��� � �
[1-6]
P �P
where H(w) is the modulation transfer function of the laser, � is the electrical frequency, �R is
the relaxation resonance frequency, � is the damping factor, (��)ST is the modified Schawlow-
Townes linewidth, ��N is the differential carrier lifetime, , h� is the optical energy, P0 is the optical
power output from the laser, � is the photodiode responsivity, and Pfiber is the optical power
coupled into the fiber.
47
0
fiber

The RIN was measured for a device with a single 400µm SOA and Dual SOAs
(350µm)
Fig 1-15a RIN for single 400�m SOA Gain section bias varied [SOA 120mA 1554.3nm]
48

Fig 1-15b RIN for Dual 350�m SOA device [SOA1&2 100mA]
As can be seen in Fig 1-15a and 1-15b, the RIN measured for devices with
single and dual SOAs are very similar. In both cases the RIN exceeds -
140dB/Hz, a specification desired for commercial SGDBR lasers. The peak of
the noise spectrum indicates a parasitic free means of measuring the
modulation bandwidth of the directly modulated SGDBR. The actual bandwidth
is limited by carrier transport[113].
49

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53

Chapter 2
MOCVD GROWTH AND FABRICATION
In this chapter, we will look at some of the epitaxial layer growth considerations
regarding the Metal Organic Chemical Vapor Deposition (MOCVD) growth of
tunable transmitters. Growth consists of a base epitaxial structure which
undergoes processing steps to etch off quantum wells and define gratings in
certain areas. Then a regrowth is performed to grow an upper InP/InGaAs
cladding over the whole structure. An optimized doping profile is critical for
high performance in both the laser and modulator sections. The basis for
characterization of the material is discussed – with the use of x-ray, PL, and
SIMs measurements. The fabrication steps are also outlined in Section 2.8.
The MOCVD technique is used to grow high quality single crystal epitaxial films
on substrates using metal-organic precursors that are transported to a heated
substrate (615C – 650C) with a carrier gas (H2). The work here was done
using a Thomas-Swan 2” horizontal flow rotating disk reactor which has very
good deposition uniformity as shown in fig 2-1.
54

Average �1540nm
Fig 2-1 Micro-Photo-luminscence(PL) Intensity and Peak wavelength uniformity across base
structure wafer S020215C
MOCVD growth of In1-xGaxAs1-yPy quaternary is very important for fabrication of
opto-electronic devices such as lasers, modulators, and detectors – all
fundamental in optical fiber communication systems. Typically this involves the
highly toxic Arsine (ArH3) and Phosphine (PH3) precursors. At UCSB,
tertiarybutylarsine (TBAs), and tertiarybutylphosphine (TBP) Group V
precursors are used as they are considerably safer, and have been shown to
be superior with respect to enabling very low III/V ratios and low growth
temperatures as well as possess similar purity. For this work, the Group III
precursors are trimethyindium(TMIn) for Indium, trimethylgallium (TMGa) for
Gallium. The material is doped using both disilane gas for silicon n-type layers,
and DiethylZinc for Zinc p-type layers.
55

The basic reactions describing the growth of InP are:
In( CH ) ( v)
TBP(
v)
���
InP(
s)
� 3CH
� TB [2.x]
3
3
� 4
where during InGaAsP growth, the group III precursors decompose as follows:
Ga( CH ) ( v)
� �� Ga(
CH ) � CH
[2.x]
3
3
3
2
3
3
2
Ga( CH ) ( v)
� �� Ga(
CH ) � CH
[2.x]
In( CH ) ( v)
� �� In(
CH ) � 2CH
[2.x]
3
3
3
3
In( CH )( v)
� �� In � CH
[2.x]
Since the TMIn source is a solid source, two bubblers are placed in series to
insure a constant source. For the liquid gallium source (TMGa), a dilution line
is utilized to achieve the desired range of material concentrations.
2.1 SEMICONDUCTOR EPITAXIAL STRUCTURE
Although SGDBR lasers have been fabricated using quantum well
intermixing(QWI)[33] and butt-joint regrowth techniques, the offset Multiple
Quantum Well (OMQW) approach has been a standard approach for the
fabrication of photonic integrated circuits[221-224]. This is due to the ease of
removal of QWs in passive regions without significant index discontinuity or
complicated regrowth processes. Next we will look at the quantum well design
and epitaxial layer structures based on both conducting sulfur-doped and semi-
insulating Fe-doped wafers.
56
3
3
3
3

2.2 QUANTUM WELL DESIGN
The quantum wells should be designed to efficiently contain the carriers and
not impede the transport across the structure. For well designed barriers, the
thermionic emission time is much larger than the diffusion and capture
times[312]. “Thermionic emission and tunneling are competing processes and
the faster one will dominate”. For barriers less than 5nm, tunneling tends to
dominate the carrier transport across the barrier. However, for wider barriers
(8nm in this case), hole transport is associated with thermionic emission while
electron transport is done mainly by tunneling. For sufficiently wide barriers
both electrons and holes are dominated by thermionic emission[312]. The
MQW structure relies on the barriers to be thick enough to provide 2-D carrier
confinement in the quantum wells – and prevent coupling of the quantum well
states – which leads to a broadening of the quantized energy levels[312].
It has been shown[225], that significant improvements in both differential gain
and threshold current density can be achieved by compressively straining the
quantum wells – at least up to approximately 1%. In this work a MQW stack of
7 Quantum wells with 8nm barriers and 6.5nm wells were used as shown in fig
2-2. Without any strain compensation in the barriers and waveguide, one is
limited to approximately 6 wells 6.5nm wide with 8.0nm barriers before
reaching the critical thickness. In the current design, the quantum wells are
57

grown with a constant x composition for the barriers and quantum wells to allow
fast switching between mass flow controller during the growth of the QWs and
the barriers. Constant x growth also has the advantage that since the growth
rate is governed by the group III flow rate, both the barrier and well grow at the
same rate. In this structure, the waveguide and barrier are grown slightly
tensile to compensate for the highly strained wells – allowing for more wells to
be grown before reaching the strain critical thickness. The target compositions
for the structure are shown in table 2-1.
Table 2-1 In1-xGaxAs1-yPy Compositions
Material In[%] As[%] Pl peak Eghh
Perpendicular
Strain
Barrier 73.5 51.3 1194 1209 -4038 (-0.3%)
QW 73.5 84.5 N/A N/A 18139 (1%)
SCH 1.226
Waveguide
76.76 50.4 1210 1226 0
1.4Q 65.14 73.77 1380 1400 -800
2.3 CONDUCTING SUBSTRATE BASE STRUCTURES
Base structures for SGDBRs have been grown for some years now – and the
best growth conditions refined considerably. These growths use an indium
pure component flow of 0.4sccm, with changes in the gallium flows to achieve
the desired quaternary compositions as governed by the segregation
coefficient Kseg. The growth was held at a constant pressure of 350 Torr.
However, the temperature of the growth is changed for the InP(615C) and
58

InGaAsP layers (650C) for best material quality. This is achieved using a three
zone infrared lamp system with a center zone at 650C and adjoining regions at
630C to improve the temperature uniformity.
As the ‘passive’ regions have the QWs etched off, we have two different layer
structures after regrowth. Fig. 2-2 shows layer structures in an active and
passive section simultaneously. A 10nm un-intentionally doped (uid)-InP stop
etch layer is grown on top of the waveguide in order to facilitate the QW wet-
etch. The base structure is grown with a Zn setback from the quantum wells
that takes into account the subsequent regrowth diffusion(80nm). This also
means that the regrowth on the active regions is on a doped material instead of
a ‘grown junction’.
Fig 2-2 Sulfur doped substrate epitaxial layer structure
59

2.4 GROWTH CHARACTERIZATION
The different layers are characterized by determining the bandgap energy from
both room temperature photoluminscence data and the lattice parameter ao
from double crystal x-ray diffraction data (XRD)[102]. According to Vegard’s
law, we can relate the binary lattice constants to the unstrained In1-xGaxAs1-yPy
quaternary lattice constant as follows[102]:
ao = 6.0584 – 0.405x – 0.190y – 0.0123xy. [2-1]
Fig. 2-3 Typical x-ray rocking curve for base structure S010406C
60

The lattice mismatch between layers can be calculated from x-ray by
examining the difference between the substrate peak and the quaternary peaks
as described by [204]:
�a� � � tan���
� cot���
as
�a
as
||
� cot���
� cot���
[2.2]
[2.3]
where � corresponds to the angle between the (hkl) plane and the (001)
reflection plane and � is the bragg angle for the hkl reflection. This lattice
mismatch is given by[205]:
�a
�
a C
o
11
C11
� 2C
12
��a
�
� ao
where ao is the lattice constant, C11 and C12 are elastic stiffness constants.
�
�
�
�
[2.4]
As many of the layers are strained in the multiquantum well structure, we must
look also at the photoluminescence data as in fig 2-4.
61

Fig 2-4 One dimensional Photoluminescence plot showing both the QW peak and WG peak
An empirically modified version of Vegard’s law expresses the band gap as a
function of In1-xGaxAs1-yPy quaternary composition [102]:
E g ( 295K)
� 1.
35 �1.
42x
� y � 0.
33xy
� ( 0.
73 � 0.
28y)
x(
1�
x)
eV [2.5]
� ( 0.
101�
0.
109x)
y(
1�
y)
� 0.
05 xy(
1�
x)(
1�
y)
The layer compositions and thickness can be fit using a model – using BEDE
software.
62

2.5 SEMI-INSULATING SUBSTRATE GROWTH
By growing a similar base structure to that discussed in the previous section –
on a Fe doped semi-insulating(SI) substrate as seen in Fig. 2-5, this enables
some of the series push-pull electrode structures discussed in chap. 4.
0.1µm 1E19 cm -3 Zn-pInGaAs
1.8µm 1E18 cm -3 Zn-p cladding InP
0.05µm setback uid InP
25nm 1.226Q uid-SCH
7 uid 6.5nm QWs 8.0nm Barriers
10 nm uid-InP stop-etch layer
0.32µm 1.4Q InGaAsP waveguide
1.8µm 1e18 cm -3 Si InP Buffer
0.05µm n-InGaAs contact layer
0.5µm 1e18 cm-3 Si InP n-buffer
100µm Fe-doped semi-i -nsulating
substrate
Fig. 2-5 Semi-insulating substrate based epitaxial layer structure
Using a semi-insulating (SI) substrate lowers the RF dielectric losses and
allows isolation between different n-contacts on the device. The n-contact
layer is typically either a InGaAsP quaternary layer with emission bandgap in
the 1.1-1.4Q range if it is required to be close to the waveguide – particularly
important if He implanting is required to isolate n-doped regions. For this work,
an InGaAs layer was used for the best contact resistivity – and placed 1.8�m
away from the waveguide so that there is only a very minimal overlap between
63

the optical mode and this highly absorptive layer. The use of an InGaAs layer
for the n-contact has been shown to improve the extinction ratio since substrate
modes are absorbed to a greater degree[220].
2.6 REGROWTH
The regrowth quality over gratings is highly dependent on surface preparation,
grating etch depth 9 and regrowth conditions. Low damage RIE (at low power
≈ 22Watts) and proper growth conditions for grating regrowth are very
important to reduce defects at the growth interface[228]. Gratings are defined
using a holographic setup 10 and etched in sampled regions in each mirror
section using a Reactive Ion Etch (RIE) system as shown in fig 2-8a&b.
Fig. 2-8a Atomic Force Microscope (AFM) image of grating burst before grating regrowth
Fig 2-8b Field Emission Scanning Electron Microscope (FESEM) image of dry-etched gratings
before regrowth
9 Typically 75nm
10 See Beck Mason Dissertation [224]
64

Previously fabricated gratings used a relatively high-power etch (500V) that
reduced the photoluminescence intensity by a factor of 100. By reducing the
RIE power and using a 1min sulfuric acid dip to remove residual organics, this
photoluminescence intensity can be improved to approximately 34% when
compared with non-grating regions[228]. Grating regrowth quality is superior if
the growth in initiated immediately as the growth temperature is reached
(615C) so that the gratings do not decompose. The growth cycle was modified
similar to the process shown in [212], to remove the bake step and reduce the
initial growth rate (1/16 growth rate) as well as use both a phosphorus (1Torr)
and arsenic (0.07Torr) overpressure during the ramp-up to growth that appears
to reduce the exposed InGaAsP desorption and increase the oxide/H2O
desorption resulting in improved growth over quaternary material surfaces.
Fig 2-9a Regrowth over gratings Poor surface preparation
Fig 2-9b Regrowth over gratings with good surface preparation and TBA with heatup
A more extensive look at the photoluminescence data under different regrowth
conditions is published in [228]. When done successfully, it is difficult to see in
65

an optical microscope where the grating bursts are after the regrowth is
performed as shown in Fig 2-9.
2.7 Zn DOPING OF InP and InGaAsP
The device doping profile needs to take into consideration the requirements of
the laser and that of the modulator section. The laser requires high Zn doping
in the cladding to improve the injection efficiency and contact resistivity,
however, this needs to taper off near the waveguide to prevent excessive free
carrier losses. Doping the waveguide with silicon helps the transport time in
the laser regions. However, control of the doping profile is critical particularly in
the modulator region where it governs both the high speed and efficiency
performance 11 . The doping profile used in the devices shown in this
dissertation is shown in section 3.1. The capacitance of the structure is
improved with a large intrinsic region and the bandwidth will be less influenced
by the reverse bias, however, this is a tradeoff with transport properties through
the structure and the efficiency as there is less electric field overlap with the
optical mode. As Zn is used as the p-type dopant in this work, understanding
of the diffusion mechanisms is important.
11 See Chapter 3 for details
66

Due to the high diffusion constant of Zinc, control of the p-doping profile is
particularly crucial. Not only does excessive P-doping of the waveguide add
considerable absorption loss – this also adversely affects the modulator
efficiency.
It is thought that the following two mechanisms account for the interactions of
Zn diffusion.
Frank-Turnbull mechanism
m�
Zni � VIn
� Zns
�
� ( m �1)
h
[2.6]
An interstitial Zn finds it’s way to an Indium vacancy site and forms a
substitutional Zn atom and a hole. M is the charge state (+2).
Kick-out mechanism
A charged interstitial Zn atom kicks out an Indium atom from its lattice site to
form a substitutional acceptor via the reaction
m�
Zni � VIn
� Zns
� I in � ( m �1)
h
�
67
[2.7]

For this reason the doping incorporation is highly dependent on the III/V ratio
and phosphorus overpressure – essentially governing the number of vacancies
in the material.
Combination of the interstitial Zn with phosphorus vacancies – often yields
neutral complexes – for this reason we have high Zn incorporation – but it is
not all electrically active. Zinc diffusion in InP thought to be dominated by
highly mobile Zn interstitals that are in equilibrium with substitutional Zn. The
diffusivity of the interstitials is on the order of 1 million times larger for the
interstitials. However, there are far more substitutional Zn impurities in
comparison with interstitials. The incorporation and activation of p-type dopant
Zn are elevated on the B and planes. The n-type dopants are
suppressed on these planes[101].
The Zn doping profile is shown for two different device runs using a Secondary
Ion Mass Spectroscopy (SIMS) technique. Run #1 and run #2 have vastly
different laser and modulator properties. The first case (fig.1.1A) has very high
bandwidth due to the large intrinsic region but poor tuning characteristics due
to added resistance in the tuning regions that heat considerably when current is
injected. The second case (fig 1.1B) has Zn doping extending into the
waveguide – that although gives better carrier transport, has higher free carrier
losses.
68

Figure 2.6A&B Secondary Ion Mass Spectroscopy for run#1 and run #2
SIMS work by Charles & Evans
As shown in fig. 2.6b, if the waveguide is doped p-type by Zn diffusion during
the regrowth we find that the effective area of the PIN junction is enhanced
considerably. During growth, since Zn readily diffuses and the quaternary is
more easily doped than InP, Zn tends to segregate at the hetro-interface
between the InP and InGaAsP waveguide. This highly conductive layer acts as
the top side of the capacitor – raising the capacitance as high as 5-10x for the
device as illustrated in fig 2-7a. This problem can be mitigated by dry-etching
off this highly conductive layer �100nm in the InGaAsP waveguide of the
modulator sections to improve the bandwidth as shown in fig 2-7b.
69

Fig. 2-7a Zn doped InGaAsP waveguide layer after dry/wet etch of shallow-ridge waveguide
structure
Fig 2-7b Passivation etch removes Zn-doped region in the modulator sections
This etch is done by RIE at low power to minimize roughness of the waveguide
– and the scattering losses which would result. Etching the waveguide makes
the ridge more confining, however can make the device more susceptible to
reflections.
2.8 TRANSMITTER FABRICATION
Fabrication of the tunable transmitters involves the steps as shown in table 2-2.
More detailed procedures are shown in Appendix C-Process. Integration of a
semiconductor optical amplifier does not add any additional steps as the SOA
region structure is the same as the gain section. However, the integration of a
high-speed modulator does add a few more steps related to the low k dielectric
and topside n-metal contacts. Although topside n-metal contacts are not
strictly necessary on S-doped substrates, they do allow for direct radio
frequency (RF) probing of the devices – without the necessity of a well
70

designed RF carrier or influences of wirebonds/ribbon bonds. Because of this,
Ni-AuGe-Ni-Au n-metal was deposited on both types of substrates. Also,
backside Ti/Pt/Au n-metal was deposited on the thinned wafers to facilitate
soldering – which has much better heat conduction than thermally conducting
epoxy.
Table 2-2 Fabrication Steps � Necessary � Desirable �Not necessary
STEPS BASIC S-Doped SI- Description
SGDBR Substrate Substrate
SGDBR + SGDBR +
Modulator Modulator
Active/Passive � � � Remove QWs in
passive regions
Sampled
Gratings � � � Grating Bursts and
holographic gratings
defined
Regrowth � � � Upper cladding is
grown over gratings
and active regions
Dry/Wet Ridge
Etch Ridge � � � Define ridge
waveguide structure
Passivation
Etch � � � Important for high
speed modulation
BCB/Dielectric
Deposition � � � Low k dielectric to
reduce capacitance
N-Metal Etch � � � Etch down to buried
InGaAs layer
N-Metal
Deposition � � � Ni Ge-Au-Ni Au
Contacts to topside
71

p-InGaAs Etch � � � Remove the InGaAs
conductive region
between sections
Isolation p+
Implant � � � Isolate the P-regions
between sections
Modulator via � � � Open through the
BCB and SiNxOy
layers for p-contact
Laser Contact
via � � � Open through the
SiNxOy for the pcontact
P-Metal � � � TiPtAu Metal
Deposited
Thinning � � � Required for
cleaving
Backside
Metal � � � N-contact or
soldering layer
Total Steps 10 13/15 14/15
The addition of the high speed modulator adds a few more process steps (see
total steps for each type) – however the yield of devices is not compromised
excessively. Detailed process development for high speed modulators is
detailed in chapter 3.
72

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[200] Chu SNG, Logan RA, Geva M, Ha NT. “Concentration dependent Zn
diffusion in InP during metalorganic vapor phase epitaxy.” Journal of
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[201] Berger P.R., S.N.G. Chu, R.A. Logan, Erin Byrne, D. Coblentz, James
Lee Ill, Nhan T. Ha, N. K. Dutta, “Substrate orientation effects on dopant
incorporation in InP grown by metalorganic chemical vapor deposition” J.
Appl. Phys. 73(8), 15, April 1993.
[202] Flemish J. R., H. Shen, K.A. Jones, M. Dutta, “Determination of the
composition of strained InGaAsP layers on InP substrates using
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(4), 15 August, 1991.
[203] Li E.H., “Material parameters of InGaAsP and InAlGaAs systems for use
in quantum well structures at low and room temperatures”, Physica E 5
(2000) 215-273.
[204] Matsui J., K. Onabe, T. Kamejima, I. Hayashi, “Lattice Mismatch Study of
LPE-Grown InGaPAs on (001)-InP Using X-Ray Double-Crystal
Diffraction”, J. Electrochemical Society. Vol 126, no4, April 1979, pg 664-
7.
[205] Asai H., K. Oe., “Energy band-gap shift with elastic strain in GaxIn1-xP
epitaxial layers on (001) GaAs substrates”, J. Appl. Phys. 54(4) April
1983.
[206] Swaminathan V., C. L. Reynolds Jr. , M. Geva, “Effect of Zn on the
electro-optical characteristics of metalorganic chemical vapour deposition
grown 1.3um InGaAsP/InP lasers” Electron. Lett. Vol 32., No.7, Mar. 28,
1996
[207] Chen C -H, U. M. Gosele, T.Y.Tan, “Dopant diffusion and segregation in
semiconductor heterostructures: Part 1. Zn and Be in III-V compound
superlattices” Applied Physics A, 1999.
[208] Camargo Silva M.T., J. E. Zucker, L. R. Carrion, C. H. Joyner, A. G.
Dentai, “Growth Optimization for p-n Junction placement in the integration
of Heterojunction Bipolar Transistors and Quantum Well Modulators on
InP” IEEE. J. Quantum Electron. Vol.6, No.1, Jan., 2000.
73

[209] Hornstra J., W. J. Bartels, “Determination of the lattice constant of
epitaxial layers of III-V Compounds” J. Crystal Growth 44(1978)513-417.
[210] Bensaada A., A. Chennouf, R.W. Cochrane, J.T. Graham, R. Leonelli,
R.A.Masut, “Misfit strain, relaxation, and bandgap shift in GaxIn1-xP/InP
epitaxial layers” J. Appl. Phys. 75 (6) 15 March 1994.
[211] Van Geelen A, T.M.F. de Smet, T. van Dongen, W.M.E.M van Gils, “Zinc
doping of InP by metal organic vapour phase diffusion (MOVPD), J.
Crystal Growth 195(1998) 79-84.
[212] Franke D., H. Roehle, “Highly reproducible and defect-free MOVPE
overgrowth of InGaAsP-based DFB gratings” J. Crystal Growth,
170(1997) 113-116.
[213] Belenky G.L., C.L. Reynolds Jr., D.V. Donetsky, G. E. Shtengel, M.S.
Hybertson, M.A. Alam, G. A. Baraff, R.K. Smith, R. F. Kazarinov, J. Winn,
L.E. Smith, “Role of p-Doping Profile and Regrowth on the Static
Characteristics of 1.3um MQW InGaAsP-InP Lasers: Experiment and
Modeling” IEEE. J. Quantum Elect., Vol. 35, No. 10, Oct. 1999.
[214] Grinberg A.A., M. A. Alam, S. K. Sputz, “Modeling of the
photoluminescence in Multi-quantum well Heterostructure Laser wafers”,
IEEE. J. Quantum Elect. Vol. 35, No. 1, Jan. 1999.
[215] Schroeter-Janssen H, Roehle H, Franke D, Bochnia R, Harde P, Grote
N. “Comparison of MOVPE-based Zn diffusion into InGaAsP/InP using
H/sub 2/ and N/sub 2/ carrier gas.” Elsevier. Journal of Crystal Growth,
vol.221, Dec. 2000, pp.70-4.
[216] Mei XB, Loi KK, Chang WSC, Tu “CW. Improved electroabsorption
properties in 1.3 mu m MQW waveguide modulators by a modified
doping profile.” Elsevier. Journal of Crystal Growth, vol.175-1762, May
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[217] Dong-Ning Wang, Venables D, Waltemyer D, Lentz J. “Investigation of
p-n junction and dopant profiles in InP-based laser by low voltage SEM.”
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[218] Swaminathan V, Reynolds CL Jr, Geva M.”Zn diffusion behavior in
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74

[219] Cheng-Yu Tai, Seiler J, Geva M. “Modeling of Zn diffusion in
InP/InGaAs materials during MOVPE growth. “ Conference Proceedings.
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Materials (IPRM'99) (Cat. No.99CH36362). IEEE. 1999, pp.245-8.
[220] Rolland C., G. Mak, W. Bardyszewski, D. Yevick, “Improved Extinction
Ratio of Waveguide Electroabsorption Optical Modulators Induced by and
InGaAs Absorbing layer”, J.of Lightwave Tech., Vol. 10, No.12, 1992.
[221] Masanovic M.L., V. Lal, J. S. Barton, E. J. Skogen, L. A. Coldren, and
D. J. Blumenthal, "Monolithically integrated Mach-Zehnder
interferometer wavelength converter and widely tunable laser in InP,"
IEEE Photonics Technology Letters, vol. 15, pp. 1117-19, 2003.
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[223] Fish G., “InGaAsP/InP based photonic integrated circuits for optical
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31, pp. 1193-1200, 1995.
75

Chapter 3
Lumped Modulator Designs
A number of different authors have fabricated lumped Mach-Zehnder
modulators, mostly using MQW structures[300-4]. For this work Franz-Kelydsh
bulk quaternary waveguides were used instead of MQWs as they tend to have
less wavelength dependence and lower optical propagation loss 12 . DBR[338-9],
DFB[306-8] and now SGDBR[309] laser integrated Mach-Zehnder modulators
have been explored as well. In this chapter, we will look at the factors that
influence the modulation efficiency, optical bandwidth, and insertion losses.
12 Fiber to fiber losses 31-40dB reported in [303]
Fig 3.1 Lumped electrode Mach-Zehnder Modulator
76

3.1 DEVICE EFFICIENCY
Mach-Zehnder modulators rely on a change in the electric field and carrier
density to change the absorption and phase in one branch of the MZ
modulator. In this work, a PN junction is formed in the first �80nm of the
waveguide by diffusion of Zn during the regrowth 13 . This gives an electric field
profile as given in fig. 3-2 for a waveguide doping level of 3e17cm -3 Si.
Fig. 3-2 Electric field profile and indexing structure for waveguide as a function of bias for
device doping structure
The device efficiency would be improved further by moving the Zn diffusion
front into the middle of the waveguide. However, this would mean more free
13 Fig 2.6a&b
77

carrier absorption due to the Zn overlap with the optical mode, and a deeper
passivation etch as outlined previously in fig. 2-7a&b. Unfortunately, this
optimization approach also maximizes the Franz-Keldysh absorption. If less
wavelength dependence is desired, it is best to use the LEO effect along with
carrier based effects with a design that improves the overlap factor (Γ) by using
a lower doping level in the waveguide – effectively spreading out the electric
field profile.
Ideally, the modulator biases will be as low as possible – to lower the
propagation loss through the device. In this work, we have a doping profile as
shown in Fig 3-3. This doping profile was chosen as a compromise between
having reasonable bandwidth and the high efficiency associated with a PN
junction and it is compatible with typical SGDBR processing.
Fig 3-3 Doping profile for device structure in passive regions
78

The semi-insulating substrate based devices have a similar structure in the
waveguide with a 0.1µm 1E18 Si doped InGaAs contact layer, and 0.5µm 1E18
Si-doped n-InP layer instead of the Sulfur doped substrate shown in fig. 3-3.
The efficiency can be measured directly by analyzing the DC extinction curves.
3.2 DC EXTINCTION CURVES
The output power intensity as a function of voltage can be derived knowing the
optical field in the two branches of the Mach-Zehnder. The light power intensity
is given by:
1
2
1
2
2
I( V , V ) � E(
V , V )
[3.1]
The optical field from the output of a Mach Zehnder is given by[301]:
�
� ���
( V
�
1)
� � � ���
( V2
)
� �
V1,
V2
) � Ei
�SRinSR
exp��
� � j��
( V1)
�L�
� exp��
� � � j���(
V ) � ���L���
�
� � 2
� � � � 2
� ��
E( out
2
where
E
i
�
E
o
( 1�
SRin
)( 1�
SRout
)
[3.2]
[3.3]
where V1 and V2 are voltages on the two MZ branches, �� is the change in optical absorption,
�� is related to the change in index where
�
2�n
�
the power splitting ratio of the input MMI. SRin = (P1/P2)in.
� . L is the length of the modulator. SRin is
79

Fiber-coupled output power was measured for a 300�m long modulator as a
function of input optical power as shown in fig. 3.3 and fit with the model from
equation [3.2].
Fig 3.3 Fiber coupled output power at � =1555nm for three different input powers (4.9mW,
11mW, 15.6mW) corresponding to SOA = (20mA, 60mA, 120mA)
Using the expression for optical field from equation 3.3, along with the change
in absorption, one can obtain the refractive index change. The Franz Keldysh
absorption can be modeled with the following model.
80

3.3 FRANZ-KELDYSH ABSORPTION
Franz-Keldysh absorption is caused by is the tilting of the bands during reverse
bias which in turn increases the tunneling probability as the electric field
increases. The phenomenon of absorption is caused by the presence of
photon assisted inter-band tunneling due to the tilted energy band diagram.
The absorption of Franz-Kelydsh based modulators [311] is given by the
following relation[346]:
�(
h�,
| E |) � Aj
| E |
� j
1/
3
�
��
�dAi(
z)
�
���
�
�
���
� dz �
�
j
��
� � �
��
where E is the electric field in V/cm. Ai are airy functions
�
A
j
j
� ( E � h�
) E
B j g
3.
55x10
�
nh�
�
�
�
4 2
�
m
j
�2
/ 3
�
�
�
4 / 3
j
2 �A( � ) �
i
j
�
�
�
�
2
[3.4]
[3.5]
[3.6]
The sum of j is over both the light and heavy hole bands where h is plank’s constant, � is the
effective mass, and � is the angular frequency, m is the mass of an electron, n is the refractive
index, and Eg is the bandgap.
Even in the absence of applied field, the built-in electric field contributes to a
static absorption coefficient is given by the Urbach tail expressed by[312]:
The internal field is given by:
o
�� ( E hv)
F �
� � A exp � /
[3.7]
i
g
bi
i
i
F � ( V �V
) / d
[3.8]
81

where Vbi is the built-in field, V is the applied voltage, and di is the intrinsic region width.
The Franz-Keldysh absorption is shown for waveguide compositions close to
the bandgap (1.4 – 1.45) in fig. 3-5 as calculated from equation 3-2.
Absorption (cm-1)
Absorption (cm-1)
400
300
200
100
1500
1000
500
Fig. 3-5 Total Franz-Keldysh Absorption as a function of wavelength for
different waveguide compositions and Electric fields.
As can be seen in Fig 3-5, considerable wavelength dependence in the
absorption will result from using a waveguide composition bandgap closer to
the operating wavelength. Verification of this model was performed by
comparison with the absorption as measured with photocurrent as the following
relation:
Quat = 1.4Q
0
1520 1540 1560 1580
Wavelength (nm)
Quat = 1.43Q
0
1520 1540 1560 1580
Wavelength (nm)
600
500
400
300
200
100
2500
2000
1500
1000
500
Quat = 1.41Q
0
1520 1540 1560 1580
Wavelength (nm)
Quat = 1.44Q
0
1520 1540 1560 1580
Wavelength (nm)
82
1000
800
600
400
200
4000
3000
2000
1000
Quat = 1.42Q
0
1520 1540 1560 1580
Wavelength (nm)
Quat = 1.45Q
0kV/cm
50kV/cm
100kV/cm
150k/cm
200kV/cm
0
1520 1540 1560 1580
Wavelength (nm)

1 � I
� � � ln�
�1�
L �
pc
( V ) ��
�
�
�
qPin
�
[3.9]
where � is the modal absorption, L is the length of the modulator, Ipc(V) is the photocurrent
generated, Pin is the input optical power. And hw/q is the phonon energy in eV.
Fig 3.6 Modal absorption data and FKE model above for a 300�m long modulator
As can be seen in fig 3.6, the Franz-Keldysh model fits the absorption data well
– as shown for three different input powers (Pin = 4.9mW, 11mW, and
15.6mW).
83

The refractive index can be changed with two different classes of effects -
Field effects consisting of the linear electro-optic effect (LEO or Pockels effect)
and the quadratic (electrorefractive or Kerr) effect, and Carrier based effects
such as carrier induced bandgap shrinkage, heating induced bandgap
shrinkage, Plasma loading, and the Burstein shift (Band-filling effect). These
effects will be outlined below:
3.4 ELECTRIC FIELD EFFECTS
The change in index due to electric field effects is given by the following
equation[305]:
1 2
3
2
� nbulk � �no
[ r(
E � Eo
) � s(
�)( E � Eo
) � ( rEo
� s(
�)
Eo
)] [3-10]
2
no is the index without field applied
Ґ -is the overlap confinement factor
E is the electric field
r is the linear coefficient
s is the quadratic coefficient
LINEAR ELECTROOPTIC EFFECT (LEO)
(Pockels effect)
This effect is due to the biaxial birefringence induced by the presence of an
electric field. It is polarization dependent – where the effect is positive for TE
84

light if the light propagates in the [-110] direction and negative if the light
propagates in the [110] direction.
3
n r41E
� nLEO �
[3.11]
2
The linear electro-optic coefficient has been measured for InGaAsP
quaternaries in [344] and can be extrapolated for different compositions as was
done in [345] using the coefficients in table 3.1.
Table 3.1 Fitting
Parameters for various
materials from [345]
�1
P44 r41
Material A0 B0 C0 D0 E0 F0
InP 8.40 6.60 -0.36 2.6 -42.06 91.32
GaP 22.28 0.92 -0.06 1.92 -83.31 16.6
GaAs 9.29 7.86 -0.21 2.12 -71.48 123.16
InAs 4.36 10.52 -1.48 2.32 -30.23 197.88
For the wavelength range, and quaternary compositions of interest, the r41
coefficient is plotted in fig. 3.7.
85

Fig 3.7 Linear electro-optic coefficient extrapolated as a function of waveguide composition and
wavelength from [345] and [344]
QUADRATIC ELECTRO-OPTIC EFFECT OR KERR EFFECT
(QEO)/ELECTROREFRACTIVE EFFECT
This effect is a third order nonlinear optical process and unlike the LEO effect,
is polarization independent. In order to determine the index change as a
function of absorption change – one uses the Kramers-Kronig transform[316].
where the Principle is given by
hc �(
�)
n(
E)
�1 � P
'
2 2 2
2 � dE
� E'
�E
P
�
�
0
� lim
�
0
a ���
0
86
E�a
�
0
�
�
�
E�a
[3.12]

�
�
� hc 1 � ( �)
�n(
� , E ) � P
dE'
[3.13]
field
2 2
� �'
�E
dE'
0
The resulting Kerr effect turns out to be fit well by a quadratic relation with
respect to electric field. The change in refractive index is related to the square
of the electric field as expressed by:
�n
ker r �
n
3
R
ker r
2
E
2
[3.14]
As the composition of the waveguide approaches that of the operating
wavelength, the quadratic dependence (due to electro-refractive effect)
becomes more pronounced where the Kerr coefficient has been empirically
approximately as[347]:
R r �
�
ker
�15
2 2
1. 5x10
exp( �8.
85 E)
cm / V
[3.15]
where � E is the difference between the photon energy of the guided light and
the quaternary material gap energy.
87

Fig 3.8 The Kerr Coeffient as a function of wavelength and waveguide composition
The Kerr coefficient as a function of wavelength and waveguide composition
wavelength are shown in fig 3.8. as extrapolated from the Adachi model [344].
3.5 CARRIER BASED EFFECTS
As the waveguide is depleted of carriers under reverse bias, the index changes
due to a few effects. As shown later, the carrier based effects are significant
for doped waveguides and important for high modulation efficiency.
PLASMA EFFECT
The Plasma effect is due to intraband free carrier absorption in both the
valence and conduction bands. Free carrier absorption in p-type material
88

(most important) consists of mostly intraband and interband absorption of
holes[2]. The free-carrier plasma reduces the index of refraction of the material.
The following formula gives the change in index for free holes and free
electrons as a function of doping.
�n
plasma
� ro�
�
2�n
2
� N
�
�me
where the hole effective mass is given by:
m
h
m
m
3/
2
hh
1/
2
hh
� m
� m
3/
2
lh
1/
2
lh
�
P
m
h
�
�
�
[3.16]
� [3.17]
where ro = 2.82E-13 cm, N is the electron density, P is the hole density, me is the electron
mass, mh is the hole effective mass, n is the refractive index, and � is the wavelength of
light[349].
89

Figure 3.9 Refractive index change due to Plasma effect as a function of Donor concentration,
and waveguide composition[349] Assuming N-doped waveguide
As can be seen from Figure 3.9, the plasma effect is significant for
higher(>1e16) waveguide doping concentrations. This relationship can be
approximated as[347]:
�
n plasma
� 3.
63x10
�21
N
[3.18]
The relationship is fairly independent of waveguide composition and
wavelength. N is the doping level (n-type) and this approximation is at 1.52um.
90

BAND-FILLING EFFECT / BURSTEIN SHIFT
This effect is brought about by the removal of carriers in the depletion region
and the resulting reduction of absorption of the region. This effect, also known
as the Burstein-Moss effect [347], has been described as bandfilling of the
conduction band in n-type semiconductors. Because of this bandfilling,
subsequent valence band electrons require greater energy to be excited into
the conduction band – resulting in less absorption. This effect is interdependent
with carrier induced bandgap shrinkage effects. When the device is reverse
biased, the index reduces as bandemptying occurs.
Assuming a parabolic band structure, the bandfilling-induced change in
absorption is given by:
��(
N,
P,
E)
� C
hh
C
lh
E�E
g
[ f
v
( E
al
) �
E�E
g
[ f
v
( E
ah
) �
f
c
( E
bl
) �1]
��
o
( E)
f
c
( E
bh
) �1]
�
[3.19]
where N, and P are the carrier densities, and E the photon energy, Eg is the bandgap energy, fv
fc is the Fermi-dirac probability functions for the valence and conduction bands respectively,
Chh and Clh are fitting constants for the light holes and heavy holes. [337]
The change in index due to the bandfilling can be evaluated from the Kramer’s
Kronig relations mentioned earlier – however, one also needs to take into
account the carrier induced bandgap shrinkage – particularly in the 0.9e17 to
91

4e17 n-type doping range. The bandfilling effect has been empirically
determined as a function of composition for 1.52�m in [347].
CARRIER INDUCED BANDGAP SHRINKAGE
Shrinkage of the waveguide bandgap occurs due to two major mechanisms.
Firstly, as the PN junction in the waveguide is depleted out, there is a change
in the bandgap due to the change in carriers based on the screening of
electrons which lower the energy of the conduction band edge and raise the
valence band. This shrinkage has been modeled analytically for InGaAsP
materials in [349]:
� An
�
� Bn
* � * / 3
� Eg
� �
*
1�
no
/ n
where A = 1.04E3, B = 2.8E-7, � is -0.19, �r is the relative dielectric constant
�
�
�
�
m
r
e
[3.20]
Additionally, due to heating in the modulator – particularly noticeable with such
large photocurrent in an integrated device, the bandgap shrinks as well, as it is
fairly sensitive to temperature.
92

3.6 TEMPERATURE INDUCED BANDGAP SHRINKAGE
The bandgap of the material will shrink with increases in temperature. This has
been expressed with the Varshni equations for unstrained materials:
�T
( T ) � E ( 0K
) �
[3.21]
Eg g
where Alpha is 4.9E-4 eV/K Beta = 327K.
���T� The change in bandgap energy with temperature has also been extrapolated
for binary data at 300K for lattice matched quaternary material and expressed
as[350]:
dE g
dT
� �1x10
�4
( 3.
18
� 0.
41y
� 0.
61y
2
)
[3.22]
The change in bandgap for the tensile strained modulator structure was
measured using a micro-photoluminescence setup as a function of temperature
as shown in fig 3.9.
93

=0.432meV/K
Fig 3-10 Temperature dependence of the waveguide composition emission wavelength
The slope of the waveguide composition vs temperature is shown in fig. 3-10.
This is a little higher than the slope shown in the literature[351] at 1.3Q of
0.333meV/K, probably due to the strain in the material. As can be seen in fig
3-10, the material is highly temperature sensitive and linear with respect to
temperature. As the modulator heats up with high optical powers, the bandgap
shrinks and the effective waveguide Q can change from 1.4Q to 1.435Q from
20-70C. Heat crosstalk is an important issue in integrated devices as the laser
benefits from low temperatures with higher gain and lower optical loss, and the
modulator benefits from the higher efficiencies at higher temperatures.
The rise in temperature with bias can be evaluated with the following model:
� T � P
[3.23]
d Zt
where Pd is the power dissipated, and the thermal impedance (Zt) is given by:
94

Z t
ln( 4h
/ w)
� [3.24]
�L�
where h is the height of the substrate, w is the width of the device and L is the length. � is the
thermal conductivity[350].
The thermal resistivity of InGaAsP has been given by [350] as:
1
�
�
1.
47
59.
78y
39.
42y
2
� T � � � K cm / W [3.25]
The refractive index of InGaAsP as a function of composition, and temperature
has been extended from a model given by Adachi, and fitted to experimental
data and given by:
n
r
�
�
�
1 �
A(
y)
f ( z)
� �
� 2
�
��
E
g
E
g
( T )
( T ) � �
o
�
�
��
3 / 2
1 ��
�
�
f ( z ) � ( ) � ( T � 300)
�
o B y
� o �T
�
�
[3.26]
where A(y) = 8.616 -3.886y [3.27]
B(y) = 6.621 + 3.461y [3.28]
2 � 1�
z � 1�
z
f ( z)
� [3.29]
2
z
z
o
E
z � [3.30]
E (T )
�
E
g
g
E
( T ) � �
Although InGaAsP data is difficult to come by, for InP near 300K
�
�T
� � �4
� 5.
16x10
�
95
o
o
[3.31]
1/K [3.32]

As one can see, the temperature dependence stems from the bandgap and the
high frequency dielectric constant terms.
3.7 ACCUMULATION OF EFFECTS
The accumulation of the aforementioned field effects and carrier effects are
plotted for a 300�m long device with three different input powers (4.9mW,
11mW, and 15.6mW) under reverse bias.
Fig. 3.11a Change in refractive index as a function of voltage with input optical power 4.9mW T
= 16C � = 1555nm
96

Fig. 3.11b Change in refractive index as a function of voltage with input optical power 11mW
T = 16C, � = 1555nm
Fig. 3.11c Change in refractive index as a function of voltage with input optical power 15.6mW
T = 16C �1555nm
97

The refractive index change was extracted from the absorption curves (fig. 3.6)
and the output power dc extinction curves (fig. 3.3) using equation 3.2 and
shown as DATA on each graph in fig. 3-11a-c. The total change is also plotted
for each case accounting for all of the effects which fit very well the observed
change in index.
A number of conclusions can be made from these plots. First of all, the
dominant effect is clearly the bandfilling effect – or in this case bandemptying
due to the n-doped waveguide. The plasma, linear and quadratic effects all
have fairly similar contributions given the doping profile that was used. The ‘rf’
change in index is the total change minus the heating portion – as under RF
modulation, the device will not heat up much. This RF line appears to line up
well with the RF Vpi data observed in the next chapter – in this case �4V.
From the change in phase due to a change in index, one can determine the
modulator arm length required to achieve a pi phase shift.
Modulator Length to achieve pi shift.
�2�L
�
��
�
� �
�n
� � �
L
�
�
�
2 | �n
|
98
[3-33]
[3-34]

For a 300�m modulator, the index change to achieve a pi shift for 1550 nm is
approximately is 0.26%.
As can be seen from the previous plots, the index and absorption are strongly
dependent on the optical power at DC – as the modulator is heated – and
experiences bandgap shrinkage. Under RF modulation, this efficiency is not
very power dependent. As the heating due to the photocurrent absorption
changes the refractive in the same direction as the other effects, at DC this
gives the appearance that the efficiency is better than it is at RF. Obviously, DC
extinction is not a very good indicator of RF performance.
3.8 HIGH SPEED DESIGN
Capacitance and carrier lifetime govern the maximum bandwidth possible for a
modulator. For a lumped modulator with an open termination port, the small-
signal modulation response is given by:
S
21
2
2
� [3.35]
1�
jwRC
assuming that the microwave attenuation is low[5]. Typically, there are three
approaches to achieving high speed operation: low impedance matching 14 ,
reducing the capacitance and distributing the modulation region 15 . The
14 See Chapter 4 Termination Section
15 with T sections as shown in Chapter 4
99

capacitance can be decreased by either increasing the intrinsic region in the
waveguide 16 , lowering the pad capacitance with low k dielectrics, or decreasing
the waveguide area[314,320] 17 . An accurate account of the capacitance in the
structure needs to take into account the junction capacitance [Cj], parallel plate
capacitance [Cpp] of the interconnect region and the fringing capacitance[Cf] for
the geometry as shown in the side-view of the modulator ridge in Fig 3-12.
Fig 3-12 Modulator end-view with different contributions of capacitance
16 Reducing the modulator efficiency and reducing the optical loss
17 Potentially increasing optical loss and or drive voltage
100

Next we will look at the minimization of these capacitances separately. In
both lumped and traveling wave devices, one would like to reduce the
capacitance per unit length.
3.5 JUNCTION CAPACITANCE MINIMIZATION
The PN junction capacitance is minimized by using a short device with a
narrow ridge. As shown in Fig. 3-13, the junction capacitance improves for
wider intrinsic region widths and lower doping levels. The material exhibits less
free carrier absorption with low doping – particularly Zn. Structures with large
intrinsic regions do not provide high electric fields so
Fig 3-13 Capacitance per unit length [pF/m] for various doping structures - 2�m ridge
101

clearly there is a tradeoff between capacitance as improved with a PIN
structure and efficiency with a PN junction. Since the devices here use a low-
doped PN junction 18 , the bandwidth varies considerably with bias as seen in
the variance in fig 3-13 of the capacitance with bias. As an illustration of this,
the small-signal modulation response is shown for a 200µm long lumped MZ at
various biases in fig. 3-14.
Fig. 3-14 Small-Signal normalized modulation response at 1555nm for a 200um long electrode
device.
18 3e17 Si as in Fig 3-9b
102

The waveguide is depleted out as the bias induced electric field increases in
the waveguide – changing the capacitance and bandwidth as shown in fig 3-
14.
In this work, the ridges were tapered down to 2.5µm (effectively 2.2µm) in the
modulator regions. This did not seem to adversely affect the insertion loss of
the modulators much – as was shown in Chap. 1. Below 2µm wide, one would
expect propagation losses to increase markedly due to light scattering.
3.6 PARASITIC CAPACITANCE MINIMIZATION
There are a number of different innovative materials that can be used for
providing a low-dielectric constant dielectric layer in the modulator section as
shown in table 3-1.
103

Table 3-1 Dielectric Materials
Material Dielectric Constant
Nanoporous silica 1.3 – 2.8
Fluorinated organic polymers 1.8 – 3.0
Fluorinated amorphous carbon 2.1 – 2.3
Non-fluorinated organic polymers 2.5 – 3.5
Cyclotene Benzocyclobutene (BCB) 2.65
SILK (Dow) 2.65
Non-fluorinated polymers 2.7 – 3.5
Inorganic polymers 2.7 – 3.5
Phase separated hybrids 2.8 – 3.0
Poly-imides 3.2 – 3.4
Fluorinated HDPCVD SiO2
3.5
Fluorinated PECVD SiO2
Thermal SiO2 3.9
Plasma deposited SiO2 4.2
Thermal silicon nitride Si3N4 7.9
Plasma silicon nitride Si-N-H 7.0 – 9.0
Application techniques, vary from LPCVD, PECVD, sputtering, to deposition of
low-K liquids by simple spin coating and multiple baking techniques, similar to
photoresist processing. These materials are helpful for a number of reasons.
First of all, the parasitic capacitance in the modulator is reduced due to the low
dielectric constant which is important for high speed. Also, the dielectrics are
useful for planarization over rough topographies on InP wafers – particularly
with n-topside contacts.
104

Although there are a number of low-k electronic material candidates for
electronic device designs such as oxide-based materials that can handle
temperatures as high as 600°C, Cyclotene BCB was chosen for fabrication as
the dielectric material has not only a low dielectric constant (2.65)– but is
easily cleaved and easily applied.
Fig 3-15a PhotoBCB planarized ridges Fig. 3-15b Dry-etchable BCB
Although dry-etchable BCB tends to have superior planarity over
photodefinable varieties (see fig 3-15ab) – the latter choice avoids excessive
overetches of the BCB that are necessary in order to remove BCB residuals
fully from the surface as shown in Fig. 3-16. The shelf life of Photo-BCB is not
very long however at room temperature 19 , so freezing it is a necessity.
19 approximately 1 week
105

voids
Fig 3-16. BCB residuals after etch
BCB scum
The reactive ion etcher (RIE) tends to leave a BCB residue scum on the
surface with the etch conditions that were used consisting of 20% CF4/ 80% O2
with either 250V (W) or 350V (W) conditions – as recommended by Dow.
Going to a lower CF4 percentage gives better selectivity between the BCB and
Silicon oxy-nitride layers – however is more susceptible to oxide scum and the
etch rate decreases dramatically.
BCB
Fig 3-17 Cyclotene 4024 PhotoBCB defined in only the modulator regions.
106

Due to adhesion problems and device heat dissipation issues – BCB was
defined to be only under the modulator section pads. This was defined using a
photolithographic stepper tool – and the rest developed off using a puddle
emersion developer DS-2100 – avoiding a 5µm BCB etch. It was found that
adhesion of the pads during wirebonding was not acceptable on the first device
run due to the BCB being etched under the pads – which had excessive
roughness and resulted in delamination during wedgebonding. Using a
different approach only etching a via to the ridge and leaving the area under
the pad unetched with a sandwiching layer of SiNyOx on top of the BCB proved
superior – not only as a thicker dielectric leaving lower parasitic capacitance –
but very good adhesion for wirebonding. See process Appendix C. Photo-BCB
does not have very good definition resolution as can be seen in fig. 17-a with
very sloped sidewalls, however it is sufficient for this application.
3.7 FRINGING CAPACITANCE
Using the basic geometry given in Fig. 3-12, one can calculate the parallel-
plate capacitance Cpp of the interconnect segment. However, in interconnect
lines where the wire thickness (t) is comparable in magnitude to the ground-
plane distance (h), fringing electric fields significantly increase the total parasitic
capacitance (fig. 3-1). It has been shown [315] that the influence of fringing
fields increases with the decreasing (w/h) ratio, and that the fringing-field
107

capacitance can be as much as 10-20 times larger than the parallel-plate
capacitance. It was mentioned earlier that the sub-micron fabrication
technologies allow the width of the metal lines to be decreased somewhat, yet
the thickness of the line must be preserved in order to ensure structural
integrity. This situation, which involves narrow metal lines with a considerable
vertical thickness, is especially vulnerable to fringing field effects.
A set of simple formulas [315] can be used to estimate the capacitance of the
interconnect structures in which fringing fields complicate the parasitic
capacitance calculation. The following two cases are considered for two
different ranges of line width (w).
�
�
� � t �
�
� �
w �
� 2�
�
2�
�
C � � � �
� for
�
h � 2h
2h
� 2h
��
�
�
ln�1
� � � � 2�
� �
�
��
t t � t ���
�
�
�
�
0.
0543t
� ( 1�
�
�w
�
C � � � �
2h
�1.
47�
for
�
h � 2h
2h
� 2h
��
�
�
ln�1
� � � � 2�
� �
� ��
t t � t ���
�
108
t
w � [3.36]
2
t
w � [3.37]
2

where t, h and w are the dimensions as shown in Fig 3-12. These formulas
permit the accurate approximation of the parasitic capacitance values to within
10% error, even for very small values of (t/h).
The other contribution of capacitance is attributed to the parasitic capacitance
of the contact pad. This contribution was measured in Fig 4-3b to be
approximately 0.2pF. Figure 3-18 shows the parasitic capacitance as a
function of dielectric thickness for different dielectrics and modulator lengths.
Parasitic Pad Capacitance (pF)
2
1.8
1.6
1.4
1.2
1
0.8
0.6
0.4
0.2
0
BCB 100um device
BCB 200um device
BCB 300um device
SiNx 100um device
SiNx 200um device
SiNx 300um device
0.5 1 1.5 2 2.5 3 3.5 4
Dielectric Thickness (µm)
Fig. 3-18. Pad Capacitance for different dielectrics and pad sizes w/fringing fields
3.8 MULTI-MODE INTERFERENCE DESIGN
Another very important element to the Mach-Zehnder design – is that of the
MMI splitters and combiners[326-331]. General MMI theory states that the
109

shortest 1x2 splitter requires a length of 3/8Lpi where the beat length of the two
lowest order modes is given by[328]:
L
�
eff eff w
2
n 4
� [3-38]
3�
o
where neff is the effective index of the mode, � is the wavelength, and weff is the equivalent width
of the MMI
Fig. 3-19a Electric Field profile for the optimized MMI design showing imaging into the two MZ
branches (Waveguide 1.4Q @ 1550nm)
Fig. 3-19b MMI with curved waveguides (Height = 9um Length = 85um, taper = 20um)
Using Beamprop, an MMI design was optimized with a center wavelength of
1550nm as shown in fig 3-19ab. MMIs have broad optical bandwidth 20 [328] –
much wider than the tuning range of the SGDBRs here (38nm). The length of
the MMI becomes very long for wide widths due to the quadratic dependence
so it is imperative to minimize the width as much as possible. A 9 µm wide
MMI was chosen to that the gap between the waveguides could be resolved
with the stepper as shown in Fig. 3-20. Note also the high angle sidewall in
20 close to 100nm for 1dB bandwidth
110

this gap due to the crystal orientation during the ridge wet etch. This sidewall is
not likely to adversely affect reflections in the device – in fact it gives a more
gradual index discontinuity.
Fig 3-20 Gap between waveguides approximately 1µm
Curved waveguides were used to extend the separation distance to 16µm as
shown in Fig 3-21 to minimize the propagation distance. The ridge was defined
using a dry etch/wet etch process where approximately 1µm of material is RIE
etched with Methane/Hydrogen/Oxygen with a subsequent 3:1 H3PO4:HCl wet-
etch to remove the rest of the InP on top of the waveguide. As the radius of
curvature is low, the sidewall roughness appears to be low as shown in fig. 3-
16.
111

Fig. 3-21 Sidewall roughness on curved waveguides and MMI taper
3.9 PHASE SHIFTER
A phase shifter electrode was integrated in one branch of the MZ in order to
facilitate changing the phase for different wavelengths. It is best to design the
waveguide structure to achieve a pi-phase shift without bias. Pi-shifted
modulators have been fabricated with one length a multiple of 0.241µm 21
longer than the other. The devices in this dissertation utilize a pi-shifted
configuration – however this is accomplished using one ridge slightly wider
(0.2µm) in the curved waveguide regions than the other to achieve the pi
shift 22 [300]. Unlike the RF sections, the phase section can be forward biased,
which gives close to 5x the index shift as reverse bias – as illustrated in fig. 3-
22.
21 for 1550nm
22 This is easier due to fabrication tolerances
112

Fig. 3-22 Fiber-coupled power for device #1 as a function of bias on phase section – in both
forward and reverse bias
As this device needs to operate over the full C-Band – in which the pi-shift will
change with wavelength, designs allowed for the use of a forward biased
electrode to achieve the pi-phase shift. This requires fairly good control over
waveguide widths/thicknesses/compositions in order to achieve from run-to-
run. By biasing this electrode however, it induces a significant amount of loss
in that waveguide as shown in fig. 3-17. Ideally the device is forward biased
slightly as very little current is required ~2mA to achieve the desired phase.
113

Fig. 3-23 Normalized Optical Loss vs. wavelength and bias for 100µm long phase electrode
The loss was measured with Device #1 23 where the laser sections are forward
biased, and the SOA is reverse biased to measure the optical power that
makes it through the phase section as a function of bias on the phase
electrode.
3.10 1 ST GENERATION DESIGN
The initial design involved the integration of a SGDBR with a passive Mach-
Zehnder modulator as demonstrated in fig 3-24.
23 see Generation 2 designs 3.10
114

Fig 3-24 Integrated SGDBR- Mach Zehnder modulator
One branch of the MZ modulator was meant for DC biasing to change the
phase for each wavelength, and the second for RF modulation (Pad #2).
Devices were fabricated with parameters as shown in Table 3-2. These
modulators uses two identical 3dB MMI splitters/combiners that are 98µm long
as described in section 3.8.
Table 3-2 1 st Generation Devices
Mach Zehnder
Lengths 550,750,950
Waveguide offset 40um
Width1 2um
Width2 2.1 to 2.2 to achieve pi shift
Curve length 185
Curve width 20um
Trench 15um
MMI length 98um
MMI width 9um
115

Although these devices were fairly long – and suffered from high capacitance
due to the problem outlined in fig. 2-7b, the DC extinction and chirp 24
characteristics looked promising as shown in fig. 3-25.
Output Power (dBm)
-5
-10
-15
-20
-25
-30
+0.8Vbias
+0.6V
+0.3V
0V Bias
-1V Bias
-2V Bias
-3V
-35
-5 -4 -3 -2 -1 0
Arm #1 DC Bias Voltage
Fig 3-25 550µm long electrode at λ = 1535nm
3.10 2 ND GENERATION DESIGNS
A number of different improvements were made to the 2 nd generation devices
to improve performance. First, SOAs were integrated before the MZ and inside
the MZ modulator to mitigate the 4-5dB insertion losses. Additionally, the gap
between the two waveguides was reduced from 37um to 16um which allowed
for shorter curved waveguide sections. A 2x2 MMI was placed at the output to
24 As will be shown in Chap 5
116

Laser Input
guide away off-state light in a controllable way as shown in fig. 3-20. The
output was curved and flared as well as a front passive detector electrode
placed on the output waveguide to reduce reflections. Also, two RF electrodes
were placed on each device so that push-pull modulation could be possible. In
addition, considerably shorter electrodes were employed to improve the high
speed performance.
1x2 splitter 2x2 combiner
Fig. 3-26 Ridge waveguide structure illustrating the 1x2 and 2x2 MMIs with curved waveguides
and output flares
The first three devices have Dual SOAs as mentioned in Chapter 1. Device 7
and 8 have electrodes at the rear of the modulator for rear resistive termination
as will be elaborated in chapter 4 and 5.
117
Output

The first 8 designs use lumped electrodes and are shown for reference:
Table 3-3 Lumped electrode MZ devices
Total device length = 3200µm
SOA
# Config MZ Electrode Length SOA Length
1 Dual 300 350
2 Dual 250 350
3 Dual 200 350
4 Single 300 400
5 Single 250 400
6 Single 200 400
7 Single 300 400
8 Single 200 400
118

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124

Chapter 4
Series Push-Pull Modulator Designs
As first demonstrated by Walker[7,455] and later Spickermann[456] in the
GaAs/AlGaAs material system, if a RF signal is applied across the two MZ
electrodes – thereby connecting the diodes in series – one achieves superior
performance in terms of optical bandwidth and zero-chirp performance with a
single RF input. This chapter will first look at the design concept and results
from lumped device designs with input side termination. Next, end-terminated
CPS transmission line electrode designs that aim to match the characteristic
impedance are explored with respect to transmission line design and device
characteristics.
SiNxOy
n-contact
MZ#1 p-contact MZ#2 p-contact
BCB
InGaAs contact layer RF SIGNAL
SI InP substrate
MZ #1
MZ #2
Figure 4-1 Series push-pull bias configuration
125
Vdc1
Vdc2 R

4.1 LUMPED SERIES PUSH-PULL BANDWIDTH
The series push-pull biasing scheme and cross-section of the modulator are
shown in fig. 4-1. The use of a SI substrate lowers the parasitic capacitance –
under the electrode pads and enables series push-pull (SPP) operation.
Ideally, there would be full isolation between the n-contact region in the
modulator and that of the rest of the chip. In the devices presented here, the n-
InP region was etched down to the SI-substrate close to the ridge and the p-
InP region was proton implanted. This leaves a narrow region below the
shallow ridge that is not effectively isolated. The resistance between the n-
contact to the modulator and that of the rest of the transmitter typically
measured approximately 150 ohms.
Lumped series push-pull devices were fabricated and tested using special
75�m pitch CPS picoprobes with integrated 50 ohm parallel resistors. In this
case, the 50 ohm termination is on the front end of the device. Later
transmission line based devices use a rear termination. The 3dB optical
bandwidth of three different devices with 200�m, 250�m and 300�m electrode
lengths are shown in fig. 4-2 and compared to the one-sided modulation. The
probe configuration is shown in fig. 4-3 for the single side and SPP
configurations.
126

GND
SIGNAL
GND
SIGNAL
Fig. 4-2 Biasing for single side and series push pull
Note that using the series push-pull electrode structure almost doubles the
optical bandwidth – the loss due to parasitics to ground on one branch of the
MZ which equates to approximately the capacitance from the n-contact to the
backside of the substrate (100�m thick). The substrate is metalized on the
backside to facilitate soldering to a carrier for good heat conduction.
Alternatively the device could be either flip-chipped without backside
metalization or epoxied to the carrier thereby removing the metallization and
reducing the parasitic capacitance at the expense of reduced thermal
conductivity. Clearly a smaller n-contact region would be beneficial.
One can see in fig 4-3b the capacitance per unit length (1225pF/m and
690pF/m for single side and SPP respectively) as the slope and pad
capacitance as the y-intercept for both single side and SPP modulation.
127

f3dB 1
�
�RC
Fig. 4-3a Comparison of single side and series push-pull 3dB optical small signal bandwidth
Fig 4-3b Capacitance vs electrode length for single side and SPP configuration DC -3V Bias
Due to the photocurrent generated in the devices –the impedance is reduced
considerably. This means that with a 50ohm parallel resistor, the single side
configuration has approximately 45.5 ohms and the SPP configuration yields
47.6 ohms. The effect of added bandwidth is also evident in the back-to-back
eye diagrams for the comparison of single side to SPP modulation as shown
for a 250�m long electrode device at 10 Gbit/s with a 2 7 -1 PRBS in fig. 4-4.
250�m single side lumped 250�m SPP
Fig. 4-4 Back to back Eyes comparing single side and series push-pull operation – with 10dB
extinction. Both at using -2V DC bias with 1.5V Vpp. OC-192 with 2 7 -1 PRBS
128

4.2 DUAL RF SERIES PUSH-PULL DEVICES
As mentioned before, the Dual RF series push-pull devices take advantage of
the improved bandwidth of the SPP electrode structure and reduced voltage by
having two of them. Figure 4-5 shows the device layout and the parasitic
conduction path between the two n-contact regions. Ideally this path would be
cut or reduced by He implanting in between the sets of electrodes. In the
current layer structure this would be difficult as the n-InP and n-InGaAs layers
are approximately 2.3µm thick – as well as the ridge on top (2µm) which is
difficult to achieve without very high implant energies. To do this a quaternary
contact would need to be placed much closer to the waveguide �0.5µm.
Fig 4-5 Dual RF Series push-pull 4 electrode structure
DATA
GND
GND
N-contact DATA
N-contact
Due to the finite conductivity of the n-layer, the conduction path prefers the
closest GND and even without He implantation, the device operates well at
10Gbit/s.
129

Fig 4-6 device layout for Dual RF SPP electrode devices
As can be seen in fig. 4-7, the swing improves considerably with Dual RF
sources. The bandwidth is compromised a bit due to the lack of isolation
between the two n-contacts – however not excessively as illustrated in the
back-to-back eyes for single SPP and Dual SPP as shown in fig. 4-7.
Both SPP
Risetime:
72ps
Falltime:
56ps
One SPP
Risetime:
65ps
Falltime:
52ps
Fig. 4-7 Optical signal levels for single and dual SPP operation and back-to-back eye
diagrams for each with Vpp = 2V with 10Gbit/s PRBS 2 7 -1 signal
130

4.3 TRAVELING WAVE MODULATORS
Numerous groups have demonstrated discrete high-speed modulators and
DFB integrated devices utilizing traveling-wave electrode structures [4,7].
Although lumped electrodes can provide fairly good performance with respect
to bandwidth, careful design of the transmission line will provide superior return
loss (S11) if the loaded transmission line is designed to match the driver [25 or
50 ohms] and/or superior bandwidth if the microwave index is matched. An
assortment of different transmission line structures have been pursued for
traveling wave electrodes. Microstrip, Coplanar waveguide (CPW), and
Coplanar strip (CPS) transmission lines are most often employed. Microstrip,
although simple is sometimes regarded as disadvantageous due to
inaccessible ground planes, difficulties in shunt connections between the strip
and ground, limitations on the substrate thickness and exhibit more radiation
with thick substrates. In the case of CPW lines – the impedance is mostly
defined by the lateral dimensions and the substrate thickness is not as
important. CPW localizes the electric field – reducing spurious coupling,
radiation and dispersion[402].
Unfortunately, both the even and odd modes can exist in CPW – which this odd
mode can be suppressed with air bridges.[408] Additionally, parallel plate
modes are supported (microstrip modes) between the CPW and the ground
plane on the bottom – which is a cause of ‘energy leakage’ from the CPW
131

[408]. As a general rule the thickness of the substrate must be > 2(2G + W) in
order to suppress the microstrip modes. In order to match the characteristic
impedance of the source, the on-chip loaded transmission lines require fairly
large unloaded characteristic impedances. Although CPW can easily be made
to match 50ohms, capacitively loaded lines require much larger unloaded
characteristic impedances to yield 50 ohms loaded – and these high values
cannot be realized in CPW easily with the current doping restraints of the
integration platform. CPW designed for index matching yields poor
characteristic impedance matching. However, Coplanar Stripline (CPS) –
which has a range of possible Zo values twice that of CPW works well for the
matching region. CPS was chosen for this reason, and the compactness of the
transmission lines suitable for further integration – such as in a photocurrent-
driven wavelength converter.
However, it is more difficult to make a 50 ohm unloaded section (for the
feedthroughs) without very narrow gaps and wide pads – leading to higher
microwave attenuation. The feedthroughs were designed at a linear taper as
this was found to be the best approach in [462]. Also, the phase difference
between the two lines will affect the matching ability at higher frequencies.
Ideally the lines should be excited with equal length feedthrough lines. The
design was chosen to have an input line at 30 degrees to allow probing away
132

from the optical waveguide – but minimize the phase difference between the
two electrodes.
4.4 TRAVELING WAVE MATCHING
The design of Traveling-Wave (TW) modulators is based on the matching of
the optical and electrical wave velocities. As has been pointed out[454], it is
the group index that should be matched – not the phase velocity. In the case
of LiNbO3 modulators, the electrical wave (neff ≈4.225) propagates slower than
the optical wave(neff ≈2.138)[303,421,424]. To perform matching, one can use
a buffer layer, phase reversal, or a shielding plane to decrease the microwave
effective index of the line[422,423]. Alternatively, one can increase the
electrode thickness, decreasing the effective index further [421]. GaAs and
InP, modulators can have electrical waves that propagate faster than the
optical wave. In order to match the index, often either capacitive coupling or
inductive coupling approaches are employed. According to work done by
Spickermann et al.[461], the inductively coupled slow wave structures have
higher attenuation loss for a given gap width – and are harder to model than
capacitively-coupled devices. LiNbO3 devices do not have a PN structure and
do not load the line substantially similarly to devices such as demonstrated by
Spickermann that rely on the electric field between the electrodes to change
the index – which usually is far less efficient than the use of a PN structure.
The devices in this dissertation use PN junctions to increase the electric field
133

overlap with the optical mode which leads to a very large capacitance per unit
length – resulting in a similar situation as the LiNbO3 where the microwave
index is much higher than the optical index.
For a capacitively loaded transmission line, the optimum loading capacitance is
given by Walker:
C
loading
n
�
2
opto �
cZ
o
n
n
2
cpw
opto
[4.1]
where c is the speed of light, nopto is the optical group index, ncpw is the electrical index and Zo is
the characteristic impedance
However, in order to fabricate high performance SGDBRs, the doping required
typically results in capacitance per unit lengths in the range of 2000pF/m to
2500pF/m for a 3�m wide ridge. The junction capacitance/length of the device
due to the PN or PIN region is considerably larger (x10) than the
capacitance/length of the coplanar line. The result of this is the line is highly
capacitively coupled which both slows the electrical wave and reduces the
characteristic impedance considerably.
First, the optical group index of the modulator section needs to be assessed.
The effective index and group index are shown in fig. 4.8 for various waveguide
compositions.
134

Effective Index
Group Index
3.35
3. 3
3. 25
3.2 1520 1530 1540 1550 1560 1570
4. 4
4. 2
4
3. 8
3. 6
3.4
Wavelength (nm)
1.3Q
1.35Q
1.4Q
1.45Q
1.3Q
1.35Q
1.4Q
1.45Q
1580
1520 1530 1540 1550 1560 1570 1580
Wavelength (nm)
Fig. 4-8 Effective Index and Group index for different waveguide compositions. Assuming a
structure where the waveguide has been etched off halfway.
One can see that not only is the group index significantly higher for waveguide
compositions at approx. 1.45 – but the dispersion increases as the operating
wavelength approaches that of the band-edge. One will obtain a superior
velocity match at the lower wavelengths and higher waveguide Q as loaded
transmission lines tend to slow the electrical wave excessively. Matching over
a wide wavelength range becomes more difficult as the waveguide composition
Q increases.
135

Next we should consider the group index of the microwave signal. The
electrical signal does not have as much dispersion as the optical signal – and is
often approximated as just the phase velocity. The dispersion has been curve
fitted from spectral domain data and is given by [415].
� r1
� � q
neff ( f ) � � eff ( f ) � � q �
�b
( 1�
aF )
f
TE
where: f = frequency; F = f/fTE
�
c
�
4h1 r1
�1
the cutoff frequency for the lowest-order TE mode
� S �
( u log�
�v)
W
�
� �
[4.2]
[4.3]
a � 10
[4.4]
u≈0.54 – 0.64q + 0.015q 2
v≈0.43 – 0.86q + 0.54q 2
q = log(S/h1)
h1 thickness of substrate
b = 1.8
�q = effective permittivity at the quasi-static limit.
136

4.5 TRANSMISSION LINE MODEL
The CPS transmission line in these series push-pull devices can be modeled
as a distributed circuit model along the device as shown in fig. 4-9. Often, the
characteristic impedance of a transmission line is approximated for low
microwave loss in equation 4.5.
Z
Lcps
� � �
[4.5]
Y
C
lossless
Zo � �
n � c ZY ����c
cps
lossless
� cps cps
[4.6]
However, the devices here experience microwave losses due to a number of
sources as outlined in section 4.7 and the transmission line model fits the data
best if the capacitive and inductive loading are accounted for in the model. The
transmission lines in the device are loaded by the depletion capacitance from
each ridge CPN1 and CPN2 in the ridge as is shown in fig. 4-9 as well as a small
amount of inductance due to the T structures. The capacitance shown in fig. 4-
9 is composed of the PN junction capacitance (significant), the CPS
metallization capacitance and the parasitic capacitance.
137
L
C

Ccps
CPN G CPN
Fig. 4-9 Device cross-section equivalent circuit for smooth CPS
This can be expressed as a distributed circuit model as shown in fig. 4-10.
Z
Y
Lcps
GPN2
GPN
R
CPN2
LT
CPN
Gn Ccps
CPara
Transmission line Equivalent circuit for T-electrode CPS line
Fig. 4-10 Transmission line distributed equivalent circuit. Gn is conductance in n-cladding
region, Cpara is parasitic capacitance to ground, LT is the T-inductance, GPN is the conductance
due to the photocurrent in the ridge Cpn is the depletion capacitance
138
LT

The equivalent circuit model for the smooth CPS line devices is the same as
given in fig 4-10 except without the inductance contribution of the Ts (LT = 0).
Given the equivalent circuit model in fig. 4-10, the characteristic impedance of
the smooth CPS transmission line can be expressed as:
Z
Y
osmooth
smooth
�
�
Z
Y
j�C
smooth
�
cps
Z R � j�L
� [4.7]
cps
Gn
�
G �
n 1
1�
� �
j�
��
CPN1
C
j�C
cps
PN 2
( R � j�Lcps
)
Gn
�
G �
n 1
1�
� �
j�
��
CPN1
C
1
� C
PN 2
p
�
�
��
1
� C
The T structures have some additional inductance as shown in the equivalent
circuit in fig. 4.10
Y
T
n
PN
PN1
PN
p
�
�
��
�C�C� PN 2
p
[4.8]
[4.9]
1
� j�
CcpsT
�
[4.10]
1
1
1
� 2 j�LT
�
�
G
G � j�C
G � j�
Z
oT
�
Z
Y
T
�
j�C
cpsT
( R � j
�
1
G
� 2 j�LT
�
G
1
1
� j�C
n
PN
139
� LcpsT
)
[4.11]
PN1
�
G
PN
1
� j�
�C�C� PN 2
p

4.6 CHARACTERISTIC IMPEDANCE COMPARISON
The CPS lines used in this dissertation were modeled using ADS software. As
the lines are considerably capacitively loaded, this means we need to design a
transmission line that has a much larger impedance unloaded – in order to
obtain 50ohms loaded. Figure 4.11 shows the unloaded characteristic
impedance for two CPS structures, one with 50 �m Ts and one with a smooth
CPS line 16�m apart with 8�m wide strips.
4.11 Unloaded Characteristic Impedance for smooth CPS and 50�m T structures from devices
as shown in table 4.2. 1.5�m thick Au
As can be seen in fig. 4-12a, narrow lines increase the characteristic
impedance by increasing the inductance – at the expense of microwave loss.
140

A much higher characteristic impedance is possible with the Ts as shown in fig
4-12b for a given electrode width. The width of the T electrodes was chosen at
8µm as a compromise with microwave loss – shown in fig 4-12b as design G.
Characteristic Impedance
50
45
40
35
30
16um spacing 1000pF/m loading
2
4
6
8
10
12
15
25
0 5 10 15 20 25 30 35 40
Frequency (GHz)
Loaded Characteristic Impedance
52
50
48
46
44
42
40
Design G 5um
DesignG 8um
DesignG 15um
38
0 5 10 15 20 25 30 35 40
Frequency (GHz)
Fig 4-12a Characteristic Impedance for different CPS line widths given a 16um spacing
Fig 4-12b Characteristic impedance for T-section electrodes vs electrode width
From S parameters and the resulting [ABCD] matrix, the characteristic
impedance was extracted for different biases for device #9. After testing the
characteristic impedance of the different devices it was clear that they don’t fit
the characteristics shown at low frequencies in fig 4-12b. After analyzing the
expected conductance in the structure, it was obvious that the n-epilayer
conductance for this structure is considerably higher than that of previous
structures done on lower doped or dielectric substrates.
141

Based on Hall measurements:
The conductivity of the n-layer between the ridges is given by
� �
InP(
n)
� n = (1.6E-19 C)(1800cm 2 /Vs)(1E18 1/cm 3 ) = 288 S/cm (0.032S/cm
q n
in Spickermann)
where the Conductance is
Area
GInP( n)
� 2�
InP(
) = 1.469 S (compare with 0.01S in Spickermann)
Length
Where the Length is 16µm; Area = (2.3µm*314µm) for Device #9
Data was taken comparing the characteristic impedance of T structures,
smooth CPS lines and lumped rear terminated electrode devices. The fit from
the model shown in equations 4.9 and 4.11 are also shown assuming for the Ts
the capacitance per unit length of the transmission line is CT = 2.737e-11F/m,
Inductance per unit length is LT = 1.736e-6 H/m and for the smooth CPS Cs =
4.602e-11F/m and Ls = 7.3068e-7H/m with Rpn = 500ohms, R = 0.2 ohms, Cpn
= 0.5pF, Cpara = 0.5pF, LT = 5e-12.
142

Fig 4.13 Lower 4 lines extracted from Device #7 (single side 300�m long electrode)
Middle 4 lines extracted from Device #18 (Smooth CPS 500�m long electrode)
Top 4 lines extracted from Device #7 (CPS Ts 250�m long electrode)
As can be seen in fig 4-13, the characteristic impedance improves for higher
reverse biases on the electrodes – where the depletion region is increased and
the capacitance/unit length decreases. Also, clearly one can see a large
benefit of using a T electrode over the smooth CPS lines in terms of better
characteristic impedance matching as it is much closer to 50 ohms.
143

4.7 RF LOSS MECHANISMS
High frequency losses stem from three different mechanisms:
1. Dielectric losses
2. Ohmic/conductor losses
3. Radiation loss
These losses can be minimized using a number of approaches such as the use
of deep trenches between electrodes or thick dielectric layers below the
electrodes – separating them from the substrate. By careful design of the
electrodes, minimization of longitudinal substrate currents may also reduce the
overall microwave attenuation[435]. Most work is done on Semi-insulating InP
and GaAs – where the bulk of the electrical attenuation comes from the
conductor and radiation losses at least below 20GHz [450]. However, typical
SGDBR design is performed on n-InP conducting substrates with lossy InGaAs
contact layers. In this case, the dielectric losses are very high and the lines
become highly dispersive. Also, the capacitance between the two lines
increases dramatically - effectively loading the line – and dropping the
characteristic impedance significantly. Dielectric loss is given by the following
relationship:
�
D
q�
�
�
r
eff
tan�
�
g
144
(Np/m) *27.3 for dB/lamda [4.12]

For a doped semiconductor – the loss tangent can be expressed as[450]:
2�f�"
( f ) � � '(
f )
tan�
( f ) � [4.13]
2�f�
'(
f ) � �"
( f )
where �’ and �” are the real and imaginary parts of the complex dielectric
permittivity and �’ and �” are the respective parts of the complex conductivity.
Taking the conductivity from the Drude model, we have � = �s/(1-j2�f�m) where
the conductivity can be extracted from Hall measurements.
The attenuation drops linearly with increasing metal thickness – up to the point
where the metal depth is 3x skin depth. As the dimensions of the transmission
line increase, the attenuation decreases. There seems to be an optimum w/d
point of approximately 0.40 for InP with 0.25um of gold. If w = 80um that
corresponds to d = 177.8 [405]
In order to match the velocities of the electrical and optical waves, one can
manipulate a few parameters – electrode thickness, coplanar gap width, and
distributed capacitance along the line. The electrode thickness highly affects
the microwave loss in the structure as shown in fig. 4-14.
145

Microwave Index
20
18
16
14
12
10
8
6
2
0 2 4 6 8 10 12 14 16
Electrode Thickness (um)
Loss (dB/cm) 5um
Loss (dB/cm) 10um
Loss (dB/cm) 20um
nload (5um)
nload (10um)
nload (20um)
Fig. 4-14 Microwave index and loss for loaded CPS line [2000pF/m loading] with different
electrode widths varying from 5-20 microns
Clearly an electrode thickness exceeding 2µm is preferable to reduce both the
microwave index and loss. For the work shown here, the p-metal thickness is
approximately 1.5�m. The loaded-microwave index drops significantly with
electrode thickness – although as we have shown before, we would like 3.7-
4.2. This does not take into account the change in effective index when the
area between the center conductor and ground are removed – or BCB is
placed below the contacts. Although thickening the electrode improves the
index match, the characteristic impedance is reduced. In order to match the
characteristic impedance and index simultaneously, the capacitance per unit
length of the line must be reduced.
146
16
14
12
10
8
6
4
Loss (dB/cm)

Table 4-1
Material Relative
Permittivity
Loss Tangent (tan � ) Typical
Resistivity
InP n 12.4,12.6 5x10-5 20E-4 ohm-cm
InP SI 12.4 1.5E7 ohm-cm
GaAs 12.85 5x10-4
High resistivity Si 11.9 1x10-4 at 30GHz 4000ohm-cm
Standard Si 11.9 4x10-3 at 30GHz 1000ohm-cm
BCB 2.65
Aluminum Nitride 8.9
Conductor loss can be estimated from the unloaded Q factor
�w
�
u �
� �
Z f
� 2 �
[4.14]
Q o
Good up to 2 GHz
above 2 GHz one must keep in mind the dielectric attenuation is mostly
dependent upon the substrate thickness. The conductor attenuation coefficient
is minimized at a particular w/s. This conductor attenuation decreases with
dielectric constant. The dispersion is lower for smaller waveguide dimensions.
The coplanar waveguide design takes into account the dielectric that the lines
reside, the thickness of the metal layer to achieve a 50 ohm line. As the lines
are deposited on a multilayer dielectric – not just on the InP surface, one needs
to take into consideration the effective dielectric constant that insues. This can
be calculated analytically using the conformal mapping technique.[407]
147

As discussed earlier, conductor losses are reduced by using wider and thicker
electrodes. However, the characteristic impedance is improved by going to
thinner and narrower electrodes. A compromise was made using 8�m wide
electrodes. The gap between the two ridges was designed to be fairly close
(16�m) to shorten the curved waveguides and reduce propagation losses.
LiNbO3 traveling wave modulators usually use CPW transmission lines as
without careful attention to the electrode gap widths have experienced large RF
losses in CPS structures [453]. It has been found [453] that leakage of the CPS
modes into substrate modes may occur at fairly low frequencies (11 and
22GHz). This leakage is due to the geometry of the device (gaps 0.5mm to
1mm on substrates 0.25-0.5mm thick)
The electrical attenuation was measured for different biases – without bias on
the laser or SOA. The loss was extracted from the ABCD parameters [see
appendix] after measuring the S parameters of the device. These values
compare closely with other similar EAM devices that report losses in the range
15-20dB/mm at 40GHz. The microwave loss results with bias are shown in fig.
4-15. As can be seen, the microwave loss decreases considerably with
148

everse bias. This has been attributed to loss due to undepleted material in the
PN junctions 25 .
Fig 4-15 Microwave loss as a function of frequency and bias from Device #9
As the PN junction depletes out, the loss becomes dominated by ohmic losses
due to the skin depth in the electrodes.
25 Spickermann Dissertation pp 133
149

4.8 CPS T-ELECTRODE DEVICES
Much work has been done to velocity match traveling wave Mach Zehnder
structures using T-sections that both increase the characteristic Impedance
and the length – thereby reducing the capacitance per unit length – and
providing better matching that results in higher bandwidth.
MZ Phase electrode
SGDBR Laser
Modulator n-contact
Fig. 4-16. T-Electrode SPP-MZ-SOA-SGDBR Transmitter Layout
The device layout of these devices is shown in fig. 4-16 above.
150
GeAuNiAu
n-contact
Semiconductor Optical
Amplifier

By distributing the capacitance using fins, one can lower the capacitance per
unit length at will. However, as the InP/InGaAsP material has considerable
optical loss, and the mismatch becomes greater between the optical and
electrical waves at longer lengths, the Ts designed in this work did not lengthen
the device much. The periodicity of the tabs is related to the cutoff frequency
for a given phase velocity and width[304].
f
cutoff
v
�
2d
where d is the spacing of the fins (period) and vphase is the phase velocity.
These Ts are 50µm long with 10µm spacing between as shown in fig. 4-17.
phase
Fig 4-17 TW electrode structure with 50µm Ts with 10µm gaps
151
[4.15]

This approach is only practical when the capacitance per unit length is already
small. For highly capacitively loaded lines – the device length that is required in
order to improve the bandwidth is very large leading to excessive microwave
and optical insertion losses.
In this work a few different CPS transmission line electrode designs were
explored as shown in table 4-2.
Table 4-2 Transmission Line based electrode MZ devices using Dual RF Series Push-pull
drive
Total
Active Electrode
SOA SOA electrode Length Electrode T length
# Config Length length (um) Width (number)
9 Single 400 250 313 8 50(5)
10
Dual 575 400 490.5 8 50(8)
11 Dual 490 500 610 8 50(10)
12 Dual 380 600 734.5 8 50(12)
16 Single 600 400 490.5 8 50(8)
17 Single 500 500 500 15 N/A
18 Single 500 500 500 5 N/A
19 Single 500 500 560 8 100(5)
20 Single 500 500 610 8 50(10)
21 Single 400 600 730.75 8 50(12)
152

4.9 TRAVELING WAVE BANDWIDTH
The bandwidth of a traveling wave modulator is governed by the difference in
the optical and electrical waves and the overlap factor of these two modes, the
frequency dependent attenuation along the device, the termination impedance,
and the length of the device. Accounting for both the attenuation and the
optical-electrical matching and assuming that the device is terminated with the
characteristic impedance – the bandwidth can be approximated as[4]:
( ) � �
B f e
�l
/ 2
� 2 ��l
� 2 ��l
��
�sinh
� sin
2 � � � 2 ��
� � � � ��
2
2
��l
� ��l
�
�
�
� 2 �
�
�
� 2 �
�
2�f
where � � [ n� � no
] [4.16]
c
It is clear from the previous equation that both the attenuation and index
matching are very important to achieve high bandwidth. Although simple, the
above equation does not take into account mismatches in the characteristic
impedance – which is important as it is difficult to reach 50 ohms with such high
loading capacitance.
The small-signal modulation response S21 can be modeled accounting for the
opto-electrical velocity mismatch, microwave attenuation, and impedance
mismatch[5].
153

�L( j�
� � ) �
exp
1
exp( ( ) 1
1 ) � �
T �
�
L j�
� � � � �
��
L �
�
S 21 �
�
� �L
exp( �2�
� L)
� �L�S
exp( �2�
� L)
L(
j�
� � � )
( j�
� � �
2
[4.17]
where the amplitude transmission into the modulator is : T � 1�
� , The
reflection coefficients at the source and the load are given by:
( Z s � Z m )
� S �
[4.18]
( Z � Z )
s
m
( Z L � Z m )
� L �
[4.19]
( Z � Z )
L
Zm is the characteristic impedance of the modulator, and Zs and ZL are the
impedances of the source and load respectively. The propagation constant is
given as:
and the optical Beta coefficient is:
�
�
2�n
� � � � � �
�
m
�
� [4.20]
�o
2�n
�
o � [4.21]
where no is the optical group index, and � is the optical wavelength
4.10 MEASURED BANDWIDTH
Taking the fit data from the characteristic impedance for device 9 as shown in
fig. 4.16, the attenuation coefficient, and microwave index, and assuming the
154
S

optical group index is 3.92 at 1555nm, the model expressed in equation [4.11]
fits the experimental data well. Fig 4-17 shows the small-signal frequency
response of two devices, (9 and 16) which have 250µm and 400µm long
electrodes. Both are terminated at the rear with 50 ohms.
Fig. 4-18 Small-signal frequency response for two T-electrode devices at 1555nm -3V bias for a
250�m electrode device (device #9) and 400�m device (device #10)
The bandwidth was also explored with a low matching resistor. This was done
with a resistor ladder structure that was fabricated on the carrier – adjacent to
the device as shown in fig. 4-19.
155

Phase electrode
Termination
Resistor
RF Signal Input
n-contact bias
Fig. 4-19 Transmission line electrode configuration Device 17
The S21 response with a 35 ohm termination resistor with wirebonds as in fig 4-
19, is shown in fig. 4-20. This is compared with the bandwidth of a lumped
electrode device with the same length and the 50 ohm terminated data. As one
might expect, the bandwidth is extended out to close to 40GHz – and some
peaking is observed due to the reflections/standing wave along the electrode
structure.
156

Fig 4.20 Small-signal response for Device #9 as a function of termination resistor
As can be seen in fig 4.20, the data fits the traveling wave bandwidth model
very well for the 50ohm termination. In this case, a 50ohm probe was used at
the end of the device with a 50ohm termination. The 35ohm termination data
was obtained using wirebonds to the carrier in which a resistor ladder was
used. One sees a little more peaking than the model predicts due to the
wirebond impedance discontinuity which leads to a reflection at the end of the
device. Note also that there is very little difference in the bandwidth observed
between the counter-propagating and co-propagating bias configurations.
Most likely the traveling wave effect would be more noticeable for longer
157

devices with low termination resistors as has been seen in the literature,
however, this could not be done due to the restraints of the resistor ladder.
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164

Chapter 5
COMPARISON OF DEVICE DESIGNS
This chapter compares the performance of the devices outlined in the previous
two chapters with respect to modulation efficiency, bandwidth and chirp
characteristics. This work was done on the test setup as shown in fig. 5-1 with
devices soldered down to RF Aluminum Nitride carriers.
Fig. 5-1 Test setup and RF carrier
165

A number of authors have proposed figures of merit for modulators of both the
EAM and Mach-Zehnder varieties in Table 5.1. These merits consider the
electrical bandwidth, drive power requirements, low return loss, optical insertion
loss, and suitability for integration with lasers. Wavelength and temperature
dependence are also a consideration. The contrast ratio is important for large
signal modulation. Often Mach-Zehnder and EAM devices, are compared as
Vpi with V10dB. This is not usually a fair comparison however as the MZ
modulator does not need to driven so hard.
Table 5.1- Figures of Merit Based on
R. G. Walker [7]
M.K.Chin[9]
� 2R
� fo�
�
�50
� R �
� V
GHz-um/V
� �
� T ; ;
� o
Optimizing the device for the largest
�
� �2 ��
�E
Modulation Bandwidth and
generator voltage.
R is shunt resistance
Change in Transmission
Bandwidth requires the consideration of a
VDrive
number of parameters as in Table 5.2. The bandwidth is a function of the
capacitance(Length), S11(Zo), and Termination/Resistance. The drive voltage
is a function of the Insertion loss/Power, Doping(Length), Waveguide
composition, and termination.
166

Table 5.2 Main Design Parameters
Bandwidth
Doping
Device Length
Device Width
Passivation
Dielectric
constant
Waveguide[Q]
Bandwidth Insertion
Loss
Drive Voltage Wavelength
Dependence
�� � � �
� � � N/A
� � � N/A
� � � N/A
� N/A N/A N/A
�for TW � � �
For analog applications, the modulation depth is low and RF gain is of utmost
concern, the modulator maximum phase shift per unit length per volt is a fairly
� �
good metric. [3] One can also compare devices simply as the change in
VaL
transmission divided by the voltage to make this change (dB/V).[8] – This
value changes quite a bit over the modulation extinction curve, however could
be compared at Vpi/2. Also, MZ modulators often are compared with phase
shift efficiency and the chirp parameter.
167

Phase shift efficiency
��
�neff
2�
� � �
[5.1]
VL �V
�neff
( real)
2��
Chirp Parameter � chirp �
�
[5.2]
�n
( imag)
��L
For InGaAs MQW structures efficiencies of approximately 12 deg/Vmm have
been reported [3].
In a real digital communications application, a device would need to exceed a
set of specifications typically for output power, drive voltage, RF extinction, and
chirp parameter. It would be best to compare different devices with respect to
a particular output power and extinction (10dB RF). This is difficult to compare.
As in this case as each design has a different output power and none of them
meet a 10dBm fiber coupled output power specification. The output power has
a large influence on the required drive voltage as illustrated in fig. 5-2.
168
eff

Fig 5-2 Reduction in drive voltage for high output power. Assumes 0dBm max power required
with -20dBm off state power
For a given output power specification (in this case 0dBm), if the transmitter
can output more power than necessary, the drive voltage can be significantly
lower than Vpi. In fact, for a doubling of output power either by increasing the
gain of the SOAs/SGDBR or reducing the insertion loss in the modulator, the
drive voltage is only ½ of the Vpi value. This point is rarely considered in
figures of merit. As the laser is integrated – and cannot be simply swapped out
for a higher power laser, high output power is very important.
Because of this, Vpi overestimates the actual drive voltage requirements – in
some cases considerably. However, it does give a metric to compare different
169

modulator designs more independently from the output power so DC extinction
was measured for each device to base a comparison.
5.1 MODULATION EFFICIENCY
The different devices are compared with respect to Vpi as a function of input
power into each branch of the MZ (measured from photocurrent as seen in
Chapter 3) and wavelength. As can be seen in fig. 5-1A&B, the drive voltage is
highly dependent on optical power. As the optical power increases, photon
induced carrier recombination occurs in the modulator electrode region – which
generates heat and the localized waveguide bandgap shrinks – giving
enhanced absorption and index change.
170

Fig. 5.3 Comparison of different modulator designs with respect to Vpi as a function of optical
power into modulator branch. λ = 1554nm SOA = 100mA
Devices with Dual SOAs have more optical power incident to the modulator –
giving both more heating in the modulator and more optical power due to the
improvement in the saturation power. The lowest Vpi was exhibited for a
500µm device with Ts and Dual SOAs 26 .
26 Device 20
171

Clearly this is a non-linear effect as the shorter devices – which heat up more -
are more efficient modulators when normalized with respect to length. Also,
the lowest values – were for T electrode devices with Dual SOAs which yielded
considerably less photocurrent at the peak.
The extinction curves were measured for different wavelengths as illustrated in
fig. 5-4.
Fig. 5-4 DC extinction curve for a 200µm long lumped MZ for three different wavelengths
172

Figure 5-4 shows the wavelength dependence of the DC extinction curves for a
200um long device. As there is more gain for centered wavelengths, the
curve peak is higher at 1564nm. Note that the residual power at 0V is highly
dependent on the coupling of the fiber (angle and distance) so some of the
variance is due to coupling and some is due to the different extinction due to
phase bias differences.
In Fig 5-5, the DC extinction Vpi was measured for different optical powers
going into the modulator at different wavelengths.
Fig 5-5 Vpi for 200µm long device as a function of optical input power to one branch and
wavelength
173

The SOA bias was varied from 20mA to 150mA. The logarithmic optical power
dependence is considerably larger than the wavelength dependence of Vpi –
indicating a large change in temperature within the device since the optical
power is high in this integrated device.
5.2 RF EXTINCTION OF DEVICES
The RF extinction was measured at 10Gbit/s with a PRBS 2 7 -1 as a function of
wavelength for various peak-to-peak voltage outputs with the ‘0’ level at the null
of the modulator characteristic. Although the DC extinction for each device
has been demonstrated in the previous section, under RF modulation not as
much heating takes place and photocurrent competition occurs – making the
modulation not as efficient. Also, in order to have high speed, a parallel
resistor is used – in which less current flows through the device. As can be
seen in fig 5-6ab, there is much more wavelength dependence with a single-
ended drive.
174

Fig 5-6a 250µm Single ended drive RF extinction at 10Gbit/s PRBS 2 7 -1 for various
wavelengths
Fig. 5-6b 250µm lumped series push-pull with 50 ohms in parallel
175

As the operation of the device is limited by the worst channel, there is a factor
of two improvement by using series-push-pull for a wide wavelength range. As
there is more photocurrent generated at lower wavelengths even though there
is also more index shift and absorption – the drive is opposing this
photocurrent, and the RF extinction tends to be superior for the longer
wavelengths. In the series push-pull case, one branch opposes the
photocurrent and one is in the same direction – giving less wavelength
sensitivity. Clearly the devices are temperature sensitive. Taking the change
in index as a function of voltage from fig. 3-6 one can estimate the drive voltage
for different optical powers as shown in fig. 5.7.
Fig 5.7 RF Vpi at 10Gbit/s – model taken from DC extinction data in fig 3-6 for different optical
powers.
176

The modeled data is compared to piecewise measured RF Voltages using
10Gbit/s BERT with maximum of Vpp of 2V. As can be seen on the plots, at RF
the modulator does not heat up as much and the voltage is higher than one
would expect from looking at DC characteristics alone. Also shown on the plot
is a comparison of Vpi with different terminations 27 . As one would expect,
unterminated the drive voltage is much lower due to reflections off the end of
the stub – and more interaction with the modulator. Also, using a low
termination of 25 ohms degrades the Vpi but it is not linear related to the
bandwidth enhancement of low termination.
27 Performed on device #20
177

5.3 BANDWIDTH COMPARISON
The main trends of the bandwidth are shown in fig. 5-8. Extrapolated curves
go through the single side modulation, and the SPP modulation from data in fig.
4-3 with 50 ohm front side termination.
Fig 5-8 Small-signal optical response for different modulator designs. 50ohm termination
Devices with Ts and end-50ohm termination have even better performance due
to the improved characteristic impedance mismatch and traveling wave design.
Using low termination resistors would enhance the bandwidth further as was
discussed in Chap 4.
178

5.4 CHIRP MEASUREMENTS
Although direct modulation of the SGDBR offers a simple solution as a
transmitter, as noted before, the chirp parameter is positive and ranges from
approximately 3-9[330] over the C-Band. Chirp in Externally Modulated
Lasers(EML) is caused by the electro-optic effect in the modulator, electrical
crosstalk between the laser sections and the modulator and from reflected
optical power[505]. Residual feedback from the output facet induces chirp and
relaxation oscillations – resulting in a lower transmission distance (between
repeaters). In order to provide an adequate transmission distance –
particularly at higher bit rates, one desires a chirp parameter that is slightly
negative – usually in the range of 0 to -1 for 10Gbit/s operation[515] as was
shown in the introduction.
5.5 CHIRP MEASUREMENT TECHNIQUES
Chirp can be measured with a few different techniques. One of the easiest
ways is the Gated-Delayed Self-Heterodyne (GDSH) technique. This setup
consists of a modulated laser with a gated sinusoidal signal where the output is
connected to a fiber interferometer similar to that of the linewidth measurement
in Chap. 1. One arm has a delay of approximately 3.5µs. Both signals are
combined and measured using an RF spectrum analyzer.
179

For a Mach-Zehnder modulator, the chirp can be expressed from the intensity
and phase of the output signal.
2
2
Where � �1���2�cos( �1
��
2
4
i E
I � [5.3]
�1�
sin�
�
1 � � sin�
2
� � tan �
�
[5.4]
�cos�1
� � cos�
2 �
2
�'1
��
�'
2 ��
( �'1
��'
2 ) cos( �1
� �2
)
� � [5.5]
� � ( �'
��'
) sin( � � � )
2
V1
� � V2
��
( V1
�V2
� �
��
( V �V
) sin
1
2
1
2
1
) cos��(
V1
�V2
) sin�t
��Vb
�
��( V �V
) sin�t
��V
�
1
2
2
b
[5.6]
so for the small-signal regime[516]
2
V1
� � V2
� � [5.7]
� V �V
)
( 1 2
However, the self-heterodyne method only gives the magnitude of the chirp
parameter. Alternatively, one can measure the magnitude and sign of the chirp
parameter using a network analyzer as discussed in [514,520] where the
resonant frequencies of the fiber frequency response as measured from the
network analyzer[514].
2 c � 2 �
fu L � �1�
2u
� arctan( �)
2
� [5.8]
2D�
� � �
where � is the chirp parameter, � is the wavelength, u is the number of the minimum in the
response, D is the Dispersion parameter of the fiber, L is the length of the fiber
180

Chirp Parameter
2
0
-2
-4
-6
Alpha 1525nm
Alpha 1545nm
(Power (dBm) 1525nm
Power (dBm) 1545nm
-8
-35
-4 -3.5 -3 -2.5 -2 -1.5 -1 -0.5 0
Arm #1 DC Bias (V)
Fig. 5-9 Chirp Parameter as a function of DC extinction curve for pi-shifted case with 550µm
device (1.38um waveguide) Generation1 device
As can seen by fig. 5-9, with a one-sided modulation scheme – with a pi-shift
configuration, the device exhibits negative chirp. In the small-signal regime, the
chirp parameter can be expressed as[541]:
-5
-10
-15
-20
-25
-30
Output Power (dBm)
d
I
dI
�
� � 2
[5.9]
where I is the intensity and � is phase. This means that at the maximum in the
output power curve, we have high chirp since the change in intensity is
minimal. It is the chirp in the ‘on’ state that affects the transmission
performance – and this is often refered to as the 3dB rule[515]. When the
device is ‘off’ – the chirp parameter is slightly positive and increasingly negative
as the modulation depth is increased close to the maximum output power. A
push-pull scheme lowers this negative chirp so that the modulation depth can
181

e increased further. This was measured for a 250µm long device as shown in
fig. 5-10.
Fig 5-10 small-signal chirp parameter for inverting and non-inverting operation for parallel pushpull
and single-sided drive
Although examining the small-signal chirp parameter is illustrative of the
dynamic device operation, performance under large signal operation is
preferred. A time-resolved or dynamic measurement of the chirp parameter
can be achieved using a setup utilizing an optical filter which is either a Fabry-
Perot etalon, waveguide grating router, Mach-Zehnder interferometer[542] or
Optical Spectrum Analyzer monochromator and high-speed oscilloscope[505].
By using this setup – one can see dynamically how the frequency shifts on the
rising and falling edge during modulation at 10Gbit/s.
182

Fig 5-11 Dynamic chirp measurement demonstrating how a chirp parameter of -0.7 can be
achieved by adjusting the gain in the two branches of the MZ performed at 10 Gbit/s
The alpha parameter can be changed by varying the power in the two
branches – either with gain using SOAs or loss with the passive sections. It is
fairly easy to achieve 0 chirp using a SPP electrode structure, however if
negative chirp is desired, it is easier to use a single sided drive modulation as
shown in fig 5-11.
183

5.7 LINEARIZATION OF MODULATORS
For analog modulation, a high degree of linearization is desired to maintain
the dynamic range of a photonic link. Mach-Zehnder modulators generally
exhibit nonlinear transfer characteristics. Fiber optic RF links require
sufficient linearity for applications such as satellite communications, radar,
CATV, and others. A number of different techniques have been employed to
achieve linearity using external modulators. Approaches have included the
use of directional coupler modulators[524], Mach Zehnder modulators[525],
and Electro-absorption modulators[526] and combinations of these in dual
parallel and series configurations[5]. Alternatively, electrical linearization can
be performed using optical negative feedback, phase-shift modulation, feed-
forward or pre-distortion techniques although an all optical scheme avoids
complicated electrical circuits and their frequency responses.
The combination of multiple modulators allows for modification of the nonlinear
response of either – to cancel out the nonlinearity of the total response. A
series combination will decrease the bandwidth. The null in the third-derivative
occurs at Vpi for a Mach-Zehnder and at a bias with significant modulation
efficiency for a Franz-Keldysh EAM[2]. This leads to the increase in SFDR
without sacrificing power – for high gain links.
184

For narrowband applications (< 1 octave) the Spur free dynamic range is
governed by only the odd order inter-modulation terms- however for broadband
applications all harmonics and orders of inter-modulation distortion must be
considered. [2]. Spur-free dynamic range (SFDR) is defined as the range of
input powers over which the output power at the carrier frequency is above the
noise floor while the third-order distortion products remain below the noise
floor.
Fig 5.12 Fundamental signals (500MHz) and 3 rd order distortion signals
When a two-tone RF signal is applied, the bias voltage is given by:
� V � � m (cos�
t � cos�
t)
� [5.10]
V b 1 o 1
2
where mo is the modulation depth and Vb is the dc bias voltage [5].
185

The inter-modulation distortion is highly dependent on the bias conditions as
shown in fig. 5-13.
Detected Power (dBm)
0
-20
-40
-60
-80
Fundamental
Second harmonic
Third harmonic
Optical Power
0 0.5 1 1.5 2 2.5 3
Reverse bias (V)
Fig 5-13. Detected average optical power and RF power of fundamental and distortion
products for 0 dBm modulation power.
The narrowband SFDR was measured for device #5, at 500MHz with a
optimized phase section bias.
186

Output RF Power, dBm
20
0
-20
-40
-60
-80
-100
-120
-140
-160
�=1551.30 nm
f=0.5GHz
fundamental
SFDR=112dB-Hz 2/3
IIP3=25.2dBm
3rd-order distortion
-140 -120 -100 -80 -60 -40 -20 0 20
Input RF Power, dBm
Fig. 5-14 Spur-free dynamic range measured on device #5 SOA = 100mA Gain = 100mA
Phase = 1.2mA Modulator bias = -1V
These results demonstrate that fairly good linearity can be obtained for
optimized bias points.
187

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192

Chapter 6
CONCLUSIONS AND FUTURE WORK
Although the device performance shown here rivals some of the best discrete
components, there are always improvements that could be done to enhance
the performance further. Higher output power will improve the transmitter
characteristics markedly as not only will the heating in the modulator make it
more efficient, but the required drive voltage will be reduced considerably. A
less conservative SGDBR design would give higher output power and more
efficiency in the modulator. Alternatively, using longer SOAs or enhancing the
saturation power by widening the ridges would improve the output power.
Using a QWI integration platform would improve the gain of the material with
centered wells – which improves the laser at the expense of the SOA
saturation characteristics. Using the quantum well-intermixing platform would
allow for more than 2 bandgaps - providing low loss waveguides in the passive
regions and higher efficiency in the modulator electrode region.
Although much better matching of Zo than many CPW TW EAMs, neither the
characteristic impedance or the microwave index are completely matched –
and this can be improved further by reducing the capacitance of the PN
junction as the parasitics are quite low using BCB and a SI substrate. Using a
193

slightly lower doped waveguide would give a better confinement factor as the
depletion region would move out and overlap more with the optical mode.
Also, integration of the termination resistor would make testing easier – and
remove the electrical reflections inherent in wirebonds/ribbonbonds.
Based on Vpi RF and bandwidth measurements, rear termination seems to be
superior due to the reduced microwave reflections at the input. This along with
a low resistance termination – gives an enhancement in the S21 characteristics
without reducing the drive voltage as much. More modeling of the microwave
properties of these devices – and loaded transmission lines – that give the
desired index and characteristic impedance is needed.
As can be seen in fig. 6-1, the current power management shows that we have
approximately 20mW output from the SGDBR untuned – which is amplified
approximately 5-7dB depending on the SOA length – then is attenuated around
5dB. This means that the SOA compensates mostly for the insertion loss of
the modulator. Unfortunately, with the lensed fiber this 20mW quickly becomes
5mW fiber coupled. The integration of a mode converter would make this
design more in-line with typical supplier requirements of 10mW fiber coupled
output power.
194

SGDBR
�20mW
SOA
�60mW
+5dB -5dB
�20mW
Fig. 6-1 Power management through the device.
�5mW fiber
coupled
Going to a shorter electrode will have a prohibitively high drive voltage >8V RF.
This doesn’t seem to be a good option unless the device is coupled with more
output power or a higher Q waveguide used to make the device more efficient
although it is more wavelength dependent. Another possibility is the use of
shallow MQW in the waveguide – which also most likely will need to be
designed carefully to prevent excessive wavelength dependence and optical
losses. Although not desirable in terms of complexity, a butt-joint regrowth is
always possible to reduce the capacitance in the modulator. This is always a
compromise with efficiency – however if the layers are not doped highly the
propagation loss could be reduced – so that the device could be longer.
In the foreseeable future, tunable Mach-Zehdner based transmitters could be
found useful in more complex photonic integrated circuits such as photocurrent
driven wavelength converters.
195

6.1 WAVELENGTH CONVERTERS
Tunable wavelength converters represent a novel class of highly sophisticated
photonic integrated circuits that are crucial in the function of future optical
networks. They allow for the manipulation of wavelengths in WDM optical
switches, routers and add/drop multiplexers. Many different implementation of
non-tunable wavelength converters have been proposed using cross phase
modulation(XPM) in SOAs and fiber[2,3], and cross absorption
modulation(XAM) of SOAs [1], Many of these architectures have been
demonstrated to perform the significant feature of digital signal regeneration –
including improvements in extinction ratio, signal to noise ratio, pulse width etc.
More recently, monolithically-integrated tunable all-optical wavelength
converters (TAO-WC)[4] have been demonstrated and have shown promise to
allow for the conversion of one wavelength to another without requiring the
signal to pass through electronics. One further extension of the work in this
thesis – is in the integration of the Mach-Zehnder-SOA-SGDBR with a photo-
detector to provide wavelength conversion over a wide tuning range[717].
196

� 2
Semiconductor Optical Amplifier
Mach-Zehnder Modulator
�1
SGDBR Laser
Franz-Keldysh
MZ optical waveguide
Fig. 6-1. Device layout for the 1 st demonstration of an OEIC wavelength converter using the
Mach-Zehnder-SOA-SGDBR transmitter and Franz-Keldysh detector.
The first implementation of wavelength converters used a Franz-Keldysh
detector that does not involve quantum wells. This type of detector gives a
fairly linear response. Improvements on the design of wavelength converters
will focus on decreasing the optical input requirements with the integration of
SOAs before the detector – and investigation of the linearity and efficiency of
using integrated active quantum-well detectors.
197

Fig. 6-2. Photocurrent Generated in the Franz-Kelydsh detector as a function of optical power
Nonetheless, first results using this configuration seem promising as the
extinction ratio is sufficient to provide

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204

Appendix A Material Properties
Material Properties of InP
Lattice Constant (ao) 5.8687Å
C11 [102] Elastic-stiffness constants 1.022x10 12 dyn/cm 2
C12 [102] 0.576x10 12 dyn/cm 2
C44 [105] 0.46x10 12 dyn/cm 2
� Eo / �P
Hydrostatic pressure coefficient [105]
11x10 -12 eV/dyncm -2
a (Hydrostatic deformation potential) [102] -8.0eV
b (Shear deformation potential) [102] -1.55eV
Density 4.81 g/cm3
Melting Pt. 1335K
Debye Temperature 321K
Direct Energy Gap (300K) 1.34eV
Band gap at 0K Eg(0) 1.421eV
Effective mass (conduction band) 0.075 (mo)
Effective mass(heavy hole) 0.56 (mo)
Effective mass (light hole) 0.12 (mo)
Effective mass (split-off) 0.12 (mo)
Refractive Index near band gap 3.41
Dielectric Constant � (0) n-substrate 12.61
Dielectric Constant Semi-insulating sub [431] 13.3
Dielectric Constant He implanted epi on SI[431] 15.5
High Frequency Dielectric Constant � (�) 9.61
Coefficient of Thermal Expansion (1e-6/C) 4.75
Thermal Conductivity (W/cm K) 0.68
Spin Orbit splitting 0.11eV
Ionic Bonding
Conductivty (ohm-cm)
42%
Base Wafer Specs – Inpact
Doping conc. Thickness Resistivity
(ohm-cm)
S doped 5.5e18 center
(2-8
wafer
across
330-370um 500
Fe doped 1e16 330-370um 1.5E7 4.9E4
205
Etch Pit
density
(cm -2 )

THERMAL PROPERTIES OF SELECTED ELECTRONIC MATERIALS
FROM CRC HANDBOOK
Material
Thermal
Conductivity
(W/cm K)
Density
(g/cm 3 )
Heat Capacity
(J/kg/K)
BCB 0.29
SILK 0.19
AlN 1.7-1.9 3.25 819.7
Al2O3 0.36 3.9
BeO 2.6 2.85
Gold 3.15 19.3
Air 0.00026
Aluminum 2.47
Brass (70Cu-30Zn) 1.15
Copper 3.98 8.96 384.56
Diamond 25
Epoxy 0.0019
Magnesium 1.7
Platinum 0.734
Silver 4.28
Silicon 1.41
Solder (63Sn- 0.5
37Pb)
Sapphire (a-axis) 0.32
Sapphire (c-axis) 0.35
Silicon Carbide 0.9
Silicon Dioxide
(amorphous)
0.014
Silicon Nitride 0.16 - 0.33 2.843
Titanium 0.157
Tungsten 1.78
Zinc 1.13
206

Property Units
Electrical
Resistivity
Dielectric
Constant
Dielectric
Loss
Coefficient of
Thermal
Expansion
Bending
Strength
Hardness
(Knoop)
Youngs
Modulus
AIuminum
Nitride (AlN)
Al2O3
BeO
Ohm-cm >10 14 >10 14 >10 14
8.9 9.8 6.7
0.0001 - 0.001 0.0002 0.0003
10-6/ o C 4.6 8.2 8.5
MPa 290 380 230
GPa 11.8 14.1 9.8
GPa 331 372 345
207

APPENDIX B DEIMBEDDING MICROWAVE INDEX
TRL based calibration was done, using picoprobe 75um probes and a
calibration substrate. Usually, traveling wave electrode structures desire
matching of both the characteristic impedance and the microwave index. The
phase velocity was extracted from S parameter data on the transmission lines.
This was done using an ABCD matrix[304]
�
�
�
A B
C D
�
�
�
�
�
�
�
�
�
( 1�s
11
)( 1�s
2s
22
21
) �s
Z0r is the impedance reference – 50ohms.
The propagation constant is given by[304]
12
s
21
Z
0r
( 1�s
)( 1�s
2s
) �s
1 ( 1�s11
)( 1�s
22 ) �s12s21
( 1�s11
)( 1�s
22 ) �s12s21
ZoR
2s
21
2s
21
[A-1]
� �
2
� � � � �
1 A D A D
� �
�
�
�
� � �
� ln � � � �1�
� � � � j�
L � � 2
��
� �
� 2 �
��
11
22
21
12
s
21
�
�
�
�
�
[A-2]
Alpha is the attenuation coefficient (Np/unit length); Beta is the phase constant (rad/unit length)
208

The characteristic impedance is given by[304]:
Z o
�
B
C
�
D � A �
2B
( A � D)
2 �
4
=
Z o
R � j�L
� �
�
R �
G �
From the above equations we can determine the phase velocity:
v phase
2�f
�
�
209
j�L
j�C
[A-3]
[A-4]

APPENDIX C PROCESS
The process involves 12 different stepper plates as shown in the following
table.
Step
#
Mask Levels
1 Active/Passive
2 Sampled Grating
Layer
3 Ridges
4 Isolation etch
5 InGaAs etch
7 n-metal etch
8 n-metal dep
9 BCB
10 Via Big
11 Via Small
12 P-Metallization
Modulator Process Semi-insulating Substrate
Description Comments
Base Structure Growth - Growth #
1 Cleave into 4 quarters - Clean
Acetone/Isopropanol/N2
2 PECVD Silicon Nitride Deposition
Pre-clean PECVD 30 min SiN clean
Deposit 1000Å SiNx [SiN10]
Measure on Ellipsometer
Record index (n)_________________
Record Thickness___________
3 Photolithography Step #1
(Active/Passive)
Hot plate bake [60 sec. at 110C] Make sure the alignment is correct
Spin coat HMDS [60 sec. at 4krpm] for the ridges!!!
Photoresist coat
210

SPR 950 [60 sec. at 4krpm] Focus = __-16__ Exp = 1sec
Hot plate bake [60 sec. at 95C]
Stepper Program __MZ2/1 Translate Origin x = -11 ; y = 2.5
Expose Resist Pattern Pass Shift x = 19.25; y = 3.00
Develop resist [30 sec. in MF – 701]
DI rinse
After Develop Inspection
4 RIE#3 (Etch 1000Å SiNx)
30 sec 250V O2 10mT descum in chamber
250V with CF4(20sccm) & O2(1.8sccm) at
10mT for 7min (32Watts)
3 min PR burn 200V O2 10mT
Nitride Mask Pattern
Flood Expose - 2min develop MF-701.
Acetone/Isopropanol/N2
Measure on Dektak
Nitride Thickness:
5 Wet Etch (approx 1500A)
Etch in H3PO4:HCl (3:1) - until bubbles
subside - they tend to stick
(1min to make sure it is through)
InP Cap Thickness:
Rinse in DI for 2min under a stream of water
Mix H2SO4:H2O2:H2O (1:1:10) Etch for additional 1/2 intervals
Let cool to room temperature (>30min) until you reach the desired
Etch 1min depth
Time Thickness
6 Strip Nitride
BOE Dip 20 minutes Measure total height
7 PECVD Silicon Nitride Deposition
Pre-clean PECVD 30 min SiN clean
Deposit 200Å SiNx w/ reference Si
sample
Measure on Ellipsometer
Record index (n) Record Thickness
211

8 Photolithography #2 (Sampled Grating
Bursts)
Dehydration Bake 2min at 105C
Spin coat HMDS [1 min at 4krpm] Translate Origin
SPR 950 [1 min at 4krpm] x = -11; y = 2.5
Hot plate bake [1 min. at 95C]
Expose Sample Stepper Program MZ2/2 Pass Shift
Exposure __1.5sec__Focus -10 x = 18.4624; y = 3.9507
Develop resist [25sec in MF-701]
DI rinse
9 RIE#3 (Etch 200Å SiNx)
45 sec 200V O2 10mT descum in chamber
2.5 min etch [10mT] 200V [CF4 = 20 sccm] [O2 = 1.8sccm]
3 min PR burn 200V O2 10mT 20 Watts
Flood expose – develop 3min
Acetone/Isopropanol/DI
Measure on Dektak
Record readings_______________
10 Strip Resist
Expose Sample under aligner
Develop in MF701 for 2 min. Note: if the water sticks
Acetone/Isopropanol/DI to the sample the 3001 will
Stripper @ 90C 10minutes also.
DI Rinse
11 Photolithography #3 (Grating)
(test sample 1st, then live sample)
Turn laser on to warm up for 1 hr
minimum
Notes: this step is tricky. Sometimes
the PR will not stick properly
Hot plate bake [1min at 105C]
Spin coat HMDS [1 min at 5krpm]
Spin on SPR 3001 [1 min at 5krpm] Should give approx 700A PR
thickness
Hot plate bake [1min at 95C]
Load sample on the vacuum holder
Adjust angular alignment on the holder
Grating pattern exp. [35mJ]
Develop resist 25sec MF701
DI rinse with slight agitation in DI beaker
N2 Dry very carefully
Mount sample on test stage D = 23 inches
Place turning mirror in laser path
Align reflection to pass through both slits For 1545nm center, the L should be
close to .95.
Read L from ruler distance between
212

incident and diffracted beams
Calculate the grating period
Period ______________nm
12 PEII
10 sec. at 100W
13 AFM to Insure that the gratings are okay
14 Dry Etch Gratings (RIE II)
Clean chamber w/O2 20 sccm
30 min at 125mT and 500V
Precoat with MHA 4/20/10 sccm
15 min at 75mT and 200V
Load sample
Etch using MHA 4/20/10 sccm
7.5 min at 75mT and 170V
O2 Plasma 20 sccm 5 min at 200V
15 Strip Resist
ACE/ISO/N2/ If gratings are good the area
Stripper 10min 80C will appear blue
AFM to examine the gratings
H2SO4 dip 1 min
16 SiN Removal
BOE (10:1) dip for 20 min.
DI rinse – 5min
UV- Ozone 1 hour
BOE (10:1) dip for 15 sec.
DI rinse – 5min
17 Regrowth
18 SiNx Deposition
Pre-clean PECVD 30 min SiN clean
Load sample
Deposit 1000Å SiNx
Measure thickness
Measure refractive index
19 Photolithography (Ridge)
Bake [ 105C 1min]
HMDS [ 1min 4krpm]
SPR 950 [ 1min 4krpm]
Bake [ 95C 1min]
Expose on stepper MZ2/2 Focus -16
Develop in MF701 25 sec Exposure 1.5sec
20 Descum (RIE#3)
15 sec. at 200V and O2 at 10 mT
21 RIE 3 (Etch 1000Å SiNx)
213

Preclean Chamber O2 20 sccm
20 min at 50mT and 500V
Load sample
Etch SiNx CF4/O2 20sccm/1.8sccm
7 min at 250V
Descum O2 20 sccm
5 min at 10mT and 200V
22 Strip Resist
Flood expose/ Develop in MF701 2min
ACE/ISO/N2/
Stripper 10min 80C
23 Dry Etch Ridge (RIE II)
Clean w/O2 20 sccm
20 min at 125mT and 500V
Precoat with MHA 4/20/10 sccm
15 min at 75mT and 500V
Load sample
Etch using MHA 4/20/10 sccm Measure on Dektak
10min at 75mT and 500V
O2 Descum 5 min at 125mT and 300V
24 Wet Etch (selective clean up etch) Time based on exp.
Mix H3PO4:HCl (3:1)
Wet etch sample until bubbles stop (2min) Measure on Dektak
DI rinse with beam / N2 blow dry
25 Passivation Etch – Lower C in modulators /Detectors
Bake [105C 2min]
HMDS [4000RPM 1min]
220-7 [4000RPM 1min]
Bake [115C 1min]
Pattern with BCB layer 4.5sec exposure
Develop 1.5minutes MF701
26 Passivation Dry Etch (RIE II)
Clean chamber w/O2 20 sccm
30 min at 125mT and 500V
Precoat with MHA 4/20/10 sccm
15 min at 75mT and 200V
Load sample
Etch using MHA 4/20/10 sccm
7.5 min at 75mT and 170V
O2 Plasma 20 sccm 5 min at 200V
27 SiNx Removal
BOE (10:1) dip for 15 min.
214

DI rinse – 5min
28 SiNx Deposition
Pre-clean PECVD 30 min SiN clean
Load sample
Deposit 2000Å SiNx
Measure thickness
Measure refractive index
29 Photolithography [n-etch]
Bake [105C 2min]
HMDS [4000RPM 1min]
220-7 [4000RPM 1min]
Bake [115C 90sec]
Exposure MZ2/netch 4.5sec
30 RIE #3 SiNx Etch (2000A)
Preclean Chamber O2 20 sccm
20 min at 50mT and 500V
Load sample
Etch SiNx CF4/O2 20sccm/1.8sccm
7 min at 250V
Descum O2 20 sccm
5 min at 10mT and 200V
31 Strip Resist
Flood expose/ Develop in MF701 2min
ACE/ISO/N2/
Stripper 10min 80C
32 RIE #2 Etch
Clean chamber w/O2 20 sccm
30 min at 125mT and 500V
Precoat with MHA 4/20/10 sccm
15 min at 75mT and 200V
Load sample
Etch using MHA 4/20/10 sccm
(15 + 20 + 6) min at 75mT and 450V
O2 Plasma 20 sccm 5 min at 200V
33 InP Wet Etch
Mix H3PO4:HCl (3:1)
Wet etch sample until bubbles stop (2min)
Dektak
34 Photolithography N-Metal
Bake [105C 2min]
HMDS [4000RPM 1min]
PMGI SF11 [4000RPM 1min]
Bake [165C 1min]
215
Need to etch through QWs and
waveguide in certain areas – then a
wet etch can proceed.

4110 [4000RPM 1min]
Bake [95C 30sec]
4110 [4000RPM 1min]
Bake [95C 1min]
Exposure MZ2/netch 2.5sec Exposure
35 Deposition Preparation
30sec RIE #3 250V or PE II
UV Ozone 10min
BOE Dip (1:10) 30 sec.
5 min DI rinse
HCl (1:10 H20) 15 sec
2 min DI rinse
Load Immediately
36 N Metal Deposition
Load sample in E-Beam #1
Evaporate Ni/AuGe/Ni/Au
1 pellet of AuGe deposited to
(50/200/200/10000)
completion
Lift-off with Acetone and pipette
Remove SF11 with Stripper 10min 80C
DI Rinse
37 Anneal – Strip Annealer
430-450C 30sec Test the contacts – make sure they
are good.
38 SiNx Deposition
Pre-clean PECVD 30 min SiN clean
Load sample
Deposit 2000Å SiNx
Measure thickness
Measure refractive index
39 Photolithography - Isolation Etch
Bake [105C 2min]
HMDS [4000RPM 1min]
220-7 [3000RPM 1min]
Bake [115C 90sec]
Exposure Isolation Etch MZ2/2 3.75sec exposure
Develop 1.5min MF-701
40 RIE #2 Etch
Clean chamber w/O2 20 sccm Want to etch through 500A n+
InGaAs layer and 0.5um n-InP layer
30 min at 125mT and 500V
Precoat with MHA 4/20/10 sccm
15 min at 75mT and 200V
Load sample
Etch using MHA 4/20/10 sccm
216

10 min at 75mT and 450V
O2 Plasma 20 sccm 5 min at 200V
41 Photolithography (Implant)
Bake [105C 2min]
HMDS [4000RPM 1min]
PMGI SF15 [4000RPM 1min]
Bake [165C 1min]
220-7 [3000RPM 1min]
Bake [115C 90sec]
Expose MZ2/2 4.5sec Exposure
Develop MF-701 90 sec.
DUV Exposure 5 minutes
Develop SAL-101 1 min.
Deep UV Flood exposure 100 sec.
Develop SAL-101 30sec
Dektak PR thickness
42 Implant (Isolation Implant )
Implant with H per table
20keV 4x10E13cm-2
55keV 5x10E13cm-2
110keV 7x10E13cm-2
175keV 9x10E13cm-2
43 Strip Resist
Flood expose/ Develop in MF-701 2min
ACE/ISO/N2
Stripper 10min 80C
44 SiNx Removal
BOE (10:1) dip for 15 min.
DI rinse – 5min
45 SiNx Deposition (3000A)
Pre-clean PECVD 30 min SiN clean
Load sample
Deposit 3000Å SiNx
Measure thickness
Measure refractive index
46 Photo-lithography InGaAs Etch
Bake [105C 1min]
220-3 [4000RPMs 1min]
Bake [115C 90sec]
Exposure MZ2/ 2.5sec
Develop MF-701 1min
47 RIE 3 (Etch 3000Å SiNx)
217

Preclean Chamber O2 20 sccm
20 min at 50mT and 500V
Load sample and align laser monitor
Etch SiNx CF4/O2 20sccm/1.8sccm
21 min at 250V
Descum O2 20 sccm
5 min at 10mT and 250V
48 Strip Resist
Flood expose/ Develop in MF-701 2min
ACE/ISO/N2
Stripper 10min 80C
49 Remove InGaAs contact layer
Etch InP Cap off
H3PO4:HCl 3:1 10sec
DI Rinse 2min
H3PO4/H2O2/H20 (3:1:50) 1min
DI Rinse
50 Deposit BCB
Spin BCB Adhesion Promoter [5000RPM
1min]
Photo-BCB [5000RPMs 1min]
Cure [70C 1min]
Exposure MZ2/BCB2 5sec
Develop on Spinner
Puddle BCB developer 1min
Spin [1 min 4000RPMs]
Puddle BCB developer 1min
Spin [1 min 4000RPMs]
Bake in Programable oven program #2
51 Deposit SiNx (1000A)
Pre-clean PECVD 30 min SiN clean
Load sample
Deposit 1000Å SiNx
Measure thickness
Measure refractive index
52 BCB Via #1
Bake [105C 1min]
HMDS [4000RPMs 1min]
220-7 [4000RPMs 1min]
Bake [115C 1min]
Expose MZ2/Via 3.75sec X = 3.9498; Y = -2.9607
Develop MF-701 1.5min
53 RIE #3 Etch SiN + BCB
218

Preclean Chamber O2 20 sccm
20 min at 50mT and 500V Etch widens out with O2 so another
Load sample Patterning is required
Etch SiNx CF4/O2 20sccm/1.8sccm
7 min at 250V
Etch BCB CF4/O2 4sccm/16sccm 20mT
350V 10 + 10 + 5 + 3 min
54 BCB Via #2
Bake [105C 1min]
HMDS [4000RPMs 1min]
220-3 [4000RPMs 1min]
Expose MZ2/Via 2sec X = 3.9498; Y = -2.9607
Develop MF-701 1min
55 RIE #3 Etch SixNy
Preclean Chamber O2 20 sccm
20 min at 50mT and 500V Etch widens out with O2 so another
Load sample patterning is required
Etch SixNy CF4/O2 20sccm/1.8sccm
16 min at 350V
56 Laser Via
Bake [105C 1min]
220-7 [4000RPMs 1min]
Bake [115C 90sec]
Exposure MZ2/Via2 1.2sec exp
Develop MF-701 35sec
57 RIE #3 O2
Preclean Chamber O2 20 sccm
20 min at 50mT and 500V
Load sample
Etch O2 20sccm 300V
5 + 2.5 + 2.5 + 1 + 1.5 + 2 + 2
58 RIE #3 SiNy etch
Etch SixNy CF4/O2 20sccm/1.8sccm
18 min at 350V
59 Strip Resist
Flood expose/ Develop in MF-701 2min
ACE/ISO/N2
Stripper 10min 80C
60 Photolithography – P-Metal
Bake [105C 2min]
HMDS [4000RPM 1min]
PMGI SF11 [4000RPM 1min]
Bake [165C 1min]
219

4110 [4000RPM 1min]
Bake [95C 30sec]
4110 [4000RPM 1min]
Bake [95C 1min]
Exposure MZ2/pmetal 2.5sec Exposure
Develop AZ400K [1:4 1min]
DUV 5min
Develop SAL-101 1min
PEII [30sec 100W]
UV Ozone 10min
BOE [1:10 30sec]
DI Rinse 5min
61 P-Metal Deposition
Load sample in E-Beam #3
Evaporate Ti/Pt/Au (200/400/5000) No Angle
Load into E-Beam #1
Evaporate Ti/Au (200/10000) Angled Spinning
Lift-off with Acetone and pipette
Remove SF11 with Stripper 10min 80C
DI Rinse
62 Photolithography – SiNy on nmetal Etch
Bake [105C 2min]
HMDS [4000RPM 1min]
220-7 [3000RPM 1min]
Bake [115C 90sec]
Exposure N-metal Dep MZ2/nmetal 3.75sec exposure
Develop 1.5min MF-701
63 RIE #3 SiNy Etch
Preclean Chamber O2 20 sccm
20 min at 50mT and 500V
Load sample
Etch SiNx CF4/O2 20sccm/1.8sccm
21 min at 350V
Descum O2 20 sccm
5 min at 10mT and 200V
Verify that the contacts are good before
thinning wafer
64 Wafer Thinning
Attach sample to Silicon wafer with
crystalbond
Use wood pieces to insure that the sample
is flat. Place vacuum on top of sample and
Let cool to room temperature
Lap sample to 100um using 12um grit in
220

a figure 8 pattern. Dismount the silicon wafer
for metalization.
65 N-Type metallization
Place sample in Ebeam #3
Deposit Ti/Pt/Au 200/400/5000 for back side
Contact
66 Metal Anneal
Strip annealer
30 sec at 390C
67 Cleave samples into Bars
Heat the silicon wafer to 130C and gently
slide the samples off the silicon
Clean in Acetone/Isopropanol
Scribe and cleave into laser bars
68 AR Coating
69 Test Devices
221