THE RADIO HANDBOOK.pdf - N3UJJ
THE RADIO HANDBOOK.pdf - N3UJJ
THE RADIO HANDBOOK.pdf - N3UJJ
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This book is revised and brought upto date (at irregular intervals) asnecessitated by technical progress.<strong>THE</strong>R AD I 0<strong>HANDBOOK</strong>Fifteenth EditionThe Standard of the Field -for aduanced amateurspractical radiomenpractical engineerspractical techniciansWILLIAM I. ORR, W6SAIEditor, 1 Sth Edition$7.50 per copy atyour dealer in U.S.A.(Add 10% on directorders to publisher)Published and Distributed to the Radio Trade bySUMMERLAND, CALIFORNIA, U.S.A.(Distributed to the Book and News Trades and Libraries by the Baker & Taylor Co., Hillside, N. J.)
<strong>THE</strong> <strong>RADIO</strong> <strong>HANDBOOK</strong>FIFTEENTH EDITIONCopyright, 1959, byEditors ond Engineers, Ltd.Summerland, California, U.S.A.Copyright under Pan-American ConventionAll Translation Rights ReservedPrinted in U.S.A.The "Radio Handbook" in Spanish or Italian is available from us at $8.25postpaid. French and Dutch editions in preparation.Outside North America, if more convenient, write: (Spanish) Marcombo, S.A., Av.Jose Antonio, 584, Barcelona, Spain; (Italian) Edizione C.E.L.I., Via Gandino 1,Bologna, Italy; (french or Dutch) P. H. Brans, Ltd., 28 Prins Leopold St., Borgerhout,Antwerp, Belgium.Other Outstanding Books from the Same Publisher(See Announcements at Back of Book)<strong>THE</strong> <strong>RADIO</strong>TELEPHONE LICENSE MANUAL<strong>THE</strong> SURPLUS <strong>RADIO</strong> CoNvERSION MANUALS<strong>THE</strong> WoRLD's <strong>RADIO</strong> TuBES (<strong>RADIO</strong> TuBE VADE MECUM)<strong>THE</strong> WoRLD's EQUIVALENT TuBES (EQUIVALENT TuBE VADE MEcuM)<strong>THE</strong> WoRLD's TELEVISION TuBES (TELEVISION TuBE VADE MECUM)
<strong>THE</strong> <strong>RADIO</strong> <strong>HANDBOOK</strong>1 Sth EditionTable of ContentsChapter One. INTRODUCTION TO <strong>RADIO</strong> ................................................1 -1 Amateur Radio ......................................................................1-2 Station and Operator Licenses ..............................................1-3 The Amateur Bands ................................................................1-4 Starting Your Study ................................................................1111121214Chapter Two. DIRECT CURRENT CIRCUITS ....................................................2-1 The Atom ..............................................................................2-2 Fundamental Electrical Units and Relationships ......................2-3 Electrostatics - Capacitors .................................................... 302-4 Magnetism and Electromagnetism .......................................... 352-5 RC and RL Transients ............................................................ 38Chapter Three. ALTERNATING CURRENT CIRCUITS ....................................3-1 Alternating Current ................................................................3-2 Resonant Circuits ..................................................................3-3 Nonsinusoidal Waves and Transients .................................... 583-4 Transformers .......................................................................... 613-5 Electric Filters .......................................................................... 63Chapter Four. VACUUM TUBE PRINCIPLES.................................................. 674-1 Thermionic Emission ................................................................ 674-2 The Diode .............................................................................. 714-3 The Triode .............................................................................. 724-4 Tetrode or Screen Grid Tubes ................................................ 774-5 Mixer and Converter Tubes .................................................... 794-6 Electron Tubes at Very High Frequencies ................................ 804-7 Special Microwave Electron Tubes .......................................... 814-8 The Cathode-Ray Tube ..................................................... ..... 844-9 Gas Tubes ..............................................•............................... 874-10 Miscellaneous Tube Types ...................................................... 88Chapter Five. TRANSISTORS AND SEMI-CONDUCTORS.............................. 905-1 Atomic Structure of Germanium and Silicon .......................... 905-25-35-45-55-6Mechanism of Conduction ......................................................The TransistorTransistor Characteristics ........................................................Transistor Circuitry ..................................................................Transistor Circuits909294961033212122414153
Chapter Eleven. ELECTRONIC COMPUTERS.................................................. 1 9411-1 Digital Computers ··-···---------···-----·------··············-··-······---·······-- 19511-2 Binary Notation -------···-----····---········---···------···-··--··------------------ 19511-3 Analog Computers ------····---······------------·····--··------··-·----------··---- 19711-4 The Operational Amplifier ·------·····-----··---·-···-··--··--·--------·····---- 19911-5 Solving Analog Problems --········-·····--··-··-----······--··--------·---··--·- 20011-6 Non-linear Functions ·-------····-----··-----·---····-····----··-----··--··-------· 20211-7 Digital Circuitry --···-·---····----··--··--··-·---·-·------··--········-··· 204Chapter Twelve. <strong>RADIO</strong> RECEIVER FUNDAMENTALS ..........--·-----·---·----·--·----- 20712-1 Detection or Demodulation 20712-2 Superregenerative Receivers --···----·--------··-------···-----------·---·------ 20912-312-4Superheterodyne Receivers ·-----··--·----···---·------·-···--·------·-----···· 210Mixer Noise and Images --··-------··-----------·--··--·····-·-----------·····-- 21212-5 R-F Stages ·-----·-----··------··----··----····-·····---·--··--·-···------···----·-···· 21 312-6 Signal-Frequency Tuned Circuits ··----·-----·············----··--···---·------ 21612-712-81-F Tuned Circuits -----··--·-···-·--·-----·-----·----·-·--··--····------·------·-·-· 218Detector, Audio, and Control Circuits ---·--·---·-··-···----··---···---·-· 22512-9 Noise Suppression ----------------··---·--------------------·---·---·-··---·-·------ 22712-10 Special Considerations in U-H-F Receiver Design ··--··--····------ 23112-1112-12Receiver Adjustment ·------------·-----·------··-----·------··---·--------·---··---- 235Receiving Accessories ----···----·-----·------·-----------·-·-·-·---··---··------· 236Chapter Thirteen. GENERATION OF <strong>RADIO</strong> FREQUENCY ENERGY .............. 23913-1 Self-Controlled Oscillators ·-----·-----··---·--····-·····-···-------------------- 23913-2 Quartz Crystal Oscillators ---·-·-----·---------·---------··-···----·-----·---··· 24413-313-413-513-613-713-813-913-1013-1113-1213-1313-1413-15Crystal Oscillator Circuits ·-----··----··----------············---···---·--·---··-· 247Radio Frequency Amplifiers ···----·-----·----·---·-··--········----··--··-··--- 251Neutralization of R.F. Amplifiers ·------·--·-··-------·----·-··--··---------· 252Neutralizing Procedure ------------·--------·-··---··------------·-------·-·----·-· 255Grounded Grid Amplifiers ··----··-----·--·-···---··--·----··----·-·---------- 258Frequency Multipliers ·------·-----·-----·----------·-----··--·---·-··--··---------- 258Tank Circuit Capacitances ---···---···-----·---····----··--·--·-----··--··--···· 261L and Pi Matching Networks ---------------·-·----------------------·--------- 265Grid Bias ---·----------··-----·---··---··----·-----·-----------·-----··-··-----------··-- 267Protective Circuits for Tetrode Transmitting Tubes ----------··---·-· 269lnterstage Coupling -·-----·----------------------------·-----·---------·--------- 270Radio-Frequency Chokes ··----··--·--··-··-··-·-·---·-······-·--··--··-·-------· 272Parallel and Push-Pull Tube Circuits ----·---·-··--····---·-·-------------- 273Chapter Fourteen. R-F FEEDBACK·----··-------------··-·---···---···--····-··----------·------ 27414-1 R-F Feedback Circuits --------··---·----------·-----·-----·----------------------- 27414-2 Feedback and Neutralization of a Two-Stage R-F Amplifier .... 27714-3 Neutralization Procedure in Feedback-Type Amplifiers --····-- 279Chapter Fifteen. AMPLITUDE MODULATION .........................-----------------·--- 28215-1 Sidebands ---·-------------------·--····----··--····--···-----··---·-------------------- 28215-2 Mechanics of Modulation ··----·-----------·-----·-----------·------·----··---- 28315-3 Systems of Amplitude Modulation ---·---------··-··--····-----------------· 28515-4 Input Modulation Systems ---·------------·-·-----------·-----·---------··---· 29215-5 Cathode Modulation -----------·----··------·-----·-----··----·-----------·---··-· 29715-6 The Doherty and the Termon-Woodyard Modulated Amplifiers .. 29815-7 Speech Clipping ·----·-----·-----·---------·-·---·-·----··---···-···-----·-·------·· 30015-8 The Bias-Shift Heising Modulator 3075
Chapter Sixteen. FREQUENCY MODULATION AND REDIOTELETYPETRANSMISSION .................................................................. 31216-1 Frequency Modulation ............................................................ 31216-2 Direct FM Circuits .................................................................. 31516-3 Phase Modulation .................................................................. 31916-4 Reception of FM Signals ........................................................ 32116-5 Radio Teletype ........................................................................ 326Chapter Seventeen. SIDEBAND TRANSMISSION ........................................ 32717-1 Commercial Applications of SSB .............................................. 32717-2 Derivation of Single-Sideband Signals .................................... 32817-3 Carrier Elimination Circuits .......................•............................ 33217-4 Generation of Single-Sideband Signals .................................. 33417-5 Single Sideband Frequency Conversion Systems .................... 34017-6 Distortion Products Due to Nonlinearity of R-F Amplifiers ...... 34417-7 Sideband Exciters .................................................................. 34617-8 Reception of Single Sideband Signals .................................... 35117-9 Double Sideband Transmission .............................................. 353Chapter Eighteen. TRANSMITTER DESIGN.................................................... 35618-1 Resistors ................................................................................ 35618-2 Capacitors .............................................................................. 35818-3 Wire and Inductors ................................................................ 36018-4 Grounds .................................................................................. 36218-5 Holes, Leads and Shafts .......................................................... 36218-6 Parasitic Resonances .............................................................. 36418-7 Parasitic Oscillation in R-F Amplifiers .................................... 36518-8 Elimination of V-H F Parasitic Oscillations ............................ 36618-9 Checking for Parasitic Oscillations .......................................... 368Chapter Nineteen. TELEVISION AND BROADCAST INTERFERENCE .............. 37119-1 Types of Television Interference .............................................. 37119-2 Harmonic Radiation ................................................................ 37319-3 Low-Pass Filters ...................................................................... 37619-4 Broadcast Interference ............................................................ 37919-5 HI-FI Interference .................................................................. 3 86Chapter Twenty. TRANSMITTER KEYING AND CONTROL ............................ 38720-1 Power Systems ........................................................................ 38720-2 Transmitter Control Methods .................................................. 39120-3 Safety Precautions .................................................................. 39320-4 Transmitter Keying .................................................................. 39520-5 Cathode Keying .................................................................... 39720-6 Grid Circuit Keying ................................................................ 39820-7 Screen Grid Keying ................................................................ 39920-8 Differential Keying Circuits .................................................... 400Chapter Twenty-One. RADIATION, PROPAGATION AND TRANSMISSIONLINES .................................................................................. 40321-1 Radiation from an Antenna .................................................... 40321-2 General Characteristics of Antennas ...................................... 40421-3 Radiation Resistance and Feed-Point Impedance .................... 40721-4 Antenna Directivity ................................................................ 41021-5 Bandwidth .......................•...................................................... 41 36
Chapter Thirty-Two. POWER SUPPLlES .................................................... 68432-1 Power Supply Requirements .................................................. 68432-2 Rectification Circuits ................, ............................................. 68932-3 Standard Power Supply Circuits ............................................ 69032-4 Selenium and Silicon Rectifiers .............................................. 69532-5 I 00 Watt Mobile Power Supply ............................................ 69732-6 Transistorized Power Supplies ................................................ 70332-7 Two Transistorized Mobile Supplies ........................................ 70632-8 Power Supply Components ...................................................... 70732-9 Special Power Supplies .......................................................... 70932-10 Power Supply Design ............................................................ 71232-11 300 Volt, 50 Ma. Power Supply .............................................. 71 532-12 500 Volt, 200 Milliampere Power Supply .............................. 71632-13 I 500 Volt, 425 Milliampere Power Supply ............................ 71732-14 A Dual Voltage Transmitter Supply ........................................ 71832-15 A Kilowatt Power Supply ........................................................ 71 8Chapter Thirty-Three. WORKSHOP PRACTICE .............................................. 72033-1 Tools ...................................................................................... 72033-2 The Material ........................................................................ 723-A33-3 TVI-Proof Enclosures ............................................................ 724-A33-4 Enclosure Openings .............................................................. 725-A33-5 Summation of the Problem .................................................. 725-A33-6 Construction Practice ............................................................ 726-A33-7 Shop Layout ......................................................................... 729-AChapter Thirty-Four. ELECTRONIC TEST EQUIPMENT............................. 721-B34-1 Voltage, Current and Power ................................................ 721-B34-2 Measurement of Circuit Constants ........................................ 727 -B34-3 Measurements with a Bridge ............................................... 728-B34-4 Frequency Measurements ............••........................................ 729-B34-5 Antenna and Transmission Line Measurements.................... 730-B34-6 A Simple Coaxial Reflectometer ........................................ 73234-734-834-934-10Measurements on Balanced Transmission Lines ...................... 734A "Balanced" SWR Bridge .................................................... 736The Antennascope .................................................................. 738A Silicon Crystal Noise Generator .......................................... 740Chapter Thirty-Five. <strong>RADIO</strong> MA<strong>THE</strong>MATICS AND CALCULATIONS ............ 7429
FOREWORD TO <strong>THE</strong> FIFTEENTH EDITIONOver two decades ago the historic first edition of the <strong>RADIO</strong> <strong>HANDBOOK</strong> waspublished as a unique, independent, communications manual written especially for theadv meed radio amateur and electronic engineer. Since that early issue, great pains havebeen taken to keep each succeeding edition of the <strong>RADIO</strong> <strong>HANDBOOK</strong> abreast ofthe rapidly expanding field of electronics.So quickly has the electron invaded our everyday affairs that it is now no longerpossible to segregate one particular branch of electronics and define it as radio communications;rather, the transfer of intelligence by electrical means encompasses morethan the vacuum tube, the antenna, and the tuning capacitor.Included in this new, advanced Fifteenth Edition of the <strong>RADIO</strong> <strong>HANDBOOK</strong> arefresh chapters covering electronic computers, r.f. feedback amplifiers, and high fidelitytechniques, plus greatly expanded chapters dealing with semi-conductors and specialvacuum tube circuits. The other chapters of this Handbook have been thoroughlyrevised and brought up to date, touching briefly on those aspects in the industrialand military electronic fields that are of immediate interest to the electronic engineerand the radio amateur. The construction chapters have been completely re-edited. Allnew equipments described therein are of modern design, free of TVI problems andvarious unwanted parasitic oscillations. An attempt has been made not to duplicateitems that have been featured in contemporary magazines. The transceiver makes itsmajor bow in this edition of the <strong>RADIO</strong> <strong>HANDBOOK</strong>, and it is felt that thiscomplete, inexpensive, compact "radio station" design will become more popularduring the coming years.The writing and preparation of this Handbook would have been impossible withoutthe lavish help that was tended the editor by fellow amateurs and sympathetic electronicorganizations. Their friendly assistance and helpful suggestions were freelygiven in the true amateur spirit to help make the 15th edition of the <strong>RADIO</strong> HANDBOOK an outstanding success.The editor and publisher wish to thank these individuals and companies whoseunselfish support made the compilation and publication of this book an interestingand inspired task.-WILLIAM I. ORR, W6SAI, 3A2AF, EditorE. P. Alvernaz, W6DMN,Jennings Radio Co.Kenneth Bay, W2GSJ,General Electric Co.Orrin H. Brown, W6HB,Eitel-McCullough, Inc.Wm. E. Bruring, W9ZSO,E. F. Johnson, Inc.Thomas Consalvi, W3EOZ,Barker & Williamson, Inc.Cal Hadlock, WlCTW,National Co., Inc.Jo E. Jennings, W6EI,Jennings Radio Co.AI Kahn, W8DUS,Electrovoice, Inc.Ken Klippel, WOSQO,Collins Radio Co.Roger Mace, W8MWZ,Heath Co.E. R. Mullings, W8VPN,Heath Co.Edw. A. Neal, W2JZK,General Electric Co.Edw. Schmeichel, W9YFV,Chicago-StandardTransformer Co.Wesley Schum, W9DYV,Central Electronics, Inc.Aaron Self, W8FYR,Continental Electronics& Sound Co.Harold Vance, K2FF,Radio Corporation ofAmericaJ. A. Haimes, Semi-conductorDivision, Radio Corporationof AmericaSpecial thanks are due CollinsRadio Co. for permission toreprint portions of theirSideband Report CTR-113by Warren Bruene, WOTTKBud Radio Co., Inc.California Chassis Co., Inc.Cardwell Condenser Co., Inc.Centralab, Inc.Comeli-Dubilier ElectricCo., Inc.Cowan Publishing Corp.International BusinessMachines Co., Inc.Marion ElectricalInstrument Co., Inc.Miller Coil Co., Inc.Raypar, Inc.Raytheon Mfg. Co., Inc.Sarkes-Tarzian, Inc.Sprague Electric Co.Triad Transformer Co.Bob Adams, W6A VAFrank Clement, W6KPCAI Cline, W6LGUTemple Ehmsen, W7VSTed Gillett, W6HXBill Glaser, W60KGBill Guimont,W6YMDTed Henry, W6UOUHerbert Johnson, W7GRAJames Lee, W6VATEarl Lucas, W2JTBill Mauzey, W6WWQKen Pierce, W6SLQDon Stoner, W6TNSBob Thompson, K6SSJKarl Trovinger, \V6KMKBill Vandermay, W7DETDick West, W6IUGEdward Willis, W6TSJoseph J asgur(photography)B. A. Ontiveros, W6FFF(drafting)Del Rairigh, W6ZAT
CHAPTER ONEIntroductionto RadioThe field of radio is a division of the muchlarger field of electronics. Radio itself is sucha broad study that it is still further brokendown into a number of smaller fields of whichonly shortwave or high-frequency radio is coveredin this book. Specifically the field of communicationon frequencies from 1.8 to 450 megacyclesis taken as the subject matter for thiswork.The largest group of persons interested inthe subject of high-frequency communication isthe more than 150,000 radio amateurs locatedin nearly all countries of the world. Strictlyspeaking, a radio amateur is anyone interestedin radio non-commercially, but the term is ordinarilyapplied only to those hobbyists possessingtransmitting equipment and a license fromthe government.It was for the radio amateur, and particularlyfor the serious and more advanced amateur,that most of the equipment described inthis book was developed. However, in eachequipment group simple items also are shownfor the student or beginner. The design principlesbehind the equipment for high-frequencyradio communication are of course the samewhether the equipment is to be used for commercial,military, or amateur purposes, theprincipal differences 1 yin g in constructionpractices, and in the tolerances and safetyfactors placed upon components.With the increasing complexity of high-ftequencycommunication, resulting primarily fromincreased utilization of the available spectrum,it becomes necessary to delve more deeplyinto the basic principles underlying radiocommunication, both from the standpoint ofequipment design and operation and from thestandpoint of signal propagation. Hence, it willbe found that this edition of the <strong>RADIO</strong> HANDBOOK has been devoted in greater proportion11to the teaching of the principles of equipmentdesign and signal propagation. It is in responseto requests from schools and agencies of theDepartment of Defense, in addition to persistentrequests from the amateur radio fraternity,that coverage of these principles has been expanded.1-1 Amateur RadioAmateur radio is a fascinating hobby withmany phases. So strong is the fascination offeredby this hobby that many executives, engineers,and military and commercial operatorsenjoy amateur radio as an avocation eventhough they are also engaged in the radio fieldcommercially. It captures and holds the interestof many people in all walks of life, and inall countries of the world where amateur activitiesare permitted by law.Amateurs have rendered much public servicethrough furnishing communications to andfrom the outside world in cases where disasterhas isolated an area by severing all wire communications.Amateurs have a proud record ofheroism and service in such occasion. Manyexpeditions to remote places have been keptin touch with home by communication with amateurstations on the high frequencies. The amateur'sfine record of performance w i t h the"wireless" equipment of World War I has beensurpassed by his outstanding service in WorldWar II.By the time peace came in the Pacific inthe summer of 1945, many thousand amateuroperators were serving in the allied armedforces. They had supplied the army, navy,marines, coast guard, merchant marine, civilservice, war plants, and civilian defense organizationswith trained personnel for radio,
12 Introduction to Radio<strong>THE</strong> R AD I 0radar, wire, and visual communications andfor teaching. Even now, at the time of thiswriting, amateurs are being called back intothe expanded defense forces, are returning todefense plants where their skills are criticallyneeded, and are being organized into communi·cation units as an adjunct to civil defensegroups.1-2 Station and Operator LicensesEvery radio transmitting station in theUnited States no matter how low its powermust have a license from the federal govern·ment before being operated; some classes ofstations must have a permit from the governmenteven before being constructed. And everyoperator of a transmitting station must havean operator's license before operating a trans·mitter. There are no exceptions. Similar lawsapply in practically every major country."Classes of Amateur There are at present sixOperator Licenses classes of amateur oper·ator licenses which havebeen authorized by the Federal Communica·tions Commission. These c 1 asses differ inmany respects, so each will be discussedbriefly.(a) Amateur Extra Class. This class of licenseis available to any U. S. citizen who atany time has held for a period of two years ormore a valid amateur license, issued by theFCC, excluding licenses of the Novice andTechnician Classes. The examination for thelicense includes a code test at 20 words perminute, the usual tests covering basic amateurpractice and general amateur regulations, andan additional test on advanced amateur practice.All amateur privileges are accorded theholders of this operator's license.(b) General Class. This class of amateurlicense is equivalent to the old Amateur ClassB license, and accords to the holders all ama·teur privileges except those which may be setaside for holders of the Amateur Extra Classlicense. This class of amateur operator's licenseis available to any U. S. citizen. Theexamination for the license includes a codetest at 13 words per minute, and the usual ex·aminations covering basic amateur practiceand general amateur regulations.(c) Conditional Class. This class of ama·teur license and the privileges accorded by itare equivalent to the General Class license.However, the license can be issued only tothose whose residence is more than 125 milesairline from the nearest location at which FCCexaminations are held at intervals of not morethan three months for the General Class ama·teur operator license, or to those who for anyof several specified reasons are unable to appearfor examination.(d) Technician Class. This is a new classof license which is available to any citizen ofthe United States. The examination is the sameas that for the General Class license, exceptthat the code test is at a speed of 5 words perminute. The holder of a Technician class licenseis accorded all authorized amateur privilegesin the amateur frequency bands above220 megacycles. and in the 50-Mc. band.(e) Novice Ctass. This is a new class oflicense which is available to any U. S. citizenwho has not previously held an amateur licenseof any class issued by any agency ofthe U. S. government, military or civilian. Theexamination consists of a code test at a speedof 5 words per minute, plus an examination onthe rules and regulations essential to beginner'soperation, inCluding sufficient elemen·tary radio theory for the understanding of thoserules. The Novice Class of license affordsseverely restricted privileges, is valid for onlya period of one year (as contrasted to all otherclasses of amateur licenses which run for aterm of five years), and is not renewable.All Novice and Technician class examinationsare given by volunteer examiners, as regularexaminations for these two classes arenot given in FCC offices. Amateur radio clubsin the larger cities have established examining committees to assist would-be amateursof the area in obtaining their Novice and Technicianlicenses.1-3 The Amateur BandsCertain small segments of the radio frequen·cy spectrum between 1500 kc. and 10,000 Me.are reserved for operation of amateur radiostations. These segments are in general agree·ment throughout the world, although certainparts of different amateur bands may be usedfor other purposes in various geographic re·gions. In particular, the 40-meter amateur bandis used legally (and illegally) for short wavebroadcasting by many countries in Europe,Africa and Asia. Parts of the SO-meter bandare used for short distance marine work in Europe,and for broadcasting in South America.The amateur bands available to American ra·dio amateurs are!160 Meters The 160-meter band is di(1800 Kc.-2000 Kc.) vided in to 25-kilocyclesegments on a regionalbasis, with day and night power limitations,and is available for amateur use provided nointerference is caused to the Loran (LongRange Navigation) stations operating in thisband. This band is least affected by the 11-
<strong>HANDBOOK</strong>Amateur Bands 1 3year solar sunspot cycle. The Maximum UsableFrequency (MUF) even during the yearsof decreased sunspot activity does not usuallydrop below 4 Me., therefore this band is notsubject to the violent fluctuations found onthe higher frequency bands. DX contacts onon this band are limited by the ionosphericabsorption of radio signals, which is quitehigh. During winter nighttime hours the absorptionis often of a low enough value to permittrans-oceanic contacts on this band. Onrare occasions, contacts up to 10,000 mileshave been made. As a usual rule, however,160-meter amateur operation is confined toground-wave contacts or single-skip contactsof 1000 miles or less. Popular before WorldWar II, the 160-meter band is now only sparselyoccupied since many areas of the countryare blanketed by the megawatt pulses of theLoran chains.80 Meters The SO-meter band is the(3500 Kc.-4000 Kc.) most popular amateurband in the continentalUnited States for I o c a! "rag-chewing" andtraffic nets. During the years of minimum sunspotactivity the ionospheric absorption onthis band may be quite low, and long distanceDX contacts are possible during the winternight hours. Daytime operation, in general, islimited to contacts of 500 miles or less. Duringthe summer months, local static and highionospheric absorption limit long distance contactson this band. As the sunspot cycle advancesand the MUF rises, increased ionosphericabsorption will tend to degrade thelong distance possibilities of this band. Atthe peak of the sunspot cycle, the SO-meterband becomes useful only for short-haul communication.mitters. In Europe and Asia the band is in achaotic state, and amateur operation in this regionis severely hampered.20 Meters At the present time,(14,000 Kc.-14,350 Kc.) the 20-meter band isby far the most popularband for long distance contacts. High enoughin frequency to be almost obliterated at thebottom of the solar cycle, the band neverthelessprovides good DX contacts during yearsof minimal sunspot activity. At the presenttime, the band is open to almost all parts ofthe world at some time during the year. Duringthe summer months, the band is active untilthe late evening hours, but during the wintermonths the band is only good for a fewhours during day light. Extreme DX contactsare usually erratic, but the 20-meter band isthe only band available for DX operation theyear around during the bottom of the DX cycle.As the sunspot count increases and the MUFrises, the 20-meter band will become open forlonger hours during the winter. The maximumskip distance increases, and DX contacts arepossible over paths other than the Great Circleroute. Signals can be heard the "long paths,"180 degrees opposite to the Great Circle path.During daylight hours, absorption may becomeapparent on the 20-meter band, and all signalsexcept very short skip may disappear. On theother hand, the band will be open for worldwideDX contacts all night long. The 20-meterband is very susceptible to "fade-outs"caused by solar disturbances, and all exceptlocal signals may completely disappear forperiods of a few hours to a day or so.40 Meters The 40-meter band is high(7000 Kc.-7300 Kc.) enough in frequency to beseverely affected by the11-year sunspot cycle. During years of minimumsolar activity, the MUF may drop below7 Me., and the band will become very erratic,with signals dropping completely out duringthe night hours. Ionospheric absorption of signalsis not as large a problem on this band asit is on 80 and 160 meters. As the MUF graduallyrises, the skip-distance will increase on40 meters, especially during the winter months.At the peak of the solar cycle, the daylightskip distance on 40 meters will be quite long,and stations within a distance of 500 miles orso of each other will not be able to hold communication.DX operation on the 40-meter bandis considerably hampered by broadcasting stations,propaganda stations, and jamming trans-IS Meters(21,000 Kc.-21,450 Kc.)This is a relativelynew b and for radioamateurs since it hasonly been available for amateur operationsince 1952. Not too much is known about thecharacteristics of this band, since it has notbeen occupied for a full cycle of solar activity.However, it is reasonable to assume thatit will have characteristics similar to both the20 and 10-meter amateur bands. It should havea longer skip distance than 20 meters for agiven time, and sporadic-E (short-skip) shouldbe apparent during the winter months. Duringa period of low sunspot activity, the AIUF willrarely rise as high as 15 meters, so this bandwill be "dead" for a large part of the year.During the next few years, 15-meter activityshould pick up rapidly, and the band shouldsupport extremely long DX contacts. Activityon the 15-meter band is limited in some areas,
1 4 Introduction to Radio<strong>THE</strong> R AD I 0since the older model TV receivers have a21 Me. i-f channel, which falls directly in the15-meter band. The interference problemsbrought about by such an unwise choice ofintermediate frequency often restrict operationon this band by amateur stations unfortunateenough to be situated near such an obsoletereceiver.10- Meters During the peak of the(28,000 Kc .. 29,700 Kc.) sunspot cycle, the 10-meter band is withoutdoubt the most popularamateur band. The combination of long skipand low ionospheric absorption make reliableDX contacts with low powered equipment possible.The great width of the band (1700 kc.)provides room for a large number of amateurs.The long skip (1500 miles or so) prevents nearbyamateurs from hearing each other, thusdropping the interference level. During the win·ter months, sporadic-E (short skip) signalsup to 1200 miles or so will be heard. The 10-meter band is poorest in the summer months,even during a sunspot maximum. Extremelylong daylight skip is common on this band, andand in years of high MUF the 10-meter bandwill support intercontinental DX contacts duringdaylight hours.The second harmonic of stations operatingin the 10-meter band falls directly into televisionchannel 2, and the higher harmonics of10-meter transmitters fall into the higher TVchannels. This harmonic problem seriouslycurtailed amateur 10-meter operation duringthe late 40's. However, with the new circuittechniques and TVI precautionary measuresstressed in this Handbook, 10-meter operationshould cause little or no interference to nearbytelevision receivers of modern design.Six Meters(SO Mc .• S4 Me.)At the peak of the sunspotcycle, the MUF occasionallyrises high enough to permitDX contacts up to 10,000 miles or so on6 meters. Activity on this band during such aperiod is often quite high. Interest in this bandwanes during a period of lesser solar activity,as contacts, as a rule, are restricted to shortskipwork. The proximity of the 6-meter bandto television channel 2 often causes interferenceproblems to amateurs located in areaswhere channel 2 is active. As the sunspot cycleincreases, activity on the 6-meter band willincrease.The Y·H-F Bands(Two Meters and "Up")The v-h-f bands arethe least affected bythe vagaries of thesunspot cycle and the Heaviside layer. Theirpredominant use is for reliable communicationover distances of 150 miles or less. Thesebands are sparsely occupied in the rural sectionsof the United States, but are quite heavilycongested in the urban areas of high population.In recent years it has been found that v-h-fsignals are propagated by other means than byline-of-sight transmission. "Scatter signals,"Aurora reflection, and air-mass boundary bendingare responsible for v-h-f communication upto 1200 miles or so. Weather conditions willoften affect long distance communication onthe 2-meter band, and all the v-h-f bands areparticularly sensitive to this condition.The other v-h-f bands have had insufficientoccupancy to provide a clear picture of theircharacteristics. In general, they behave muchas does the 2-meter band, with the weathereffects becoming more pronounced on the higherfrequency bands.1-4 Starting Your StudyWhen you start to prepare yourself for theamateur examination you will find that the circuitdiagrams, tube characteristic curves, andformulas appear confusing and difficult of understanding.But after a few study sessionsone becomes sufficiently familiar with thenotation of the diagrams and the basic conceptsof theory and operation so that the acquisitionof further knowledge becomes easierand even fascinating.As it takes a considerable time to becomeproficient in sending and receiving code, it isa good idea to intersperse technical study sessionswith periods of code practice. Manyshort code practice sessions benefit one morethan a small number of longer sessions. Alternatingbetween one study and the other keepsthe student from getting "stale" since eachtype of study serves as a sort of respite fromthe other.When you have practiced the code longenough you will be able to follow the gist ofthe slower sending stations. Many stationssend very slowly when working other stationsat great distances. Stations repeat their callsmany times when calling other stations beforecontact is established, and one need not haveachieved much code proficiency to make outtheir calls and thus determine their location.The Code The applicant for any class of amateuroperator license must be ableto send and r e c e i v e the Continental Code(sometimes called the International MorseCode). The speed required for the sending andreceiving test may be either 5, 13, or 20 wordsper minute, depending upon the class of license,assuming an average of five charactersto the word in each case. The sending and re-
<strong>HANDBOOK</strong> Learning the Code 15A N·--·-·c pD Q-··FG ··-· --·T·---J-·-K--XM-· ·----8 0 2-··· --- ·--· ··--- .......--·-347····-s-••• 6 ......--···E • R·-·5 •••••H •••• u ··- ---··8I •• v·--9···- ----·w-··-I2JL ..... y-·--0ITWAITDOUBLEERRORFRACTIONENDENDINTERNAT.z--··PERIOD(,)·-·-·-COMMA(,)--··--INTERROGATION (7l··--··QUOTATION MARK (")COLON (: J ·-··-· ---···SEMICOLON (;)-·-·-·PAREN<strong>THE</strong>SIS ( >-·--·------MEANS ZERO, AND IS WRITTEN IN THISWAY TO DISTINGUISH IT FROM <strong>THE</strong> LETTER '0".OFTEN IS TRANSMITTED INSTEAD AS ONELONG DASH (EQUIVALENT TO S DOTS)SIGN (AS)DASH (BREAK) ·-···(ERASE SIGN) -···- ••••••••BAR(/)OF MESSAGE (AR)·-·-· -··-·OF TRANSMISSION (SK)DISTRESS SIG. (SOS)···---··· ···-·-Figure 1The Contin&ntol (or International Morse) Cocle is usee/ for substontiollr. all non-automatic rocliocommunication. DO NOT memorize from the printecl page; cocle is a onguoge of SOUND, one/must not be leornecl visually; learn by listening as exploinecl In the text.cetvtng tests run for five minutes, and oneminute of errorless transmission or receptionmust be accomplished within the five-minuteinterval.If the code test is failed, the applicant mustwait at least one month before he may againappear for another test. Approximately 30% ofamateur applicants fail to pass the test. Itshould be expected that nervousness and ex·citement will at least to some degree tempo·racily lower the applicant's code ability. Thebest prevention against this is to master thecode at a little greater than the required speedunder ordinary conditions. Then if you slowdown a little due to nervousness during a testthe result will not prove fatal.Memorizingthe CodeThere is no shortcut to code pro·ficiency. To memorize the al·phabet entails but a few eve·nings of diligent application, but considerabletime is required to build up speed. The exacttime required depends upon the individual'sability and the regularity of practice.While the speed of learning will naturallyvary greatly with different individuals, about70 hours of practice (no practice period to beover 30 minutes) will usually suffice to bringa speed of about 13 w.p.m.; 16 w.p.m. requiresabout 120 hours; 20 w.p.m., 175 hours.Since code reading requires that individualletters be recognized instantly, any memoriz·ing scheme which depends upon orderly se·quence, such as learning all "dab" lettersand all "dit" letters in separate groups, is tobe discouraged. Before beginning with a codepractice set it is necessary to memorize thewhole alphabet perfectly. A good plan is tostudy only two or three letters a day and todrill with those letters until they become partof your consciousness. Mentally translate eachday's letters into their sound equivalentwherever they are seen, on signs, in papers,indoors and outdoors. Tackle two additionalletters in the code chart each day, at the sametime reviewing the characters already learned.Avoid memorizing by routine. Be able tosound out any letter immediately without somuch as hesitating to think about the letterspreceding or following the one in question.Know C, for example, apart from the sequenceABC. Skip about among all the characterslearned, and before very long sufficient letterswill have been acquired to enable you to spellout simple words to yourself in "dit dabs."This is interesting exercise, and for tl:mt rea·son it is good to memorize all the vowels firstand the most conunon consonants next.Actual code practice should start only whenthe entire alphabet, the numerals, period, com·
16 Introduction to Radio·--·- ·-·-a----3inra..-··che---·0··--iif1--·--Figure 2These code characters are used in languagesother than English. They may occasionallybe encountered so it is well to know them.rna, and question mark have been memorizedso thoroughly that any one can be soundedwithout the slightest hesitation. Do not botherwith other punctuation or miscellaneous signalsuntil later.SoundNot SightEach letter and figure must bememorized by its so u n d ratherthan its appearance. Code is asystem of sound communication, the same asis the spoken word. The letter A, for example,is one short and one long sound in combinationsounding like dit dab, and it must be rememberedas such, and not as "dot dash."Practice Time, patience, and regularity arerequired to learn the code properly.Do not expect to accomplish it within a fewdays.Don't practice too long at one stretch; itdoes more harm than good. Thirty minutes ata time should be the limit.Lack of regularity in practice is the mostcommon cause of lack of progress. Irregularpractice is very little better than no practiceat all. Write down what you have heard; thenforget it; do not look back. If your mind dwellseven for an instant on a signal about whichyou have doubt, you will miss the next fewcharacters while your attention is diverted.While various automatic code machines,phonograph records, etc., will give you practice,by far the best practice is to obtain astudy companion who is also interested inlearning the code. '-"ben you have both memorizedthe alphabet you can start sending toeach other. Practice with a key and oscillatoror key and buzzer generally proves superiorto all automatic equipment. Two such setsoperated between two rooms are fine-or betweenyour house and his will be just thatmuch better. Avoid talking to your partnerwhile practicing. If you must ask him a ques-<strong>THE</strong> R AD I 0tion, do it in code. It makes more interestingpractice than confining yourself to randompractice material.When two co-learners have memorized thecode and are ready to start sending to eachother for practice, it is a good idea to enlistthe aid of an experienced operator for the firstpractice session or two so that they will getan idea of how properly formed characterssound.During the first practice period the speedshould be such that substantially solid copycan be made without strain. Never mind if thisis only two or three words per minute. In thenext period the speed should be increasedslightly to a point where n e a r 1 y all of thecharacters can be caught only through consciouseffort. When the student becomes proficientat this new speed, another slight increasemay be made, progressing in this manneruntil a speed of about 16 words per minuteis attained if the object is to pass the amateur13-word per minute code test. The margin of3 w.p.m. is recommended to overcome a possibleexcitement factor at examination time.Then when you take the test you don't have toworry about the "jitters" or an "off day."Speed should not be inc r e a s e d to a newlevel until the student finally makes solidcopy with ease for at 1 east a five-minuteperiod at the old level. How frequently increasesof speed can be made depends uponindividual ability and the amount of practice.Each increase is apt to prove disconcerting,but remember "you are never learning whenyou are comfortable."A number of amateurs are sending codepractice on the air on schedule once or twiceeach week; excellent practice can be obtainedafter you have bought or constructed your receiverby taking advantage of these sessions.If you live in a medium-size or large city,the chances are that there is an amateur radioclub in your vicinity which offers free codepractice lessons periodically.Skill When you listen to someone speakingyou do not consciously think how hiswords are spelled. This is also true when youread. In code you must train your ears to readcode just as your eyes were trained in schoolto read printed matter. With enough practiceyou acquire skill, and from skill, speed. Inother words, it becomes a habit, somethingwhich can be done without conscious effort.Conscious effort is fatal to speed; we can'tthink rapidly enough; a speed of 25 words aminute, which is a common one in commercialoperations, means 125 characters per minuteor more than two per second, which leavesno time for conscious thinking.
<strong>HANDBOOK</strong>Learning the Code 17P.erfect Formationof CharactersWhen transmitting on thecode practice set to yourpartner, concentrate on thequality of your sending, not on your speed.Your partner will appreciate it and he couldnot copy you if you speeded up anyhow.If you want to get a reputation as having anexcellent "fist" on the air, just remember thatspeed alone won't do the trick. Proper executionof your letters and spacing will makemuch more of an impression. Fortunately, asyou get so that you can send evenly and accurately,your sending speed will automaticallyincrease. Remember to try to see how evenlyyou can send, and how fast you can receive.Concentrate on making signals properly withyour key. Perfect formation of characters isparamount to everything else. Make every sig·nal right no matter if you have to practice ithundreds or thousands of times. Never allowyourself to vary the slightest from perfect formationonce you have learned it.If possible, get a good operator to listen toyour sending for a short time, asking him tocriticize even the slightest imperfections.Timing It is of the utmost importance tomaintain uniform spacing in charactersand combinations of characters. Lack ofuniformity at this point probably causes beginnersmore trouble than any other single factor.Every dot, every dash, and every spacemust be correctly timed. In other words, accuratetiming is absolutely essential to intelligibility,and timing of the spaces betweenthe dots and dashes is just as important asthe lengths of the dots and dashes themselves.The characters are timed with the dot as a"yardstick." A standard dash is three timesas long as a dot. The spacing between partsof the same letter is e qua 1 to one dot; thespace between letters is equal to three dots,and that between words equal to five dots.The rule for spacing between letters andwords is not strictly observed when sendingslower than about 10 words per minute for thebenefit of someone learning the code and desiringreceiving practice. When sending at,say, 5 w.p.m., the individual letters should bemade the same as if the sending rate wereabout 10 w.p.m., except that the spacing betweenletters and words is greatly exaggerated.The reason for this is obvious. The letter L,for instance, will then sound exactly the sameat 10 w.p.m. as at 5 w.p.m., and when thespeed is increased above 5 w.p.m. the studentwill not have to become familiar with whatmay seem to him like a new sound, althoughit is in reality only a faster combination ofdots and dashes. At the greater speed he willmerely have to learn the identification of thesame sound without taking as long to do so...~ ~A~.~ ·~ ~ ~BT! i i :: :--~~0Figure 3b:::xX:i±.b::il, r::: :·--·~-·~ c~~ ~N EDiagram illustrating relative lengths ofclashes one/ spaces referred to the clurationof a clot. A clash is exactly equal in durationto three clots; spaces between ports of aletter equal one clot; those between letters,three clots; space between words, five dots.Note that a slight increase between two partsof a letter will make it so unci I ike twoletters.Be particularly careful of letters like B.Many beginners seem to have a tendency toleave a longer space after the dash than thatwhich they place between succeeding dots,thus making it sound like TS. Similarly, makesure that you do not leave a longer space afterthe first dot in the letter C than you do betweenother parts of the same letter; otherwiseit will sound like NN.Sending vs.ReceivingOnce you have memorized thecode thoroughly you should concentrateon increasing your receivingspeed. True, if you have to practicewith another newcomer who is learning thecode with you, you will both have to do somesending. But don't attempt to practice sendingjust for the sake of increasing your sendingspeed.When transmitting on the code practice setto your partner so that he can get receivingpractice, concentrate on the quality of yoursending, not on your speed.Because it is comparatively easy to learnto send rapidly, especially when no particularcare is given to the quality of sending, manyoperators who have just received their licensesget on the air and send mediocre or worse codeat 20 w.p.m. when they can barely receivegood code at 13. Most oldtimers remember theirown period of initiation and are only too gladto be patient and considerate if you tell themthat you are a newcomer. But the surest wayto incur their scorn is to try to impress themwith your "lightning speed," and then to requestthem to send more slowly when theycome back at you at the same speed.Stress your copying ability; never stressyour sending ability. ~t should be obvious thatthat if you try to send faster than you can re·cei ve, your ear will not recognize any mistakeswhich your hand may make.
18 Introduction to Radio<strong>THE</strong> R AD I 0fingers to become tense. Send with a full, freearm movement. Avoid like the plague any fin·ger motion other than the slight cushioningeffect mentioned above.Stick to the regular hand key for learningcode. No other key is satisfactory for this pur·pose. Not until you have thoroughly masteredboth sending and receiving at the maximumspeed in which you are interested should youtackle any form of automatic or semi-automatickey such as the Vibroplex ("bug") or an elec·tronic key.Figure 4PROPER POSITION OF <strong>THE</strong> FINGERS FOROPERATING A TELEGRAPH KEYThe fingers hold the knob and act as a cushion.The hand rests lightly on the key. Themuscles of the forearm provide the power,the wrist acting as the fulcrum. The powershould not come from the fingers, but ratherfrom the forearm muscles.Using the Key Figure 4 shows the proper posi·tion of the hand, fingers andwrist when manipulating a telegraph or radiokey. The forearm should rest naturally on thedesk. It is preferable that the key be placedfar enough back from the edge of the table(about 18 inches) that the elbow can rest onthe table. Otherwise, pressure of the tableedge on the arm will tend to hinder the circu·lation of the blood and weaken the ulnar nerveat a point where it is close to the surface,which in turn will tend to increase fatigueconsiderably.The knob of the key is grasped lightly withthe thumb along the edge; the index and thirdfingers rest on the top towards the front or faredge. The hand moves with a free up and downmotion, the wrist acting as a fulcrum. Thepower must come entirely from the arm mus·des. The third and index fingers will bendslightly during the sending but not because ofdeliberate effort to manipulate the finger mus·des. Keep your finger muscles just tightenough to act as a cushion for the arm motionand let the slight movement of the fingers takecare of itself. The key's spring is adjusted tothe individual wrist and should be neither toostiff nor too loose. Use a moderately stiff ten·sion at first and gradually lighten it as youbecome more proficient. The separation be·tween the contacts must be the proper amountfor the desired speed, being somewhat under1/16 inch for slow speeds and slightly closertogether (about 1/32 inch) for faster speeds.Avoid extremes in either direction.Do not allow the muscles of arm, wrist, orDifficulties Should you experience difficultyin increasing your code speedafter you have once memorized the characters,there is no reason to become discouraged. Itis more difficult for some people to learn codethan for others, but there is no justificationfor the contention sometimes made that "somepeople just can't learn the code." It is not amatter of intelligence; so don't feel ashamedif you seem to experience a little more thanthe usual difficulty in learning code. Your re·action time may be a little slower or your co·ordination not so good. If this is the case,remember you can still learn the code. Youmay never learn to send and receive at 40w.p.m., but you can learn sufficient speed forall non-commercial purposes and even for mostcommercial purposes if you have patience,and refuse to be discouraged by the fact thatothers seem to pick it up more rapidly.When the sending operator is sending justa bit too fast for you (the best speed for prac·tice), you will occasionally miss a signal or asmall group of them. When you do, leave ablank space; do not spend time futilely tryingto recall it; dismiss it, and center attentionon the next letter; otherwise you'll miss more.Do not ask the sender any questions until thetransmission is finished.To prevent guessing and get equal practiceon the less common letters, depart occasional·ly from plain language material and use a jum·ble of letters in which the usually less com·monly used letters predominate.As mentioned before, many students put agreater space after the dash in the letter Bthan between other parts of the same letter soit sounds like TS. C, F, Q, V, X, Y and Zoften give similar trouble. Make a list of wordsor arbitrary combinations in which these let·ters predominate and practice them, both send·ing and receiving until they no longer give youtrouble. Stop everything e l s e and stick atthem. So long as they give you trouble you arenot ready for anything else.Follow the same procedure with letterswhich you may tend to confuse such as F andL, which are often confused by beginners.
<strong>HANDBOOK</strong>Learning the Code 19Figure 5<strong>THE</strong> SIMPLEST CODE PRACTICESET CONSISTS OF A KEY AND ABUZZERThe buzzer is adjusted to give asteady, high-pitched whine. If desired,the phones may be omitted,In which case the buzzer should bemounted firmly on a sounding board.Crystal, magnetic, or dynamic ear·phones may be used. Additionalsets of phones should be connectedin parallel, not in series.BUZZER,...-- ---,..---------;;---, : ~~~pp~~~~~~g~~TER_ 1.0 TO 4.0 VOLTS , I -----J;ow••~~j;'IT ~ L ____ j 0 ~.:~R 4L____.,.?i ___._- - -- - - -- or BATTERY I : o PHONES.KEY<strong>THE</strong>SE PARTS REQUIREDONLY IF HEADPHONEOPERATION IS DESIREDKeep at it u n t i I you always get them rightwithout having to stop even an instant to thinkabout it.If you do not instantly recognize the soundof any character, you have not learned it; goback and practice your alphabet further. Youshould never have to omit writing down everysignal you hear except when .the transmissionis too fast for you.Write down what you hear, not what youthink it should be. It is surprising how oftenthe word which you guess will be wrong.Copying Behind All good operators copy severalwords behind, that is,while one word is being received, they arewriting down or typing, say, the fourth or fifthprevious word. At first this is very difficult,but after sufficient practice it will be foundactually to be easier than copying close up.It also results in more accurate copy and enablesthe receiving operator to capitalize and10 K0.5W1, 51/.CK-7221 =COLLECTOR2= BASE3= EMITTERFigure 6SIMPLE TRANSISTOR CODEPRACTICE OSCILLATOR2000 n.PHONESAn inexpensive Raytheon CK-722 transistorrequires only a single 1Vz-volt flashlightbattery for power. The inductance of the earphonewindings forms part of the oscillatorycircuit. The pitch of the note moy be changedby varying the value of the two capacitorsconnected across the earphones.punctuate copy as he goes along. It is not recommendedthat the beginner attempt to do thisuntil he can send and receive accurately andwith ease at a speed of at least 12 words aminute.It requires a considerable amount of trainingto dissociate the action of the subconsciousmind from the direction of the consciousmind. It may help some in obtaining this trainingto write down two columns of short words.Spell the first word in the first column out loudwhile writing down the first word in the secondcolumn. At first this will be a bit awkward,but you will rapidly gain facility with practice.Do the same with all the words, and then reversecolumns.Next try speaking aloud the words in the onecolumn while writing those in the other column;then reverse columns.After the foregoing can be done easily, trysending with your key the words in one columnwhile spelling those in the other. It won'tbe easy at first, but it is well worth keepingafter if you intend to develop any real codeproficiency. Do not attempt to catch up. Thereis a natural tendency to close up the gap, andyou must train yourself to overcome this.Next have your code companion send you aword either from a list or from straight text;do not write it down yet. Now have him sendthe next word; after receiving this secondword, write down the first word. After receivingthe third word, write the second word; andso on. Never mind how slowly you must go,even if it is only two or three words per minute.Stay behind.It will probably take quite a number of practicesessions before you can do this with anyfacility. After it is relatively easy, then trystaying two words behind; keep this up untilit is easy. Then try three words, four words,and five words. The more you practice keepingreceived material in mind, the easier itwill be to stay behind. It will be found easierat first to copy material with which one isfairly familiar, then gradually switch to lessfamiliar material.
20 Introduction to RadioAutomatic CodeMachinesThe two practice sets whichare deso;ribed in this chapterare of most value when youhave someone with whom to practice. Automa·tic code machines are not recommended to any·one who can possibly obtain a companion withwhom to practice, someone who is also interestedin learning the code. If you are unableto enlist a code partner and have to practiceby yourself, the best way to get receivingpractice is by the use of a tape machine (auto·matic code sending machine) with severalpractice tapes. Or you can use a set of phonographcode practice records. The records areof use only if you have a phonograph whoseturntable speed is readily adjustable. The tapemachine can be rented by the month for a rea·sonable fee.Once you can copy about 10 w.p.m. you canalso get receiving practice by listening to slowsending stations on your receiver. Many amateurstations send slowly particular! y whenworking far distant stations. When receivingconditions are particularly poor many commer·cial stations also send slowly, sometimes re·peating every word. Until you can copy around10 w.p.m. your receiver isn't much use, andeither another operator or a machine or recordsare necessary for getting receiving practiceafter you have once memorized the code.Code PracticeSetsIf you don't feel too foolishdoing it, you can secure ameasure of code practice withthe help of a partner by sending "dit-dah"messages to each other while riding to work,eating lunch, etc. It is better, however, to usea buzzer or code practice oscillator in con·junction with a regular telegraph key.As a good key may be considered an invest·ment it is wise to make a well-made key yourfirst purchase. Regardless of what type codepractice set you use, you will need a key, andlater on you will need one to key your transmitter.If you get a good key to begin with,you won't have to buy another one later.The key should be rugged and have fairlyheavy contacts. Not only will the key standup better, but such a key will contribute tothe "heavy" type of sending so desirable forradio work. Morse (telegraph) operators usea "light" style of sencling and can send some·what faster when using this light touch. But,in radio work static and interference are oftenpresent, and a slightly heavier dot is desirable.If you use a husky key, you will lindyourself automatically sending in this manner.To generate a tone simulating a code signalas heard on a receiver, either a mechanicalbuzzer or an audio oscillator may be used. Figure5 shows a simple code-practice set usinga buzzer which may be used directly simplyby mounting the buzzer on a sounding board,or the buzzer may be used to feed from one tofour pairs of conventional high-impedancephones.An example of the audio-oscillator type ofcode-practice set is illustrated in figures 6and 7. An inexpensive Raytheon CK-722 trans·istor is used in place of the more expensive,power consuming vacuum tube. A single "penlite"1?5-volt cell powers the unit. The coilsof the earphones form the ind uc ti ve portionof the resonant circuit. 'Phones having animpedance of 2000 ohms or higher should beused. Surplus type R-14 earphones also workwell with this circuit.Figure 7The circuit of Figure 6 is used in thisminiature transistorized code Practiceoscillator. Components are mounted in asmall plastic case. The transistor isattached to a three terminal phenolicmounting strip. Sub-miniature jacks areused for the key and phones connections.A hearing aid earphone may also be used,as shown. The phone is stored in theplastic case when not in use.
CHAPTER TWODirect Current CircuitsAll naturally occurring matter (excludingartifically produced radioactive substances) ismade up of 92 fundamental constituents calledelements. These elements can exist either inthe free state such as iron, oxygen, carbon,copper, tungsten, and aluminum, or in chemicalunions commonly called compounds. Thesmallest unit which still retains all the originalcharacteristics of an element is the atom.Combinations of atoms, or subdivisions ofcompounds, result in another fundamentalunit, the molecule. The molecule is the smallestunit of any compound. All reactive elementswhen in the gaseous state also existin the molecular form, made up of two or moreatoms. The nonreactive gaseous elementshelium, neon, argon, krypton, xenon, andradon are the only gaseous elements that everexist in a stable monatomic state at ordinarytemperatures.2-1 The AtomAn atom is an extremely small unit ofmatter- there are literally billions of themmaking up so small a piece of material as aspeck of dust. To understand the basic theoryof electricity and hence of radio, we must gofurther and divide the atom into its maincomponents, a positively charged nucleus anda cloud of negatively charged particles that;surround the nucleus. These particles, swirlingaround the nucleus in elliptical orbits at anincredible rate of speed, are called orbitalelectrons.It is upon the behavior of these electronswhen freed from the atom, that depends thestudy of electricity and radio, as well asallied sciences. Actually it is possible to subdividethe nucleus of the atom into a dozen or21so different particles, but this further subdivisioncan be left to quantum mechanics andatomic physics. As far as the study of elec·tronics is concerned it is only necessary forthe reader to think of the normal atom as beingcomposed of a nucleus having a net positivecharge that is exactly neutralized by the oneor more orbital elecw>ns surrounding it.The atoms of different elements differ inrespect to the charge on the positive nucleusand in the number of electrons revolvingaround this charge. They range all the wayfrom hydrogen, having a net charge of oneon the nucleus and one orbital electron, touranium with a net charge of 92 on the nucleusand 92 orbital electrons. The number of orbitalelectrons is called the atomic number of theelement.Action of theElectronsFrom the above it must not bethought that the electrons revolvein a haphazard manneraround the nucleus. Rather, the electrons inan element having a large atomic number aregrouped into rings having a definite number ofelectrons. The only atoms in which these ringsare completely filled are those of the inertgases mentioned before; all other elementshave one or more uncompleted rings of electrons.If the uncompleted ring is nearly empty,the element is metallic in character, beingmost metallic when there is only one electronin the outer ring. If the incomplete ring lacksonly one or two electrons, the element isusually non-metallic. Elements with a ringabout half completed will exhibit both nonmetallicand metallic characteristics; carbon,silicon, germanium, and arsenic are examples.Such elements are called semi-conductors.In metallic elements these outer ring elec·trans are rather loosely held. Consequently,
22 Direct Current Circuits<strong>THE</strong> R AD I 0there is a conunuous helter-skelter movementof these electrons and a continual shiftingfrom one atom to another. The electrons whichmove about in a substance are called freeelectrons, and it is the ability of these electronsto drift from atom to atom which makespossible the electric current.If the free electrons are numerous and loosely held, theelement is a good conductor.Conductors andlnsulotorsOn the other hand, if there are few free electrons,as is the case when the electrons in anouter ring are tightly held, the element is apoor conductor. If there are virtually no freeelectrons, the element is a good insulator.2-2 Fundamental ElectricalUnits and RelationshipsElectromotive Force: The free electrons inPotential Difference a conductor move con·stantly about and changetheir position in a haphazard manner. Toproduce a drift of electrons or electric currentalong a wire it is necessary that there be adifference in "pressure" or potential betweenthe two ends of the wire. This potential differencecan be produced by connecting asource of electrical potential to the ends ofthe wire.As will be explained later, there is an excessof electrons at the negative terminal ofa battery and a deficiency of electrons at thepositive terminal, due to chemical action.When the battery is connected to the wire, thedeficient atoms at the positive terminal attractfree electrons from the wire in order for thepositive terminal to become neutral. Theattracting of electrons continues through thewire, and finally the excess electrons atthe negative terminal of the battery are attractedby the positively charged atoms at theend of the wire. Other sources of electricalpotential (in addition to a battery) are: anelectrical generator (dynamo), a thermocouple,an electrostatic generator (static machine), aphotoelectric cell, and a crystal or piezoelectricgenerator.Thus it is seen that a potential differenceis the result of a difference in the number ofelectrons between the two (or more) points inquestion. The force or pressure due to apotential difference is termed the electromotiveforce, usually abbreviated e.m.f. orE. M. F. It is expressed in units called volts.It should be noted that for there to be apotential difference between two bodies orpoints it is not necessary that one have apositive charge and the other a negativecharge. If two bodies each have a negativecharge, but one more negative than the other,the one with the lesser negative charge willact as though it were positively charged withrespect to the other body. It is the algebraicpotential difference that determines the forcewith which electrons are attracted or repulsed,the potential of the earth being taken as thezero reference point.The ElectricCurrentThe flow of electrons along aconductor due to the applicationof an electromotive force constitutesan electric current. This drift is inaddition to the irregular movements of theelectrons. However, it must not be thoughtthat each free electron travels from one endof the circuit to the other. On the contrary,each free electron travels only a short distancebefore colliding with an atom; this collisiongenerally knocking off one or more electronsfrom the atom, which in turn move a shortdistance and collide with other atoms, knockingoff other electrons. Thus, in the generaldrift of electrons along a wire carrying anelectric current, each electron travels only ashort distance and the excess of electrons atone end and the deficiency at the other arebalanced by the source of the e.m.f. When thissource is removed the state of normalcy returns;there is still the rapid interchange offree electrons between atoms, but there is nogeneral trend or "net movement" in eitherone direction or the other.Ampere andCoulombThere are two units of measurementassociated with current,and they are often confused.The rate of flow of electriciry is stated inamperes. The unit of quantity is the coulomb.A coulomb is equal to 6.28 x 10 18 electrons,and when this quantity of electrons flows bya given point in every second, a current ofone ampere is said to be flowing. An ampereis equal to one coulomb per second; a coulombis, conversely, equal to one ampere-second.Thus we see that coulomb indicates amount,and ampere indicates rate of flow of electriccurrent.Older textbooks speak of current flow asbeing from the positive terminal of the e.m.f.source through the conductor to the negativeterminal. Nevertheless, it has long been anestablished fact that the current flow in ametallic conductor is the electronic flow fromthe negative terminal of the source of voltagethrough the conductor to the positive terminal.The only exceptions to the electronic directionof flow occur in gaseous and electrolytic conductorswhere the flow of positive ions towardthe cathode or negative electrode constitutesa positive flow in the opposite direction to theelectronic flow. (An ion is an atom, molecule,
<strong>HANDBOOK</strong>Resistance 23or particle which either lacks one or moreelectrons, or else has an excess of one ormore electrons.)In radio work the terms "electron flow" and"current" are becoming accepted as beingsynonymous, but the older terminology is stillaccepted in the electrical (industrial) field.Because of the confusion this sometimescauses, it is often safer to refer to the directionof electron flow rather than to the directionof the "current." Since electron flowconsists actually of a passage of negativecharges, current flow and algebraic electronflow do pass in the same direction.Resistance The flow of current in a materialdepends upon the ease withwhich electrons can be detached from theatoms of the material and upon its molecularstructure. In other words, the easier it is todetach electrons from the atoms the more freeelectrons there will be to contribute to theflow of current, and the fewer collisions thatoccur between free electrons and atoms thegreater will be the total electron flow.The opposition to a s tea d y elecrron flowis called the resistance of a material, and isone of its physical properties.The u n i t of resistance is the ohm. Everysubstance has a specific resistance, usuallyexpressed as ohms per mil-foot, which is deter·mined by the material's molecular structureand temperature. A mil-foot is a piece ofmaterial one circular mil in area and one footlong. Another measure of resistivity frequentlyused is expressed in the units microhms percentimeter cube. The resistance of a uniformlength of a given substance is directly pro·portional to its length and specific resistance,and inversely proportional to its cross-section·al area. A wire with a certain resistance for agiven length will have twice as much resist·ance if the length of the wire is doubled. Fora given length, doubling the cross-sectionalarea of the wire will halve t h e resistance,while doubling the diameter will reduce theresistance to one fourth. This is true sincethe cross-sectional area of a wire varies asthe square of the diameter. The relationshipbetween the resistance and the linear dimen·sions of a conductor may be expressed by thefollowing equation:r IR=AWhereR = resistance in ohmsr = resistivity in Ohms per mil- footl = length of conductor in feetA = cross-sectional area in circular milsTABLE OF RESISTIVITYKes st vlty 1nOhms per Temp. Coeff. ofCircular resistance per ccMaterial Mil-Foot at 20• C.Aluminum 17 0.0049Brass 45 0.003 to 0.007Cadmium 46 0.0038Chromium 16 0.00Copper 10.4 0.0039Iron 59 0.006Silver 9.8 0.004Zinc 36 0.0035Nichrome 650 0.0002Constantan 295 0.00001Manganin 290 0.00001Monel 255 0.0019FIGURE 1The resistance also depends upon temperature,increasing with increases in temperaturefor most substances (including most metals),due to increased electron acceleration andhence a greater number of impacts betweenelectrons and atoms. However, in the case ofsome substances such as carbon and glass thetemperature coefficient is negative and theresistance decreases as the temperature increases.This is also true of electrolytes. Thetemperature may be raised by the external applicationof heat, or by the flow of the currentitself. In the latter case, the temperarure israised by the heat generated when the electronsand atoms collide.Conductors ondInsulatorsIn the molecular structure ofmany materials such as glass,porcelain, and mica all electronsare tightly held within their orbits andthere are comparatively few free electrons.This type of substance will conduct an electriccurrent only with great difficulty and isknown as an insulator. An insulator is said tohave a high electrical resistance.On the other hand, materials that have alarge number of free electrons are known asconductors. Most metals, those elements whichhave only one or two electrons in their outerring, are good conductors. Silver, copper, andaluminum, in that order, are the best of thecommon metals u sed as conductors and aresaid to have the greatest conductivity, or low·est r e s i stance to the flow of an electriccurrent.FundamentalElectrical UnitsThese units are the volt,the ampere, and the ohm.They were mentioned in thepreceding paragraphs, but were not completelydefined in terms of fixed, known quantities.The fundamental unit of current, or rate offlow of electricity is the ampere. A current ofone ampere will deposit silver from a specifiedsolution of silver nitrate at a rate of1.118 milligrams per second.
24 Direct Current Circuits <strong>THE</strong> R AD I 0-figure 2TYPICAL RESISTORSShown above are various types of resistors used In electronic circuits. The larger units arepower resistors. On the left is a variable power resistor. Three preclsion·type resistors areshown in the center with two small composition resistors beneath them. At the right Is acomposition-type potentiometer, used for audio clrculfty.The international standard for the ohm isthe resistance offered by a uniform column ofmercury at 0° C., 14.4521 grams in mass, ofconstant cross-sectional area and 106.300centimeters in length. The expression megohm(1,000,000 ohms) is also sometimes usedwhen speaking of very large values of resistance.A volt is the e.m.f. that will produce a currentof one ampere through a resistance ofone ohm. The standard of electromotive forceis the Weston cell which at 20° C. has apotential of 1.0183 volts across its terminals.This cell is used only for reference purposesin a bridge circuit, since only an infinitesimalamount of current may be drawn from it withoutdisturbing its characteristics.Ohm's Law The relationship between theelectromotive force (voltage),the flow of current (amperes), and the resistancewhich impedes the flow of current (ohms),is very clearly expressed in a simple buthighly valuable law known as Ohm's law.This law states that the current in amperes isequal to the voltage in volts divided by theresistance in ohms. Expressed as an equation:EI=-RRESISTANCE'®@Figure 3SIMPLE SERIES CIRCUITSAt (A) the battery is in series with a singleresistor. At (8) the battery is in series withtwo resistors, the resistors themselves beingIn series. The arrows indicate the direction ofelectron flow.If the voltage (E) and resistance (R) areknown, the current (I) can be readily found.If the voltage and current are known, and theresistance is unknown, the resistance (R) isEequal to - . When the voltage is the un-Iknown quantity, it can be found by multiplyingI x R. These three equations are all securedfrom the original by simple transposition.The expressions are here repeated for quickreference:EI=RER=IE = IR
<strong>HANDBOOK</strong>Resistive Circuits25AFigure 4SIMPLE PARALLELCIRCUITFigureSSERIES-PARALLELCIRCUITThe two resistors Rt and R 2 are sa/rJ to be inparallel since the flow of current is offeredtwo parallel paths. An electron leaving pointA will pass either through Rt or R2, but notthrough both, to reach the positive terminalof the battery. II a Iorge number of electronsare cons/clerecl, the greater number will passthrough whichever of the two resistors hasthe /ower res/stance.where I is the current in amperes,R is the resistance in ohms,E is the electromotive force in volts.Appl icotion ofOhm's LowAll electrical circuits fall intoone of three classes: seriescircuits, parallel circuits, andseries-parallel circuits. A series circuit isone in which the current flows in a singlecontinuous path and is of the same value atevery point in the circuit (figure 3). In a parallelcircuit there are two or more currentpaths between two points in the circuit, asshown in figure 4. Here the current divides atA, part g~ing through R 1 and part through R 2,~d comb1nes at B to_return to the battery.F1gure 5 shows a series-parallel circuit. Thereare two paths between points A and B as inthe parallel circuit, and in addition there aretwo resistances in series in each branch ofthe parallel combination. Two other examplesof series-parallel arrangements appear in figure6. The way in which the current splits toflow through the parallel branches is shown bythe arrows.In every circuit, each of the parts has someresistance: the batteries or generator, the connectingconductors, and the apparatus itself.Thus, if each part has some resistance, nomatter how litde, and a current is flowingthrough it, there will be a voltage drop acrossit. In other words, there will be a potentialdifference between the two ends of the circuitelement in question. This drop in voltage isequal to the product of the current and theresistance, hence it is called the IR drop.The source of voltage has an internal resistance,and when connected into a circuitso that current flows, there will be an IR dropin the source just as in every other part of thecircuit. Thus, if the terminal voltage of thesource could be measured in a way that wouldcause no current to flow, it would be foundto be more than the voltage measured when acurrent flows by the amount of the IR dropIn this type ol circuit the resistors are arrangedIn series groups, one/ these ser/esedgroups are then placed In paro//e/.in the source. The voltage measured with nocurrent flowing is termed the no load voltage;that measured with current flowing is the loadvoltage. It is apparent that a voltage sourcehaving a low internal resistance is most desirable.Res I stancesin SeriesThe current flowing in a seriescircuit is equal to the voltageimpressed divided by the totalresistance across which the voltage is impressed.Since the same current flows throughevery part of the circuit, it is merely necessaryto add all the individual resistances toobtain the total resistance. Expressed as aformula:Rtotal = R, + R• + R, + • • • + RN ·Of course, if the resistances happened to beall the same value, the total resistance wouldbe the resistance of one multiplied by thenumber of resistors in the circuit.Resi,stoncesin ParallelConsider two resistors, one of100 ohms and one of 10 ohms,connected in parallel as in figure4, with a voltage of 10 volts appliedacross each resistor, so the current througheach can be easily calculated.E = 10 voltsR = 100 ohmsE = 10 voltsR = 10 ohmsEI=R10I, = --= 0.1 ampere10010I 2 =- = 1.0 ampere10Total current =I, + I 2 = 1.1 ampereUntil it divides at A, the entire current of1.1 amperes is flowing through the conductorfrom the battery to A, and again from B throughthe conductor to the battery. Since this is morecurrent than flows through the smaller resistorit is evident that the resistance of the parallelcombination must be less than 10 ohms, theresistance of the smaller resistor. We can findthis value by applying Ohm's law.
26 Direct Current Circuits<strong>THE</strong> R AD I 0EE ~ 10 voltsI ~ 1.1 amperesR~I10R ~ _, 9.09 ohms1.1The resistance of the parallel combination is9.09 ohms.Mathematically, we can derive a simpleformula for finding the effective resistance oftwo resistors connected in parallel.This formula is:R1Rz+ +® ®R, x R 2 R, __R, +R 2where R is the unknown resistance,R 1is the resistance of the first resistor,R 2is the resistance of the second resistor.If the effective value required is known,and it is desired to connect one unknown resistorin parallel with one of known value,the following transposition of the above formulawill simplify the problem of obtainingthe unknown value:R, xRR ---2- R,- Rwhere R is the effective value required,R 1is the known resistor,R 2is the value of the unknown resistancenecessary to give R whenin parallel with R,.The resultant value of placing a number ofunlike resistors in parallel is equal to the reciprocalof the sum of the reciprocals of thevarious resistors. This can be expressed as:R11 1 1-+-+-+·R, R 2R,1RnThe effective value of placing any numberof unlike resistors in parallel can be determinedfrom the above formula. However, itis commonly used only when there are threeor more resistors under consideration, sincethe simplified formula given before is moreconvenient when only two resistors are beingused.From the above, it also follows that whentwo or more resistors of the same value areplaced in parallel, the effective resistance ofthe paralleled resistors is equal to the valueof one of the resistors divided by the numberof resistors in parallel.The effective value of resistance of two orFigure 6O<strong>THE</strong>R COMMON SERIES-PARALLELCIRCUITSmore resistors connected in parallel is alwaysless than the value of the lowest resistance inthe combination. It is well to bear this simplerule in mind, as it will assist greatly in approximatingthe value of paralleled resistors.Resistors InSeries ParallelTo find the total resistance ofseveral resistors connected inseries-parallel, it is usuallyeasiest to apply either the formula for seriesresistors or the parallel resistor formula first,in order to reduce the original arrangement toa simpler one. For instance, in figure 5 theseries resistors should be added in eachbranch, then there will be but two resistors inparallel to be calculated. Similarly in figure 7,although here there will be three parallel resistorsafter adding the series resistors ineach branch. In figure 6B the paralleled resistorsshould be reduced to the equivalentseries value, and then the series resistancevalues can be added.Resistances in series-parallel can be solvedby combining the series and parallel formulasinto one similar to the following (refer tofigure 7):1R ~ ------------1 1 1---+ -----1-------R, + R 2R, + R 4 R 5 + R 6 + R 7Voltage Dividers A voltage divider is exactlywhat its name implies:a resistor or a series of resistors connectedacross a source of voltage from whichvarious lesser values of voltage may be obtainedby connection to various points alongthe resistor.A voltage divider serves a most useful purposein a radio receiver, transmitter or amplifier,because it offers a simple means ofobtaining plate, screen, and bias voltages ofdifferent values from a common power supply
<strong>HANDBOOK</strong>Voltage Divider27R5RsR71Rl+ AFigure 7ANO<strong>THE</strong>R TYPE OFSERIES-PARALLEL CIRCUITsource. It may also be used to obtain very lowvoltages of the order of .01 to .001 volt witha high degree of accuracy, even though ameans of measuring such voltages is lacking.The procedure for making these measurementscan best be given in the following example.Assume that an accurately calibrated voltmeterreading from 0 to 150 volts is available,and that the source of voltage is exactly 100volts. This 100 volts is then impressed througha resistance of exactly 1,000 ohms. It will,then, be found that the voltage along variouspoints on the resistor, with respect to thegrounded end, is exactly proportional to theresistance at that point. From Ohm's law, thecurrent would be 0.1 ampere; this current remainsunchanged since the original value ofresistance (1,000 ohms) and the voltage source(100 volts) are unchanged. Thus, at a 500-ohm point on the resistor (half its entire resistance),the voltage will likewise be halvedor reduced to 50 volts.The equation (E ~I x R) gives the proof:E ~ 500 x 0.1 ~50. At the point of 250 ohmson the resistor, the voltage will be one-fourththe total value, or 25 volts (E ~ 250 x 0.1 ~ 25).Continuing with this process, a point can befound where the resistance measures exactly1 ohm and where the voltage equals 0.1 volt.It is, therefore, obvious that if the originalsource of voltage and the resistance can bemeasured, it is a simple matter to predeterminethe voltage at any point along the resistor,provided that the current remains constant,and provided that no current is taken from thetap-on point unless this current is taken intoconsideration.Voltage DividerCalculationsProper design of a voltagedivider for any type of radioequipment is a relativelysimple matter. The first consideration is theamount of "bleeder current" to be drawn.In addition, it is also necessary that the desiredvoltage and the exact current at each tapon the voltage divider be known.Figure 8 illustrates the flow of current in asimple voltage divider and load circuit. Thelight arrows indicate the flow of bleeder current,while the heavy arrows indicate the flowof the load current. The design of a combinedFigure 8SIMPLE VOLTAGE DIVIDERCIRCUITThe arrows Indicate the manner In which thecurrent flow rllvtJ .. between tlte voltage Jlv/Jer/tse/1 anJ tlte uterna/ loaJ circuit.bleeder resistor and voltage divider, such asis commonly used in radio equipment, is illustratedin the following example:A power supply delivers 300 volts and isconservatively rated to supply all needed currentfor the receiver and still allow a bleedercurrent of 10 milliamperes. The following voltagesare wanted: 75 volts at 2 milliamperesfor the detector tube, 100 volts at 5 milliamperesfor the screens of the tubes, and250 volts at 20 milliamperes for the plates ofthe tubes. The required voltage drop across R,is 75 volts, across R 2 25 volts, across R 3150volts, and across R 4it is 50 volts. Thesevalues are shown in the diagram of figure 9.The respective current values are also indicated.Apply Ohm's law:E 75R, ~- ~- ~ 7,500 ohms.I .01E 25R 2 ~- ~--~ 2,083 ohms.I .012E 150R 3 ~ -~ -- ~ 8,823 ohms.I .017E 50R 4 ~-~- ~ 1,351 ohms.I .037RTota/ ~ 7,500 + 2,083 + 8,823 +1,351 ~ 19,757 ohms.A 20,000-ohm resistor with three sliding tapswill be of the approximately correct size, andwould ordinarily be used because of the difficultyin securing four separate resistors of theexact odd values indicated, and because noadjustment would be possible to compensatefor any slight error in estimating the probablecurrents through the various taps.When the sliders on the resistor once areset to the proper point, as in the above ex-
28 Direct Current Circuits10+2+5+20 MA.50 VOL.TS DROP R4tII 250V. 20MA.I - ~10+2+ 5 MA.150 VOLTS DROP { R3 III300 VOLTSt1001/. SMA.I -+--{ t10+ 2 MA.25 VOLTS DROP R~- -~BLEEDER CURRENT, 10 MA{75 VOLTS DROPR1lIII 75 v., 2 MA.1 t1 f_j _j~----POWER SUPPLY ---t-~--LOAD ~Figure 9MORE COMPLEX VOLTAGE DIVIDERThe method for computing the values of theresistors Is cliscussecl In the accompanying text.ample, the voltages will remain constant atthe values shown as long as the current remainsa constant value.Disadvantages of One of the serious disadvanVoltage Dividers tages of the voltage dividerbecomes evident when thethe current drawn from one of the taps changes.It is obvious that the voltage drops are interdependentand, in turn, the individual dropsare in proportion to the current which flowsthrough the respective sections of the dividerresistor. The only remedy lies in providing aheavy steady bleeder current in order to makethe individual currents so small a part of thetotal current that any change in current willresult in only a slight· change in voltage. Thiscan seldom be realized in practice because ofthe excessive values of bleeder current whichwould be required.Kirchhoff's Laws Ohm's law is all that isnecessary to calculate thevalues in simple circuits, such as the precedingexamples; but in more complex problems,involving several loops or more thanone voltage in the same closed circuit, theuse of Kirchhoff's laws will greatly simplifythe calculations. These laws are merely rulesfor applying Ohm's law.Kirchhoff's first law is concerned with netcurrent to a point in a circuit and states that:At any point in a circuit the currentflowing toward the point is equal tothe current flowing away from thepoint.Stated in another way: if currents flowing tothe point ate considered positive, and thoseflowing from the point are considered nega·<strong>THE</strong> R AD I 0,--2 AMPSR1l-2 AMPSA ,\_,._R2buCMP5+~1111,20 vnLTSFigure 10ILLUSTRATING KIRCHHOFF'SFIRST LAWThe current flowing toward point "A" Is equalto the current flowing away lrom point "A...tive, the sum of all currents flowing towardand away from the point- taking signs intoaccount- is equal to zero. Such a sum isknown as an algebraic sum; such that the lawcan be stated thus: The algebraic sum of allcurrents entering and leaving a point is zero.Figure 10 illustrates this first law. Since theeffective resistance of the network of resistorsis 5 ohms, it can be seen that 4 amperes flowtoward point A, and 2 amperes flow awaythrough the two 5-ohm resistors in series. Theremaining 2 amperes flow away through the 10-ohm resistor. Thus, there ate 4 amperes flowingto point A and 4 amperes flowing away fromthe point. If R is the effective resistance ofthe network (5 ohms), R 1 = 10 ohms, R 2 = 5ohms, R, = 5 ohms, and E = 20 volts, we canset up the following equation:E E E------=0R R 1 R 2 + R,20 20 20------=05 10 5 + 54-2-2=0Kirchhoff's second law is concerned withnet voltage drop around a closed loop in acircuit and states that:In any closed path or loop in a circuitthe sum of the IR drops must equalthe sum of the applied e.m.f.'s.The second law also may be convenientlystated in terms of an algebraic sum as: Thealgebraic sum of all voltage drops around aclosed path or loop in a circuit is zero. Theapplied e.m. f.'s (voltages) are consideredpositive, while IR drops taken in the directionof current flow (including the internal dropof the sources of voltage) are considerednegative.Figure 11 shows an example of the applicationof Kirchhoff's laws to a comparativelysimple circuit consisting of three resistors and
<strong>HANDBOOK</strong>Kirchoff's Laws 291.2.20H~2 OHMS" '. / T- 3 VOLTS- /~ +~30HMS+-=- 3 VOLTSSET VOLT ACE DROPS AROUND EACH LOOP EQUAL TO ZERO.112(0HMS)+2 (!d -12) +3= 0 (FIRST LOOP)-6+2 (12-!1)+3!2=0 (sECOND LOOP)SIMPLIFY2!1+211 -212+3=03. EQUATE41~+3 =!2411+3 211+6--2- --5-4. SIMPLIFY2011+15= 411+12II =--ft AMPERE5. RE-SUBSTITUTE-_g_+32 II2= - 1 T- = : = 1 i- AMPERE212-2!1+312-6=05 l2- 2 I 1 -6 = 021~+6=12Figure 11ILLUSTRATING KIRCHHOFF'SSECOND LAWThe voltage c/rop around any closed loop In anetwork Is equal to Z'ero.two batteries. First assume an arbitrary directionof current flow in each closed loop of thecircuit, drawing an arrow to indicate the assumeddirection of current flow. Then equatethe sum of all IR drops plus battery dropsaround each loop to zero. You will need oneequation for each unknown to be determined.Then solve the equations for the unknown currentsin the general manner indicated in figure11. If the answer comes out positive the directionof current flow you originally assumedwas correct. If the answer comes out negative,the current flow is in the opposite direction tothe arrow which was drawn originally. This isillustrated in the example of figure 11 wherethe direction of flow of I, is opposite to thedirection assumed in the sketch.Power in In order to cause electronsResistive Circuits to flow through a conductor,constituting a current flow,it is necessary to apply an electromotive force(voltage) across the circuit. Less power isexpended in creating a small current flowthrough a given resistance than in creatinga large one; so it is necessary to have a unitof power as a reference.The unit of electrical power is the watt,which is the rate of energy consumption whenan e.m.f. of 1 volt forces a current of 1 amperethrough a circuit. The power in a resistivecircuit is equal to the product of the voltageapplied across, and the current flowingin, a given circuit. Hence: P (watts) ~ E(volts) xI (amperes).Since it is often convenient to expresspower in terms of the resistance of the circuitand the current flowing through it, a sub::;titutionof IR for E (E "' IR) in the above formulagives: P ~ IR x I or P ~ I 2 R. In terms of voltageand resistance, P ~ E 2 /R. Here, I ~ E/Rand when this is substituted for I the originalformula becomes P ~ E x E/R, or P ~ E 2 /R.To repeat these three expressions:P ~ EI, P ~ I 2 R, and P = E 2 /R,where P is the power in watts,E is the electromotive force in volts,andI is the current in amperes.To apply the above equations to a typicalproblem: The voltage drop across a cathoderesistor in a power amplifier stage is 50 volts;the plate current flowing through the resistoris 150 milliamperes. The number of watts theresistor will be required to dissipate is foundfrom the formula: P ~ EI, or 50 x .150 ~ 7.5watts (.150 amperes is equal to 150 milliamperes).From the foregoing it is seen thata 7.5-watt resistor will safely carry the requiredcurrent, yet a 10- or 20-watt resistorwould ordinarily be used to provide a safetyfactor.In another problem, the conditions being~imilar to tfiose above, but with the resistance(R ~ 333 1 /, ohms), and current being the knownfactors, the solution is obtained as follows:P = I"R = .0225 x 333.33 = 7.5. If only the voltageand resistance are known, P = E 2 /R =2500/333.33 = 7.5 watts. It is seen that allthree equations give the same results; theselection of the particular equation dependsonly upon the known factors.Power, Energy It is important to rememberand Work that power (expressed in watts,horsepower, etc.), representsthe rate of energy consumption or the rate ofdoing work. But when we pay our electric billFigure 12MATCHING OFRESISTANCESDL+To deliver the great .. t amount of power to theload, the load resistance RL should be equal tothe Internal res/stance of the &attel'y R1.
30 Direct Current Circuits<strong>THE</strong> R AD I 0is said to have a certain capacitance. Theenergy stored in an electrostatic field is expressedin joules (watt seconds) and is equalto CE'/2, where Cis the capacitance in farads(a unit of capacitance to be discussed) and Eis the potential in volts. The charge is equalto CE, the charge being expressed in coulombs.Figure 13TYPICAL CAPACITORSThe two large units are high value filter capac/.tors. Shown beneath these are various types ofby-pass capacitors for r·f anJ auclio application.to the power company we have purchased aspecific amount of energy or work expressedin the common units of kilowatt-hours. Thusrate of energy consumption (watts or kilowatts)multiplied by time (seconds, minutes or hours)gives us total energy or work. Other units ofenergy are the watt-second, BTU, calorie, erg,and joule.Heating Effect Heat is generated when asource of voltage causes acurrent to flow through a resistor (or, for thatmatter, through any conductor). As explainedearlier, this is due to the fact that heat isgiven off when free electrons collide with theatoms of the material. More heat is generatedin high resistance materials than in those oflow resistance, since the free electrons muststrike the atoms harder to knock off otherelectrons. As the heating effect is a functionof the current flowing and the resistance ofthe circuit, the power expended in heat isgiven by the second formula: P = I 2 R.2-3 Electrostatics - CapacitorsElectrical energy can be stored in an electrostaticfield. A device capable of storingenergy in such a field is called capacitor(in earlier usage the term condenser wasfrequently used but the IRE standards call forthe use of capacitor instead of condenser) andCapacitance and Two metallic plates sepCapacitors arated from each other bya thin layer of insulatingmaterial (called a dielectric, in this case),becomes a capacitor. When a source of d-epotential is momentarily applied across theseplates, they may be said to become charged.If the same two plates are then joined to·gether momentarily by means of a switch, thecapacitor will discharge.When the potential was first applied, electronsimmediately flowed from one plate to theother through the battery or such source ofd-e potential as was applied to the capacitorplates. However, the circuit from plate toplate in the capacitor was incomplete (the twoplates being separated by an insulator) andthus the electron flow ceased, meanwhile establishinga shortage of electrons on one plateand a surplus of electrons on the other.Remember that when a deficiency of elec·trons exists at one end of a conductor, thereis always a tendency for the electrons to moveabout in such a manner as to re-establish astate of balance. In the case of the capacitorherein discussed, the surplus quantity of electronson one of the capacitor plates cannotmove to the other plate because the circuithas been broken; that is, the battery or d-e po·tential was removed. This leaves the capacitorin a charged condition; the capacitor platewith the electron deficiency is positivelycharged, the other plate being negative.In this condition, a considerable stressexists in the insulating material (dielectric)which separates the two capacitor plates, dueto the mutual attraction of two unlike potentialson the plates. This stress is known aselectrostatic energy, as contrasted with electromagneticenergy in the case of an inductor.This charge can also be called potentialenergy because it is capable of performingwork when the charge is released through anexternal circuit. The charge is proportional tothe voltage but the energy is proportional tothe voltage squared, as shown in the followinganalogy.The charge represents a definite amount ofelectricity, or a given number of electrons.The potential energy possessed by theseelectrons depends not only upon their number,but also upon their potential or voltage.Compare the electrons to water, and twocapacitors to standpipes, a 1 llfd. capacitor to
<strong>HANDBOOK</strong>Capacitance 31,..-~1 LeECJROSTATIC.IV"~~~L~~~RONS1 micro-microfarad~ 1/1,000,000 of a micro·farad, or .000001 microfarad, or w-• microfarads.+~CHARGING CURRENTFigure 14SIMPLE CAPACITORIllustrating the Imaginary lines of Ioree repre~sentln!l the paths along whic:h the repelling lorc:eol the electrons would oct on a free electronlocated between the two capacitor plates.a standpipe having a cross section of 1 squareinch and a 2 pJd. capacitor to a standpipe havinga cross section of 2 square inches. Thecharge will represent a given volume of water,as the "charge" simply indicates a certainnumber of electrons. Suppose the water isequal to 5 gallons.Now the potential energy, or capacity fordoing work, of the 5 gallons of water will betwice as great when confined to the 1 sq. in.standpipe as when confined to the 2 sq. in.standpipe. Yet the volume of water, or "charge"is the same in either case.Likewise a 1 /-(fd. capacitor charged to 1000volts possesses twice as much potentialenergy as does a 2 llfd. capacitor charged to500 volts, though the charge (expressed incoulombs: Q ~ CE) is the same in either case.The Unit of Capacitance:The FaradIf the external circuit ofthe two capacitor plates iscompleted by joining theterminals together with a piece of wire, theelectrons will rush immediately from one plateto the other through the external circuit andestablish a state of equilibrium. This latterphenomenon explains the discharge of a capacitor.The amount of stored energy in a chargedcapacitor is dependent upon the charging potential,as well as a factor which takes intoaccount 'the size of the plates, dielectricthickness, nature of the dielectric, and thenumber of plates. This factor, which is determinedby the foregoing, is called the capacitanceof a capacitor andis expressed in farads.The farad is such a large unit of capacitancethat it is rarely used in radio calculations,and the following more practical unitshave, therefore, been chosen.1 microfarad~ 1/1,000,000 of a farad, or.000001 farad, or w-• farads.1 micro-microfarad ~ one-millionth of onemillionthof a farad, or 10- 12 farads.If the capacitance is to be expressed inmicrofarads in the equation given for energystorage, the factor C would then have to bedivided by 1,000,000, thus:c X E 2Stored energy in joules ~ -------2 X 1,000,000This storage of energy in a capacitor is oneof its very important properties, particularlyin those capacitors which are used in powersupply filter circuits.DielectricMaterialsAlthough any substance which hasthe characteristics of a good insulatormay be used as a dielectricmaterial, commercially manufactured ca·pacitors make use of dielectric m~terialswhich have been selected because their characteristicsare particularly suited to the job athand. Air is a very good dielectric material,but an air-spaced capacitor does not have ahigh capacitance since the dielectric constantof air is only slightly greater than one. Agroup of other commonly used dielectric mateialsis listed in figure 15.Certain materials, such as bakelite, lucite,and other plastics dissipate considerableenergy when used as capacitor dielectrics.TABLE OF DIELECTRIC MATERIALSDIELE:CTRIJ POWER SOFTENINGMATERIAL CONSTANT FACTOR POINTIOMC. 10 MC. FAHRENHEITI IANILINE-FORMALDEHYDERESIN3.40.004 260°BARIUY TITANATE 1200 1.0CASTOR OIL 4.67CELLULOSE ACETATE 3. 7 0.04 180°GLASS, WINDOW 6-6 POOR 2000°GLASS, PYREX 4.5 0.02KEL-F FLUORO<strong>THE</strong>NE 2.5 0.6----~METHYL- METHACRYLATE2.6 0.007 160°LUCITEMICA 5.4 0.0003MYCALEX MYKROY 7.0 0.002 650°PHENOL- FORMALDEHYDE,LOW-LOSS YELLOW 5.0 0.015 270°PHENOL- FORMALDEHYDEBLACK BAKELITE5.5 0.03 350°PORCELAIN 7.0 o.oos -~~POLYETHYLENE2~- r-¥o*}- 220°POLYSTYRENE _2.55 -- 1750QUARTZ, FUSED 4.2 r--o.ooo2·- I 2eoOo-RUBBER, HARD-EBONITE 2.8 0.007 150°STEATITE 6.1 0.003 2700"'SULFUR 3.8 0.003 23&•TEFLON 2. I 0.02TITANIUM DIOXIDE 100- t75 0.0006 r-27oo-;-TRANSFORMER OIL 2.2 0.003UREA -F0Rt4ALDEHYDE 5.0 0.05 260"'\IINYL RESINS 4.0 0.02 200"'WOOD, MAPLE 4.4 POORFIGURE 15
32 Direct Current Circuits<strong>THE</strong> R AD I 0This energy loss is expressed in terms of thepower factor of the capacitor, which representsthe portion of the input volt-ampereslost in the dielectric material. Other materialsincluding air, polystyrene and quartz have avery low power factor.The new ceramic dielectrics such as steatite(talc) and titanium dioxide products areespecially suited for high frequency and hightemperature operation. Ceramics based upontitanium dioxide have an unusually high dielectricconstant combined with a low powerfactor. The temperature coefficient with respectto capacity of units made with thismaterial depends upon the mixture of oxides,and coefficients ranging from zero to over-700 parts per million per degree Centigrademay be obtained in commercial production.Mycalex is a composition of minute micaparticles and lead borate glass, mixed andfired at a relatively low temperature. It is hardand brittle, but can be drilled or machinedwhen water is used as the cutting lubricant.Mica dielectric capacitors have a very lowpower factor and extremely high voltage breakdownper unit of thickness. A mica and copperfoil"sandwich" is formed under pressure toobtain the desired capacity value. The effectof temperature upon the pressures in the"sandwich" causes the capacity of the usualmica capacitor to have large, non-cyclic variations.If the copper electrodes are plateddirectly upon the mica sheets, the temperaturecoefficient can be stablized at about 20 partsper million per degree Centigrade. A processof this type is used in the manufacture of"silver mica" capacitors.Paper dielectric capacitors consist of stripsof aluminum foil insulated from each other bya thin layer of paper, the whole assembly beingwrapped in a circular bundle. The cost of sucha capacitor is low, the capacity is high inproportion to the size and weight, and thepower factor is good. The life of such a capacitorisdependent upon the moisture penetrationof the paper dielectric and upon the appliedd-e voltage.Air dielectric capacitors are used in transmittingand receiving circuits, principallywhere a variable capacitor of high resetabilityis required. The dielectric strength is high,though somewhat less at radio frequenciesthan at 60 cycles. In addition, corona dischargeat high frequencies will cause ionizationof the air dielectric causing an increasein power loss. Dielectric strength may be increasedby increasing the air pressure, as isdone in hermetically sealed radar units. Insome units, dry nitrogen gas may be used inplace of air to provide a higher dielectricstrength than that of air.Likewise, the dielectric strength of an "air"capacitor may be increased by placing theunit in a vacuum chamber to prevent ionizationof the dielectric.The temperature coefficient of a variableair dielectric capacitor varies widely and isoften non-cyclic. Such things as differentialexpansion of various parts of the capacitor,changes in internal stresses and differenttemperature coeffidents of various parts contributeto these variances.DielectricConstantThe capacitance of a capacitor isdetermined by the thickness andnature of the dielectric materialbetween plates. Certain materials offer agreater capacitance than others, dependingupon their physical makeup and chemical constitution.This property is expressed by aconstant K, called the dielectric constant.(K = 1 for air.)DielectricBreakdownIf the charge becomes too greatfor a given thickness of a certaindielectric, the capacitor will breakdown, i.e., the dielectric will puncture. It isfor this reason that capacitors are rated in themanner of the amount of voltage they willsafely withstand as well as the capacitancein microfarads. This rating is commonly expressedas the d-e working voltage.Calculation of The capacitance of two parallelCapacitance plates is given with good accuracyby the following formula:
<strong>HANDBOOK</strong>Capacitive Circuits 33PARALLEL CAPACITORS1crSERIES CAPACITORSCAPACITORS IN SERIES-PARALLELFigure 17CAPACITORS IN SERIES, PARALLEL,AND SERIES-PARALLELAC~0.2248xKx-,twhere C ~ capacitance in micro-microfarads,K ~ dielectric constant of spacingmaterial,A ~ area of dielectric in square inches,t ~ thickness of dielectric in inches.lcIThis formula indicates that the capacitanceis directly proportional to the area of theplates and inversely proportional to the thicknessof the dielectric (spacing between theplates). This simply means that when the areaof the plate is doubled, the spacing betweenplates remaining constant, the capacitancewill be doubled. Also, if the area of the platesremains constant, and the plate spacing isdoubled, the capacitance will be reduced tohalf.The above equation also shows that capacitanceis directly proportional to the dielectricconstant of the spacing material. An air-spacedcapacitor that has a capacitance of 100 p.p.fd.in air would have a capacitance of 467 p.p.fd.when immersed in castor oil, because the dielectricconstant of castor oil is 4.67 times asgreat as the dielectric constant of air.Where the area of the plates is definitelyset, when it is desired to know the spacingneeded to secure a required capacitance,A X 0.2248 X Kt~-------cwhere all units are expressed just as in thepreceding formula. This formula is not confinedto capacitors having only square orrectangular plates, but also applies when theplates are circular in shape. The only changewill be the calculation of the area of suchcircular plates; this area can be computed bysquaring the radius of the plate, then multiplyingby 3.1416, or "pi." Expressed as anequation:A~ 3.1416 x r 2where r ~ radius in inchesThe capacitance of a multi-plate capacitorcan be calculated by taking the capacitance ofone section and multiplying this by the numberof dielectric spaces. In such cases, however,the formula gives no consideration to theeffects of edge capacitance; so the capacitanceas calculated will not be entirely accurate.These additional capacitances will bebut a small part of the effective total capacitance,particularly when the plates are reasonablylarge and thin, and the final result will,therefore, be within practical limits of accuracy.Capacitors inParallel andin SeriesEquations for calculating capacitancesof capacitors in parallelconnections are the sameas those for resistors in series.C ~ C 1 + C 2 + ... + CnCapacitors in series connection ~re ca_lculatedin the same manner as are resistors mparallel connection.The formulas are repeated: (I) For two ormore capacitors of unequal capacitance inseries:c~----1 1-+-+c,c. c,1 1 1or-~-+-+C C, C 2C,(2) Two capacitors of unequal capacitance inseries:C 1X C 2c~--c, +c.(3) Three capacitors of equal capacitance inseries:c,C ~- where C, is the common capacitance.3( 4) Three or more capacitors of equal capacitancein series.c~Value of common capacitanceNumber of capacitors in series(5) Six capacitors in series parallel:
34 Direct Current Circuits<strong>THE</strong> R A 0 I 01 1c~---+---+---1 1 1 1-+- -+- +c, c2 c, c. c, c.Capacitors in A-C When a capacitor is conandD-C Circuits nected into a direct-currentcircuit, it will blockthe ~i.
<strong>HANDBOOK</strong>Magnetism35the capacitor connects to the positive terminalof the next capacitor in the series combination.The method of connection for electrolytic capacitorsin series is shown in figure 18. Electrolyticcapacitors have very low cost permicrofarad of capacity, but also have a largepower factor and high leakage; both dependentupon applied voltage, temperature and the ageof the capacitor. The modern electrolytic capacitoruses a dry paste electrolyte embeddedin a gauze or paper dielectric. Aluminium foiland the dielectric are wrapped in a circularbundle and are mounted in a cardboard or metalbox. Etched electrodes may be employed toincrease the effective anode area, and thetotal capacity of the unit.The capacity of an electrolytic capacitor isaffected by the applied voltage, the usage ofthe capacitor, and the temperature and humidityof the environment. The capacity usually dropswith the aging of the unit. The leakage currentand power factor increase with age. At highfrequencies the power factor becomes so poorthat the electrolytic capacitor acts as a seriesresistance rather than as a capacity.2-4 Magnetismand ElectromagnetismThe common bar or horseshoe magnet isfamiliar to most people. The magnetic fieldwhich surrounds it causes the magnet to attractother magnetic materials, such as ironnails or tacks. Exactly the same kind of magneticfield is set up around any conductorcarrying a curtent, but the field exists onlywhile the current is flowing.Magnetic Fields Before a potential, or voltage,is applied to a conductorthere is no external field, because thereis no general movement of the electrons inone direction. However, the electrons do progressivelymove along the conductor when ane.m.f. is applied, the direction of motion dependingupon the polarity of the e.m.f. Sinceeach electron has an electric field about it, theflow of electrons causes these fields to buildup into a resultant external field which acts ina plane at right angles to the direction inwhich the current is flowing. This field isknown as the magnetic field.The magnetic field around a current-carryingconductor is illustrated in figure 19. Thedirection of this magnetic field depends entirelyupon the direction of electron drift orcurrent flow in the conductor. When the flowis toward the observer, the field about theconductor is clockwise; when the flow is awayfrom the observer, the field is counter-clockwise.This is easily remembered if the lefthand is clenched, with the thumb outstretchedELECTRON - OR I FT--._SWITCH- +Figure 19I.EFT·HAND RULEShowing the direction of the magnetic lines ofIoree produced around a conrluctor carrying anelectric current.and pointing in the direction of electron flow.The fingers then indicate the direction of themagnetic field around the conductor.Each electron adds its field to the total externalmagnetic field, so that the greater thenumber of electrons moving along the conductor,the stronger will be the resulting field.One of the fundamental laws of magnetismis that like poles repel one another and unlikepoles attract one another. This is true of current-carryingconductors as well as of permanentmagnets. Thus, if two conductors are placedside by side and the current in each is flowingin the same direction, the magnetic fields willalso be in the same direction and will combineto form a larger and stronger field. If the currentflow in adjacent conductors is in oppositedirections, the magnetic fields oppose eachother and tend to cancel.The magnetic field around a conductor maybe considerably increased in strength by windingthe wire into a coil. The field around eachwire then combines with those of the adjacentturns to form a total field through the coilwhich is concentrated along the axis of thecoil and behaves externally in a way similarto the field of a bar magnet.If the left hand is held so that the thumbis outstretched and parallel to the axis of acoil, with the fingers curled to indicate thedirection of electron flow around the turns ofthe coil, the thumb then points in the directionof the north pole of the magnetic field.The MagneticCircuitIn the magnetic circuit, theunits which correspond to current,voltage, and resistancein the electrical circuit are flux, magnetomotiveforce, and reluctance.Flux, FluxDensityAs a current is made up of a driftof electrons, so is a magneticfield made up of lines of force andthe total number of lines of force in a givenmagnetic circuit is termed the flux. The fluxdepends upon the material, cross section, andlength of the magnetic circuit, and it variesdirectly as the curtent flowing· in the circuit.
36 Direct Current Circuits<strong>THE</strong> R AD I 0The unit of flux is the maxwell, and the symbolis the Greek letter cp (phi).Flux density is the number of lines of forceper unit area. It is expressed in gauss if theunit of area is the square centimeter (1 gauss= 1 line of force per square centimeter), orin lines per square inch. The symbol for fluxdensity is B if it is expressed in gausses, orB if expressed in lines per square inch.MagnetomotiveForceThe force which produces aflux in a magnetic circuitis called magnetomotive force.It is abbreviated m.m.f. and is designated bythe letter F. The unit of magnetomotive forceis the gilbert, which is equivalent to 1.26 x NI,where N is the number of turns and I is thecurrent flowing in the circuit in amperes.The m.m.f. necessary to produce a givenflux density is stated in gilberts per centimeter(oersteds) (H), or in ampere-turns perinch (H).Reluctance Magnetic reluctance correspondsto electrical resistance, and isthe property of a material that opposes thecreation of a magnetic flux in the material.It is expressed in rels, and the symbol is theletter R. A material has a reluctance of 1 relwhen an m.m.f. of 1 ampere-turn (NI) generatesa flux of 1 line of force in it. Combinationsof reluctances are treated the same as resistancesin finding the total effective reluctance.The specific reluctance of any substanceis its reluctance per unit volume.Except for iron and its alloys, most commonmaterials have a specific reluctance verynearly the same as that of a vacuum, which,for all practical purposes, may be consideredthe same as the specific reluctance of air.Ohm's Law for The relations between flux,Magnetic Circuits magnetomotive force, andreluctance are exactly thesame as the relations between current, voltage,and resistance in the electrical circuit.These can be stated as follows:F F
<strong>HANDBOOK</strong>Inductance 37original direction, the flux passes through atypical hysteresis loop as shown.Residual Magnetism; The magnetism remainingRetentivity in a material after themagnetizing force is removedis called residual magnetism. Retentivityis the property which causes a magneticmaterial to have residual magnetism afterhaving been magnetized.Hysteresis;Coercive ForceHysteresis is the characteristicof a magnetic systemwhich causes a loss of powerdue to the fact that a negative magnetizingforce must be applied to reduce the residualmagnetism to zero. This negative force istermed coercive force. By "negative" magnetizingforce is meant one which is of theopposite polarity with respect to the originalmagnetizing force. Hysteresis loss is apparentin transformers and chokes by the heating ofthe core.Inductance If the switch shown in figure 19is opened and closed, a pulsatingdirect current will be produced. When it isfirst closed, the current does not instantaneouslyrise to its maximum value, but buildsup to it. While it is building up, the magneticfield is expanding around the conductor. Ofcourse, this happens in a small fraction of asecond. If the switch is then opened, the currentstops and the magnetic field contractsquickly. This expanding and contracting fieldwill induce a current in any other conductorthat is part of a continuous circuit which itcuts. Such a field can be obtained in the wayjust mentioned by means of a vibrator interruptor,or by applying a.c. to the circuit inplace of the battery. Varying the resistance ofthe circuit will also produce the same effect.This inducing of a current in a conductor dueto a varying current in another conductor notin acutal contact is called electromagnetic induction.Self-inductance If an alternating current flowsthrough a coil the varyingmagnetic field around each turn cuts itself andthe adjacent turn and induces a voltage in thecoil of opposite polarity to the applied e. m.f.The amount of induced voltage depends uponthe number of turns in the coil, the currentflowing in the coil, and the number of linesof force threading the coil. The voltage soinduced is known as a counter-e.m.f. or backe.m. f., and the effect is termed self-induction.When the applied voltage is building up, thecounter-e.m.f. opposes the rise; when the appliedvoltage is decreasing, the counter-e.m.f.is of the same polarity and tends to maintainthe current. Thus, it can be seen that selfinductiontends to prevent any change in thecurrent in the circuit.The storage of energy in a magnetic fieldis expressed in joules and is equal to (LI 2 )/2.(A joule is equal to 1 watt-second. L is definedimmediately following.)The Unit ofInductance;The HenryInductance is usually denoted bythe letter L, and is expressed inhenrys. A coil has an inductanceof 1 henry when a voltage of 1volt is induced by a current change of 1 ampereper second. The henry, while commonlyused in audio frequency circuits, is too largefor reference to inductance coils, such asthose used in radio frequency circuits; millihenryor microhenry is more commonly used,in the following manner:1 henry = 1,000 millihenrys, or 10 3 millihenrys.1 millihenry= 1/1,000 of a henry, .001 henry,or w-• henry.1 microhenry = 1/1,000,000 of a henry, or.000001 henry, or w-o henry.1 microhenry = 1/1,000 of a millihenry, .001or w-• millihenrys.1,000 microhenrys = 1 millihenry.Mutual Inductance When one coil is near another,a varying current inone will produce a varying magnetic fieldwhich cuts the turns of the other coil, inducinga current in it. This induced current is alsovarying, and will therefore induce another currentin the first coil. This reaction betweentwo coupled circuits is called mutual induction,and can be calculated and expressed in henrys.The symbol for mutual inductance is M. Twocircuits thus joined are said to be inductivelycoupled.The magnitude of the mutual inductance dependsupon the shape and size of the two circuits,their positions and distances apart, andthe premeability of the medium. The extent toFigure 21MUTUAL INDUCTANCEThe quantity M represents the mutual inductancebetween the two coifs L1 and L2.
38 Direct Current Circuits<strong>THE</strong> R AD I 0WHEREINDUCTANCE OFSINGLE- LAYERSOLENOID COILSR2 N2L = gR+ 10LR =RADIUS OF COIL TO CENTER OF WIREL =LENGTH OF COILN = NUMBER OF TURNSMICROHENFUESFigure 22FORMULA FORCALCULATING INDUCTANCEThrough the use ol the equation and the sketchshown above the Inductance of single-layersolenoid coils can be calcufatecl with an accuracyof about one per cent for the types ofcoils normally used In the h-1 and v·h·l range.which two inductors are coupled is expressedby a relation known as coefficient of coupling.This is the ratio of the mutual inductance ac·tually present to the maximum possible value.The formula for mutual inductance is L =L, + L 2 + 2M when the coils are poled so thattheir fields add. When they are poled so thattheir fields buck, then L = L, + L 2- 2M(figure 21).Inductors inParallelInductors in parallel are com·bined exactly as are resistors inparallel, provided rhat they arefar enough apart so that the mutual inductanceis entirely negligible.Inductors inSeriesInductors in series are additive,just as are resistors in series,again provided that no mutualinductance exists. In this case, the total in·ductance L is:L = L, + L 2 + •••••••• etc.Where mutual inductance does exist:L = L, + L 2 +2M,where M is the mutual inductance.This latter expression assumes that thecoils are connected in such a way that all fluxlinkages are in the same direction, i.e., ad·ditive. If this is not the case and the mutuallinkages subtract from rhe self-linkages, thefollowing formula holds:L = L 1 + L 2 - 2M,where M is the mutual inductance.Core Material Ordinary magnetic cores can·not be used for radio frequen·cies because the eddy current and hysteresislosses in the core material becomes enormousas rhe frequency is increased. The principaluse for conventional magnetic cores is in theaudio-frequency range below approximately15,000 cycles, whereas at very low frequencies(50 to 60 cycles) their use is mandatory ifan appreciable value of inductance is desired.An air core inductor of only 1 henry in·ductance would be quite large in size, yetvalues as high as 500 henrys are commonlyavailable in small iron core chokes. The in·ductance of a coil with a magnetic core willvary with the amount of current (both a-c andd-e) which passes through the coil. For thisreason, iron core chokes that are used in powersupplies have a certain inductance rating at apredetermined value of d-e.The premeability of air does not changewith flux density; so the inductance of ironcore coils often is made less dependent uponflux density by making part of the magneticpath air, instead of utilizing a closed loop ofiron. This incorporation of an air gap is necessaryin many applications of iron core coils,particularly where the coil carries a considerabled-e component. Because the permeabilityof air is so much lower than that of iron, theair gap need comprise only a small fraction ofthe magnetic circuit in order to provide a substantialproportion of the total reluctance.Iron Cared Inductors Iron-core inductors mayat Radio Frequencies be used at radio frequen·cies if rhe iron is in avery finely divided form, as in the case of thepowdered iron cores used in some types of r·fcoils and i-f transformers. These cores aremado of extremely small particles of iron. Theparticles are treated with an insulating materialso that each particle will be insulated fromthe others, and the treated powder is moldedwith a binder into cores. Eddy current lossesare greatly reduced, with the result that thesespecial iron cores are entirely practical in circuitswhich operate up to 100 Me. in frequency.2-5 RC and RL TransientsA voltage divider may be constructed asshown in £i g u r e 23. Kirchhoff's and Ohm'sLaws hold for such a divider. This circuit isknown as an RC circuit.Time Constant· When switch S in figure 23 isRC and RL placed in position 1, a volt-Circuits meter across capacitor C willindicate the manner in whichthe capacitor will become charged through theresistor R from battery B. If relatively largevalues are used for R and C, and if a v·t volt·meter which draws negligible current is used
<strong>HANDBOOK</strong>Time Constant 39to measure the voltage e, the rate of charge ofthe capacitor may actually be plotted with theaid of a stop watch.~~ 80 f==--·-f---..,..~--'"'+----1---1< I~LU60,---~7 -- ~-~~ ~ 40~-7 --- ---t----w oY- t 2 3a: TIMEt, IN TERMS OF TIME CONSTANT RCw~ 100~~
40 Direct Current Circuits'05 R (INCLUDING D.C. RESISTANCEc" '""'"'" c:w[a:~100~a:a:::> 80
CHAPTER THREEAlternating Current CircuitsThe previous chapter has been devoted toa discussion of circuits and circuit elementsupon which is impressed a current consistingof a flow of electrons ia one direction. Thistype of unidirectional current flow is calleddirect current, abbreviated d. c. Equally as importantin radio and communications work,and power practice, is a type of current flowwhose direction of electron flow reversesperiodically. The reversal of flow may takeplace at a low rate, in the case of power systems,or it may take place millions of timesper second in the case of communicationsfrequencies. This type of current flow iscalled alternating current, abbreviated a. c.3-1 Alternating CurrentFrequency of anAlternating CurrentAn alternating current isone whose amplitude ofcurrent flow periodicallyrises from zero to a maximum in one direction,decreases to zero, changes its direction,rises to maximum in the opposite direction,and decreases to zero again, This completeprocess, starting from zero, passing throughtwo maximums in opposite directions, and returningto zero again, is called a cycle. Thenumber of times per second that a currentpasses through the complete cycle is calledthe frequency of the current. One and onequarter cycles of an alternating current waveare illustrated diagrammatically in figure I.41Frequency Spectrum At present the usable frequencyrange for alternatingelectrical currents extends over the enormousfrequency range from about 15 cycles persecond to perhaps 30,000,000,000 cycles persecond. It is obviously cumbersome to use afrequency designation in c.p.s. for enormouslyhigh frequencies, so three common units whichare multiples of one cycle per second havebeen established.~l+r---------------------------------::!1C((31 0 ~---------------TIME~IDIRECT CURRENTALTERNATING CURRENTFigure IALTERNATING CURRENTAND DIRECT CURRENTGraphical comparison between un/Jrectlonal( clirect} current and alternating current as plottedagainst time.
42 Alternating Current Circuits <strong>THE</strong> R AD I 0These units are:(1) the kilocycle (abbr., kc.), 1000 c.p.s.(2) the Megacycle (abbr., Me.), 1,000,000c.p.s. or 1000 kc.(3) the kilo-Megacycle (abbr., kMc.),1,000,000,000 c.p.s. or 1000 Me.With easily handled units such as these wecan classify the entire usable frequency rangeinto frequency bands.The frequencies falling between about 15and 20,000 c.p.s. are called audio frequencies,abbreviated a. f., since these frequencies areaudible to the human ear when converted fromelectrical to acoustical signals by a loudspeakeror headphone. Frequencies in thevicinity of 60 c.p. s. also are called power frequencies,since they are commonly used todistribute electrical power to the consumer.The frequencies falling between 10,000c.p.s. (10 kc.) and 30,000,000,000 c.p.s. (30kMc.) are commonly called radio frequencies,abbreviated r.f., since they are commonly usedin radio communication and allied arts. Theradio-frequency spectrum is often arbitrarilyclassified into seven frequency bands, eachone of which is ten times as high in frequencyas the one just below it in the spectrum (exceptfor the v-1-£ band at the bottom end ofthe spectrum). The present spectrum, withclassifications, is given below.Frequency10 to 30 kc.30 to 300 kc.300 to 3000 kc.3 to 30 Me.30 to 300 Me.300 to 3000 Me.3 to 30 kMc.30 to 300 kMc.Generation ofAlternating CurrentClassificationVery-low frequenciesLow frequenciesMedium frequenciesHigh frequenciesVery-high frequenciesUltra-high frequenciesSuper-high frequenciesExtremely-highfrequenciesAbbrev.v.l.f1.£.m.f.h. f.v.h.f.u.h.f.s.h.f.e.h.f.Faraday discovered that ifa conductor which formspart of a closed circuit ismoved through a magnetic field so as to cutacross the lines of force, a current will flowin the conductor. He also discovered that, if aconductor in a second closed circuit is broughtnear the first conductor and the current in thefirst one is varied, a current will flow in thesecond conductor. This effect is known asinduction, and the currents so generated areinduced currents. In the latter case it is thelines of force which are moving and cuttingthe second conductor, due to the varying currentstrength in the first conductor.A current is induced in a conductor if thereis a relative motion between the conductorand a magnetic field, its direction of flow dependingupon the direction of the relativeFigure 2<strong>THE</strong> ALTERNATORSeml·schematlc representation of the simplestform of the alternator.motion between the conductor and the field,and its strength depends upon the intensity ofthe field, the rate of cutting lines of force, andthe number of turns in the conductor.Alternators A machine that generates an alternatingcurrent is called an alternatoror a-c generator. Such a machine in itsbasic form is shown in figure 2. It consists oftwo permanent magnets, M, the opposite polesof which face each other and are machined sothat they have a common radius. Betweenthese two poles, north (N) and south (S),a substantially constant magnetic field exists.If a conductor in the form of C is suspendedso that it can be freely rotated between thetwo poles, and if the opposite ends of conductorC are brought to collector rings, therewill be a flow of alternating current when conductorC is rotated. This current will flow outthrough the collector rings R and brushes Bto the external circuit, X-Y.The field intensity between the two polepieces is substantially constant over the entirearea of the pole face However, when theconductor is moving parallel to the lines offorce at the top or bottom of the pole faces,no lines are being cut. As the conductor moveson across the pole face it cuts more and morelines of force for each unit distance of travel,until it is cutting the maximum number oflines when opposite the center of the pole.Therefore, zero current is induced in the conductorat the instant it is midway betweenthe two poles, and maximum current is inducedwhen it is opposite the center of thepole face. After the conductor has rotatedthrough 180° it can be seen that its positionwith respect to the pole pieces will be exactlyopposite to that when it started. Hence, thesecond 180° of rotation will produce an alternationof current in the opposite direction tothat of the first alternation.The current does not increase directly asthe angle of rotation, but rather as the sineof the angle; hence, such a current has themathematical form of a sine wave. Although
<strong>HANDBOOK</strong>The Sine Wave43LINES Of FORCEi+-----1 CYCLE------ILINES OF FORCE(UNIFORM DENSITY)Figure 3OUTPUT OF <strong>THE</strong> ALTERNATORGraph showing sine-wave output cu,ent ol thealternator olligure 2.most electrical machinery does not produce astrictly pure sine curve, the departures areusually so slight that the assumption can beregarded as fact for most practical purposes.All that has been said in the foregoing paragraphsconcerning alternating current also isapplicable to alternating voltage.The rotating arrow to the left in figure 3represents a conductor rotating in a constantmagnetic field of uniform density. The arrowalso can be taken as a vector representing thestrength of the magnetic field. This means thatthe length of the arrow is determined by thestrength of the field (number of lines of force),which is constant. Now if the arrow is rotatingat a constant rate (that is, with constantangular velocity), then the voltage developedacross the conductor will be proportional tothe rate at which it is cutting lines of force,which rate is proportional to the verticaldistance between the tip of the arrow and thehorizontal base line.If EO is taken as unity or a voltage of 1,then the voltage (vertical distance from tip ofarrow to the horizontal base line) at point Cfor instance may be determined simply byreferring to a table of sines and looking up thesine of the angle which the arrow makes withthe horizontal.When the arrow has traveled from A to pointE, it has traveled 90 degrees or one quartercycle. The other three quadrants are not shownbecause their complementary or mirror relationshipto the first quadrant is obvious,It is important to note that time units arerepresented by degrees or quadrants. The factthat AB, BC, CD, and DE are equal chords(forming equal quadrants) simply means thatthe arrow (conductor or vector) is travelingat a constant speed, because these points onthe radius represent the passage of equalunits of time.The whole picture can be represented inanother way, and its derivation from the foregoingis shown in figure 3. The time base isrepresented by a straight line rather than by+w0:::>f.....Ja.::li
44 Alternating Current Circuits <strong>THE</strong> R AD I 0notation are ~sed in discussions of alternatingcurrent cucults. However, trigonometric tablesare much more readily available in terms ofdegrees than radians, so the following simpleconversions are useful.2rr radians = 1 cycle = 360°TT radians = 'l, cycle = 180°1T- radians = Y. cycle = 90°21T-radians = ';. cycle = 60°31T-radians = '1. cycle = 45°411 radian =-cycle= 57.3°2TTWhen the conductor in the simple alternatorof figure 2 has made one complete revolutionit has generated one cycle and has rotatedthrough 2rr radians. The expression 2rrfthen represents the number of radians in onecycle multiplied by the number of cycles persecond (the frequency) of the alternatingvoltage or current. The expression then representsthe number of radians per second throughwhich the conductor has rotated. Hence 2rrfrepresents the angular velocity of the rotatingconductor, or of the rotating vector whichrepresents any alternating current or voltage,expressed in radians per second.In technical literature the expression 2rr fis often replaced by w, the lower-case Greekletter omega. Velocity multiplied by timegives the distance travelled. so 2rrft (or wt)represents the angular distance through whichthe rotating conductor or the rotating vectorhas travelled since the reference time t = 0.In the case of a sine wave the reference timet = 0 represents that instant when the voltageor the current, whichever is under discussion,also is equal to zero.Instantaneous Value The instantaneous voltageor current is proporof Voltage orCurrenttional to the sine of theangle through which therotating vector has travelled since referencetime t = 0. Hence, when the peak value of thea-c wave amplitude (either voltage or curren.tamplitude) i~ known, and the angle throughwhrch the rotatrng vector has travelled isestablished, the amplitude of the wave atthis instant can be determined through useof the following expression:e = Emax sin 2rrft,WHERE:++-J'----'--4"-"-.LlL.!...Ll- 8 = TT9 (<strong>THE</strong>.TA):: PHASE ANGLE= 2 TT FTA=~ RADIANS OR 90°RADIANS 0 R 180 oC = 3 ~ RADIANS OR 270"D = 2 TT RADIANS OR 360"1 RADIAN= 57.324 DEGREESFigure 5ILLUSTRATING RADIAN NOTATIONThe radian is a unit of phase angle, equal to57.324 degrees. It Is commonly used In mathematicalrelationships involving phase anglessince such relationships are slmplifiecl whenradian notation Is used.where e = the instantaneous voltageE = maximum crest value of voltage,f = frequency in cycles per second, andt =period of time which has elapsedsince t = 0 expressed as a fractionof one second.The instantaneous current can be found fromthe same expression by substituting j for eand Imax for Emax.It is often easier to visualize the process ofdetermining the instantaneous amplitude byignoring the frequency and considering onlyone cycle of the a-c wave. In this case, for asine wave, the expression becomes:e = Emax sinewhere e represents the angle through whichthe vector has rotated since time (and amplitude)were zero. As examples:when e = 30°sin e = 0.5so e = 0.5 Emaxwhen e = 60°sine= 0.866so e = 0.866 Emaxwhen e = 90°sin e = 1.0so e = Emaxwhen e = 1 radiansin e = 0.8415so e = 0.8415 Emax
<strong>HANDBOOK</strong>A-CRelationships45Effective Valueof anAI ternating CurrentThe instantaneous valueof an alternating currentor voltage varies continuouslythroughout the cycle.So some value of an a-c wave must be chosento establish a relationship between the effectivenessof an a-c and a d-e voltage or current,The heating value of an alternatingcurrent has been chosen to establish the referencebetween the effective values of a.c. andd. c. Thus an alternating current will have aneffective value of I ampere when it producesthe same heat in a resistor as does 1 ampereof direct current.The effective value is derived by taking theinstantaneous values of current over a cycle ofalternating current, squaring these values.taking an average of the squares, and thentaking the square root of the average. By thisprocedure, the effective value becomes knownas the root mean square or r.m. s. value. Thisis the value that is read on a-c voltmeters anda-c ammeters. The r.m.s. value is 70.7 (forsine waves only) per cent of the peak or maximuminstantaneous value and is expressed asfollows:Eeff. or Er.m.s. = 0.707 X Emax orleff. or lr.m.s. = 0.707 X lmax.The following relations are extremely usefulin radio and power work:Er.m.s. = 0.707 X Emax, andEmax= 1.414 X Er.m.s.Figure 6FULL-WAVE RECTIFIEDSINE WAVEWaveform obtained at the output of a lui/waverectifier being feel with a sine wave one/ having100 per cent rectilication efficiency. Eachpulse has the same shape as one-half cycle ofa sine wave. This type of current Is known aspulsating direct current.It is thus seen that the average value is 63.6per cent of the peak value.Relationship BetweenPeak, R.M.S. orEffective, andAverage ValuesTo summarize the threemost significant valuesof an a·c sine wave: thepeak value is equal to1.41 times the r.m.s. oreffective, and the r.m.s. value is equal to0. 707 times the peak value; the average valueof a full-wave rectified a-c wave is 0.636times the peak value, and the avera~e valueof a rectified wave is equal to 0.9 Urnes ther.m.s. value.R.M.S. = 0. 707 x PeakAverage = 0.636 x PeakRectified Alternating If an alternating currentCurrent or Pulsat- is passed through a rectiingDirect Current fier, it emerges in theform of a current ofvarying amplitude which. flows in one di_r~ctiononly. Such a current IS known as rect'f'eda. c. or pulsating d. c. A typical wave form _of apulsating direct current as would be obtrunedfrom the output of a full-wave rectifier isshown in figure 6.Measuring instruments designed for d-eoperation will not read the peak for instantaneousmaximum value of the pulsating d-e outputfrom the rectifier; they will read only theaverage value. This can be explained by assumingthat it could be possible to cut offsome of the peaks of the waves, using the cutoffportions to fill in the spaces that are open,thereby obtaining an average d-e value. Amilliammeter and voltmeter connected to theadjoining circuit, or across the output o_f therectifier, will read this average value. It IS relatedto peak value by the following expression:Eavg = 0.636 X EmaxAverage = 0.9 x R.M.S.R.M.S. = 1.11 x AveragePeakPeak= 1.414 X R.M.S.= 1.57 X AverageApplying Ohm's Law Ohm's law appliesto Alternating Current equally to direct or alternatingcurrent, providedthe circuits under consideration arepurely resistive, that is: circuits which_ haveneither inductance (coils) nor capacitance(capacitors). Problems which involve tubefilaments, drop resistors, electric lamps,heaters or similar resistive devices can besolved from Ohm's law, regardless of whetherthe current is direct or alternating. When acapacitor or coil is made a part of the circuit,a property common to either, called reactance,must be taken into consideration. Ohm's lawstill applies to a-c circuits containing reactance,but additional considerations are involved;these will be discussed in a laterparagraph.
46 Alternating Current Circuits <strong>THE</strong> R AD I 0EE1 I1+-•o·-jCURRENT~ VOLTAGE BY 90°(CIRCUIT CONTAINING PURE INDUCTANCE ONLY)Figure 7LAGGING PHASE ANGLEShowing the manner In which the current lagsthe voltage In an o-c circuit containing pureinductance only. The lag Is equal to one-quartercycle or 90 degrees.CURRENT LEADING VOLTAGE BY 90°(CIRCUIT CONTAIN IN
<strong>HANDBOOK</strong>Reactance47wave (either peak or r.m.s.) for both voltageand current are used in the calculations.However, in calculations involving alternatingcurrents the voltage and current are notnecessarily in phase. The current through thecircuit may lag behind the voltage, in whichcase the current is said to have lagging phase.Lagging phase is caused by inducuve reactance.If the current reaches its maximum valueahead of the voltage (figure 8) the current issaid to have a leading phase. A leading phaseangle is caused by capacitive reactance.In an electrical circuit containing reactanceonly, the current will either lead or lag thevoltage by 90°. If the circuit contains inductivereactance only, the current will lag thevoltage by 90°. If only capacitive reactance isin the circuit, the current will lead the voltageby 90°.Inductive and capaC!tl ve reactance have exactly oppositeeffects on the phase relationReactancesin Combinationbetween current and voltage in a circuit.Hence when they are used in combinationtheir effects tend to neutralize. The combinedeffect of a capacitive and an inductive reactanceis often called the net reactance of acircuit. The net reactance (X) is found by subtractingthe capacitive reactance from the inductivereactance, X= XL - Xc.The result of such a combination of purereactances may be either positive, in whichcase the positive reactance is greater so thatthe net reactance is inductive, or it may benegative in which case the capacitive reactanceis greater so that the net reactance iscapacitive. The net reactance may also bezero in which case the circuit is said to beresonant. The condition of resonance will bediscussed in a later section. Note that inductivereactance is always taken as being positivewhile capacitive reactance is alwaystaken as being negative.Impedance; CircuitsContaining Reactanceand ResistancePure reactances introducea phase angle of90° between voltage andcurrent; pure resistanceintroduces no phase shift between voltage andcurrent. Hence we cannot add a reactance anda resistance directly. When a reactance and aresistance are used in combination the resultingphase angle of current flow with respectto the impressed voltage lies somewherebetween plus or minus 90° and 0° dependingupon the relative magnitudes of the reactanceand the resistance.The term impedance is a general term whichcan be applied to any electrical entity whichimpedes the flow of current. Hence the termmay be used to designate a resistance, a pureY-AXISFigure 9Operation on the vector (+A) by the quantity (-1)causes vector to rotate through 180 degrees.reactance, or a complex combination of bothreactance and resistance. The designation forimpedance is Z. An impedance must be definedin such a manner that both its magnitudeand its phase angle are established. Thedesignation may be accomplished in either oftwo ways- one of which is convertible intothe other by simple mathematical operations.The "J" Operator The first method of designatingan impedance isactually to specify both the resistive and thereactive component in the form R + jX. In thisform R represents the resistive component inohms and X represents the reactive component.The "j" merely means that the X componentis reactive and thus cannot be added directlyto the R component. Plus jX means that thereactance is positive or inductive, while ifminus jX were given it would mean that thereactive component was negative or capacitive.In figure 9 we have a vector (+A) 1 ying alongthe positive X-axis of the usual X-Y coordinatesystem. If this vector is multiplied by thequantity (-1), it becomes (-A) and its positionnow lies along the X-axis in the negativedirection. The operator (-1) has caused thevector to rotate through an angle of 180 degrees.Since (-1) is equal to ( y-1 x R>. thesame result may be obtained by operating onrhe vector with the operator (y-1 x y-1 ).However if the vector is operated on but onceby the operator ( yCTI, it is caused to rotateonly 90 degrees (figure 10). Thus the operator( y -1)rotates a vector by 90 degrees. For convenience,this operator is called the j operator.In like fashion, the operator ( -j) rotates thevector of figure 9 through an angle of 270degrees, so that the resulting vector (-jA)falls on the (-Y) axis of the coordinate system.
48 Alternating Current Circuits <strong>THE</strong> R AD I 0Y-AXIS-- _r-(+A) X (-v::i J ROTATES""< VECTOR THROUGH 90°' \+jA ~go• \Figure 10Operation on the vector (+A) by the quantity (/)causes vector to rotate through 90 degrees.Polar NotationThe second method of ~epresentingan impedance 1s tospecify its absolute magnitude and the phaseangle of current with respect to voltage, inthe form Z L 0. Figure 11 shows graphicallythe relationship between the two common waysof representing an impedance.The construction of figure 11 is called animpedance diagram. Through the use of sucha diagram we can add graphically a resistanceand a reactance to obtain a value for the resultingimpedance in the scalar form. Withzero at the origin, resistances are plotted tothe right, positive values of reactance (inductive)in the upward direction, and negativevalues of reactance (capacitive) in the downwarddirection.Note that the resistance and reactance aredrawn as the two sides of a right triangle,with the hypotenuse representing the resultingimpedance. Hence it is p o s sib 1 e to determinemathematically the value of a resultantimpedance through the familiar right-trianglerelationship- the square of the hypotenuse isequal to the sum of the squares of the othertwo sides:z• = R 2 + x•or IZI = y'R 2 + X 2Note also that the angle () included betweenR and Z can be determined from any of thefollowing trigonometric relationships:IZIXsin()=-Rcos 0=IZItan()= XOne common problem is that of determiningthe scalar magnitude of the impedance, IZI,RFigure II<strong>THE</strong> IMPEDANCE TRIANGLEShowing the graphical construction of a trianglefor obtaining the net (scalar) impedance re·suiting from the connection of a resistance anda reactance in series. Shown also alongside Isthe alternative mathematical procedure lorobtaining the values associated with the triangle.and the phase angle (), when resistance andreactance are known; hence, of convertingfrom the Z = R + jX to the I Zl L () form. In thiscase we use two of the expressions just given:IZI = y'R 2 + X 2XXtan () =-, (or()= tan-•-)RRThe inverse problem, that of convertingfrom the IZI L () to the R + jX form is donewith the following relationships, both of whichare obtainable by simple division from thetrigonometric expressions just given for determiningthe angle ():R = IZI cos()jX = I Zl j sin ()By simple addition these two expressions maybe combined to give the relationship betweenthe two most common methods of indicatingan impedance:R + jX = IZI (cos () + j sin (})In the case of impedance, resistance, or reactance,the unit of measurement is the ohm;hence, the ohm may be thought of as a unit ofopposition to current flow, without referenceto the relative phase angle between the appliedvoltage and the current which flows.Further, since both capacitive and inductivereactance are functions of frequency, impedancewill vary with frequency. Figure 12shows the manner in which IZ\ will vary withfrequency in an RL series circuit and in anRC series circuit.Series RLC Circuits In a series circuit containingR, L, and C, the im-
<strong>HANDBOOK</strong>Impedance49pedance is determined as discussed before exceptthat the reactive component in the expressionsbecomes: (The net reactance- thedifference between XL and Xc-) Hence (XL -Xc) may be substituted for X in the equations.Thus:JZI = y'R 2 +(XL- Xc) 2(XL- Xc)e = tan_,_....:._ __ _RA series RLC circuit thus may present animpedance which is capacitively reactive ifthe net reactance is capacitive, inductivelyreactive if the net reactance is inductive, orresistive if the capacitive and inductive reactancesare equal.Addition ofComplex QuantitiesThe addition of complexquantities (for example,impedances in series) isquite simple if the quantities are in the rectangularform. If they are in the polar formthey only can be added graphically, unlessthey are converted to the rectangular form bythe relationships previously given. As an exampleof the addition of complex quantitiesin the rectangular form, the equation for theaddition of impedances is:(R, + jX,) + (Rl + jX2)= (R, + R2) + j(X, + X2)For example if we wish to add the impedances(10 + j50) and (20- j30) we obtain:(10 + j50) + (20- j30)= (10 + 20) + j(50 + (-30))= 30 + j(S0-30)= 30 + j20Multiplieation andDivision ofComplex QuantitiesIt is often necessary insolving certain types ofcircuits to multiply or dividetwo complex quantities.It is a much simplier mathematicaloperation to multiply or divide complex quantitiesif they are expressed in the polar form.Hence if they are given in the rectangularform they should be converted to the polarform before multiplication or division is begun.Then the multiplication is accomplished bymultiplying the IZI terms together and addingalgebraically the L e terms, as:c tz,t LO,) c IZ2I L0 2 ) = tz,JIZ2\ cLe, + Le,>For example, surpose that the two impedances120\ L43° and 132\ L-23° are to be multiplied.Then:c tzot L 43°) Cl32l L-zn = 120 ·321( L43° + L-23°)= 640 L 20°Figure 12IMPEDANCE AGAINST FRE.QUENCYFOR R-L AND R-C CIRCUITSThe Impedance of an R-C circuit approachesinfinity as the frequency approaches zero (d. c.},while the impedance of a series R-L circuitapproaches Infinity as the frequency approachesInfinity. The impedance of an R-C circuit approachesthe lmpec/once of the series resistoras the Frequency approaches Infinity, while the/mpec/once of a series R-L circuit approachesthe lmpeclonce of the resistor as the frequencyapproaches zero.Division is accomplished by dividing thedenominator into the numerator, and subtractingthe angle of the denominator fromthat of the numerator, as:\Z,I Le,\Z2\ L 02~ (LO -l. e)IZll 1 2For example, suppose that an impedance ofISO I L 67° is to be divided by an impedanceof \101 L45°. Then:\50\L67° \50\~--= -(L67°-L45°) = I5ICL22°)\10 I L 45 ° \10 IOhm's Law forComplex QuantitiesThe simple form of Ohm'sLaw used for d-e circuitsmay be stated in a moregeneral form for application to a-c circuitsinvolving either complex quantities or simpleresistive elements. The form is:E1=zin which, in the general case, I, E, and Z arecomplex (vector) quantities. In the simplecase where the impedance is a pure resistancewith an a-c voltage applied, the equationsimplifies to the familiar I == E/R. In any casethe applied voltage may be expressed eitheras peak, r.m.s., or average; the resulting
50 Alternating Current Circuits <strong>THE</strong> R AD I 011200~10r0L TS + j 100j 1 T _j 3000Figure 13SERIES R-L-C CIRCUITcurrent always will be in the same type ofunits as used to define the voltage.In the more general case vector algebramust be used to solve the equation. And,since either division or multiplication is involved,the complex quantities should be expressedin the polar form. As an example,take the case of the series circuit shown infigure 13 with 100 volts applied. The impedanceof the series circuit can best be obtainedfirst in the rectangular form, as:200 + j(l00-300) = 200-j200Now, to obtain the current we must convertthis impedance to the polar form.\Z\ = y 200 2 + ( -200) 2v 40,000 + 40,000y80,000= 282 n_,X _, -200 _,e = tan - = tan --= tan -1R 200= -45°,Therefore Z = 282 L-45°Note that in a series circuit the resulting impedancetakes the sign of the largest react·ance in the series combination.Where a slide-rule is being used to makethe computations, the impedance may be foundwithout any addition or subtraction operationsby finding the angle e first, and then usingthe trigonometric equation below for obtain·ing the impedance. Thus:X -200e = tan-' - = tan-'-- = tan-• -1R 200= -45°RoThen \Z\ = -- cos -45 = 0. 707cos e200\Z\ = --= 28200.707Since the applied voltage will be the referencefor the currents and voltages within the cir·cuit, we may define it as having a zero phaseangle: E = 100 L0°. Then:100 LO 0 0 0I= =0.354LO -(-45)282 L-45°= 0.35 4 L45° amperes.This same current must flow through all threeelements of the circuit, since they are inseries and the current through one must alreadyhave passed through the other two.Hence the voltage drop across the resistor(whose phase angle of course is 0°) is:E=IRE = (0.354 L 45°) (200 L 0°)= 70.8 L 45° voltsThe voltage drop across the inductive react·ance is:E =I XLE = (0.354 L45°) (100 L 90°)= 35.4 L 135° voltsSimilarly, the voltage drop across the capacitivereactance is:E =I XcE = (0.354 L 45°) (300 [-90°)= 106.2 L-45°Note that the voltage drop across the capacitivereactance is greater than the supplyvoltage. This condition often occurs in aseries RLC circuit, and is explained by thefact that the drop across the capacitive react·ance is cancelled to a lesser or greater extentby the drop across the inductive reactance.It is often desirable in a problem such asthe above to check the validity of the answerby adding vectorially the voltage drops acrossthe components of the series circuit to makesure that they add up to the supply voltageorto use the terminology of Kirchhoff's SecondLaw, to make sure that the voltage dropsacross all elements of the circuit, includingthe source taken as negative, is equal to zero.In the general case of the addition of anumber of voltage vectors in series it is bestto resolve the voltages into their in-phaseand out-of-phase components with respect tothe supply voltage. Then these componentsmay be added directly. Hence:ER = 70.8 L 45°= 70.8 (cos 45° + j sin 45°)= 70.8 (0.707 + j0.707)=50+ j50
<strong>HANDBOOK</strong>Vector Algebra s 1VOLTAGE DROP ACROSSXt. 0 35.4~"90"DROP ACROSS RESISTOR.,~ 70.8~~INE VOLTAGE=1QO &_± 1BQO ------
52 Alternating Current Circuits <strong>THE</strong> R AD I 0Then:[z[6cos- 33.7°6--~ 7.21 ohms0.8326 -j4 ~ 7.21 L-33.7°24 L-90°7.21 L- 33. 7o ~ 3-33 L-56.3o~ 3-33 (cos- 56.3° + j sin- 56.3°)~ 3-33 [0.5548 + j (-0.832)]~ 1.85- j 2.77Th p& IhE1Ic !'a2IEz=E1~R1+Rz20 lEz.=E1~XC1 +XC2CaEz=E1 ~IE 2 = E 1 ____h:_L__L1+L2® ® ©Figure 16SIMPLE A·C VOLTAGE DIVIDERS2Equivalent Series Through the series of opCircuit erations in the previousparagraph we have converteda circuit composed of two impedances inparallel into an equivalent series circuit composedof impedances in series. An equivalentseries circuit is one which, as far as the terminalsare concerned, acts identically to theoriginal parallel circuit; the current throughthe circuit and the power dissipation of theresistive elements are the same for a givenvoltage at the specified frequency.We can check the equivalent series circuitof figure 15 with respect to the original circuitby assuming that one volt a. c. (at thefrequency where the capacitive reactance inthe parallel circuit is 4 ohms) is applied tothe terminals of both.In the parallel circuit the current throughthe resistor will be 'f. ampere (0.166a.) whilethe current through the capacitor will be j %ampere ( + j 0.25 a.). The total current will bethe sum of these two currents, or 0.166 +j 0.25 a. Adding these vectorially we obtain:[I[~ y0.166 2 + 0.25 2 ~ y0.09 ~ 0.3 a.The dissipation in the resistor will be 1 2 /6 "'0.166 watts.In the case of the equivalent series circuitthe current will be:E 1III ~TZ\~ 3-33 ~o.3 a.And the dissipation in the resistor will be:W ~ I 2 R ~ 0.3 2 X 1.85~ 0.9 X 1.85= 0.166 wattsSo we see that the equivalent series circuitchecks exactly with the original parallel circuit.Parallel RLCCircuitsIn solving a more complicatedcircuit made up of more thantwo impedances in parallel wemay elect to use either of two methods ofsolution. These methods are called the admittancemethod and the assumed-voltage method.However, the two methods are equivalentsince both use the sum-of-reciprocals equation:1 1--~-+-+- ••.••Z, 0 , Z, Z 2 Z,In the admittance method we use the relationY ~ 1/Z, where Y ~ G + jB; Y is called theadmittance, defined above, G is the conductanceor R/Z 2 and B is the susceptance or-X/Z 2 • ThenYtot ~ 1/Ztot ~ Y, + Y 2 + Y, • • • •In the assumed-voltage method we multiplyboth sides of the equation above by E, theassumed voltage, and add the currents, as:E E E E--~-+-+- ••. ~Iz +Iz +IzZtot z, z2 z, ' 2 'Then the impedance of the parallel combinationmay be determined from the relation:AC VoltageDividersZtot ~ E/Iz totVoltage dividers for use withalternating current are quite similarto d-e voltage dividers. However,since capacitors and inductors opposethe flow of a-c current as well as resistors,voltage dividers for alternating voltages maytake any of the configurations shown in figure16.Since the impedances within each dividerare of the same type, the output voltage is inphase with the input voltage. By using combinationsof different types of impedances, thephase angle of the output may be shifted inrelation to the input phase angle at the sametime the amplitude is reduced. Several dividersof this type are shown in figure 17.Note that the ratio of output voltage to inputvoltage is equal to the ratio of the outputimpedance to the total divider impedance.This relationship is true only if negligiblecurrent is drawn by a load on the output terminals.
<strong>HANDBOOK</strong>Resonant Circuits 53Figure 18SERIES RESONANT CIRCUITEz= E1 __ x_c_XL-XC© @ E3=E1 v.;:o~Xc~";;='C:Y'R•+(Xc Xc)2E 4 =E!XL-XC""VR• +(XL Xc)2Figure 17COMPLEX A-C VOLTAGE DIVIDERS3-2 Resonant CircuitsA series circuit such as shown in figure 18is said to be in resonance when the appliedfrequency is such that the capacitive reactanceis exactly balanced by the inductive reactance.At this frequency the two reactanceswill cancel in their effects, and the impedanceof the circuit will be at a minimum so th'atmaximum current will flow. In fact, as shownin figure 19 the net impedance of a seriescircuit at resonance is equal to the resistancewhich remains in the circuit after the react·ances have been cancelled.Resonant Frequency Some resistance is alwayspresent in a circuit be·cause it is possessed in some degree by boththe inductor and the capacitor. If the fre·quency of the alternator E is varied fromnearly zero to some high frequency, there willbe one particular frequency at which the in·ductive reactance and capacitive reactancewill be equal. This is known as the resonantfrequency, and in a series circuit it is thefrequency at which the circuit current will bea maximum. Such series resonant circuits arechiefly used when it is desirable to allow acertain frequency to pass through the circuit(low impedance to this frequency), while atthe same time the circuit is made to offerconsiderable opposition to currents of otherfrequencies.If the values of inductance and capacitanceboth are fixed, there will be only one resonantfrequency.If both the inductance and capacitance aremade variable, the circuit may then be changedor tuned, so that a number of combinationsof inductance and capacitance can resonate atthe same frequency. This can be more easilyunderstood when one considers that inductivereactance and capacitive reactance travel inopposite directions as the frequency is changed.For example, if the frequency were to remainconstant and the values of inductance andcapacitance were then changed, the followingcombinations would have equal reactance;Frequency is constant at 60 cycles.L is expressed in henrys.C is expressed in microfarads (. 000001farad.)L.2652.6526.5265.002,650.00Frequencyof ResonanceXL c Xc100 26.5 1001,000 2.65 1,00010,000 .265 10,000100,000 .0265 100,0001,000,000 .00265 1,000,000From the formula for reson·ance, 2TrfL : l/brfC, the resonantfrequency is determined:1 f: __ _21r y'LCwhere f: frequency in cycles,L : inductance in henrys,C : capacitance in farads.It is more convenient to express L and Cin smaller units, especially in making radiofrequencycalculations; f can also be ex·pressed in megacycles or kilocycles. A veryuseful group of such formulas is:25,330 25,330 25,330f 2 :---or L :---or C =---LC f 2 C f'Lwhere f = frequency in megacycles,L =inductance in microhenrys,C = capacitance in micromicrofarads.
54 Alternating Current Circuits <strong>THE</strong> R AD I 0w+uz(§"'0..:::;;Iu "'z~
<strong>HANDBOOK</strong>Circuit Q 55sistance and L-to-C ratio are the importantconsiderations. The curves B and C in figure20 show the effect of adding increasing valuesof resistance to the circuit. It will be seenthat the peaks become less and less prominentas the resistance is increased; thus, it can besaid that the selectivity of the circuit isthereby decreased. Selectivity in this casecan be defined as the ability of a circuit todiscriminate against frequencies adjacent tothe resonant frequency.Voltage Across Coiland Capacitor inSeries CircuitBecause the a.c. or r-fvoltage across a coil andcapacitor is proportionalto the reactance (for agiven current), the actual voltages across thecoil and across the capacitor may be manytimes greater than the terminal voltage of thecircuit. At resonance, the voltage across thecoil (or the capacitor) is Q times the appliedvoltage. Since the Q (or merit factor) of aseries circuit can be in the neighborhood of100 or more, the voltage across the capacitor,for example, may be high enough to causeflashover, even though the applied voltage isof a value considerably below that at whichthe capacitor is rated.Circuit Q- Sharp· An extremely importantness of Resonance property of a capacitor oran inductor is its factorof-merit,more generally called its Q. It is thisfactor, Q, which primarily determines thesharpness of resonance of a tuned circuit.This factor can be expressed as the ratio ofthe reactance to the resistance, as follows:2nfLQ=R,where R = total resistance.Skin Effect The actual resistance in a wireor an inductor can be far greaterthan the d-e value when the coil is used in aradio-frequency circuit; this is because thecurrent does not travel through the entirecross-section of the conductor, but has a tendencyto travel closer and closer to the surfaceof the wire as the frequency is increased. Thisis known as the skin effect.The actual current-carrying portion of thewire is decreased, as a result of the skineffect, so that the ratio of a-c to d-e resistanceof the wire, called the resistance ratio,is increased. The resistance ratio of wires tobe used at frequencies below about 500 kc.may be materially reduced through the use oflitz wire. Litz wire, of the type commonly usedto wind the coils of 455-kc. i-f transformers,may consist of 3 to 10 strands of insulatedwire, about No. 40 in size, with the individualstranas connected together only at the ends ofthe coils.Variation of Qwith FrequencyExamination of the equationfor determining Q might giverise to the thought that eventhough the resistance of an inductor increaseswith frequency, the inductive reactance doeslikewise, so that the Q might be a constant.Actually, however, it works out in practicethat the Q of an inductor will reach a relativelybroad maximum at some particular frequency.Hence, coils normally are designed in such amanner that the peak in their curve of Q withfrequency will occur at the normal operatingfrequency of the coil in the circuit for whichit is designed.The Q of a capacitor ordinarily is muchhigher than that of the best coil. Therefore,it usually is the merit of the coil that limitsthe overall Q of the circuit.At audio frequencies the core losses in aniron-core inductor greatly reduce the Q fromthe value that would be obtained simply bydividing the reactance by the resistance. Obviouslythe core losses also represent circuitresistance, just as though the loss occurredin the wire itself.ParallelResonanceIn radio circuits, parallel resonance(more correctly termed antiresonance)is more frequentlyencountered than series resonance; in fact, itis the basic foundation of receiver and transmittercircuit operation. A circuit is shown infigure 21.The "Tank" In this circuit, as contrasted withCircuit a circuit for series resonance, L(inductance) and C (capacitance)are connected in parallel, yet the combinationcan be considered to be in series with theremainder of the circuit. This combinationof L and C, in conjunction with R, the resistancewhich is principally included in L, issometimes called a tank circuit because it effectivelyfunctions as a storage tank when incorporatedin vacuum tube circuits.Contrasted with series resonance, there aretwo kinds of current which must be consideredin a parallel resonant circuit: ( 1) the line current,as read on the indicating meter M,, (2)the circulating current which flows within theparallel L-C-R portion of the circuit. Seefigure 21.At the resonant frequency, the line current(as read on the meter M,) will drop to a verylow value although the circulating current inthe L-C circuit may be quite large. It is interestingto note that the parallel resonant circuitacts in a distinctly opposite manner tothat of a series resonant circuit, in which the
56 Alternating Current Circuits <strong>THE</strong> R A 0 I 0r-----------~~~M~'----------~Figure 21PARALLEL-RESONANT CIRCUITThe inductance L ancl capacitance C comprisethe reactive elements of the parallel-resonant(anti-reso11ant) tank circuit, and the resistanceR indicates the sum of the r-f resistance of thecoil and capacitor, plus the resistance coupledinto the circuit from the external load. In mostcases the tuning capacitor has much lower r-fresistance than the coil and con therefore beignored in comparison with the coil resistanceand the coupled-In res/stance. The InstrumentM1 indicates the "line current" which keepsthe circuit in a state of oscillation- this currentIs the same as the fundamental componentof the plate current of a Class C amplifier whichmight be feeding the tonic circuit. The in ..strument M2 indicates the utanlc current" whichis equal to the line current multiplied by theoperating Q of the tonic circuit.current is at a maximum and the impedance isminimum at resonance. It is for this reasonthat in a parallel resonant circuit the principalconsideration is one of impedance rather thancurrent. It is also significant that the impedancecurve for parallel circuits is very nearlyidentical to that of the current curve for seriesresonance. The impedance at resonance isexpressed as:(2rrfL)"Z=-Rwhere Z = impedance in ohms,L = inductance in henrys,f = frequency in cycles,R = resistance in ohms.Or, impedance can be expressed as a functionof Q as:Z = 2rrfLQ,Lshowing thjt the impedance of a circuit isdirectly proportional to its effective Q atresonance.The curves illustrated in figure 20 can beapplied to parallel resonance. Reference to thecurve will show that the effect of adding resistanceto the circuit will result in both abroadening out and lowering of the peak of thecurve. Since the voltage of the circuit isdirectly proportional to the impedance, andsince it is this voltage that is applied to thegrid of the vacuum tube in a detector or amplifiercircuit, the impedance curve must havea sharp peak in order for the circuit to beselective. If the curve is broad-topped inshape, both the desired signal and the interferingsignals at close proximity to resonancewill give nearly equal voltages on the grid ofthe tube, and the circuit will then be nonselective:i.e., it will tune broadly.Effect of L/C Ratioin Parallel CircuitsIn order that the highestpossible voltage can bedeveloped across a parallelresonant circuit, the impedance of thiscircuit must be very high. The impedance willbe greater with conventional coils of limitedQ when the ratio of inductance-to-capacitanceis great, that is, when L is large as comparedwith C. When the resistance of the circuit isvery low, XL will equal Xc at maximum impedance.There are innumerable ratios of Land C that will have equal reactance, at agiven resonant frequency, exactly as in thecase in a series resonant circuit.In practice, where a certain value of inductanceis tuned by a variable capacitanceover a fairly wide range in frequency, theL/ C ratio will be small at the lowest frequencyend and large at the high-frequency end.The circuit, therefore, will have unequal gainand selectivity at the two ends of the band offrequencies which is being tuned. Increasingthe Q of the circuit (lowering the resistance)will obviously increase both the selectivityand gain.Cireulating Tank The Q of a circuit hasCurrent at Resonance a definite bearing onthe circulating tankcurrent at resonance. This tank current isvery nearly the value of the line current multipliedby the effective circuit Q. For example:an r-f line current of 0.050 amperes, with acircuit Q of 100, will give a circulating tankcurrent of approximately 5 amperes. From thisit can be seen that both the inductor and theconnecting wires in a circuit with a high Qmust be of very low resistance, particularly inthe case of high power transmitters, if heatlosses are to be held to a minimum.Because the voltage across the tank atresonance is determined by the Q, it is possibleto develop very high peak voltagesacross a high Q tank with but little line current.If a parallel resonant circuit is coupled to anothercircuit, such as an antennaEffect of Couplingon Impedanceoutput circuit, the impedance and the effectiveQ of the parallel circuit is decreased as thecoupling becomes closer. The effect of closer(tighter) coupling is the same as though an
<strong>HANDBOOK</strong> Circuit Impedance 57ffi D ill D ill 0 ffiOLOOSE COUPLING MEDIUM COUPLING CRITICAL COUPLING OVERCOUPLI NGHIGHQ MEDIUMQ LOWQ LOWQ,Ui_ 'liJl lJ~ leuf'~- 1'- f-- f@ © @Figure 22EFFECT OF COUPLING ON CIRCUIT IMPEDANCE AND Qactual resistance were added in series withthe parallel tank circuit. The resistance thuscoupled into the tank circuit can be consideredas being reflected from the output orload circuit to the driver circuit.The behavior of coupled circuits dependslargely upon the amount of coupling, as shownin figure 22. The coupled current in the secondarycircuit is small, varying with frequency,being maximum at the resonant frequency ofthe circuit. As the coupling is increasedbetween the two circuits, the secondary resonancecurve becomes broader and the resonantamplitude increases, until the reflectedresistance is equal to the primary resistance.This point is called the critical couplingpoint. With greater coupling, the secondaryresonance curve becomes broaderand developsdouble resonance humps, which become morepronounced and farther apart in frequency asthe coupling between the two circuits isincreased.Tonk CircuitFlywheel EffectWhen the plate circuit of aClass B or Class C operatedtube is connected to a parallelresonant circuit tuned to the same frequencyas the exciting voltage for the amplifier,the plate current serves to maintain thisL/C circuit in a state of oscillation.The plate currentis supplied in short pulseswhich do not begin to resemble a sine wave,even though the grid may be excited by a sinewavevoltage. These spurts of plate currentare converted into a sine wave in the platetank circuit by virtue of the "Q" or "flywheeleffect" of the tank.If a tank did not have some resistancelosses, it would, when given a "kick" with asingle pulse, continue to oscillate indefinite! y.With a moderate amount of resistance or "friction"in the circuit the tank will still haveinertia, and continue to oscillate with decreasingamplitude for a time after being givena "kick." With such a circuit, almost puresine-wave voltage will be developed acrossthe tank circuit even though power is suppliedto the tank in short pulses or spurts, so longas the spurts are evenly spaced with respectto time and have a frequency that is the sameas the resonant frequency of the tank.Another way to visualize the action of thetank is to recall that a resonant tank withmoderate Q will discriminate strongly againstharmonics of the resonant frequency. The distortedplate current pulse in a Class C amplifiercontains not only the fundamental frequency(that of the grid excitation voltage)but also higher harmonics. As the tank offerslow impedance to the harmonics and high impedanceto the fundamental (being resonant tothe latter), only the fundamental - a sinewavevoltage- appears across the tank circuitin substantial magnitude.Loaded andUnloaded QConfusion sometimes exists asto the relationship between theunloaded and the loaded Q of thetank circuit in the plate of an r-f power amplifier.In the normal case the loaded Q of thetank circuit is determined by such factors asthe operating conditions of the amplifier, band~width of the signal to be emitted, permissiblelevel of harmonic radiation, and such factors.The normal value of loaded Q for an r-f amplifierused for communications service is fromperhaps 6 to 20. The unloaded Q of the tankcircuit determines the efficiency of the outputcircuit and is determined by the losses in thetank coil, its leads and plugs and jacks if any,and by the losses in the tank capacitor whichordinarily are very low. The unloaded Q of agood quality large diameter tank coil in thehigh-frequency range may be as high as 500
58 Alternating Current Circuits<strong>THE</strong> R AD I 0to 800, and values greater than 300 are quitecommon.Tank Circuit Since the unloaded Q of a tankEfficiency circuit is determined by theminimum losses in the tank,while the loaded Q is determined by usefulloading of the tank circuit from the externalload in addition to the internal losses in thetank circuit, the relationship between the twoQ values determines the operating efficiencyof the tank circuit. Expressed in the form ofan equation, the loaded efficiency of a tankcircuit is:Tank efficiency = 1 _ Ql X 100Quwhere Qu = unloaded Q of the tank circuitQ 1 = loaded Q of the tank circuitAs an example, if the unloaded Q of thetank circuit for a class C r-f power amplifieris 400, and the external load is coupled to thetank circuit by an amount such that the loadedQ is 20, the tank circuit efficiency will be:eff. = (1 - 20/400) x 100, or (1 - 0.05) x 100,or 95 per cent. Hence 5 per cent of the poweroutput of the Class C amplifier will be lostas heat in the tank circuit and the remaining95 per cent will be delivered to the load.Figure 23COMPOSITE WAVE-FUNDAMENTALPLUS THIRD HARMONICFUNDAMENTAL PLUS JRD HARMONICFUNDAMENTAL PLUS 3RD AND5TH HARMONICS( E)Figure 24THIRD HARMONIC WAVE PLUSFIFTH HARMONIC3-3 Nonsinusoidal Wavesand TransientsPure sine waves, discussed previously, arebasic wave shapes. Waves of many differentand complex shape are used in electronics,particularly square waves, saw-tooth waves,and peaked waves.Wave Composition Any periodic wave {one thatrepeats itself in definitetime intervals) is composed of sine waves ofdifferent frequencies and amplitudes, addedtogether. The sine wave which has the samefrequency as the complex, periodic wave iscalled the fundamental. The frequencies higherthan the fundamental are called harmonics,and are always a whole number of times higherthan the fundamental. For example, the frequencytwice as high as the fundamental iscalled the second harmonic.The Square W·ave Figure 23 compares a squarewave with a sine wave (A)of the same frequency. If another sine wave(B) of smaller amplitude, but three times thefrequency of (A), called the third harmonic, isadded to (A), the resultant wave (C) morenearly approaches the desired square wave.Figure 25RESULTANT WAVE, COMPOSED OFFUNDAMENTAL, THIRD, FIFTH,AND SEVENTH HARMONICSThis resultant curve (figure 24) is added toa fifth harmonic curve {D), and the sides ofthe resulting curve (E) are steeper than before.This new curve is shown in figure 25 after a7th harmonic component has been added to it,making the sides of the composite wave evensteeper. Addition of more higher odd harmonicswill bring the resultant wave nearer and nearerto the desired square wave shape. The squarewave will be achieved if an infinite number ofodd harmonics are added to the original sinewave.
<strong>HANDBOOK</strong>Nonsinusoidal Waves59FUNDAMENTAL PLUS 3RDAND 5TH HARMONICSFigure 26COMPOSITION OF A SAWTOOTH WAVE@The Sawtooth Wave In the same fashion, asawtooth wave is made upof different sine waves (figure 26). The additionof all harmonics, odd and even, producesthe sawtooth wave form.The Peaked Wave Figure 27 shows the compositionof a peaked wave.Note how the addition of each sucessive harmonicmakes the peak of the resultant higherand the sides steeper.Other Waveforms The three preceeding examplesshow how a complexperiodic wave is composed of a fundamentalwave and different harmonics. The shape ofthe resultant wave depends upon the harmonicsthat are added, their relative amplitudes, andrelative phase relationships. In general, thesteeper the sides of the waveform, the moreharmonics it contains.AC Transient Circuits If an a-c voltage is substitutedfor the d-e inputvoltage in the RC Transient circuits discussedin Chapter 2, the same principles maybe applied in the analysi~ of the transientbehavior. An RC coupling circuit is designedto have a long time constant with respect tothe lowest frequency it must pass. Such acircuit is shown in figure 28. If a nonsinusoidalvoltage is to be passed unchangedthrough the coupling circuit, the time constantFigure 27COMPOSITION OF A PEAKED WAVEmust be long with respect to the period of thelowest frequency contained in the voltagewave.RC Dlfferentiatorand IntegratorAn RC voltage divider thatis designed to distort the inputwaveform is known as adifferentiator or integrator, depending uponthe locations of the output taps. The outputfrom a differentiator is taken across the resistance,while the output from an integratoris taken across the capacitor. Such circuitswill change the shape of any complex a-cwaveform that is impressed upon them. Thisdistortion is a function of the value of thetime constant of the circuit as compared tothe period of the waveform. Neither a differentiatornor an integrator can change the
60 Alternating Current Circuits <strong>THE</strong> R A 0 I 0IOOV. ~ E=O.I.UFIOOC' C.P.S. L--------.1~--_~M OUTPUTVOLTAGER xC= 50000 .l.ISECONOSPERIOD OF e = 1000 ..USECONOSe = IOOV.(••--)1000 C.P.S.C=o,I~F} ec= INTEGRATOR OUTPUT---- -OU:~:TVWAVE FORML-----4-R-=-10-K-{J} eR = OIFFERENTIATOR OUTPUTFigure 28R-C COUPLING CIRCUIT WITHLONG TIME CONSTANTLEIe 0 OF GENERATOR-- -IOOV.E=10ov.(PEAK)1000'\..eo~ - +IZ'V.- +75V.OUTPUT OF;~F~I::.:;IATOR (eR)© ®Figure 29R-C DIFFERENTIATOR ANDINTEGRATOR ACTION ONA SINE WAVEshape of a pure sine wave, they will merelyshift the phase of the wave (figure 29). Thedifferentiator output is a sine wave leadingthe input wave, and the integrator output is asine wave which lags the input wave. The sumof the two outputs at any instant equals theinstantaneous input voltage.Square Wave Input If a square wave voltage isimpressed on the circuit offigure 30, a square wave voltage output maybe obtained across the integrating capacitorif the time constant of the circuit allows thecapacitor to become fully charged. In thisparticular case, the capacitor never fullycharges, and as a result the output of theintegrator has a smaller amplitude than theinput. The differentiator output has a maximumvalue greater than the input amplitude, sincethe voltage left on the capacitor from theprevious half wave will add to the input voltage.Such a circuit, when used as a differentiator,is often called a peaker. Peaks oftwice the input amplitude may be produced.Figure 30R·C DIFFERENTIATOR ANDINTEGRATOR ACTION ONA SQUARE WAVESawtooth Wave Input If a back-to·back sawtoothvoltage is appliedto an RC circuit having a time constant onesixththe period of the input voltage, the resultis shown in figure 31. The capacitorvoltage will closely follow the input voltage,if the time constant is short, and the integra·tor output closely resembles the input. Theamplitude is slightly reduced and there is aslight phase lag. Since the voltage across thecapacitor is increasing at a constant rate, thecharging and discharging current is constant.The output voltage of the differentiator, there·fore, is constant during each half of the saw·tooth input.Miscellaneous Various voltage waveformsInputs other than those representedhere may be applied to shortRC circuits for the purpose of producingacross the resistor an output voltage with anamplitude proportional to the rate of change ofthe input signal. The shorter the RC time con·stant is made with respect to the period of theinput wave, the more nearly the voltage across
<strong>HANDBOOK</strong>Transformers61e = 1oo v.(PEAK)1000 c.P.s."'"®------- +100eouT-- -100®e'" ~Yr---t--v~/-T-e
62 Alternating Current Circuits <strong>THE</strong> R AD I 0® © @Figure 33Amplifier Jeflclent In low frequency response will distort square wave applied to the Input circuit, asshown. A 60-cyc/e square wave may be used.A: Drop In gain ot /ow frequencies8: Leading phase shift ot /ow frequenciesC: Lagging phase shift ot /ow frequenciesD: Accentuated low frequency gainTypes afTransformersTransformers are used in alternating-currentcircuits to transferpower at one voltage and impedanceto another circuit at another voltageand impedance. There are three main classificationsof transformers: those made for usein power-frequency circuits, those made foraudio-frequency applications, and those madefor radio frequencies.The Transformation In a perfect transformer allRatio the magnetic flux linesproduced by the primarywinding link every turn of the secondary winding.For such a transformer, the ratio of theprimary and secondary voltages is exactlythe same as the ratio of the number of turnsin the two windings:where NpNsEpNpNsEpEs= number of turns in the primarywinding= number of turns in the secondarywinding= voltage across the primary windingE s = voltage across the secondarywindingIn practice, the transformation ratio of atransformer is somewhat less than the turnsratio, since unity coupling does not existbetween the primary and secondary windings.Ampere Turns (NI) The current that flows inthe secondary winding as aresult of the induced voltage must produce aflux which exactly equals the primary flux.The magnetizing force of a coil is expressedas the product of the number of turns in thecoil times the current flowing in it:Npxlp=Nsxls,'where I p = primary currentIs = secondary currentNp Isor--=-Ns lpIt can be seen from this expression thatwhen the voltage is stepped up, the currentis stepped down, and vice-versa.Leakage Reactance Since unity coupling doesnot exist in a practicalFigure 34Output waveshape of amp/Iller having deficiencyIn .high-frequency respon ... Tested with 10-kc.square wave.Figure 35Output waveshape of amplifier having limitedlow-frequency and high-frequency response.Tested with l-lc:c. square wave.
<strong>HANDBOOK</strong>Electric Filters63INPUTVOLTAGEOUTPUTVOLTAGEFigure 36IMPEDANCE-MATCHING TRANSFORMERTire reflected Impedance ZP varies directly Inproportion to the secondary /ooJ Z L, anJdirectly In proportion to tire square ol tireprimary-to-secottclary turns ratio.transformer, part of the flux passing from theprimary circuit to the secondary circuit fol·lows a magnetic circuit acted upon by theprimary only. The same is true of the secondaryflux. These leakage fluxes cause leakagereactance in the transformer, and tend tocause the transformer to have poor voltageregulation. To reduce such leakage reactance,the primary and secondary windings shouldbe in close proximity to each other. The moreexpensive transformers have interleaved wind·ings to reduce inherent leakage reactance.Impedance In the ideal transformer, theTransformation impedance of the secondaryload is reflected back into theprimary winding in the following relationship:where ZpNZsZp ~ N 2 Zs, or N ~ .jZp/Zs=reflected primary impedance~ turns ratio of transformer~ impedance of secondary loadThus any specific load connected to thesecondary terminals of the transformer willbe transformed to a different specific valueappearing across the primary terminals of thetransformer. By the proper choice of turnsratio, any reasonable value of secondary loadimpedance may be "reflected" into the primarywinding of the transformer to produce thedesired transformer primary impedance. Thephase angle of the primary "reflected" impedancewill be the same as the phase angleof the load impedance. A capacitive second·ary load will be presented to the transformersource as a capaciry, a resistive load willpresent a resistive "reflection" to the primarysource. Thus the primary source "sees" atransformer load entirely dependent upon thesecondary load impedance and the turns ratioof the transformer (figure 36).The Auto The type of transformer in figureTransformer 37, when wound with heavy wireover an iron core, is a commondevice in primary power circuits for the purposeof increasing or decreasing the line volt·Figure 37<strong>THE</strong> AUTO· TRANSFORMERSchematic diagram ol an auto-transformershowing tire met/rod ol connecting it to tire //neand to tire load. Wlten only a small amount olstep up or step down Is required, the autotransformermay be much smaller physicallythan would be a transformer with a separatesecondary winding. Continuously variableauto-translormers (Variac and Powerstof} orewidely used commercially.age. In effect, it is merely ~ contin~ous windingwith taps taken at vartous Ptnts al~ngthe winding, the input voltage betng app~tedto the bottom and also to one tap on the wmding.If the output is taken from this sametap, the voltage ratio will be 1-to-1; i.e., theinput voltage will be the same as the outputvoltage. On the other hand, if the outpu~ tapis moved down toward the common termtnal,there will be a step-down in the turns ratiowith a consequent step-down in voltage. Theinitial setting of the middle input tap is chosenso that the number of turns will have sufficientreactance to keep the no-load primarycurrent at a reasonably low value.3-5 Electric FiltersThere are many applications where it isdesirable to pass a d-e component withoutpassing a superimposed a-c component, or toL-SECTIONSELEMENTARY FILTER SECTIONST-NETWORK~--F ~---YL _L_ITliFigure 38Complex 1//ters may be made up lrom these basic11/ter sect/on•.
64 Alternating Current Circuits <strong>THE</strong> R AD I 0LOW-PASS SHUNT-DERIVED FILTER(sERIES-ARM RESONATED)l..Lr2I2HIGH-PASS SERIES-DERIVED FILTER(S!iUNT-ARM RESONATED)C1r~z0~z"~z0~z"~"'~
<strong>HANDBOOK</strong> Filter Design 65LOWPAS5R =LOAD RESISTANCEf'2 =CUT-OFF FREQUENCYfeo= FREQUENCY OF VERYHIGH ATTENUATIONLK=-RTTf'2CK= -- 1 -TTf'2RTT-SECTION FILTER DESIGNCONSTANT KM-0.6TERMINATING HALF-SECTIONS~o 2C2I rz-C2 oL1=o.6LK=MLKc, = 0.207 CK= '4~ 2 CKC2=o.e CK=MCKlL_ I I ,,~ ~ Lf-f'a:FREQUENCYFREQUENCYR= LOAD RESISTANCEf1 =CUT-OFF FREQUENCYfoo =FREQUENCY OF VERYHIGH ATTENUATIONHIGH PASSC1 = CKL2 = LKil~!
66 Alternating Current CircuitsThe chart of figure 40 gives design dataand procedure on the pi-section type of filter.M-derived sections with an M of 0.6 will befound to be most satisfactory as the inputsection (or half-section) of the usual filtersince the input impedance of such a sectionis most constant over the pass band of thefilter section.Simple filters may use either L, T, or 11 sections.Since the 11 section is ~he more commonly"used type figure 40 gives design dataand characteristics for this type of filter.A PUSH-PULL 250-TH AMPLIFIER WITH TVI SHIELD REMOVEDUse of harmonic filters in power leads and antenna circuit reduces radiation of TVI-producing harmonicsof typical push-pull amplifier. Shielded enclosure completes harmonic reduction measures.
CHAPTER FOURVacuum Tube PrinciplesIn the previous chapters we have seen themanner in which an electric current flowsthrough a metallic conductor as a result of anelectron drift. This drift, which takes placewhen there is a difference in potential betweenthe ends of the metallic conductor, is in additionto the normal random electron motionbetween the molecules of the conductor.The electron may be considered as a minutenegatively charged particle, having a mass of9 X 10- 28 gram, and a charge of 1. 59 X 10- 19coulomb. Electrons are always identical,regardless of the source from which they areobtained.An electric current can be caused to flowthrough other media than a metallic conductor.One such medium is an ionized solution, suchas the sulfuric acid electrolyte in a storagebattery. This type of current flow is calledelectrolytic conduction. Further, it was shownat about the turn of the century that an electriccurrent can be carried by a stream of freeelectrons in an evacuated chamber. The flowof a current in such a manner is said to takeplace by electronic conduction. The study ofelectron tubes (also called vacuum tubes, orvalves) is actually the study of the control anduse of electronic currents within an evacuatedor partially evacuated chamber.Since the current flow in an electron tubetakes place in an evacuated chamber, theremust be located within the enclosure both asource of electrons and a collector for theelectrons which have been emitted. The electronsource is called the cathode, and theelectron collector is usually called the anode.Some external source of energy must be appliedto the cathode in order to impart sufficientvelocity to the electrons within thecathode material to enable them to overcomethe surface forces and thus escape into thesurrounding medium. In the usual types of67electron tubes the cathode energy is appliedin the form of heat; electron emission from aheated cathode is called thermionic emission.In another common type of electron tube, thephotoelectric cell, energy in the form of lightis applied to the cathode to cause photoelectricemission.4-1 Thermionic EmissionElectron Emission of electrons from theEmission cathode of a thermionic electrontube takes place when the cathodeof the tube is heated to a temperature sufficientlyhigh that the free electrons in theemitter have sufficient velocity to overcomethe restraining forces at the surface of thematerial. These surface forces vary greatlywith different materials. Hence different typesof cathodes must be raised to different temperaturesto obtain adequate quantities of electronemission. The several types of emittersfound in common types of transmitting andreceiving tubes will be described in the followingparagraphs.Cathode Types The emitters or cathodes asused in present-day thermionicelectron tubes may be classified intotwo groups: the directly-heated or filamenttype and the indirectly-heated or heater-cathodetype. Directly-heated emitters may be furthersubdivided into three important groups, allof which are commonly used in modern vacuumtubes. These classifications are: the puretungstenfilament, the thoriated-tungstenfilament, and the oxide-coated filament.The Pure Tung·sten FilamentPure tungsten wire was usedas the filament in nearly allthe earlier transmitting and
6 8 Vacuum Tube Principles <strong>THE</strong> R AD I 0Figure 1ELECTRON TUBE TYPESThe new General Electric ceramic triode(6BY4)is shown alongside a conventionalminiature tube (6265) and an octal-based receiving tube (25L6). The ceramictube is designed for rugged service and features extremely low lead inductance.receiving tubes. However, the thermionic efficiencyof tungsten wire as an emitter (thenumber of milliamperes emission per watt offilament heating power) is quite low, the filamentsbecome fragile after use, their life israther short, and they are susceptible to burnoutat any time. Pure tungsten filaments mustbe run at bright white heat (about 2500° Kelvin).For these reasons, tungsten filamentshave been replaced in all applications whereanother type of filament could be used. Theyare, however, still universally employed inlarge water-cooled tubes and in certain large,high-power air-cooled triodes where anotherfilament type would be unsuitable. Tungstenfilaments are the most satisfactory for highpower,high-voltage tubes where the emitteris subjected to positive ion bombardmentcaused by the residual gas content of thetubes. Tungsten is not adversely affected bysuch bombardment.The Thoriated- In the course of experiTungsten Filament ments made upon tungstenemitters, it was found thatfilaments made from tungsten having a smallamount of thoria (thorium oxide) as an impurityhad much greater emission than thosemade from the pure metal. Subsequent developmenthas resulted in the highly efficient carburizedthoriated-tungsten filament as used invirtually all medium-power transmitting tubestoday.Thoriated-tungsten emitters consist of atungsten wire containing from 1% to 2% thoria.The activation process varies between differentmanufacturers of vacuum tubes, butit is essentially as follows: (l) the tube isevacuated; (2) the filament is burned for ashort period at about 2800° Kelvin to cleanthe surface and reduce some of the thoriawithin the filament to metallic thorium; (3)the filament is burned for a longer period atabout 2100° Kelvin to form a layer of thoriumon the surface of the tungsten; ( 4) thetemperature is reduced to about 1600° Kelvinand some pure hydrocarbon gas is admittedto form a layer of tungsten carbide on thesurface of the tungsten. This layer of tungstencarbide reduces the rate of thorium evaporationfrom the surface at the normal operatingtemperature of the filament and thus increasesthe operating life of the vacuum tube. Thoriumevaporation from the surface is a naturalconsequence of the operation of the thoriatedtungstenfilament. The carburized layer on thetungsten wire plays another role in acting asa reducing agent to produce new thorium fromthe thoria to replace that lost by evaporation.This new thorium continually diffuses tothe surface during the normal operation ofthe filament. The last process, (5), in theactivation of athoriated tungsten filament consistsof re-evacuating the envelope and thenburning or ageing the new filament for a considerableperiod of time at the normal operatingtemperature of approximately 1900° K.One thing to remember about any type offilament, particularly the thoriated type, isthat the emitter deteriorates practically asfast when "standing by" (no plate current) asit does with any normal amount of emissionload. Also, a thoriated filament may be eithertemporarily or permanently damaged by aheavy overload which may strip the surfacelayer of thorium from the filament.ReactivatingThoriated-TungstenFilamentsThoriated-tungsten filaments(and only thoriatedtungstenfilaments) whichhave lost emission asa result of insufficient filament voltage, asevere temporary overload, a less severe extendedoverload, or even normal operation
<strong>HANDBOOK</strong> Types of Emitters 69Figure 2V-H-F and U-H-F TUBE TYPESThe tube to the left In this photograph Is a 955 "acorn" triode. The 6F4 acorn triode Is very similar Inappearance to the 955 but has two leads brought out each for the grid and for the plate connection. Thesecond tube Is a 446A "lighthouse" triode. The 2C40, 2C43, and 2C44 ore more recent examples of thesame type tube and are essentially the some In external appearance. The third tube from the left Is a2C39 ''oil can•• tube. This tube type is essentially the Inverse of the lighthouse variety since the cathodeand heater connections come out the small end and the plate Is the large finned radiator on the large encl.The use of the finned plate radiator makes the oilcan tube capable of approximately 10 times as muchplate dissipation as the lighthouse type. The tube to the right is the 4Xl50A beam tetroc/e. This tube, acomparatively recent release, Is capable ol somewhat greater power output than any of the other tubetypes shown, one/ is ratec/ lor lu/1 output at 500 Me. one/ at rec/ucec/ output at frequencies greater than1000 Me.may quire frequently be reactivated to theiroriginal characteristics by a process similarto that of the original activation. However,only filaments which have not approached tooclose to the end of their useful life may besuccessfully reactivated.The actual process of reactivation is relativelysimple. The tube which has gone"flat" is placed in a socket to which only thetwo filament wires have been connected. Thefilament is then "flashed" for about 20 to 40seconds at about I Y, times normal rated voltage.The filament will become extremely brightduring this time and, if there is still somethoria left in the tungsten and if the tube didnot originally fail as a result of an air leak,some of this thoria will be reduced to metallicthorium. The filament is then burned at 15 to25 per cent overvoltage for from 30 minutes to3 to 4 hours to bring this new thorium to thesurface.The tube should then be tested to see if itshows signs of renewed life. If it does, but isstill weak, the burning process should be continuedat about 10 to 15 per cent overvoltagefor a few more hours. This should bring itback almost to normal. If the tube checks stillvery low after the first attempt at reactivation,the complete process can be repeated as alast effort.The OxideCoated FilamentThe most efficient of allmodern filaments is theoxide-coated type which consistsof a mixture of barium and strontiumoxides coated upon a nickel alloy wire orstrip. This type of filament operates at a dullredto orange-red temperature (1050" to 1170°K) at which temperature it will emit largequanuues of electrons. The oxide-coatedfilament is somewhat more efficient than thethoriated-tungsten type in small sizes and itis considerably less expensive to manufacture.For this reason all receiving tubes and quitea number of the low-powered transmittingtubes use the oxide-coated filament. Anotheradvantage of the oxide-coated emitter is itsextremely long life- the average rube can beexpected to run from 3000 to 5000 hours, andwhen loaded very lightly, tubes of this typehave been known to give 50,000 hours of lifebefore their characteristics changed to anygreat extent.Oxide filaments are unsatisfactory for useat high continuous plate voltages because: (I)their activity is seriously impaired by thehigh temperature necessary to de-gas the high·voltage tubes and, (2) the positive ion born·bardment which takes place even in the bestevacuated high-voltage tube causes destruc·tion of the oxide layer on the surface of thefilament.Oxide-coated emitters have been found capableof emitting an enormously large currentpulse with a high applied voltage for a veryshort period of time without damage. Thischaracteristic has proved to be of great value
70 Vacuum Tube Principles<strong>THE</strong> R AD I 0Figure 3CUT-AWAY DRAWING OF A 6C4 TRIODEin radar work. For example, the relativelysmall cathode in a microwave magnetron maybe called upon to deliver 25 to 50 amperes atan applied voltage of perhaps 25,000 volts fora period in the order of one microsecond.After this large current pulse has been passed,plate voltage normally will be removed for1000 microseconds or more so that the cathodesurface may be restored in time for the nextpulse of current. If the cathode were to besubjected to a continuous current drain of thismagnitude, it would be destroyed in an exceedinglyshort period of time.The activation of oxide-coated filamentsalso varies with tube manufacturers but consistsessentially in heating the wire which hasbeen coated with a mixture of barium andstrontium carbonates to a temperature of about1500° Kelvin for a time and then applying apotential of 100 to 200 volts through a protectiveresistor to limit the emission current.This process reduces the carbonates to oxidesthermally, cleans the filament surface offoreign materials, and activates the cathodesurface.Reactivation of oxide-coated filaments isnot possible since there is always more thansufficient reduction of the oxides and diffusionof the metals to the surface of the filament tomeet the emission needs of the cathode.The HeaterCathodeThe heater type cathode was developedas a result of the requirementfor a type of emitterwhich could be operated from alternating currentand yet would not introduce a-c ripplemodulation even when used in low-level stages.It consists essentially of a small nickel-alloycylinder with a coating of strontium and bariumoxides on its surface similar to the coatingused on the oxide-coated filament. Insidethe cylinder is an insulated heater elementconsisting usually of a double spiral of tungstenwire. The heater may operate on any volt-age from 2 to 117 volts, although 6. 3 is themost common value. The heater is operatedat quite a high temperature so that the cathodeitself usually may be brought to operatingtemperature in a matter of 15 to 30 seconds.Heat-coupling between the heater and thecathode is mainly by radiation, although thereis some thermal conduction through the insulatingcoating on the heater wire, as thiscoating is also in contact with the cathodethimble.Indirectly heated cathodes are employed inall a-c operated tubes which are designed tooperate at a low level either for r-f or a-f use.However, some receiver power tubes useheater cathodes (6L6, 6V6, 6F6, and 6K6-GT)as do some of the low-power transmitter tubes(802, 807, 815, 3E29, 2E26, 5763, etc.). Heatercathodes are employed almost exclusivelywhen a number of tubes are to be operated inseries as in an a.c.-d.c. receiver. A heatercathode is often called a uni-potential cathodebecause there is no voltage drop along itslength as there is in the directly-heated orfilament cathode.The BombardmentCathodeA special bombardment cathodeis employed in many ofthe new high powered televisiontransmitting klystrons (Eimac 3K 20,000LA). The cathode takes the form of a tantalumdiode, heated to operating temperature by thebombardment of electrons from a directlyheated filament. The cathode operates at apositive potential of 2000 volts with respectto the filament, and a d-e bombardment currentof 0.66 amperes flows between filamentand cathode. The filament is designed tooperate under space-charge limited conditions.Cathode temperature is varied by changing thebombardment potential between the filamentand the cathode.The EmissionEquationThe emission of electrons froma heated cathode is quite similarto the evaporation of moleculesfrom the surface of a liquid. The moleculeswhich leave the surface are those havingsufficient kinetic (heat) energy to overcomethe forces at the surface of the liquid. As thetemperature of the liquid is raised, the averagevelocity of the molecules is increased,and a greater number of molecules will acquiresufficient energy to be evaporated. The evaporationof electrons from the surface of a thermionicemitter is similarly a function of averageelectron velocity, and hence is a functionof the temperature of the emitter.Electron emission per unit area of emittingsurface is a function of the temperature Tin degrees Kelvin, the work function of theemitting surface b (which is a measure of the
<strong>HANDBOOK</strong>Thermionic Emission71BOO6CB6TYPE 6W4-GTEF= 6.3VOLTS_tiL~lL =-=---~JHEATERFigure 4CUT-AWAY DRAWING OF A 6CB6 PENTODEfl)wa:wa..::;600< 400.J.J~wf-- 200
72Vacuum Tube Principles<strong>THE</strong> R AD I 0OXIDE COATEDfzwtttt::>uwf
<strong>HANDBOOK</strong>Triode Characteristics 731/~~ L _L I~ 4~ri-t.H-7-tl/+r~+i~++.r~~~~ 1/v100 200 300 400PLATE VOLTS (Ep)Figure 8NEGATIVE-GRID CHARACTERISTICS OpVS. Ep CURVES) OF A TYPICALTRIODEAverage plate characteristics of this typeare most commonly usee/ in determining theClass A operating characteristics of atriode amp/ ifier stage.tive charge on this grid will effectively repelthe negatively charged electrons (like chargesrepel; unlike charges attract) back into thespace charge surrounding the cathode. Hence,the number of electrons which are able to passthrough the grid mesh and reach the plate willbe reduced, and the plate current will be reducedaccordingly. If the charge on the gridis made sufficiently negative, all the electronsleaving the cathode will be repelled back toit and the plate current will be reduced to zero.Any d-e voltage placed upon a grid is calleda bias (especially so when speaking of a controlgrid). The smallest negative voltage whichwill cause cutoff of plate current at a particularplate voltage is called the value of cutoffbias (figure 7).Amplification The amount of plate current in aFactor triode is a result of the net fieldat the cathode from interactionbetween the field caused by the grid bias andthat caused by the plate voltage. Hence, bothgrid bias and plate voltage affect the platecurrent. In all normal tubes a small change ingrid bias has a considerably greater effectthan a similar change in plate voltage. Theratio between the change in grid bias and thechange in plate current which will cause thesame small change in plate current is calledthe amplification factor or p. of the electrontube. Expressed as an equation:~Epf.l=-L\Eg500with ip constant (L\ represents a small increment).The p. can be determined experimentally bymaking a small change in grid bias, thusslightly changing the plate current. The platecurrent is then returned to the original valueby making a change in the plate voltage. Theratio of the change in plate voltage to thechange in grid voltage is the p. of the tubeunder the operating conditions chosen for thetest.Current Flowin a TriodeIn a diode it was shown thatthe electrostatic field at thecathode was proportional tothe plate potential, Ep, and that the totalcathode current was proportional to the threehalvespower of the plate voltage. Similiarly,in a triode it can be shown that the field atthe cathode space charge is proportional tothe equivalent voltage (Eg + Ep/p.), wherethe amplification factor, p., actually representsthe relative effectiveness of grid potential andplate potential in producing a field at thecathode.It would then be expected that the cathodecurrent in a triode would be proportional tothe three-halves power of (Eg + Ep/f.£). Thecathode current of a triode can be representedwith fair accuracy by the expression:E , 12Cathode current = K ( E g +-;-)where K is a constant determined by elementgeometry within the triode.Plate Resistance The plate resistance of avacuum tube is the ratio of achange in plate voltage to the change in platecurrent which the change in plate voltageproduces. To be accurate, the changes shouldbe very small with respect to the operatingvalues. Expressed as an equation:L\EpRp=-MpEg = constant, L\ = smallincrementThe plate resistance can also be determinedby the experiment mentioned above. By notingthe change in plate current as it occurs whenthe plate voltage is changed (grid voltageheld constant), and by dividing the latter bythe former, the plate resistance can be deter·mined. Plate resistance is expressed in Ohms.Transconductance The mutual conductance,also referred to as transconductance,is the ratio of a change in theplate current to the change in grid voltagewhich brought about the plate current change,the plate voltage being held constant. Expressedas an equation:
74 Vacuum Tube Principles<strong>THE</strong> R AD I 0(lp)...::l;..:z0030 00w~ 200::>
<strong>HANDBOOK</strong>Triode Load Line75!P(MA.) E_l'__0 3005 25010 20015 15020 10025 5030Figure 12TRIODE TUBE CONNECTED FOR DETER·MINATION OF PLATE CIRCUIT LOADLINE, AND OPERATING PARAMETERSOF <strong>THE</strong> CIRCUITFigure IIThe static loacl line for a typical triodetube with a plate loacl resistance of 10,000ohms.garding dynamic load lines, the reader isreferred to theRadiotron Designer's Handbook,4th edition, distributed by Radio Corporationof America.Application of Tube As an example of the ap·Characteristics plication of tube characteristics,the constantsof the triode amplifier circuit shown in figure12 may be considered. The plate supply isEP300 volts, and the plate load is 8,000 ohms.If the tube is considered to be an open circuitno plate current will flow, and there is novoltage drop across the plate load resistor,RL. The plate voltage on the tube is therefore300 volts. If, on the other hand, the tube isconsidered to be a short circuit, maximumpossible plate current flows and the full 300volt drop appears across RL. The plate voltageis zero, and the plate current is 300/8,000,or 37.5 milliamperes. These two extreme conelitions define the load line on the IP vs. Epcharacteristic curve, figure 13.For this application the grid of the tube isreturned to a steady biasing voltage of - 4volts. The steady or quiescent operation of thetube is determined by the intersection of theload line with the -4 volt curve at point Q.By projection from point Q through the plate31.5Figure 13APPLICATION OF lp VS. EpCHARACTERISTICS OFVACUUM TUBE1;;ww~ ~t--------184 VOLT PLATE SWINGPLATE VOLTS(EP)
76 Vacuum Tube PrinciplesT~current axis it is found that the value of platecurrent with no signal applied to the grid is12.75 milliamperes. By projection from pointQ through the plate voltage axis it is foundthat the quiescent plate vltage is 198 volts.This leaves a drop of 102 volts across RLwhich is borne out by the relation 0.01275 x8,000 = 102 volts.An alternating voltage of 4 volts maximumswing about the normal bias value of -4 voltsis applied now to the grid of the triode amplifier.This signal swings the grid in a positivedirection to 0 volts, and in a negative directionto -8 volts, and establishes the operatingregion of the tube along the load line betweenpoints A and B. Thus the maxima and minimaof the plate voltage and plate current areestablished. By projection from points A andB through the plate current axis the maximuminstantaneous plate current is found to be18.25 milliamperes and the minimum is 7.5milliamperes. By projections from points A andB through the plate voltage axis the minimuminstantaneous plate voltage swing is found tobe 154 volts and the maximum is 240 volts.By this graphical application of the lp vs.Ep characteristic of the 6SN7 triode the operationof the circuit illustrated in figure 12 be-r---1:;f:::CG-PII;t:::cc.-1\I L __ _<strong>THE</strong>----,I II-;f::: CP-KIII-- __ jR AD I 0Figure 15SCHEMATIC REPRESENTATIONOFINTERELECTRODECAPACITANCET-comes apparent. A voltage variation of 8 volts(peak-to-peak) on the grid produces a variationof 84 volts at the plate.EPT-Figure 14POLARITY REVERSAL BETWEEN GRIDAND PLATE VOLTAGESPolarity Inversion When the signal voltage appliedto the grid has itsmaximum positive instantaneous value theplate current is also maximum. Reference tofigure 12 shows that this maximum plate currentflows through the plate load resistor RL,producing a maximum voltage drop across it.The lower end of RL is connected to the platesupply, and is therefore held at a constantpotential of 300 volts. With maximum voltagedrop across the load resistor, the upper end ofRL is at a minimum instantaneous voltage.The plate of the tube is connected to this endof RL and is therefore at the same minimuminstantaneous potential.This polarity reversal between instantaneousgrid and plate voltages is further clarified bya consideration of Kirchhoff's law as it appliesto series resistance. The sum of the IRdrops around the plate circuit must at all timesequal the supply voltage of 300 volts. Thuswhen the instantaneous voltage drop acrossRL is maximum, the voltage drop across thetube is minimum, and their sum must equal300 volts. The variations of grid voltage,platecurrent and plate voltage about their steadystate values is illustrated in figure 14.lnterelectrodeCapacitanceCapacitance always exists betweenany two pieces of metalseparated by a dielectric. Theexact amount of capacitance depends upon thesize of the metal pieces, the dielectric betweenthem, and the type of dielectric. Theelectrodes of a vacuum tube have a similarcharacteristic known as the interelectrodecapacitance, illustrated in figure 15. Thesedirect capacities in a triode are: grid-tocathodecapacitance, grid-to-plate capacitance,and plate-to-cathode capacitance. The interelectrodecapacitance, though very small, hasa coupling effect, and often can cause unbalancein a particular circuit. At very high
<strong>HANDBOOK</strong>Tetrodes andPentodes7710TYPE 24-Aesc=gov.eG-O
78 Vacuum Tube Principles<strong>THE</strong>R AD I 0GRIOA THODECoRIOC ... THODEREMOTE CUT-OFFGRIDFigure 18SHARP CUT-OFFGRIDREMOTE CUTOFF GRID STRUCTUREGRID VOLTSFigure 19ACTION OF A REMOTE CUTOFFGRID STRUCTUREtive wtth respect to the mm1mum plate voltage.The secondary electrons that would travelto the screen if there were no suppressor arediverted back to the plate. The plate currentis, therefore, not reduced and the amplificationpossibilities are increased (figure 17).Pentodes for audio applications are designedso that the suppressor increases thelimits to which the plate voltage may swing;therefore the consequent power output andgain can be very great. Pentodes for radiofr~:quencyservice function in such a mannerthat the suppressor allows high voltage gain,at the same time permitting fairly high gainat low plate voltage. This holds true even ifthe plate voltage is the same or slightly lowerthan the screen voltage.Remote CutoffTubesRemote cutoff tubes (variablemu) are screen grid tubes inwhich the control grid structurehas been physically modified so as tocause the plate current of the tube to drop offgradually, rather than to have a well definedcutoff point {figure 18). A non-uniform controlgrid structure is used, so that the amplificationfactor is different for different parts of thecontrol grid.Remote cutoff tubes are used in circuitswhere it is desired to control the amplificationby varying the control grid bias. The characteristiccurve of an ordinary screen grid tubehas considerable curvature near the plate currentcutoff point, while the curve of a remotecutoff tube is much more linear (figure 19).The remote cutoff tube minimizes crosstalkinterference that would otherwise beproduced. Examples of remote cutoff tubesare: 6BD6, 6K7, 6SG7 and 6SK7.Beam PowerTubesA beam power tube makes useof another method for suppressingsecondary emission. In this tubethere are four electrodes: a cathode, a grid, ascreen, and a plate, so spaced and placed thatsecondary emission from the plate is suppressedwithout actual power loss. Becauseof the manner in which the electrodes arespaced, the electrons which travel to theplate are slowed down when the plate voltageis low, almost to zero velocity in a certainregion between screen and plate. For thisreason the electrons form a stationary cloud,or space charge. The effect of this spacecharge is to repel secondary electrons emittedfrom the plate and thus cause them to returnto the plate. In this way, secondary emissionis suppressed.Another feature of the beam power tube isthe low current drawn by the screen. Thescreen and the grid are spiral wires wound sothat each turn in the screen is shaded fromthe cathode by a grid turn. This alignment ofthe screen and the grid causes the electronsto travel in sheets between the turns of thescreen so that very few of them strike thescreen itself. This formation of the electronstream into sheets or beams increases thecharge density in the screen-plate region andassists in the creation of the space charge inthis region.Because of the effective suppressor actionprovided by the space charge, and because ofthe low current drawn by the screen, the beampower tube has the advantages of high poweroutput, high power-sensitivity, and high efficiency.The 6L6 is such a beam power tube,designed for use in the power amplifier stagesof receivers and speech amplifiers or modulators.Larger tubes employing the beam-powerprinciple are being made by various manufacturersfor use in the radio-frequency stagesof transmitters. These tubes feature extremelyhigh power-sensitivity (a very small amountof driving power is required for a large output),good plate efficiency, and low grid-toplatecapacitance. Examples of these tubesare 813, 4-250A, 4X150A, etc.Grid-Screen The grid- screen mu factor (p. 5g)Mu Factor is analogous to the amplificationfactor in a triode, except thatthe screen of a pentode or tetrode is sub-
<strong>HANDBOOK</strong>Mixer and Converter Tubes 79stituted for the plate of a triode. /Lsg denotesthe ratio of a change in grid voltage to achange in screen voltage, each of which willproduce the same change in screen current.Expressed as an equation:Is g = constant, 11 = smallincrementThe grid-screen mu factor is important indetermining the operating bias of a tetrodeor pentode tube. The relationship between control-gridpotential and screen potential determinesthe plate current of the tube as well asthe screen current since the plate current isessentially independent of the plate voltagein tubes of this type. In other words, whenthe tube is operated at cutoff bias as determinedby the screen voltage and the gridscreenmu factor (determined in the same wayas with a triode, by dividing the operatingvoltage by the mu factor) the plate currentwill be substantially at cutoff, as will be thescreen current. The grid-screen mu factor isnumerically equal to the amplification factorof the same tetrode or pentode tube whenit is triode connected.The following equation is theexpression for total cathode current in a triode tube. The expressionfor the total cathodeCurrent Flowin Tetrodesand Pentodescurrent of a tetrode and a pentode tube is thesame, except that the screen-grid voltage andthe grid-screen JL- factor are used in place ofthe plate voltage and /L of the triode.Cathode current = K (Eg + Esg J 12/LsgCathode current, of course, is the sum of thescreen and plate current, plus control grid currentin the event that the control grid is positivewith respect to the cathode. It will benoted that total cathode current is independentof plate voltage in a tetrode or pentode. Also,in the usual tetrode or pentode the plate currentis substantially independent of platevoltage over the usual operating range- whichmeans simply thar the effective plate resistanceof such tubes is relatively high. However,when the plate voltage falls below thenormal operating range, the plate currentfalls sharply, while the screen current rises tosuch a value that the total cathode currentremains substantially constant. Hence, thescreen grid in a tetrode or pentode will almostinvariably be damaged by excessive dissipationif the plate voltage is removed while thescreen voltage is still being applied from alow-impedance source.The Effect ofGrid CurrentThe current equations show howthe total cathode current intriodes, tetrodes, and pentodesis a function of the potentials applied to thevarious electrodes. If only one electrode ispositive with respect to the cathode (such aswould be the case in a triode acting as aclass A amplifier) all the cathode current goesto the plate. But when both screen and plateare positive in a tetrode or pentode, the cathodecurrent divides between the two elements.Hence the screen current is taken from thetotal cathode current, while the balance goesto the plate. Further, if the control grid in atetrode or pentode is operated at a positivepotential the total cathode current is dividedbetween all three elements which have a positivepotential. In a tube which is receiving alarge excitation voltage, it may be said thatthe control grid robs electrons from the outputelectrode during the period that the grid ispositive, making it always necessary to limitthe peak-positive excursion of the controlgrid.Coefficients ofTetrodes andPentodesIn general it may be statedthat the amplification factorof tetrode and pentode tubesis a coefficient which is notof much use to the designer. In fact the amplificationfactor is seldom given on the designdata sheets of such tubes. Its value is usuallyvery high, due to the relatively high plateresistance of such tubes, but bears littlerelationship to the stage gain which actuallywill be obtained with such tubes.On the other hand, the grid-plate transconductanceis the most important coefficient ofpentode and tetrode tubes. Gain per stage canbe computed directly when the Gm is known.The grid-plate transconductance of a tetrodeor pentode tube can be calculated through useof the expression:MPG -m-
80 Vacuum Tube Principles<strong>THE</strong> R AD I 0CATHODEDOSCILLATOR GRID~.SCREEN GRIDrl J) PLATEMETAL SHELL-1EslFILAME~: l': • SUPPRESSOR AND SHELLSIGNAL GRtDFigure 20GRID STRUCTURE OF 6SA7CONVERTER TUBEeludes at least one stage for changing thefrequency of the incoming signal to the fixedfrequency of the main intermediate amplifierin the receiver. This frequency changingprocess is accomplished by selecting thebeat-note difference frequency between alocally generated oscillation and the incomingsignal frequency. If the oscillator signal issupplied by a separate tube, the frequencychanging tube is called a mixer. Alternatively,the oscillation may be generated by additionalelements within the frequency changer tube.In this case the frequency changer is commonlycalled a converter tube.ConversionConductanceThe conversion conductance(Gc)is a coefficient of interest in thecase of mixer or converter tubes,or of conventional triodes, tetrodes, or pentodesoperating as frequency changers. Theconversion conductance is the ratio of achange in the signal-grid voltage at the inputfrequency to a change in the output current atthe converted frequency. Hence Gc in a mixeris essentially the same as transconductancein an amplifier, with the exception that theinput signal and the output current are on differentfrequencies. The value of Gc in conventionalmixer tubes is from 300 to 1000micromhos. The value of Gc in an amplifiertube operated as a mixer is approximately 0.3the Gm of the tube operated as an amplifier.The voltage gain of a mixer stage is equal toGcZL where ZL is the impedance of the plateload into which the mixer tube operates.The Diode Mixer The simplest mixer tube isthe diode. The noise figure,or figure of merit, for a mixer of this type isnot as good as that obtained with other morecomplex mixers; however, the diode is usefulas a mixer in u-h-f and v-h-f equipment wherelow interelectrode capacities are vital to circuitoperation. Since the diode impedance isFigure 21SHOWING <strong>THE</strong> EFFECT OF CATHODELEAD INDUCTANCEThe degenerative action ol cathode lead in.cluctance tends to reduce the effective grid~tocathoclevoltage with respect to the voltageavailable across the Input tuned circuit. Cathodelead ,.ncluctance also Introduces unclesir·able coupling between the Input and the outputcircuits.low, the local oscillator must furnish cons~derabl~power to the diode mixer. A gooddiode m1xer has an overall gain of about 0. 5.The Triode Mixer A triode mixer has bettergain and a better noise figurethan the diode mixer. At low frequencies, thegain and noise figure of a triode mixer closelyapproaches those figures obtained when thetube is used as an amplifier. In the u-h-f andv-h-f range, the efficiency of the triode mixerdeteriorates rapidly. The optimum local oscillatorvoltage for a triode mixer is about 0. 7 aslarge as the cutoff bias of the triode. Verylittle local oscillator power is required by atriode mixer.Pentode Mixers and The most common multiConverter Tubes grid converter tube forbroadcast or shortwaveuse is the penta grid converter, typified bythe 6SA7, 6SB7-Y and 6BA7 tubes (figure 20).Operation of these converter tubes and pen to demixers will be covered in the Receiver FundamentalsChapter.4-6 Electron Tubes at VeryHigh FrequenciesAs the frequency of operation of the usualtype of electron tube is increased above about20 Me., certain assumptions which are validfor operation at lower frequencies must be reexamined.First, we find that lead inductancesfrom the socket connections to the actualelements within the envelope no longer arenegligible. Second, we find that electron
<strong>HANDBOOK</strong>The Klystron 81transit time no longer may be ignored; anappreciable fraction of a cycle of input signalmay be required for an electron to leave thecathode space charge, pass through the gridwires, and travel through the space betweengrid and plate.Effects ofLead InductanceThe effect of lead inductanceis two-fold. First, asshown in figure 21, thecombination of grid-lead inductance, gridcathodecapacitance, and cathode lead inductancetends to reduce the effective grid-cathodesignal voltage for a constant voltage at thetube terminals as the frequency is increased.Second, cathode lead inductance tends tointroduce undesired coupling between thevarious elements within the tube.Tubes especially designed for v-h-f andu-h-f use have had their lead inductancesminimized. The usual procedures for reducinglead inductance are: ( 1) using heavy leadconductors or several leads in parallel (examplesare the 6SH7 and 6AK5), (2) scalingdown the tube in all dimensions to reduceboth lead inductances and interelectrodecapacitances (examples are the 6AK5, 6F4,and other acorn and miniature tubes), and (3)the use of very low inductance extensions ofthe elements themselves as external connections(examples are lighthouse tubes such asthe 2C40, oilcan tubes such as the 2C29, andmany types of v-h-f transmitting tubes).Effect of When an electron tube is opTransit Time erated at a frequency highenough that electron transittime between cathode and plate is an appreciablefraction of a cycle at the input frequency,several undesirable effects take place.First, the grid takes power from the inputsignal even though the grid is negative at alltimes. This comes about since the grid willhave changed its potential during the timerequired for an electron to pass from cathodeto plate. Due to interaction, and a resultingphase difference between the field associatedwith the grid and that associated with a movingelectron, the grid presents a resistance toan input signal in addition to its normal"cold" capacitance. Further, as a result ofthis action, plate current no longer is in phasewith grid voltage.An amplifier stage operating at a frequencyhigh enough that transit time is appreciable:(a) Is difficult to excite as a result of gridloss from the equivalent input grid resistance,(b) Is capable of less output since transconductanceis reduced and plate current isnot in phase with grid voltage.The effects of transit time increase with thesquare of the operating frequency, and theyincrease rapidly as frequency is increasedabove the value where they become just appreciable.These effects may be reduced byscaling down tube dimensions; a procedurewhich also reduces lead inductance. Further,transit-time effects may be reduced by theobvious procedure of increasing electrode potentialsso that electron velocity will be increased.However, due to the law of electronmotionin an electric field, transit time isincreased only as the square root of the ratioof operating potential increase; therefore thisexpedient is of limited value due to otherlimitations upon operating voltages of smallelectron tubes.4-7 Special MicrowaveElectron TubesDue primarily to the limitation imposed bytransit time, conventional negative-grid electrontubes are capable of affording worthwhileamplification and power output only up to adefinite upper frequency. This upper frequencylimit varies from perhaps 100 Me. for conventionaltube types to about 4000 Me. forspecialized types such as the lighthouse tube.Above the limiting frequency, the conventionalnegative-grid tube no longer is practicable andrecourse must be taken to totally differenttypes of electron tubes in which electrontransit time is not a limitation to operation.Three of the most important of such microwavetube types are the klystron, the magnetron, andthe travelling wave tube.The Power Klystron The klystron is a typeof electron tube in whichelectron transit time is used to advantage,Such tubes comprise, as shown in figure 22,a cathode, a focussing electrode, a resonatorconnected to a pair of grids which affordvelocity modulation of the electron beam(called the "buncher"), a drift space, andanother resonator connected to a pair of grids(called the "catcher"). A collector for theexpended electrons may be included at theend of the tube, or the catcher may also performthe function of electron collection.The tube operates in the following manner:The cathode emits a stream of electrons whichis focussed into a beam by the focussingelectrode. The stream passes through thebuncher where it is acted upon by any fieldexisting between the two grids of the bunchercavity. When the potential between the twogrids is zero, the stream passes through withoutchange in velocity. But when the potentialbetween the two grids of the buncher is increasinglypositive in the direction of electron
82 Vacuum Tube Principles<strong>THE</strong>R AD I 0+600-2~\1.+f!I00-2!>00V.f111.5 VACFigure 22TWO-CAVITY KLYSTRON OSCILLATORA conventional two-cavity klystron Is shownwith a feedback loop connected between thetwo cavities so that the tube may be used ason oscillator.motion, the velocity of the electrons in thebeam is increased. Conversely, when the fieldbecomes increasingly negative in the directionof the beam (corresponding to the other halfcycle of the exciting voltage from that whichproduced electron acceleration) the velocityof the electrons in the beam is decreased.When the velocity-modulated electron beamreaches the drift space, where there is no field,those electrons which have been sped up onone half-cycle overtake those immediatelyahead which were slowed down on the otherhalf-cycle. In this way, the beam electrons becomebunched together. As the bunched groupspass through the two grids of the catchercavity, they impart pulses of energy to thesegrids. The catcher grid-space is charged todifferent voltage levels by the passing electronbunches, and a corresponding oscillating fieldis set up in the catcher cavity. The catcher isdesigned to resonate at the frequency of thevelocity-modulated beam, or at a harmonic ofthis frequency.In the klystron amplifier, energy deliveredby the buncher to the catcher grids is greaterthan that applied to the buncher cavity by theinput signal. In the klystron oscillator a feedbackloop connects the two cavities. Couplingto either buncher or catcher is provided bysmall loops which enter the cavities by way ofconcentric lines.The klystron is an electron-coupled device.When used as an oscillator, its output voltageis rich in harmonics. Klystron oscillators ofvarious types afford power outputs rangingfrom less than 1 watt to many thousand watts.Operating efficiency varies between 5 and 30per cent. Frequency may be shifted to someextent by varying the beam voltage. Tuning isFigure 23REFLEX KLYSTRON OSCILLATORA conventional reflex klystron oscillator ofthe type commonly usecJ as a local oscillatorin superheterodyne receivers operating aboveabout 2000 Me. is shown above. Frequencymoclulation of the output frequency of the oscillator,or a-f-c operation in a receiver, may beobtainecJ by varying the negative voltage on therepeller electrocle.carried on mechanically in some klystrons byaltering (by means of knob settings) the shapeof the resonant cavity.The Reflex Klystron The two-cavity klystronas described in the precedingparagraphs is primarily used as a transmittingdevice since quite reasonable amountsof power are made available in its output circuit.However, for applications where a muchsmaller amount of power is required- powerlevels in the milliwatt range- for low-powertransmitters, receiver local oscillators, etc.,another type of klystron having only a singlecavity is more frequently used.The theory of operation of the single-cavityklystron is essentially the same as the multicavitytype with the exception that the velocity-modulatedelectron beam, after having leftthe "buncher" cavity is reflected back intothe area of the buncher again by a repellerelectrode as illustrated in figure 23. Thepotentials on the various electrodes are adjustedto the value such that proper bunchingof the electron beam will take place just as aparticular portion of the velocity-modulatedbeam reenters the area of the resonant cavity.Since this type of klystron has only one circuitit can be used only as an oscillator and not asan amplifier. Effective modulation of the frequencyof a single-cavity klystron for FMwork can be obtained by modulating the repellerelectrode voltage.
<strong>HANDBOOK</strong>The Magnetron 83ANODE®-;- MAGNET: BATTERYREYElETFigure 24TUBUlATIONCUTAWAY VIEW OFWESTERN ELECTRIC 416-B/6280VHF PLANAR TRIODE TUBEThe 416-B, clesignecl by the BellTelephone Laboratories is intencleclfor amplifier or frequency multiplierservice in the 4000 me region. Employinggricl wires having a diameterequal to fifteen wavelengths of light,the 416-B has a tronsconc/uctonce of50,000. Spacing between gricl one/cathode is .0005•, to reduce transittime effects. Entire tube is gofc/ p/atecl.The Magnetron The magnetron is an s-h-foscillator tube normally employedwhere very high values of peak poweror moderate amounts of average power arerequired in the range from perhaps 700 Me.to 30,000 Me. Special magnetrons were devel_opedfor wartime use in radar equipmentswhtch had peak power capabilities of severalmillion watts (megawatts) output at frequenciesin the vicinity of 3000 Me. The normalduty cycle of operation of these radar equipmentswas approximately I/10 of one percent (the tube operated about I/1000 of thetime and rested for the balance of the operatingperiod) so that the average power outputof these magnetrons was in the vicinity of1000 watts.®Figure 25SIMPLE MAGNETRON OSCILLATORAn external tank circuit Is usetl with this typeof magnetron oscillator for operation In thelower u-h-1 range.In its simplest form the magnetron tube is afilament-type diode with two half-cylindricalplates or anodes situated coaxially with respectto the filament. The construction isillustrated in figure 25A. The anodes of themagnetron are connected to a resonant circuitas illustrated on figure 25B. The tube is surround~dby an electromagnet coil which, inturn, 1s connected to a low-voltage d-e energizingsource through a rheostat R for controllingthe strength of the magnetic field. Thefield coil is oriented so that the lines ofmagnetic force it sets up are parallel to theaxis of the electrodes.Under the influence of the strong magneticfield, electrons leaving the filament are deflectedfrom their normal paths and move incircular orbits within the anode cylinder. Thiseffect results in a negative resistance whichsustains oscillations. The oscillation frequencyis very nearly the value determined byL and C. In other magnetron circuits, the frequencymay be governed by the electron rotation,no external tuned circuits being employed.Wavelengths of less than I centimeterhave been produced with such circuits.More complex magnetron tubes employ noexternal tuned circuit, but utilize instead oneor more resonant cavities which are integralwith the anode structure. Figure 26 shows amagnetron of this type having a multi-cellular
84Vacuum Tube Principles<strong>THE</strong> R AD I 0--CATHODE LEADS"H00EANODE STR ... PS,.t,NOOE SLOCK"Figure 26MODERN MULTI-CAVITY MAGNETRONlllustratec:l Is an external-anode strapped mag•netron of the type commonly usee/ in radar equip·ment for the 10-cm, range. A permanent magnetof the general type used with such a magnetronis shown in the right-hand portion of the drawing,with the magnetron in place between the polepieces of the magnet.anode of eight cavities. It will be noted, also,that alternate cavities (which would operate atthe same polarity when the tube is oscillating)are strapped together. Strapping was found toimprove the efficiency and stability of highpowerradar magnetrons. In most radar applicationsof magnetron oscillators a powerfulpermanent magnet of controlled characteristicsis employed to supply the magnetic fieldrather than the use of an electromagnet.The Travelling The Travelling Wave TubeWave Tube (figure 27) consists of a helixlocated within an evacuatedenvelope. Input and output terminations areaffixed to each end of the helix. An electronbeam passes through the helix and interactswith a wave travelling along the helix to producebroad band amplification at microwavefrequencies.When the input signal is applied to the gunend of the helix, it travels along the helix wireat approximately the speed of light. However,the signal velocity measured along the axisof the helix is considerably lower. The electronsemitted by the cathode gun pass axiallythrough the helix to the collector, located atthe output end of the helix. The average velocityof the electrons depends upon the potentialof the collector with respect to the cathode.When the average velocity of the electrons isgreater than the velocity of the helix wave,the electrons become crowded together in thevarious regions of retarded field, where theyimpart energy to the helix wave. A power gainof 100 or more may be produced by this tube.4-8 The Cathode-Ray TubeThe Cathode-Ray Tube The cathode-ray tubeis a special type ofFigure 27<strong>THE</strong> TRAVELLING WAVE TUBEOperation ol this tube is the result of interactionbetween the electron beam and wavetravelling along the helix.electron tube which permits the visual observationof electrical signals. It may be incorporatedinto an oscilloscope for use as a testinstrument or it may be the display device forradar equipment or a television receiver.A cathode-ray tube always ineludes an electron gun for producinga stream of electrons, aOperation ofthe CRTgrid for controlling the intensity of the electronbeam, and a luminescent screen for convertingthe impinging electron beam into visiblelight. Such a tube always operates in conjunctionwith either a built-in or an externalmeans for focussing the electron stream into anarrow beam, and a means for deflecting theelectron beam in accordance with an electricalsignal.The main electrical difference betweentypes of cathode-ray tubes lies in the meansemployed for focussing and deflecting theelectron beam. The beam may be focussedand/ or deflected either electrostatically ormagnetically, since a stream of electrons canbe acted upon either by an electrostatic or amagnetic field. In an electrostatic field theelectron beam tends to be deflected toward thepositive termination of the field (figure 28).In a magnetic field the stream tends to bedeflected at right angles to the field. Further,an electron beam tends to be deflected so thatit is normal (perpendicular) to the equipotentiallines of an electrostatic field- and it tends tobe deflected so that it is parallel to the linesof force in a magnetic field.Large cathode-ray tubes used as kinescopesin television receivers usually are both focusedand deflected magnetically. On the other hand,the medium-size CR tubes used in oscilloscopesand small television receivers usuallyare both focused and deflected electrostatically.But CR tubes for special applicationsmay be focused magnetically and deflectedelectrostatically or vice versa.There are advantages and disadvantages to
<strong>HANDBOOK</strong>The Cathode Ray Tube 85Figure 28TYPICAL ELECTROSTATICCATHODE-RAY TUBEboth types of focussing and deflection. How·ever, it may be stated that electrostatic deflec·tion is much better than magnetic deflectionwhen high-frequency waves are to be displayedon the screen; hence the almost universal useof this type of deflection for oscillographicwork. But when a tube is operated at a highvalue of accelerating potential so as to obtaina bright display on the face of the tube as fortelevision or radar work, the use of magneticdeflection becomes desirable since it is rela·tively easier to deflect a high-velocity electronbeam magnetically than electrostatically.However, an ion trap is required with mag·netic deflection since the heavy negative ionsemitted by the cathode are not materially de·fleeted by the magnetic field and hence wouldburn an "ion spot" in the center of the luminescentscreen. With electrostatic deflectionthe heavy ions are deflected equally as wellas the electrons in the beam so that an ionspot is not formed.Construction of The construction of a typicalElectrostatic CRT electrostatic-focus, electrostatic-deflectioncathode-raytube is illustrated in the pictorial diagram offigure 28. The indirectly heated cathode K releasesfree electrons when heated by theenclosed filament. The cathode is surroundedby a cylinder G, which has a small hole in itsfront for the passage of the electron stream.Although this element is not a wire mesh asis the usual grid, it is known by the samename because its action is similar: it controlsthe electron stream when its negative potentialis varied.Next in order, is found the first acceleratinganode, H, which resembles another disk orcylinder with a small hole in its center. Thiselectrode is run at a high or moderately highpositive voltage, to accelerate the electronstowards the far end of the tube.The focussing electrode, F, is a sleevewhich usually contains two small disks, eachwith a small hole.After leaving the focussing electrode, theelectrons pass through another acceleratinganode, A, which is operated at a high positivepotential. In some tubes this electrode is oper·ated at a higher potential than the first acceleratingelectrode, H, while in other tubes bothaccelerating electrodes are operated at thesame potential.The electrodes which have been describedup to this point constitute the electron gun,which produces the free electrons and focussesthem into a slender, concentrated, rapidlytravelingstream for projecting onto the view·ing screen.Electrostatic To make the tube useful, meansDeflection must be provided for deflectingthe electron beam along two axesat right angles to each other. The more com·mon tubes employ electrostatic deflectionplates, one pair to exert a force on the beamin the vertical plane and one pair to exert aforce in the horizontal plane. These platesare designated as B and C in figure 28.Standard oscilloscope practice with smallcathode-ray tubes calls for connecting one ofthe B plates and one of the C plates togetherand to the high voltage accelerating anode.With the newer three-inch tubes and with five·inch tubes and larger, all four deflecting platesare commonly used for deflection. The positivehigh voltage is grounded, instead of the negativeas is common practice in amplifiers, etc.,in order to permit operation of the deflectingplates at a d-e potential at or near ground.An Aquadag coating is applied to the insideof the envelope to attract any secondary elec·trons emitted by the flourescent screen.In the average electrostatic-deflection CRtube the spot will be fairly well centered if allfour deflection plates are returned to the po·tential of the second anode (ground). However,for accurate centering and to permit movingthe entire trace either horizontally orvertically to permit display of a particularwaveform, horizontal and vertical centeringcontrols usually are provided on the front ofthe oscilloscope.After the spot is once centered, It IS necessaryonly to apply a positive or negative voltage(with respect to ground) to one of theungrounded or "free" deflector plates in orderto move the spot. If the voltage is positivewith respect to ground, the beam will beattracted toward that deflector plate, while ifnegative the beam and spot will be repulsed.The amount of deflection is directly propor·tiona! to the voltage (with respect to ground)that is applied to the free electrode.With the larger-screen higher-voltage tubesit becomes necessary to place deflecting voltageon both horizontal and both vertical plates.This is done for two reasons: First, the amountof deflection voltage required by the high-
86 Vacuum Tube Principles<strong>THE</strong> R AD I 0Figure 29TYPICAL ELECTROMAGNETICCATHODE-RAY TUBEvoltage tubes is so great that a transmittingtube operating from a high voltage supplywould be required to attain this voltage withoutdistortion. By using push-pull deflectionwith two tubes feeding the deflection plates,the necessary plate supply voltage for the deflectionamplifier is halved. Second, a certainamount of de-focussing of the electron streamis always present on the extreme excursions indeflection voltage when this voltage is appliedonly to one deflecting plate. When the deflectingvoltage is fed in push-pull to bothdeflecting plates in each plane, there is no defocussingbecause the average voltage actingon the electron stream is zero, even though thenet voltage (which causes the deflection)acting on the stream is twice that on eitherplate.The fact that the beam is deflected by amagnetic field is important even in an oscilloscopewhich employs a tube using electrostaticdeflection, because it means that precautionsmust be taken to protect the tube fromthe transformer fields and sometimes even theearth's magnetic field. This normally is doneby incorporating a magnetic shield around thetube and by placing any transformers as farfrom the tube as possible, oriented to the positionwhich produces minimum effect upon theelectron stream.Construction of Electro- The electromagneticmagnetic CRTcathode-ray tube allowsgreater definition thandoes the electrostatic tube. Also, electromagneticdefinition has a number of advantageswhen a rotating radial sweep is requiredto give polar indications.The production of the electron beam in anelectromagnetic tube is essentially the sameas in the electrostatic tube. The grid structureis similar, and controls the electron beam inan identical manner. The elements of a typicalelectromagnetic tube are shown in figure 29.The focus coil is wound on an iron core whichmay be moved along the neck of the tube tofocus the electron beam. For final adjustment,Figure 30Two pairs of coils arranged for electro·magnetic deflection in two directions.the current flowing in the coil may be varied.A second pair of coils, the deflection coilsare mounted at right angles to each otheraround the neck of the tube. In some cases,these coils can rotate around the axis of thetube.Two anodes are used for accelerating theelectrons from the cathode to the screen. Thesecond anode is a graphite coating (Aquadag)on the inside of the glass envelope. The functionof this coating is to attract any secondaryelectrons emitted by the flourescent screen,and also to shield the electron beam.In some types of electromagnetic tubes, afirst, or accelerating anode is also used inaddition to the Aquadag.ElectromagneticDeflectionA magnetic field will deflectan electron beam in a directionwhich is at right anglesto both the direction of the field and the directionof motion of the beam.In the general case, two pairs of deflectioncoils are used (figure 30). One pair is forhorizontal deflection, and the other pair is forvertical deflection. The two coils in a pairare connected in series and are wound in suchdirections that the magnetic field flows fromone coil, through the electron beam to theother coil. The force exerted on the beam bythe field moves it to any point on the screenby application of the proper currents to thesecoils.The Trace The human eye retains an imagefor about one-sixteenth secondafter viewing. In a CRT, the spot can bemoved so quickly that a series of adjacentspots can be made to appear as a line, if thebeam is swept over the path fast enough. As
<strong>HANDBOOK</strong>Gas Tubes 87long as the electron beam strikes in a givenplace at least sixteen times a second, thespot will appear to the human eye as a sourceof continuous light with very little flicker.Screen Materials-"Phasphars"At least five types of luminescentscreen materialsare commonly available onthe various types of CR tubes commerciallyavailable. These screen materials are calledphosphors; each of the five phosphors is bestsuited to a particular type of application. TheP-1 phosphor, which has a green flourescencewith medium persistence, is almost invariablyused for oscilloscope tubes for visual observation.The P-4 phosphor, with white fluorescenceand medium persistence, is used ontelevision viewing tubes ("Kinescopes"). TheP-5 and P-ll phosphors, with blue fluorescenceand very short persistence, are usedprimarily in oscilloscopes where photographicrecording of the trace is to be obtained. TheP-7 phosphor, which has a blue flash and along-persistence greenish-yellow persistence,is used primarily for radar displays whereretention of the image for several secondsafter the initial signal display is required.4-9 Gas TubesThe space charge of electrons in the vicinityof the cathode in a diode causes the plate-tocathodevoltage drop to be a function of thecurrent being carried between the cathode andthe plate. This voltage drop can be rather highwhen large currents are being passed, causinga considerable amount of energy loss whichshows up as plate dissipation.Action ofPositive IonsThe negative space charge canbe neutralized by the presenceof the proper density of positiveions in the space between the cathode andanode. The positive ions may be obtained bythe introduction of the proper amount of gas ora small amount of mercury into the envelope ofthe tube. When the voltage drop across thetube reaches the ionization potential of thegas or mercury vapor, the gas molecules willbecome ionized to form positive ions. Thepositive ions then tend to neutralize the spacecharge in the vicinity of the cathode. The voltagedrop across the tube then remains constantat the ionization potential of the gas up to acurrent drain equal to the maximum emissioncapability of the cathode. The voltage dropvaries between 10 and 20 volts, dependingupon the particular gas employed, up to themaximum current rating of the tube.Mercury VaporTubesMercury-vapor tubes, althoughvery widely used, have thedisadvantage that they must beoperated within a specific temperature range(25° to 70° C.) in order that the mercury vaporpressure within the tube shall be within theproper range. If the temperature is too low,the drop across the tube becomes too highcausing immediate overheating and possibledamage to the elements. If the temperature istoo high, the vapor pressure is too high, andthe voltage at which the tube will "flash back"is lowered to the point where destruction ofthe tube may take place. Since the ambienttemperature range specified above is withinthe normal room temperature range, no troublewill be encountered under normal operatingconditions. However, by the substitution ofxenon gas for mercury it is possible to producea rectifier with characteristics comparableto those of the mercury-vapor tube except thatthe tube is capable of operating over the rangefrom approximately -70° to 90° C. The 3B25rectifier is an example of this type of tube.Thyratron If a grid is inserted between the caTubes thode and plate of a mercury-vaporgaseous-conduction rectifier, a negativepotential placed upon the added elementwill increase the plate-to-cathode voltage droprequired before the tube will ionize or "fire."The potential upon the control grid will haveno effect on the plate-to-cathode drop after thetube has ionized. However, the grid voltagemay be adjusted to such a value that conductionwill take place only over the desiredportion of the cycle of the a-c voltage beingimpressed upon the plate of the rectifier.Voltage Regulator In a glow-discharge gas tubeTubesthe voltage drop across theelectrodes remains constantover a wide range of current passing throughthe tube. This property exists because thedegree of ionization of the gas in the tubevaries with the amount of current passingthrough the tube. When a large current ispassed, the gas is highly ionized and theinternal impedance of the tube is low. When asmall current is passed, the gas is lightlyionized and the internal impedance of the tubeis high. Over the operating range of the tube,the product (IR) of the current through the tubeand the internal impedance of the tube is verynearly constant. Examples of this type of tubeare VR-150, VR-105 and the old 874.Vacuum Tube Vacuum tubes are grouped intoClassification three major classifications:commercial, ruggedized, andpremium (or reliable). Any one of these threegroups may also be further classified for
88 Vacuum Tube Principlesmilitary duty (JAN classification). To qualifyfor JAN classification, sample lots of theparticular tube must have passed specialqualification tests at the factory. It should notbe construed that a JAN-type tube is betterthan a commercial tube, since some commercialtests and specifications are more rigid thanthe corresponding JAN specifications. TheJAN-stamped tube has merely been acceptedunder a certain set of conditions for militaryservice.Ruggedized orPremium TubesRadio tubes ate being used inincreasing numbers for industrialapplications, such ascomputing and control machinery, and in aviationand marine equipment. When a tube failsin a home radio receiver, it is merely inconvenient,but a tube failure in industrial applicationsmay bring about stoppage of some vitalprocess, resulting in financial loss, or evendanger to life.To meet the demands of these industrialapplications, a series of tubes was evolvedincorporating many special features designedto ensure a long and pre-determined operatinglife, and uniform characteristics among similartubes. Such tubes are known as ruggedized orpremium tubes. Early attempts to select re-TRIODE PLA~E FLUORESCENT ANODETRIODE CRIO ··- ·-- RAY CONTROLELECTRODECATHODESFigure 31SCHEMATIC REPRESENTATIONOF "MAGIC EYE" TUBEliable specimens of tubes trom ordinary stocktubes proved that in the long run the selectedtubes were no better than tubes picked atrandom. Long life and ruggedness had to bebuilt into the tubes by means of proper choiceand 100% inspection of all materials used inthe tube, by critical processing inspection andassembling, and by conservative ratings of thetube.Pure tungsten wire is used for heaters inpreference to alloys of lower tensile strength.Nickel tubing is employed around the heaterwires at the junction to the stem wires toreduce breakage at this point. Element structuresare given extra supports and bracing.Finally, all tubes are given a 50 hour test rununder full operating conditions to eliminateearly failures. When operated within theirratings, ruggedized or premium tubes shouldprovide a life well in excess of 10,000 hours.Ruggedized tubes will withstand severeimpact shocks for short periods, and will1006060.l.J 40<strong>THE</strong> R AD I 0IP= 2..5MA._,-- -L/0 02010 2.0 30 40 50 60EP (VOLTS)Figure 32AMPLIFICATION FACTOR 0 F TYPICAL MODETUBE DROPS RAPIDLY AS PLATE VOLTAGEIS DECREASED BELOW 20 VOLTSoperate under conditions of vibration for manyhours. The tubes may be identified in manycases by the fact that their nomenclature includesa "W" in the type number, as in 807W,5U4W, etc. Some ruggedized tubes are includedin the "5000" series nomenclature. The 5654is a ruggedized version of the 6AK5, the 5692is a ruggedized version of the 6SN7, etc.4-10 Miscellaneous Tube TypesElectronRay TubesThe electron-ray tube or magic eyecontains two sets of elements, oneof which is a triode amplifier andthe other a cathode-ray indicator. The plate ofthe triode section is internally connected tothe ray-control electrode (figure 31), so thatas the plate voltage varies in accordance withthe applied signal the voltage on the ray-controlelectrode also varies. The ray-control electrodeis a metal cylinder so placed relative to thecathode that it deflects some of the electronsemitted from the cathode. The electrons whichstrike the anode cause it to fluoresce, or giveoff light, so that the deflection caused by theray-control electrode, which prevents electronsfrom striking part of the anode, produces awedge-shaped electrical shadow on the fluorescentanode. The size of this shadow is determinedby the voltage on the ray-electrode. Whenthis electrode is at the same potential as thefluorescent anode, the shadow disappears; ifthe ray-electrode is less positive than theanode, a shadow appears the width of whichis proportional to the voltage on the ray-electrode.Magic eye tubes may be used as tuningindicators, and as balance indicators in electri·cal bridge circuits. If the angle of shadow iscalibrated, the eye tube may be used as a voltmeterwhere rough measurements suffice.
<strong>HANDBOOK</strong>Miscellaneous 89EG1:::12.6VFigure 33504030I P(MA. I20100-10 -8CHARACTERISTIC CURVES OF 12AKSSPACE-CHARGE TRIODE/vv-·4EG2 (VOLTS)r:=~:/i-21/ IControlled Series heater strings are employedWarm-up in ac-dc radio receivers and televisionsets to reduce the cost,Tubessize, and weight of the equipment.Voltage surges of great magnitude occur inseries operated filaments because of variationsin the rate of warm-up of the various tubes.As the tubes warm up, the heater resistancechanges. This change is not the same betweentubes of various types, or even between tubesof the same type made by different manufacturers.Some 6-volt tubes show an initialsurge as high as 9-volts during warm-up, whileslow-heating tubes such as the 25BQ6 areunderheated during the voltage surge on the6-volt tubes.Standardization of heater characteristics ina new group of tubes designed for series heaterstrings has eliminated this trouble. The newtubes have either 600 rna. or 400 rna. heaters,with a controlled warm-up time of approximately11 seconds. The 5U8, 6CG7, and 12BH7-A areexamples of controlled warm-up tubes.LowPlatePotentialTubesIntroduction of the 12-volt ignitionsystem 10 American automobileshas brought about the design of aseries of tubes capable of operationwith a plate potential of 12-14volts. Standard tubes perform poorly at lowplate potentials, as the amplification factorof the tube drops rapidly as the plate voltageis decreased (figure 32). Contact potentialeffects, and change of characteristics withvariations of filament voltage combine to makeoperation at low plate potentials even moreerratic.By employing special processing techniquesand by altering the electrode geometry a seriesof low voltage tubes has been developed byTung-Sol that effectively perform with all electrodesenergized by a 12-volt system. With asuitable power output transistor, this makespossible an automobile radio without a vibratorpower supply. A special space-charge tube(12K5) has been developed that delivers 40milliwatts of audio power with a 12 volt platesupply (figure 33).ForeignTubesThe increased number of importedradios and high-fidelity equipmenthave brought many foreign vacuumtubes into the United States. Many of thesetubes are comparable to, or interchangeablewith standard American tubes. A completelisting of the electrical characteristics andbase connection diagrams of all general-purposetubes made in all tube-producing countriesoutside the "Iron Curtain" is containedin the Radio Tube Vade Mecum (norld's RadioTubes) available at most larger radio partsjobbers for $5.00, or by mail from the publishersof this Handbook at $5.50 postpaid. TheEquivalent Tubes Vade Mecum (ltorld's EquivalentTubes) available at the same pricesgives all replacement tubes for a given type,both exact and near-equivalents (with pointsof difference detailed). (Data on TV and special-purposetubes if needed is contained ina companion volume Television Tubes VadeMecum).
CHAPTER FIVETransistors andSemi-ConductorsOne of the earliest detection devices usedin radio was the galena crystal, a crude exampleof a semiconductor. More modern examplesof semiconductors are the copperoxiderectifier, the selenium rectifier and thegermanium diode. All of these devices offerthe interesting property of greater resistanceto the flow of electrical current in one directionthan in the opposite direction. Typicalconduction curves for these semiconductorsare shown in Figure 1. The copper oxide rectifieraction results from the function of a thinfilm of cuprous oxide formed upon a pure copperdisc. This film offers low resistance forpositive voltages, and high resistance fornegative voltages. The same action is observedin selenium rectifiers, where a film ofselenium is deposited on an iron surface.1040enUJf5 30a.:l!~.J 2 0.J:l!00-o. 1-o. 2-o -0. •,/I 1 N34CRYISTAL DIIOETYPICAL ~TATIC \CHARACTERISTICS1/II/-so -4o 30 20 - 10VOLTSFigure lATYPICAL CHARACTERISTIC CURVEOF SEMI-CONDUCTOR DIODE}_I3905-1 Atomic Structure ofGermanium and SiliconIt has been previously stated that the electronsin an element having a large atomicnumber are grouped into rings, each ring havinga definite number of electrons. Atoms inwhich these rings are completely filled arecalled inert gases, of which helium and argonare examples. All other elements have one ormore incomplete rings of electrons. If the incompletering is loosely bound, the electronsmay be easily removed, the element is calledmetallic, and is a conductor of electric current.If the incomplete ring is tightly bound, withonly a few missing electrons, the element iscalled non-metallic and is an insulator of electriccurrent. Germanium and silicon fall betweenthese two sharply defined groups, andexhibit both metallic and non-metallic characteristics.Pure germanium or silicon may beconsidered to be a good insulator. The additionof certain impurities in carefully controlledamounts to the pure germanium will alter theconductivity of the material. In addition, thechoice of the impurity can change the directionof conductivity through the crystal, some impuritiesincreasing conductivity to positive voltages,and others increasing conductivity to negativevoltages.5-2 Mechanism ofConductionAs indicated by their name, semiconductorsare substances which have a conductivityintermediate between the high values observedfor metals and the low values observed for insulatingmaterials. The mechanism of conductionin semiconductors is different from that
Transistors 91ANODESSCHEMATIC REPRESENTATION---c::J>=.==-==IQF==~r Color Band8Gluo----ic::::Jif----.......TUBE, GERMANIUM, SILICONAND SELtNIUM DIODESFigure 1-BCOMMON DIODE COLOR CODESAND MARKINGS ARE SHOWNIN ABOVE CHARTCATHODESobserved in metallic conductors. There existin semiconductors both negatively chargedelectrons and positively charged particles,called holes, which behave as though theyhad a positive electrical charge equal in magnitudeto the negative electrical charge onthe electron. These holes and electrons driftin an electrical field with a velocity which isproportional to the field itself:vd. = p.•Ewhere vdh = drift velocity of holeEp.•magnitude of electric field= mobility of holeIn an electric field the holes will drift in adirection opposite to that of the electron andwith about one-half the velocity, since thehole mobility is about one-half the electronmobility. A sample of a semiconductor, such asgermanium or silicon, which is both chemicallypure and mechanically perfect will contain in itapproximately equal numbers of holes and electronsand is called an intrinsic semiconductor.The intrinsic resistivity of the semiconductordepends strongly upon the temperature, beingabout 50 ohm/em. for germanium at roomtemperature. The intrinsic resistivity of siliconis about 65,000 ohm/em. at the same temperature.If, in the growing of the semiconductor crystal,a small amount of an impurity, such asphosphorous, arsenic or antimony is includedin the crystal, each atom of the impurity contributesone free electron. This electron isavailable for conduction. The crystal is saidto be doped and has become electron-conductll.a&~"J®Figure 2ACUT-AWAY VIEW OF JUNCTIONTRANSISTOR, SHOWING PHYSICALARRANGEMENTFigure 2BPICTORIAL EQUIVALENT OFP·N·P JUNCTION TRANSISTOR
92 Transistors and Semi-Conductors <strong>THE</strong> <strong>RADIO</strong>E~ITTER~COLLECTORHAl TTER ~COLLECTOR~CONNECTIO8A.5E. CONNECTIONSUPPORT~BASEP-N-P TRANSISTOR ORPOINT CONTACT TRANSISTORBASE~N-P-N TRANSISTORFigure 4ELECTRICAL SYMBOLSFOR TRANSISTORSFigure 3CONSTRUCTION DETAIL OF APOINT CONTACT TRANSISTORing in nature and is called N (negative) typegermanium. The impurities which contributeelectrons are called donors. N-type germaniumhas better conductivity than pure germanium inone direction, and a continuous stream of electronswill flow through the crystal in this directionas long as an external potential of thecorrect polarity is applied across the crystal.Other impurities, such as amminum, galliumor indium add one hole to the semiconductingcrystal by accepting one electron foreach atom of impurity, thus creating additionalholes in the semiconducting crystal. The materialis now said to be hole-conducting, or P(positive! type germanium. The impuritieswhich create holes are called acceptors. P-typegermanium has better conductivity than puregermanium in one direction. This direction isopposite to that of the N-type material. Eitherthe N-type or the P-type germanium is calledextrinsic conducting type. The doped materialshave lower resistivities than the pure materials,and doped semiconductors in the resistivityrange of .01 to 10 ohm/em. are normally usedin the production of transistors.5-3 The TransistorIn the past few years an entire new technologyhas been developed for the applicationof certain semiconducting materials in productionof devices having gain properties. Thesegain properties were previously found only invacuum tubes. The elements germanium andsilicon are the principal materials which exhibitthe proper semiconducting properties permittingtheir application in the new amplifyingdevices called transistors. However,other semiconducting materials, including thecompounds indium antimonide and lead sulfidehave been used experimentally in the productionof transistors.Types of Transistors There are two basictypes of transistors, thepoint-contact type and the junction type (figure2). Typical construction detail of a pointcontacttransistor is shown in Figure 3, andthe electrical symbol is shown in Figure 4. Theemitter and collector electrodes make contactwith a small block of germanium, called thebase. The base may be either N-type or P-typegermanium, and is approximately .05" longand .03" thick. The emitter and collector electrodesare fine wires, and are spaced about.005" apart on the germanium base. The completeassembly is usually encapsulated in asmall, plastic case to provide ruggedness andto avoid contaminating effects of the atmosphere.The polarity of emitter and collectorvoltages depends upon the type of germaniumemployed in the base, as illustrated in figure 4.The junction transistor consists of a pieceof either N-type or P-type germanium betweentwo wafers of germanium of the opposite type.Either N-P-N or P-N-P transistors may bemade. In one construction called the growncrystal process, the original crystal, grownfrom molten germanium or silicon, is createdin such a way as to have the two closely spacedjunctions imbedded in it. In the other constructioncalled the fusion process, the crystalsare grown so as to make them a single conductivitytype. The junctions are then producedby fusing small pellets of special metalalloys into minute plates cut from the originalcrystal. Typical construction detail of a junctiontransistor is shown in figure 2A.The electrical schematic for the P-N-P junctiontransistor is the same as for the pointcontacttype, as is shown in figure 4.Transistor Action Presently available types oftransistors have three essentialactions which collectively are calledtransistor action. These are: minority carrierinjection, transport, and collection. Figure 2Bshows a simplified drawing of a P-N-P junction-typetransistor, which can illustrate this
<strong>HANDBOOK</strong>Transistors 93collective action. The P-N-P transistor consistsof a piece of N-type germanium on oppositesides of which a layer of P-type materialhas been grown by the fusion process.Terminals are connected to the two P-sectionsand to the N-type base. The transistor may beconsidered as two P-N junction rectifiersplaced in close juxaposition with a semiconductioncrystal coupling the two rectifierstogether. The left-hand terminal is biased inthe forward (or conducting) direction and iscalled the emitter. The right-hand terminal isbiased in the back (or reverse) direction andis called the collector The operating potentialsare chosen with respect to the base terminal,which may or may not be grounded. If anN-P-N transistor is used in place of the P-N-P,the operating potentials are reversed.The P. - Nb junction on the left is biasedin the forward direction and holes from theP region are injected into the Nb region, producingtherein a concentration of holes substantiallygreater than normally present in thematerial. These holes travel across the baseregion towards the collector, attracting neighboringelectrons, finally increasing the availablesupply of conducting electrons in thecollector loop. As a result, the collector looppossesses lower resistance whenever the emittercircuit is in operation. In junction transistorsthis charge transport is by means ofdiffusion wherein the charges move from aregion of high concentration to a region oflower concentration at the collector. The collector,biased in the opposite direction, actsas a sink for these holes, and is said to collectthem.It is known that any rectifier biased in theforward direction has a very low internal impedance,whereas one biased in the back directionhas a very high internal impedance. Thus,current flows into the transistor in a low impedancecircuit, and appears at the output ascurrent flowing in a high impedance circuit.The ratio of a change in collector current toa change in emitter current is called the currentamplification, or alpha:lca=iewhere a= current amplificationic = change in collector currentie = change in emitter currentValues of alpha up to 3 or so may be obtainedin commercially available point-contacttransistors, and values of alpha up to about0.95 are obtainable in junction transistors.Alpha CutoffFrequencyThe alpha cutoff frequency ofa transistor is that frequencyat which the grounded basecurrent gain has decreased to 0. 7 of the gainobtained a·t 1 kc. For audio transistors, thealpha cutoff frequency is in the region of 0. 7Me. to 1.5 Me. For r-f and switching transistors,the alpha cutoff frequency may be 5 Me.or higher. The upper frequency limit of operationof the transistor is determined by thesmall but finite time it takes the majority carrierto move from one electrode to another.Drift Transistors As previously noted, thesignal current 10 a conventionaltransistor is transmitted across thebase region by a diffusion process. The transittime of the carriers across this region is, thereforerelatively long. RCA has developed atechnique for the manufacture of transistorswhich does not depend upon diffusion fortransmission of the signal across the base region.Transistors featuring this new process areknown as drift transistors. Diffusion of chargecarriers across the base region is eliminated andthe carriers are propelled across the region bya ""built in" electric field. The resulting reductionof transit time of the carrier permits drifttransistors to be used at much higher frequenciesthan transistors of conventional design.The "built in" electric field is in the baseregion of the drift transistor. This field isachieved by utilizing an impurity densitywhich varies from one side of the base to theother. The impurity density is high next tothe emitter and low next to the collector. Thus,there are more mobile electrons in the regionnear the emitter than in the region near thecollector, and they will try to diffuse evenlythroughout the base. However, any displacementof the negative charge leaves a positivecharge in the region from which the electronscame, because every atom of the base materialwas originally electrically neutral. The displacementof the charge creates an electricfield that tends to prevent further electron diffusionso that a condition of equilibrium isreached. The direction of this field is such asto prevent electron diffusion from the highdensity area near the emitter to the low densityarea near the collector. Therefore, holes enteringthe base will be accelerated from the emitterto the collector by the electric field. Thusthe diffusion of charge carriers across the baseregion is augmented by the built-in electricfield. A potential energy diagram for a drifttransistor is shown in figure 5.
94 Transistors and Semi-Conductors <strong>THE</strong> <strong>RADIO</strong>DECREASINGPOTENTIAL ENERGYOF MAJORITY CARRIER.u,l. ~ ~-!---",..... 7/~ rs:• ~ ...---fv p. ~-f--EMITTER BASE I DEPL.ETION j COLLECTORREGION I RECIONIDISTANCEFigure 5POTENTIAL ENERGY DIAGRAMFOR DRIFT TRANSISTOR !2N247l32If--"v - T-"0-1--" ~ ~ ~/1--" -~~!------~.,-:6'" -/ ~1--/ ~~- - 1--"=1-tr ,...-f--l'\\0 20 30 40 so eo 70COLLECTOR VOLTS®100,---.----.---,----,---.----.---.---.5-4 TransistorCharacteristicsThe transistor produces results that may becomparable to a vacuum tube, but there is abasic difference between the two devices. Thevacuum tube is a voltage controlled devicewhereas the transistor is a current controlleddevice. A vacuum tube normally operates withits grid biased in the negative or high resistancedirection, and its plate biased in thepositive or low resistance direction. The tubeconducts only by means of electrons, and hasits conducting counterpart in the form of theN-P-N transistor, whose majority carriers arealso electrons. There is no vacuum tube equivalentof the P-N-P transistor, whose majoritycarriers are holes.The biasing conditions stated above providethe high input impedance and low output impedanceof the vacuum tube. The transistor isbiased in the positive or low resistance directionin the emitter circuit, and in the negative,or high resistance direction in the collectorcircuit resulting in a low input impedanceand a high output impedance, contrary to andopposite from the vacuum tube. A comparisonof point-contact transistor characteristics andvacuum tube characteristics is made in figure 6.The resistance gain of a transistor is expressedas the ratio of output resistance toinput resistance. The input resistance of atypical transistor is low, in the neighborhoodof 300 ohms, while the output resistance isrelatively high, usually over 20,000 ohms. Fora point-contact transistor, the resistance gainis usually over 60.The voltage gain of a transistor is theproduct of alpha times the resistance gain,and for a point-contact transistor is of the
<strong>HANDBOOK</strong>Transistor Characteristics 95+ •Cl)wa:~+:::!i
96 Transistors and Semi-Conductors<strong>THE</strong> <strong>RADIO</strong>GROUNDED BASECONNECTIONGROUNDED EMITTERCONNECTIONGROUNDED COLLECTORCONNECTIONFigure 10COMPARISON OF BASIC VACUUM TUBE AND TRANSISTOR CONFIGURATIONS5-5 Transistor Circuitry overcome these difficulties. The simple self-biasTo establish the correct operating parametersof the transistor, a bias voltage must be establishedbetween the emitter and the base. Sincetransistors are temperature sensitive devices,and since some variation in characteristics usuallyexists between transistors of a given type,attention must be given to the bias system tosystem is shown in figure 12A. The base issimply connected to the power supply througha large resistance which supplies a fixed valueof base current to the transistor. This biassystem is extremely sensitive to the currenttransfer ratio of the transistor, and must be adjustedfor optimum results with each transistor.When the supply voltage is fairly high andLFLIP-FLOP COUNTER R.F. OSCILLATOR ONE-STAGE RECEIVER~~t---1-f~CRYSTAL OSCILLATORBLOCKING OSCILLATORAUDIO AMPLIFIERDIRECT-COUPLED AMPLIFIERFigure 11TYPICAL TRANSISTOR CIRCUITS
<strong>HANDBOOK</strong>Transistor Circuitry 97-ELOAD;:Nv;;....~-4_:.R:_:E.;_S t S TOR-ER2=1oReRe = soo-1ooo .n.0®50 ..UF'----.--.--1----.J + (%~~~~~£;:AL:::;;OR)-::-Figure 12BIAS CONFIGURATIONS FOR TRANSISTORS.The voltage divider system of Cis recommended for general transistor use. Ratio of R1/Rzestablishes base bias, and emitter bias is provided by voltage drop across Re.Battery Polarity is reversed for N-P-N transistors.wide variations in ambient temperature do notoccur, the bias system of figure 12B may beused, with the bias resistor connected frombase to collector. When the collector voltageis high, the base current is increased, movingthe operating point of the transistor down theload line. If the collector voltage is low, theoperating point moves upwards along the loadline, thus providing automatic control of thebase bias voltage. This circuit is sensitive tochanges in ambient temperature, and may permittransistor failure when the transistor isoperated ncar maximum dissipation ratings.A better bias system is shown in figure 12C,where the base bias is obtained from a voltagedivider, ( R 1, R2), and the emitter is forwardbiased. To prevent signal degeneration, theemitter bias resistor is bypassed with a largecapacitance. A high degree of circuit stabilityis provided by this form of bias, providing theemitter capacitance is of the order of 50 JLfd.for audio frequency applications.Audio Circuitry A simple voltage amplifieris shown in figure 13. Directcurrent stabilization is employed in thee· nitter circuit. Operating parameters for theamplifier are given in the drawing. In this case,the input impedance of the amplifier is quitelow. When used with a high impedance drivingsource such as a crystal microphone a stepdown input transformer should be employedas shown in figure 13B. The grounded collectorcircuit of figure 13C provides a high inputimpedance and a low output impedance, muchas in the manner of a vacuum tube cathodefollower.The circuit of a two stage resistance coupledamplifier is shown in figure 14A. The inputimpedance is approximately 1100 ohms. Feedbackmay be placed around this amplifier fromthe emitter of the second stage to the base ofthe first stage, as shown in figure 14B. Adirect coupled version of the r-c amplifier isshown in figure 14C. The input impedance isof the order of 15,000 ohms, and an overallvoltage gain of 80 may be obtained with asupply potential of 12 volts.It is possible to employ N-P-N and P-N-Ptransistors in complementary symmetry circuitswhich have no equivalent in vacuum tube design.Figure 15A illustrates such a circuit. Asymmetrical push-pull circuit is shown 111-12 v.-9 V. +12 vVOL TAG£ GAIN== 80INPUT IMPEDANCE ""t' 1200 fl.~1]11"'::' 500K.10K®VOLTAGE GAtN: 0.97INPUT IMPEDANCE 'U 300 X Jl©Figure 13P-N-P TRANSISTOR VOLTAGE AMPLIFIERSA resistance coupled amplifier employing an inexpensive CK-722 transistor is shown in A. For use witha high impedance crystal microphone, a step~down translormer matches the low input impedance of thetransistor, as shown in B. The grounded collector configuration of C provides an Input impedance ofabout 300,000 ohms.
98 Transistors and Semi-Conductors <strong>THE</strong> <strong>RADIO</strong>0.1M 0.1 M.tj - CK 722010JJF10 KL---V'RNI'--'.... _B~ 47K 10K4.7K lO.UF -GAIN'V ~ -Figure 14 ® ©TWO STAGE TRANSISTOR AUDIO AMPLIFIERSThe feedback loop of B may be added to the r-c amplifier to reduce distortion, or to control theaudio response. A direct coupled amplifier is shown in C.10JJFfigure 15B. This circuit may be used to directlydrive a high impedance loudspeaker,eliminating the output transformer. A directcoupled three stage amplifier having a gainfigure of 80 db is shown in figure 15C.The transistor may also be used as a classA power amplifier, as shown in figure 16A.Commercial transistors are available that willprovide five or six watts of audio power whenoperating from a 12 volt supply. The smallerunits provide power levels of a few milliwatts.The correct operating point is chosen sothat the output signal can swing equally in thepositive and negative directions, as shown inthe collector curves of figure 16B.The proper primary impedance of the ourputtransformer depends upon the amount ofpower to be delivered to the load:E,'R.=--2P.The collector current bias is:2P.Ic-=-EeIn a class A output stage, the maximum a-cpower output obtainable is limited to 0.5 theallowable dissipation of the transistor. Theproduct I,E, determines the maximum collectordissipation, and a plot of these values is shownin figure 16B. The load line should always lieunder the dissipation curve, and should encompassthe maximum possible area between theaxes of the graph for maximum output condition.In general, the load line is tangent to thedissipation curve and passes through the supplyvoltage point at zero collector current. The d-eoperating point is thus approximately one-halfthe supply voltage.The circuit of a typical push-pull class Btransistor amplifier is shown in figure 17 A.Push-pull operation is desirable for transistoroperation, since the even-order harmonics arelargely eliminated. This permits transistors tobe driven into high collector current regionswithout distortion normally caused by nonlinearityof the collector. Cross-over distortionis reduced to a minimum by providing a slightforward base bias in addition to the normalemitter bias. The base bias is usually less than0.5 volt in most cases. Excessive base bias willboost the quiescent collector current and therebylower the overall efficiency of the stage.NPNPNP2N78NPN2N78NPN2N77PNP®Figure 15COMPLEMENTARY SYMMETRY AMPLIFIERS.N-P-N ond P-N-P transistors may be combined in circuits which hove no equivalent in vacuum tubedesign. Direct coupling between cascaded stages using a single power supply source may be employed, asin C. Impedance of power supply should be extremely low.
<strong>HANDBOOK</strong>Transistor Circuitry 99Figure 16TYPICAL CLASS-AAUDIO POWERTRANSISTOR CIRCUIT.The correct operating paint ischosen so that output signal canswing equally in a positive ornegative direction, without exceedingmaximum collector dissipation.2N 187A0®The operating point of the class B amplifieris set on the I,=O axis at the point wherethe collector voltage equals the supply voltage.The collector to collector impedance of theoutput transformer is:2E, 2Rc-c=-PoIn the class B circuit, the maximum a-cpower input is approximately equal to fivetimes the allowable collector dissipation ofeach transistor. Power transistors, such as the2N301 have collector dissipation ratings of5.5 watts and operate with class B efficiencyof about 67%. To achieve this level of operationthe heavy duty transistor relies upon efficientheat transfer from the transistor caseto the chassis, using the large thermal capacityof the chassis as a heat sink. An infinite heatsink may be approximated by mounting thetransistor in the center of a 6" x 6" copper oraluminum sheet. This area may be part of alarger chassis.The collector of most power transistors iselectrically connected to the case. For applicationswhere the collector is not grounded athin sheet of mica may be used between thecase of the transistor and the chassis.Power transistors such as the Phi/co T-1041may be used in the common collector class Bconfiguration (figure 17C) to obtain highpower output at very low distortions comparablewith those found in quality vacuum tubecircuits having heavy overall feedback. In addition,the transistor may be directly bolted tothe chassis, assuming a negative groundedpower supply Power output is of the order of10 watts, with about 0.5% total distortion.R-F Circuitry Transistors may be used forradio frequency work providedthe alpha cutoff frequency of the units issufficiently higher than the operating frequency.Shown in figure 18A is a typical i-famplifier employing an N-P-N transistor. Thecollector current is determined by a voltagedivider on the base circuit and by a bias resistorin the emitter leg. Input and output arecoupled by means of tuned i-f transformers.Bypass capacitors are placed across the biasresistors to prevent signal frequency degeneration.The base is connected to a low impedanceuntuned winding of the input transformer,and the collector is connected to a tapon the output transformer to provide propermatching, and also to make the performance ofthe stage relatively independent of variationsbetween transistors of the same type. With arate-grown N-P-N transistor such as the G.E.2N293, it is unnecessary to use neutralizationto obtain circuit stability. When P-N-P alloyADJUST R_ 1
100 Transistors and Semi-Conductors <strong>THE</strong> <strong>RADIO</strong>10 K 82. K -:-+9V.® @Figure 18TRANSISTORIZED 1-F AMPLIFIERS.Typical P·N-P transistor must be neutralired because of high collector capacitance.grown N-P-N transistor does not usually require external neutraliring circuit.RateFigure 19AUTOMATIC VOLUMECONTROL CIRCUITFOR TRANSISTORIZED1-F AMPLIFIER.A.V.C. LINE0.1 M -9 v.+:J 10.UFtransistors are used, It 1s necessary to neutralizethe circuit to obtain stability (figure 18Bl.The gain of a transistor i-f amplifier willdecrease as the emitter current is decreased.This transistor property can be used to controlthe gain of an i-f amplifier so that weak andstrong signals will produce the same audiooutput. A typical i-f strip incorporating thisautomatic volume control action is shown infigure 19.R-f transistors may be used as mixers orautodyne converters much in the same manneras vacuum tubes The autodyne circuit is shownm figure 20. Transformer T, feeds back aC2rIILFigure 20<strong>THE</strong> AUTODYNE CONVERTER CIRCUITUSING A 2N168A AS A MIXER.T3signal from the collector to the emitter causingoscillation. Capacitor C tunes the oscillatorcircuit to a frequency 4 55 kc. higher than thatof the incoming signal. The local oscillatorsignal is inductively coupled into the emittercircuit of the transistor. The incoming signalis resonated in T, and coupled via a low impedancewinding to the base circuit. Noticethat the base is biased by a voltage dividercircuit much the same as is used in audio frequencyoperation. The two signals are mixedin this stage and the desired beat frequency of455 kc. is selected by i-f transformer T, andpassed to the next stage. Collector currents of0.6 rna. to 0.8 rna. are common, and the localoscillator injection voltage at the emitter is mthe range of 0.15 to 0.25 volts, r.m.s.A complete receiver "front end" capable ofoperation up to 23 Me. is shown in figure 21.The RCA 2N24 7 drift transistor is used forthe r-f amplifier (TR1), mixer (TR2), andhigh frequency oscillator (TR3). The 2N247incorporates an interlead shield, cutting theinterlead capacitance to .003 fLpJd. If propershielding is employed between the tuned circuitsof the r-f stage, no neutralization of thestage is required. The complete assembly obtainspower from a 9-volt transistor battery.Note that input and output circuits of the transistorsare tapped at low impedance points onthe r-f coils to achieve proper impedance match.
<strong>HANDBOOK</strong>Transistor Circuitry 101r-------~------~~~------4r------939K 470 39 K 470Figure 21RF AMPLIFIER, MIXE.R,AND OSCILLATORSTAGES FORTRANSISTORIZEDHIGH FREQUENCYRECEIVER. <strong>THE</strong> RCA2N247 DRIFTTRANSISTOR ISCAPABLE OFEFFICIENT OPERATIONUP TO 23 Me.III=I,001fI - 1L - - - - - - ----'- - - -- II39 K.05TransistorOscillatorsSufficient coupling of the properphase between input and outputcircuits of the transistor will permitoscillation up to and slightly above thealpha cutoff frequency. Various forms of transistoroscillators are shown in figure 22. Asimple grounded emitter Hartley oscillator havingpositive feedback between the base and thecollector ( 2 2A) is compared to a groundedbase Hartley oscillator ( 22B). In each casethe resonant tank circuit is common to the inputand output circuits of the transistor. Selfbiasof the transistor is employed in both thesecircuits A more sophisticated oscillator employinga 2N247 transistor and utilizing avoltage divider-type bias system (figure 22C)is capable of operation up to 50 Me. or so.The tuned circuit is placed in the collector,with a small emitter-collector capacitor providingfeedback to the emitter electrode.A P-N-P and an N-P-N transistor may becombined to form a complementary Hartleyoscillator of high stability (figure 23). Thecollector of the P-N-P transistor is directlycoupled to the base of the N-P-N transistor,and the emitter of the N-P-N transistor furnishesthe correct phase reversal to sustain oscillation.Heavy feedback is maintained betweenthe emitter of the P-N-P transistor andthe collector of N-P-N transistor. The degreeof feedback is controlled by R,. The emitterresistor of the second transistor is placed at the+-9V.~-t--,~~~~~------+1~- Q05 O.IM PNP_ 2N2471N81 1N81-= Figure 23COMPLEMENTARY HARTLEYOSCILLATORP-N-P and N-P-N transistors form high stabilityoscillator. Feedback between P-N-Pemitter and N-P-N collector is controlled byR,. INBI diodes are used as amplitude limiters.Frequency of oscillation is determinedby L, c,-c,.RFCPNP-EJ®-ERFCFigure 22TYPICAL TRANSISTOR OSCILLATOR CIRCUITSA-Grounded Emitter HartleyB-Grounded Base HartleyC-2N247 Oscillator Suitable lor 50 Me. operation.©
102 Transistors and Semi-Conductors <strong>THE</strong> <strong>RADIO</strong>2N33POl NT-CONTACT TRANSISTORP01Nf-CONTACTTRANSISTORmR2C1EEE IC1 CHARGING~ ~ PERIOD--t2 I----t--~IEFigure 24NEGATIVE RESISTANCE OFPOINT-CONTACT TRANSISTORPERMITS HIGH FREQUENCYOSCILLATION (50 Mel WITHOUTWITHOUT NECESSITY OFEXTERNAL FEEDBACK PATH.center of the oscillator coil to eliminate loadingof the tuned circuit.Two germanium diodes are employed asamplitude limiters, further stabilizing amplifieroperation. Because of the low circuit impedances,it is permissible to use extremelyhigh-C in the oscillator tank circuit, effectivelylimiting oscillator temperature stability to variationsin the tank inductance.The point-contact transistor exhibits negativeinput and output resistances over part ofits operaing range, due to its unique abilityto multiply the input current. This characteristicaffords the use of oscillator circuitry havingno external feedback paths (figure 24).A high impedance resonant circuit in the baselead produces circuit instability and oscillationat the resonant frequency of the L-C circuit.Positive emitter bias is used to insure thermalcircuit stability.RelaxationOscillators0@Figure 25RELAXATION OSCILLATOR USINGPOINT-CONTACT OR SURFACEBARRIER TRANSISTORS.Transistors have almost unlimiteduse in relaxation and R-C oscillatorservice. The negative resistancecharacteristic of the point contact transistormake it well suited to such application.Surface barrier transistors are also widely usedin this service, as they have the highest alphacutoff frequency among the group of "alphaless-than-unity'"transistors. Relaxation oscillatorsused for high speed counting require transistorscapable of operation at repetition ratesof 5 Me. to 10 Me.A simple emitter controlled relaxation oscillatoris shown in figure 25, together withits operating characteristic. The emitter of thetransistor is biased to cutoff at the start of thecycle (point 1 ) . The charge on the emitter capacitorslowly leaks to ground through theemitter resistor, R,. Discharge time is determinedby the time constant of R.C. When theemitter voltage drops sufficiently low to permitthe transistor to reach the negative resistanceregion (point 2) the emitter and collector resistancesdrop to a low value, and the collector+E-E8.2 K10 K0@Figure 26TRANSISTORIZED BLOCKING OSCILLATOR
<strong>HANDBOOK</strong>Transistor Circuitry103
104 Transistors and Semi-Conductors2N10922.0 KCRYSTcl-0 K !J -PICKUPI-b- = 2JJF4 74.7.05ZP = 3001 K100..UF~~100.UF-12. v.4-50 MAFigure 29HIGH GAIN, LOW DISTORTION AUDI'O AMPLIFIER, SUITABLE FOR USEWITH A CRYSTAL PICKUP. POWER OUTPUT IS 250 MILLIWATTS.power may be obtained with a battery supplyof 12 volts. Peak current drain under maximumsignal conditions is 40 rna. Other interestingtransistor projects will be shown in laterchapters of this Handbook.
E&E TECHNI-SHEETTABLE OF WROUGHT STEEL STANDARD PIPE(BLACK OR GALVANIZED) RANDOM LENGTHS. 20-21 FEETNOMINAL SIZE O.D. I. D. POUNDSINCHES INCHES INCHES PER FOOT1/8 11 .405 .269 .2441/4 11 .540 .364 .4243/8" .675 .493 .5671/2 11 .840 .622 .8503/4 11 1.050 .824 1.1301 II 1 .31 5 1 .049 1 .6781 1 /4" 1 .660 1 .380 2.2721 1 /2" 1 .900 1. 610 2.717211 2.375 2.067 3.6522 1 /2" 2.875 2.469 5.793311 3.500 3.068 7.5753 1/2" 4.000 3.548 9 .1 09411 4.500 4.026 10.790TABLE OF ELECTRICAL METALLIC TUBING (E MT)GALVAN I ZED, 10 FOOT LENGTHSNOMINAL SIZE O.D. I. D. POUNDSINCHES INCHES INCHES PER FOOT3/8" .577 .493 .2501/2 II .706 .622 .3213/4 11 .922 .824 .4881 II 1.16 3 1.049 . 7111 1/4 11 1.508 1.308 1.0001 1/2" 1. 738 1 .61 0 1.1802" 2.195 2.067 1.500TABLE OF ALUMINUM ROD, 12 FOOT LENGTHSDIAMETERWT. #/FT.1 /8 II .o 1 53/16" .0331/4 II .059
CHAPTER SIXVacuum Tube Amplifiers6-1 Vacuum Tube ParametersThe ability of the control grid of a vacuumtube to control large amounts of plate powerwith a small amount of grid energy allows thevacuum tube to be used as an amplifier. It isthis ability of vacuum tubes to amplify anextremely small amount of energy up to almostany level without change in anything exceptamplitude which makes the vacuum tube suchan extremely valuable adjunct to modern elec·tronics and communication.Symbols for As an assistance in simplifyVacuum· Tube ing and shortening expressionsParameters involving vacuum-tube parameters,the following symbolswill be used throughout this book:Tube Constantsf-L- amplification factorRp- plate resistanceGm -transconductancef-Lsg- grid-screen mu factorGc- conversion transconductance(mixer tube)lnterelectrode CapacitancesCgk -grid-cathode capacitanceCgp -grid-plate capacitanceCpk -plate-cathode capacitancecin- input capacitance (tetrode or pentode)Cout -output capacitance ( tetrode or pentode)Electrode PotentialsEbb -d-e plate supply voltage (a positivequantity)Ecc- d-e grid supply voltage (a negativequantity)Egm -peak grid excitation voltage (~ totalpeak-to-peak grid swing)Epm -peak plate voltage (~ total peak-to-peakplate swing)ep -instantaneous plate potentialeg -instantaneous grid potentialepmin -minimum instantaneous plate voltageegmp -maximum positive instantaneous gridvoltageEP- static plate voltageEg- static grid voltageeco - cutoff biasElectrode Currentslb- average plate currentI c - average grid currentIpm- peak fundamental plate currentipmax -maximum instantaneous plate currentigmax.- maximum instantaneous grid currentlp -static plate currentI g - static grid currentOther SymbolsPi- plate power inputP 0- plate power outputPp- plate dissipationPd- grid driving power (grid plus bias losses)106
Classes of Amplifiers 107STATICTANCESTRIODEr---.I r-~-r--CCP;f: e§ ;h:CoUTc1N ;.r::Figure 1INTER ELECTRODEWITHIN A TRIODE,OR TETRODEIII..._---"'I•PENTODE OR TETRODECAPACIPENTODE,P 8- grid dissipationNp -plate efficiency( expressed as a decimal)ep -one-half angle of plate current floweg -one-half angle of grid current flowR L -load resistanceZL -load impedanceVacuum-TubeConstantsThe relationships between certainof the electrode potentialsand currents within a vacuumtube are reasonably constant under specifiedconditions of operation. These relationshipsare called vacuum-tube constants and arelisted in the data published by the manufacturersof vacuum tubes. The defining equationsfor the basic vacuum-tube constants are givenin Chapter Four.lnterelectrodeCapacitances andMiller EffectThe values of interelectrodecapacitance published invacuum-tube tables are thestatic values measured, inthe case of triodes for example, as shown infigure 1. The static capacitances are simplyas shown in the drawing, but when a tube isoperating as amplifier there is another considerationknown as Miller Effect which causesthe dynamic input capacitance to be differentfrom the static value. The output capacitanceof an amplifier is essentially the same as thestatic value given in the published tube tables.The grid-to-plate capacitance is also the sameas the published static value, but since theCgp acts as a small capacitance coupling energyback from the p 1 ate circuit to the gridcircuit, the dynamic input capacitance is equalto the static value plus an amount (frequentlymuch greater in the case of a triode) determinedby the gain of the stage, the plate loadimpedance, and the Cgp feedback capacitance.The total value for an audio amplifier stagecan be expressed in the following equation:~(dynamic)= ~(static)+ (A+ l) Cgpwhere Cgk is the grid-to-cathode capacitance,Cgp is the grid-to-plate capacitance, and A isthe stage gain. This expression assumes thatthe vacuum tube is operating into a resistiveload such as would be the case with an audiostage working into a resistance plate load inthe middle audio range.The more complete expression for the inputadmittance (vector sum of capacitance andresistance) of an amplifier operating into anytype of plate load is as follows:Input capacitance = Cgk + ( 1 + A cos e) CgpInput resistance = -~)w CgpA sineWhere: Cgk = grid-to-cathode capacitancecgp = grid-to-plate capacitanceA = voltage amplification of the tubealonee = phase angle of the plate load impedance,positive for inductiveloads, negative for capacitiveIt can be seen from the above that if theplate load impedance of the stage is capacitiveor inductive, there will be a resistive componentin the input admittance of the stage.The resistive component of the input admittancewill be positive (tending to load thecircuit feeding the grid) if the load impedanceof the plate is capacitive, or it will be negative(tending to make the stage oscillate) if theload impedance of the plate is inductive.Neutralizationaf lnterelectradeCapacitanceNeutralization of the effectsof interelectrode capacitanceis employed most frequentlyin the case of radio frequencypower amplifiers. Before the introductionof the tetrode and pentode tube, triodeswere employed as neutralized Class A amplifiersin receivers. This practice has beenlargely superseded in the present state of theart through the use of tetrode and pentodetubes in which the Cgl' or feedback capacitancehas been reduced to such a low valuethat neutralization of its effects is not necessaryto prevent oscillation and instability.6-2 Classes and Types ofVacuum-Tube AmplifiersVacuum-tube amplifiers are grouped intovarious classes and sub-classes according tothe type of work they are intended to perform.The difference between the various classes isdetermined primarily by the value of averagegrid bias employed and the maximum value of
108 Vacuum Tube Amplifiers<strong>THE</strong>R AD I 0the exciting signal to be impressed upon thegrid.Class AAmplifierA Class A amplifier is an amplifierbiased and supplied with excitationof such amplitude that plate currentflows continuously (360° of the excitingvoltage waveshape) and grid current does notflow at any time. Such an amplifier is normallyoperated in the center of the grid-voltageplate-current transfer characteristic and givesan output waveshape which is a substantialreplica of the input waveshape.Class A 1 This is another term applied to theAmplifier Class A amplifier in which gridcurrent does not flow over anyportion of the input wave cycle.Class A2AmplifierThis is a Class A amplifier operatedunder such conditions that thegrid is driven positive over a portionof the input voltage cycle, but plate currentstill flows over the entire cycle.Class AB 1 This is an amplifier operated underAmplifier such conditions of grid bias andexciting voltage that plate currentflows for more than one-half the input voltagecycle but for less than the complete cycle. Inother words the operating angle of plate currentflow is appreciably greater than 180° butless than 360°. The suffix 1 indicates that gridcurrent does not flow over any portion of theinput cycle.Class AB 2 A Class AB 2 amplifier is operatedAmplifier under essentially the same conditionsof grid bias as the ClassAB 1 amplifier mentioned above, but the excitingvoltage is of such amplitude that grid currentflows over an appreciable portion of theinput wave cycle.Class BAmplifierA Class B amplifier is biased substantiallyto cutoff of plate current(without exciting voltage) so thatplate current flows essentially over one-halfthe input voltage cycle. The operating angleof plate current flow is essentially 180°. TheClass B amplifier is almost always excitedto such an extent that grid current flows.Class C A Class C amplifier is biased to aAmplifier value greater than the value requiredfor plate current cutoff andis excited with a signal of such amplitudethat grid current flows over an appreciableperiod of the input voltage waveshape. Theangle of plate current flow in a Class C amplifieris appreciably less than 180°, or inother words, plate current flows appreciably®®Figure 2TYPES OF BIAS SYSTEMSA- Grid biasB - Cathode biasC - Grid leak biasless than one-half the time. Actually, the conventionaloperating conditions for a Class Camplifier are such that plate current flows for120° to 150° of the exciting voltage waveshape.Types ofAmplifiers©There are three general types ofamplifier circuits in use. Thesetypes are classified on the basisof the return for the input and output circuits.Conventional amplifiers are called cathode returnamplifiers since the cathode is effectivelygrounded and acts as the common return forboth the input and output circuits. The secondtype is known as a plate return amplifier orcathode follower since the plate circuit is effectivelyat ground for the input and outputsignal voltages and the output voltage orpower is taken between cathode and plate. Thethird type is called a grid-return or groundedgridamplifier since the grid is effectively atground potential for input and output signalsand output is taken between grid and plate.6-3 Biasing MethodsThe difference of potential between grid andcathode is called the grid bias of a vacuumtube. There are three general methods ofproviding this bias voltage. In each of thesemethods the purpose is to establish the gridat a potential with respect to the cathodewhich will place the tube in the desired operatingcondition as determined by its characteristics.Grid bias may be obtained from a source ofvoltage especially provided for this purpose,as a battery or other d-e power supply. Thismethod is illustrated in figure 2A, and isknown as fixed bias.A second biasing method is illustrated infigure 2B which utilizes a cathode resistoracross which an IR drop is developed as aresult of plate current flowing through it. The
<strong>HANDBOOK</strong>Amplifier Distortion 1 09cathode of the tube is held at a positive potentialwith respect to ground by the amount ofthe IR drop because the grid is at ground potential.Since the biasing voltage dependsupon the flow of plate current the tube cannotbe held in a cutoff condition by means of thecathode bias voltage developed across thecathode resistor. The value of this resistor isdetermined by the bias required and the platecurrent which flows at this value of bias, asfound from the tube characteristic curves.A capacitor is shunted across the bias resistorto provide a low impedance path to ground forthe a-c component of the plate current whichresults from an a-c input signal on the grid.The third method of providing a biasingvoltage is shown in figure 2C.' and is c~lledgrid-leak bias. During the portion of the I_n~mtcycle which causes the grid. to be positivewith respect to the cathode, gnd current flowsfrom cathode to grid, charging capacitor C~.When the grid draws current, the grid-to-~at?o.deresistance of the tube drops from an mflmtevalue to a very low value, on the order of1,000 ohms or so, making the chargin~ timeconstant of the capacitor very short. This enablesCg to charge up to essentially the fullvalue of the positive input voltage and resultsin the grid (which is connected to the low potentialplate of the capacitor) being held essentiallyat ground potent~al.During _the negativeswing of the input signal no gr_Id currentflows and the discharge path of Cg Is throughthe grid resistance which has a value of500 000 ohms or so. The discharge time consta~tfor C 8 is, therefore, v~ry lon~ in comparisonto the period of the mput signal andonly a small part of the charge on C 8 is lo~t.Thus, the bias voltage developed by the dischargeof C 8 is substantially constant an~ ~hegrid is not permitted to follow the posltiveportions of the input signal.6-4 Distortion in AmplifiersThere are three main types of distortion thatmay occur in amplifiers: ~requen_cy di_stortion,phase distortion and amphtude distortion.FrequencyDistortionFrequency distortion may occurwhen some frequency componentsof a signal are amplified more thanothers. Frequency distortion occurs at lowfrequencies if coupling capacitors b_etweenstages are too small, or may occ~r at high frequenciesas a result of the shunting effects ofthe distributed capacities in the circuit.PhaseDi stortio'hIn figure 3 an input signal consistingof a fundamental and athird harmonic is passed throughINPUTSIGNALOUTPUTSIGNAL1~1I 1 II I III I~ I 1I I I. '"'JFigure 3Illustration of the effect of phase distortionon input wave containing a third harmonicsignala two stage amplifier. Although the amplitudesof both components are amplified by identicalratios, the output waveshape is considerablydifferent from the input signal because thephase of the third harmonic signal has. beenshifted with respect to the fundamental signal.This phase shift is known as phase distortion,and is caused principally by the coupling circuitsbetween the stages of the amplifier.Most coupling circuits shift the phase of asine wave, but this has no effect on the shapeof the output wave. However, when a complexwave is passed through the same couplingcircuit, each component frequency of the waveshapemay be shifted in phase by a differentamount so that the output wave is not a faithfulreproduction of the input waveshape.Amp I itude If a signal is passed through a vacDistortion uum tube that is operating on anynon-linear part of its characteristic,amplitude distortion will occur. In such a region,a change in grid voltage ~oes. not. resultin a change in plate current which IS duectlyproportional to the cha~ge in grid voltage. ~orexample, if an amplifier is excited wlth a signalthat overdrives the tubes, the resultantsignal is distorted in amplitude, ~inc e th_etubes operate over a non-linear portion of theucharacteristic.6-5 ResistanceCapacitance CoupledAudio-Frequency Amp I ifiersPresent practice in the design of audio-frequencyvoltage amplifiers is almost exclusivelyto use resistance-capacitance coupling betweenthe low-level stages. Both triodes and
11 0 Vacuum Tube Amplifiers<strong>THE</strong> R AD I 0Figure 4STANDARD CIRCUIT FOR RESISTANCE·CAPACITANCE COUPLED TRIODE AM·PLIFIER STAGEpentodes are used; triode amplifier stageswill be discussed first.R-C CoupledTriode StagesFigure 4 illustrates the standardcircuit for a resistancecapacitancecoupled amplifierstage utilizing a triode tube with cathode bias.In conventional audio-frequency amplifier designsuch stages are used at medium voltagernRGE =-.U EGMID FREQUENCY RANGElevels (from 0.01 to 5 volts peak on the gridof the tube) and use medium-f.! triodes suchas the 6] 5 or high-f.! triodes such as the 6SF5or 6SL 7-GT. Normal voltage gain for a singlestage of this type is from 10 to 70, dependingupon the tube chosen and its operating conditions.Triode tubes are normally used in thelast voltage amplifier stage of an R-C amplifiersince their harmonic distortion with largeoutput voltage (25 to 75 volts) is less thanwith a pentode tube.Voltage Gain The voltage gain per stage ofper Stage a resistance-capacitance coupledtriode amplifier can be calculatedwith the aid of the equivalent circuitsand expressions for the mid-frequency, highfrequency,and low-frequency range given infigure 5.A triode R-C coupled amplifie; stage isnormally operated with values of cathode resistorand plate load resistor such that theactual voltage on the tube is approximatelyone-half the d-e plate supply voltage. ToA= .U RL RGRP (RL +Rc;)+RL RGRGCGK(DYNAMIC~NEXT STAGE)A HIGH FREQ. ::.. r=7.:"=1 ==:;;::==;:;=A MID FREQ.V 1 + (REQ/Xs) 2HIGH FREQUENCY RANGEREQ=--~~R~~--1+B.k_+.B.!,_RG RPXs =12TTF (CPK+CGK (OVN.O.MIC)E=-.U Ec;ITJpRc;LOW FREQUENCY RANGEA LOW FREQ.A MID FREQ.Xc aV1+(Xc/R)•""""'2""rr"" 1 •"'c"'c=--------RL RPR = Rc; + .....,R.f.L'";:+-jR~P;:--Figure 5Equivalent circuits and gain equations for a triode R-C coupled amplifier stage. In using theseequations, be sure to select the values of mu and Rp which ore proper for the static current andvoltages with which the tube will operate. These values may be obtained from curves publishedin the RCA Tube Handbook RC- 76.
<strong>HANDBOOK</strong>R-C Coupled Amplifiers 1 1 1Figure 6STANDARD CIRCUIT FOR RESISTANCE·CAPACITANCE COUPLED PENTODE AM·PLIFIER STAGEassist the designer of such stages, data onoperating conditions for commonly used tubesis published in the RCA Tube Handbook RC-16.It is assumed, in the case of the gain equationsof figure 5, that the cathode by-pass capacitor,Ck, has a reactance that is low with respectto the cathode resistor at the lowest frequencyto be passed by the amplifier stage.R-C CoupledPentode StagesCcFigure 6 illustrates the standardcircuit for a resistance·capacitance coupled pentodeamplifier stage. Cathode bias is used and thescreen voltage is supplied through a droppingresistor from the plate voltage supply. In conventionalaudio-frequency amplifier designsuch stages are normally used at low voltagelevels (from 0.00001 to 0.1 volts peak on thegrid of the tube) and use moderate·Gm pentodessuch as the 6SJ7. Normal voltage gain for astage of this type is from 60 to 250, depend·ing upon the tube chosen and its operatingconditions. Pentode tubes are ordinarily usedthe first stage of an R-C amplifier where thehigh gain which they afford is of greatest ad·vantage and where only a small voltage outputis required from the stage.The voltage gain per stage of a resistance·capacitance coupled pentode amplifier can becalculated with the aid of the equivalent circuitsand expressions for the mid-frequency,high-frequency, and low-frequency range givenin figure 7.To assist the designer of such stages, dataon operating conditions for commonly usedtypes of tubes is published in the RCA TubeHandbook RC-16. It is assumed, in the case ofthe gain equations of figure 7, that the cathodeby-pass capacitor, ck, has a reactance that islow with respect to the cathode resistor at thelowest frequency to be passed by the stage. Itis additionally assumed that the reactance ofthe screen by-pass capacitor Cd, is low withrespect to the screen dropping resistor, Rd, atthe lowest frequency to be passed by the amplifierstage.Cascade Voltage When voltage amplifier stagesAmplifier Stages are operated in such a mannerthat the output voltage of thefirst is fed to the grid of the second, and soforth, such stages are said to be cascaded.The total voltage gain of cascaded amplifierFigure 7Equivalent circuits andgain equations for a pentodeR-C coupled amplifierstage. In using these equationsbe sure to select thevalues of Gm and RP whichore proper for the staticcurrents and voltages withwhich the tube will operate.These values may beobtained from curves pub·lished in the RCA TubeHandbook RC-16.]]~}A=MID FREQUENCY RANGE~__tlDCK(OYNAMIC)HIGH FREQUENCY RANGE~~__L_JRGGM REQREQ =A Htc;H FREQ.A MID FREQ.Xs =A LOW FREO.A MID FREQ.RL1+~+~~V 1+ (REQ/Xs)«2 TTF (CPK+ CGK (DYNAMIC)LOW FREQUENCY RANGEXc =2 TTF CcR= RG + ~~+ ~:
1 1 2 Vacuum Tube Amplifiers<strong>THE</strong>R AD I 0100 1. RL= !>00000 OHMSMID-FREQUENCY GAIN= GMV1 RL100 1000 1 0000 100000FREQUENCY (c.P.S)Figure 8The variation of stage gain with frequencyin an r-c coupled pentode amplifier for vari•ous values of plate load resistance000HIGH-FREQUENCY G.4.1N = GM V1 i! COUPLING NETWORKC : CouT V1 + ( INV2 + C DISTRIBUTEDFOR COMPROMISE HIGH FREQUENCY EQUALIZATION.XLL = 0.!1 Xc AT fcRL = Xc AT f'cWHERE f C::: CUTOFF FREQUENCY OF AMPLIFIERLL = PEAKING INDUCTORFOR COMPROMISE LOW FREQUENCY EQUALIZATIONRs = RK (GM V1 RL)stages is obtained by taking the product of thevoltage gains of each of the successive stages.Sometimes the voltage gain of an amplifierstage is rated in decibels. Voltage gain isconverted into decibels gain through the useof the following expression: db= 20 logio A,where A is the voltage gain of the stage. Thetotal gain of cascaded voltage amplifier stagescan be obtained by adding the number ofdecibels gain in each of the cascaded stages.R.C AmplifierResponseA typical frequency responsecurve for an R-C coupled audioamplifier is shown in figure 8.It is seen that the amplification is poor for theextreme high and low frequencies. The reducedgain at the low frequencies is caused by theloss of voltage across the coupling capacitor.In some cases, a low value of coupling capacitoris deliberately chosen to reduce the re·sponse of the stage to hum, or to attenuatethe lower voice frequencies for communicationpurposes. For high fidelity work the product ofthe grid resistor in ohms times the couplingcapacitor in microfarads should equal 25,000.(ie.: 500,000 ohms x 0.05 p.fd = 25,000).The amplification of high frequencies fallsoff because of the Miller effect of the subsequentstage, and the shunting effect of resi·dual circuit capacities. Both of these effectsmay be minimized by the use of a low value ofplate load resistor.Grid Leak Biasfor High Mu TriodesThe correct operating biasfor a high-mu triode suchas the 6SL 7, is fairly crit·ical, and will be found to be highly variablefrom tube to tube because of minute variationsin contact potential within the tube itself. Asatisfactory bias method is to use grid leakbias, with a grid resistor of one to ten meg·ReCs=RKCKCrc. = 25 TO 50 .UFO. IN PARALLEL WITH 001 MICACeo= CAPACITANCE FROM ABOVE WiTH 001 MICA IN PARALLELFigure 9SIMPLE COMPENSATED VIDEOAMPLIFIER CIRCUITResistor RL in conjunction with coil Lserves to flatten the high-frequency respons~of the stage, while C 8and R serve to equalizethe low-frequency respo~se of this sim·pie video amplifier stage.ohms connected d i r e c t I y between grid andcathode of the tube. The cathode is grounded.Grid current flows at all times, and the effectiveinput resistance is about one-half theresistance value of the grid leak. This circuitis particularly well suited as a high gainamplifier following low output devices, suchas cry s t a I microphones, or dynamic microphones.R.C Amplifier A resistance-capacityGeneral Characteristics coupled amplifier canbe designed to providea good frequency response for almost anydesired range. For instance, such an amplifiercan be built to provide a fairly uniform amplificationfor frequencies in the audio range ofabout 100 to 20,000 cycles. Changes in thevalues of coupling capacitors and load resistorscan extend this frequency range tocover the very wide range required for videoservice. However, extension of the range canonly be obtained at the cost of reduced over·all amplification. Thus the R-C method ofcoupling allows good frequency response withminimum distortion, but low amplification.Phase distortion is less with R-C coupling
<strong>HANDBOOK</strong>Video Frequency Amplifiers 11 3than with other types, except direct coupling.The R-C amplifier may exhibit tendencies to"motorboat" or oscillate if it is used with ahigh impedance plate supply.6-6 Video-FrequencyAmplifiersA video-frequency amplifier is one whichhas been designed to pass frequencies fromthe lower audio range (lower limit perhaps 50cycles) to the middle r-f range (upper limitperhaps 4 to 6 megacycles). Such amplifiers,in addition to passing such an extremely widefrequency range, must be capable of amplifyingthis range with a minimum of amplitude,phase, and frequency distortion. Video amplifiersare commonly used in television, pulsecommunication, and radar work.Tubes used in video amplifiers must havea high ratio of Gm to capacitance if a usablegain per stage is to be obtained. Commonlyavailable tubes which have been designed foror are suitable for use in video amplifiers are:6AU6, 6AG5, 6AK5, 6CB6, 6AC7, 6AG7, and6K6-GT. Since, at the upper frequency limitsof a video amplifier the input and outputshunting capacitances of the amplifier tubeshave rather low values of reactance, lowvalues of coupling resistance along withpeaking coils or other special interstage couplingimpedances are usually used to flattenout the gain/frequency and hence the phase/frequency characteristic of the amplifier.Recommended operating conditions along withexpressions for calculation of gain and circuitvalues are given in figure 9. Only a simpletwo-terminal interstage coupling network isshown in this figure.The performance and gain-per-stage of avideo amplifier can be improved by the useof increasingly complex two-terminal interstagecoupling networks or through the useof four-terminal coupling networks or filtersbetween successive stages. The reader is re·£erred to Terman's "Radio Engineer's Handbook"for design data on such interstagecoupling networks.6-7 Other lnterstageCoupling MethodsFigure 10 illustrates, in addition to resistance-capacitanceinterstage coupling, sevenadditional methods in which coupling betweentwo successive stages of an audio-frequencyamplifier may be accomplished. Althoughresistance-capacitance coupling is most commonlyused, there are certain circuit condi·tions wherein coupling methods other thanresistance capacitance are more effective.TransformerCouplingTransformer coupling, as illus·trated in figure lOB, is seldomused at the present time betweentwo successive single-ended stages of anaudio amplifier. There are several reasons whyresistance coupling is favored over transformercoupling between two successive single-endedstages. These are: ( 1) a transformer havingfrequency characteristics comparable with aproperly designed R-C stage is very expensive;(2) transformers, unless they are very wellshielded, will pick up inductive hum fromnearby power and filament transformers; (3)the phase characteristics of step-up interstagetransformers are poor, making very difficultthe inclusion of a transformer of this typewithin a feedback loop; and (4) transformersare heavy.However, there is one circuit applicationwhere a step-up interstage transformer is ofconsiderable assistance to the designer; thisis the case where it is desired to obtain alarge amount of voltage to excite the grid of acathode follower or of a high-power Class Aamplifier from a tube operating at a moderateplate voltage. Under these conditions it is pos·sible to obtain a peak voltage on the secondaryof the transformer of a value somewhat greaterthan the d-e plate supply voltage of the tubesupplying the primary of the transformer.Push-Pull Transformer Push-pull transformerlnterstage Coupling coupling between twostages is illustrated infigure lOC. This interstage coupling arrang~mentis fairly commonly used. The system 1sparticularly effective when it is desired, as inthe system just described, to obtain a fairlyhigh voltage to excite the grids of ~ highpoweraudio stage. The arrangement 1s alsovery good when it is desired to apply feedbackto the grids of the push-pull stage byapplying the feedback voltage to the lowpotentialsides of the two push-pull second·aries.ImpedanceCouplingImpedance coupling between twostages is shown in figure lOD.This circuit arrangement is seldomused, but it offers one strong advantage overR-C interstage coupling. This advantage isthe fact that, since the operating voltage onthe tube with the impedance in the plate circuitis the plate supply voltage, it is possibleto obtain approximately twice the peak volt·age output that it is possible to obtain withR·C coupling. This is because, as has been
114 Vacuum Tube Amplifiers <strong>THE</strong> R AD I 0@ RESISTANCE-CAPACITANCE COUPLING@TRANSFORMER COUPLING@ PUSH-PULL TRANSFORMER COUPLING@ IMPEDANCE COUPLING@ IMPEDANCE-TRANSFORMER COUPLING® RESISTANCE-TRANSFORMER COUPLING=+B@ CATHODE COUPLING=@ DIRECT COUPLINGFigure 10INTERSTAGE COUPLING METHODS FOR AUDIO FREQUENCY VOLTAGE AMPLIFIERSmentioned before, the d-e plate voltage on anR-C stage is approximately one-half the platesupply voltage.Impedance-Transformerand Resistance· TransformerCoup I in gThese two circuit ar·rangements, illustratedin figures IOE and IOF,are employed when it isdesired to use transformer coupling for thereasons cited above, but where it is desiredthat the d-e plate current of the amplifierstage be isolated from the primary of the couplingtransformer. With most types of highpermeabilitywide-response transformers it isnecessary that there be no direct-current flowthrough the windings of the transformer. Theimpedance-transformer arrangement of figureIOE will give a higher voltage output fromthe stage but is not often used since the platecoupling impedance (choke) must have veryhigh inductance and very low distributed capacitancein order not to restrict the range of
<strong>HANDBOOK</strong>Phase Inverters 11 5JJ' = -JJ G~1G = RK G" (1 + ~ )RK =CATHODE RESISTORlJ = J.J OF EACH TUBERP = RP OF EACH TUB£EQUIVALENT FACTORS INDICATED ABOVE BY (') ARETHOSE OBTAINED BV USING AN AMPLIFIER WITH A PAIROF SIMILAR TUBE TYPES IN CIRCUIT SHOWN ABOVE.Figure 11Equivalent factors for o pair of simi/or tri·odes operating as o cathode-coupled audio·frequency voltage amplifier.the transformer which it and its associatedtube feed. The resistance-transformer arrange·ment of figure !OF is ordinarily quite satis·factory where it is desired to feed a trans·former from a voltage amplifier stage with nod.c.in the transformer primary.CathodeCouplingThe cathode coupling arrangementof figure lOG has been widely usedonly comparatively recently. Oneoutstanding characteristic of such a circuit isthat there is no phase reversal between thegrid and the plate circuit. All other commontypes of interstage coupling are accompaniedby a 180° phase reversal between the gridcircuit and the plate circuit of the tube.Figure 11 gives the expressions for determiningthe appropriate factors for an equivalenttriode obtained through the use of a pairof similar triodes connected in the cathode·coupled circuit shown. With these equivalenttriode factors it is possible to use the expressionsshown in figure 5 to determine thegain of the stage at different frequencies. Theinput capacitance of such a stage is less thanthat of one of the triodes, the effective grid·to-plate capacitance is very much less (it isso much less that such a stage may be usedas an r-f amplifier without neutralization), andthe output capacitance is approximately equalto the grid-to-plate c,;apacitance of one of thetriode sections. This circuit is particularlyeffective with tubes such as the 6]6, 6N7, and6SN7-GT which have two similar triodes inone envelope. An appropriate value of cathoderesistor to use for such a stage is the valuewhich would be used for the cathode resistorof a conventional amplifier using one of thesame type tubes with the values of plate voltageand load resistance to be used for thecathode-coupled stage.Inspection of the equations in figure 11shows that as the cathode resistor is madesmaller, to approach zero, the Gm approacheszero, the plate resistance approaches the Rpof one tube, and the mu approaches zero. Asthe cathode resistor is made very large the Gmapproaches one half that of a single tube ofthe same type, the plate resistance approachestwice that of one tube, and the mu approachesthe same value as one tube. But since the Gmof each tube decreases as the cathode resistoris made larger (since the plate current willdecrease on each tube) the optimum value ofcathode resistor will be found to be in thevicinity of the value mentioned in the previousparagraph.Direct Coupling Direct coupling between successiveamplifier stages (plateof first stage connected directly to the grid ofthe succeeding stage) is complicated by thefact that the grid of an amplifier stage mustbe operated at an average negative potentialwith respect to the cathode of that stage.However, if the cathode of the second amplifierstage can be operated at a potential morepositive than the plate of the preceding stageby the amount of the grid bias on the secondamplifier stage, this direct connection betweenthe plate of one stage and the grid of the succeedingstage can be used. Figure IOH illustratesan application of this principle inthe coupling of a pentode amplifier stage tothe grid of a "hot-cathode" phase inverter. Inthis arrangement the values of cathode, screen,and plate resistor in the pentode stage arechosen such that the plate of the pentode is atapproximately 0. 3 times the plate supply potential.The succeeding phase-inverter stagethen operates with conventional values ofcathode and plate resistor (same value of resistance)in its normal manner. This type ofphase inverter is described in more detail inthe section to follow.6-8 Phase InvertersIt is necessary in order to excite the gridsof a push-pull stage that voltages equal inamplitude and opposite in polarity be appliedto the two grids. These voltages may be obtainedthrough the use of a push-pull inputtransformer such as is shown in figure IOC.It is possible also, without the attendant bulkand expense of a push-pull input transformer,to obtain voltages of the proper polarity and
116 Vacuum Tube Amplifiers<strong>THE</strong> R AD I 0phase through the use of a so-called phaseinverterstage. There are a large number ofphase inversion circuits which have been developedand applied b.ut the three shown infigure 12 have been found over a period oftime to be the most satisfactory from the pointof view of the number of components requiredand from the standpoint of the accuracy withwhich the two out-of-phase voltages are heldto the same amplitude with changes in supplyvoltage and changes in tubes.All of these vacuum tube phase invertersare based upon the fact that a 180° phaseshift occurs within a vacuum tube between thegrid input voltage and the plate output voltage.In certain circuits, the fact that the grid inputvoltage and the voltage appearing across thecathode bias resistor are in phase is used forphase inversion purposes.@)"HOT CATHODE" PHASE INVERTER"Hot-Cathode" Figure 12A illustrates the hotPhase Inverter cathode type of phase inverter.This type of phase inverteris the simplest of the three types sinceit requires only one tube and a minimum ofcircuit components. It is particularly simplewhen directly coupled from the plate of apentode amplifier stage as shown in figurelOH. The circuit does, however, possess thefollowing two disadvantages: ( 1) the cathodeof the tube must run at a potential of approximately0.3 times the plate supply voltageabove the heater when a grounded commonheater winding is used for this tube as wellas the other heater-cathode tubes in a receiveror amplifier: (2) the circuit actually has aloss in voltage from its input to either of theoutput grids- about 0. 9 times the input voltagewill be applied to each of these grids.This does represent a voltage gain of about1.8 in total voltage output with respect to input(grid-to-grid output voltage) but it is stillsmall with respect to the other two phaseinverter circuits shown.Recommended component values for usewith a 6]5 tube in this circuit are shown infigure 12A. If it is desired to use another tubein this circuit, appropriate values for the operationof that tube as a conventional amplifiercan be obtained from manufacturer's tube data.The value of RL obtained should be divided bytwo, and this new value of resistance placedin the circuit as RL. The value of Rk fromtube manual tables should then be used asRkl in this circuit, and then the total of Rkland Rk 2should be equal to RL."Floating Paraphase" An alternate type ofPhase Inverter phase inverter sometimescalled the "floatingparaphase" is illustrated in figure 12B.This circuit is quite often used with a 6N7@"FLOATING PARAPHASE"PHASE INVERTER=+B300V.© CATHODE COUPLED PHASE INVERTERFigure 12THREE POPULAR PHASE-INVERTER CIRCUITS WITH RECOMMENDED VALUES FORCIRCUIT COMPONENTStube, and appropriate values for the 6N7 tubein this application are shown. The circuitshown with the values given will give a voltagegain of approximately 21 from the inputgrid to each of the grids of the succeedingstage. It is capable of approximately 70 voltspeak output to each grid.The circuit inherently has a small unbalancein output voltage. This unbalance can be eliminated,if it is required for some special application,by making the resistor R 81 a few percent lower in resistance value than Rg 3 .Cathode-Coupled The circuit shown in figurePhase Inverter 12C gives approximately onehalfthe voltage gain from theinput grid to either of the grids of the succeedingstage that would be obtained from asingle tube of the same type operating as aconventional R-C amplifier stage. Thus, witha 6SN7-GT tube as shown (two 6J5's in one
<strong>HANDBOOK</strong>Vacuum Tube Voltmeter 117.01 Rs RsD.C.INPUTRP2+Ec = EP- llf:.:+-+--------+----.::r+6,3001/,Figure 14DIRECT COUPLEDD-C AMPLIFIERFigure 13VOLTAGE DIVIDER PHASEINVERTERsame amplitude as the output of V,, but ofopposite phase.envelope) the voltage gain from the inputgrid to either of the output grids will be approximately7- the gain is, of course, 14 fromthe input to both output grids. The phasecharacteristics are such that the circuit iscommonly used in deriving push-pull deflectionvoltage for a cathode-ray tube from asignal ended input signal.The first half of the 6SN7 is used as anamplifier to increase the amplitude of the appliedsignal to the desired level. The secondhalf of the 6SN7 is used as an inverter andamplifier to produce a signal of the sameamplitude but of opposite polarity. Since thecommon cathode resistor, Rk, is not by-passedthe voltage across it is the algebraic sum ofthe two plate currents and has the same shapeand polarity as the voltage applied to the inputgrid of the first half of the 6SN7. When asignal, e, is applied to the input circuit, theeffective grid-cathode voltage of the firstsection is Ae/2, when A is the gain of thefirst section. Since the grid of the secondsection of the 6SN7 is grounded, the effect ofthe signal voltage across Rk (equal to e/2 ifRk is the proper value) is the same as thougha signal of the same amplitude but of oppositepolarity were applied to the grid. The outputof the second section is equal to - Ae/2 if theplate load resistors are the same for both tubesections.Voltage Divider A commonly used phase inPhase Inverter verter is shown in figure 13.The input section (V 1) is connectedas a conventional amplifier. The outputvoltage from V, is impressed on the voltagedivider R, ·R 6 • The values of R, and R 6are in such a ratio that the voltage impressedupon the grid of V 2 is 1/ A times the outputvoltage of V,, where A is the amplificationfactor of V,. The output of V 2 is then of the6-9 D-C Amp I ifiersDirect current amplifiers are special typesused where amplification of very slow variationsin voltage, or of d-e voltages is desired.A simple d-e amplifier consists of a singletube with a grid resil!tor across the inputterminals, and the load in the plate circuit.Basic D-CA simple d-e amplifierAmplifier Circuit circuit is shown infigure 14, wherein thethe grid of one tube is connected directly tothe plate of the pre ceding tube in such amanner that voltage changes on the grid ofthe first tube will be amplified by the system.The voltage drop across the plate couplingresistor is impressed directly upon the gridof the second tube, which is provided withenough negative grid bias to balance out theexcessive voltage drop across the couplingresistor. The grid of the second tube is thusmaintained in a slightly negative position.The d-e amplifier will provide good low frequencyresponse, with negligible phase distortion.High frequency response is limitedby the shunting effect of the tube capacitances,as in the normal resistance coupled amplifier.A common fault with d-e amplifiers of alltypes is static instability. Small changes inthe filament, plate, or grid voltages cannotbe distinguished from the exciting voltage.Regulated power supplies and special balancingcircuits have been devised to reduce theeffects of supply variations on these amplifiers.A successful system is to apply theplate potential in phase to two tubes, and toapply the exciting signal to a push-pull grid
118 Vacuum Tube Amplifiers<strong>THE</strong> R AD I 0Figure 15LOFTIN-WHITED-C AMPLIFIERFigure 16PUSH-PULL D-C AMPLIFIERWITH EI<strong>THE</strong>R SINGLE-ENDEDOR PUSH-PULL INPUTcircuit configuration. If the two tubes areidentical, any change in electrode voltage isbalanced out. The use of negacive feedbackcan also greatly reduce drift problems.The "Loftin-White" Two d-e amplifierCircuitstages may be arranged,so that their platesupplies are effectively in series, as illustratedin figure 15- This is known as a LoftinW bite amplifier. All plate and grid voltagesmay be obtained from one master power supplyinstead of separate grid and plate supplies.A push-pull version of this amplifier (figure 16)can be used to balance out the effects of slowvariations in the supply voltage.6-10 Single-ended TriodeAmplifiersFigure 17 illustrates five circuits for theoperation of Class A triode amplifier stages.Since the cathode current of a triode Class A 1(no grid current) amplifier stage is constantwith and without excitation, it is commonpractice to operate the tube with cathodebias. Recommended operating conditions inregard to plate voltage, grid bias, and loadimpedance for conventional triode amplifierstages are given in the RCA Tube Manual,RC-16.Extended Class AOperationIt is possible, under certainconditions to operate singleendedtriode amplifier stages(and pentode and tetrode stages as well) withgrid excitation of sufficient amplitude thatgrid current is taken by the tube on peaks.This type of operation is called Class A 2 andis characterized by increased plate-cucuuefficiency over straight Class A amplificationwithout grid current. The normal Class A 1amplifier power stage will operate with a platecircuit efficiency of from 20 per cent to perhaps35 per cent. Through the use of Class A 2operation it is possible to increase this platecircuit efficiency to approximately 38 to 45per cent. However, such operation requirescareful choice of the value of plate load impedance,a grid bias supply with good regulation(since the tube draws grid current onpeaks although the plate current does notchange with signal), and a driver tube withmoderate power capability to excite the gridof the Class A 2 tube.Figures 17D and 17E illustrate two methodsof connection for such stages. Tubes such asthe 845, 849, and 304TL are suitable for sucha stage. In each case the grid bias is approximatelythe same as would be used for a ClassA1 amplifier using the same tube, and asmentioned before, fixed bias must be usedalong with an audio driver of good regulationpreferablya triode stage with a 1: 1 or stepdowndriver transformer. In each case it willbe found that the correct value of plate loadimpedance will be increased about 40 per centover the value recommended by the tube manufacturerfor Class A 1 operation of the tube.Operation Characteristicsof a Triode operates in such a way asA Class A power amplifierPower Amplifier to amplify as faithfully aspossible the waveform appliedto the grid of the tube. Large power outputis of more importance than high voltageamplification, consequent! y gain characteristicsmay be sacrificed in power tube designto obtain more important power handling capabilities.Class A power tubes, such as the 45,2A3 and 6AS7 are characterized by a lowamplification factor, high plate dissipationand relatively high filament emission.The operating characteristics of a Class A
<strong>HANDBOOK</strong> Triode Amplifier Characteristics 11 9-ll@ IMPEDANCE COUPLING@ TRANSFORMER COUPLING© IMPEDANCE-TRANSFORMER COUPLINGAUTOTRANSfORMERTO CLASS C-BIAS +B LOAD©CLASS A2 MODULATOR WITH AUTO-TRANS-FORMER COUPLINGFigure 17Output coupling arrangements for slng/e .. endedCloss A triode aud/o .. frequency power amplifiers.triode amplifier employing an output transformer-coupledload may be calculated fromthe plate family of curves for the particularrube in question by employing the followingsteps:1- The load resistance should be approximate!y twice the plate resistance of therube for maximum undistorted power output.Remember this fact for a quick checkon calculations.2- Calculate the zero-signal bias voltage(Eg).-(0. 68 X Ebb)Where Ebb is the actual plate voltage ofthe Class A stage, and ll is the amplificationfactor of the tube.3- Locate the Eg bias point on the Ip vs.EP graph where the Eg bias line crossesthe plate voltage line, as shown in figure18. Call this point P.4- Locate on the plate family of curves thevalue of zero-signal plate current, IP,corresponding to the operating point, P.5- Locate 2 x I P (twice the value of I p) onthe plate current axis (Y-axis). Thispoint corresponds to the value of maximumsignal plate current, imax·6- Locate point x on the d-e bias curve atzero volts (Eg = 0), corresponding to thevalue of imax•7- Draw a straight line(x- y) through pointsx and P. This line is the load resistanceline. Its slope corresponds to the valueof the load resistance.8- Load Resistance, (in ohms)llemax- em inRL =. .lmax - lminwhere e is in volts, i is in amperes, andRL is in ohms.9- Check: Multiply the zero-signal platecurrent, Ip, by the operating plate voltage,Ep. If the plate dissipation ratingof the tube is exceeded, it is necessaryto increase the bias (Eg) on the tube sothat the plate dissipation falls withinthe maximum rating of the tube. If thisstep is taken, operations 2 through 8must be repeated with the new value ofEg.10- For maximum power output, the peak a-cgrid voltage on the tube should swing to2Eg on the negative cycle, and to zerobiason the positive cycle. At the peakof the negative swing, the plate voltagereaches emax and the plate currentdrops to imin. On the positive swing ofthe grid signal, the plate voltage dropsto emin and the plate current reachesimax· The power output of the tube is:Power Output (watts)Cimax- imio) X (emax- em in)where i is in amperes and e is in volts.11- The second harmonic distortion generatedin a single-ended Class A triode amplifier,expressed as a percentage of thefundamental output signal is:8
1 2 0 Vacuum Tube Amplifiers<strong>THE</strong> R AD I 025 0VlW200a:wQ_:;
<strong>HANDBOOK</strong>Push-Pull Amplifiers 1 21II6- The power output is:Power Output (watts)(imax- iminl + 1.41 (Ix- Iy) 2 X RLPo ~----------------~--~-----=32v:'here I x is the plate current at the pointon the load line where the grid voltage,e 8 , is equal to: E 8- 0. 7 E 8; and wherely is the plate current at the point whereeg is equal to: E 8+ 0. 7 E 8•7- The percentage harmonic distortion is:% 2d harmonic distortionP(STATIC VALUE) e MAXPLATE VOLTSFigure 20GRAPHIC DETERMINATION OF OPERATING CHARACTERISTICS OF A PENTODEPOWER AMPLIFIER"V" is the negative control gricl voltage atthe operating point Pimax- imin + 1.41 (Ix- ly) X 100Where lp is the static plate current ofof the tube.% 3d harmonic distortionimax- imin- 1.41 (IX- ly)________ _:__:.___ X 100imax- imin + 1.41 Ox- ly)The grid voltage curve that this pointfalls upon should be one that is about ~the value of E 8required to cut the platecurrent to a very low value (Point B).Point B represents imin on the plate currentaxis (y-axis). The line imax/2 shouldbe located half-way between imax andimin·3- A trial load line is constructed aboutpoint P and point A in such a way thatthe lengths A-P and P-B are approximatelyequal.4- V:'hen the most satisfactory load line hasbeen determined, the load resistance maycalculated:emax- emin5- The operating bias (Eg) is the bias atpoint P.6-12Push-Pull AudioAmplifiersA number of advantages are obtained throughthe use of the push-pull connection of two orfour tubes in an audio-frequency power amplifier.Two conventional circuits for the useof triode and tetrode tubes in the push-pullconnection are shown in figure 21. The twomain advantages of the push-pull circuit arrangementare: ( 1) the magnetizing effectof the plate currents of the output tubes iscancelled in the windings of the output transformer;(2) even harmonics of the input signal(second and fourth harmonics primarily) generatedin the push-pull stage are cancelledwhen the tubes are balanced.The cancellation of even harmonics generatedin the stage allows the tubes to be oper-+B PLATEPUSH-PULL TRIODE AND TETRODE+BS.G.+8 PLATEFIGURE 21
122 Vacuum Tube Amplifiers<strong>THE</strong> R AD I 030 025 0=-i= UAAXttp\1/·x-~~rl~::::~IIi300A250-;;:~20 01/)w"' wa.~IS:J_j:::;:0UJ 1001-
<strong>HANDBOOK</strong>Class 8Audio Amplifiers1 23where RL is expressed in ohms, EP involts, and imax in amperes.Figure 22 illustrates the above steps appliedto a push-pull Class A amplifier usingtwo 2A3 tubes.4- The average plate current is 0.636 i mw"and, multiplied by the plate voltage, Ep,will give the average watts input to theplates of the two tubes. The power outputshould be subtracted from this valueto obtain the total operating plate dissipationof the two tubes. If the platedissipation is excessive, a slightlyhigher value of RL should be chosen tolimit the plate dissipation.5- The correct value of operating bias, andthe static plate current for the push-pulltubes may be determined from the E 8vs.lp curves, which are a derivation of theEp vs. lp curves for various values ofEg.6- The E 8vs. lp curve may be ~ons_tructedin this manner: Values of gnd b1as areread from the intersection of each gridbias curve with the load line. Thesepoints are transferred to the E 8vs. Ipgraph to produce a curved line, A-B. Ifthe grid bias curves of the Ep vs. Ipgraph were straight lines, the lines ofthe E 8vs. lp graph would also be straightThis is usually not the case. A tangentto this curve is therefore drawn, startingat point A 1 , and intersecting the gridvoltage abscissa (x-axis). This intersection(C) is the operating bias pointfor fixed bias operation.7- This operating bias point may now beplotted on the original E 8vs. lp familyof curves (C 1 ), and the zero-signal currentproduced by this bias is determined.This operating bias point (C 1 ) does not fallon the operating load line, as in the case of asingle-ended amplifier.8- Under conditions of maximum power output,the exciting signal voltage swingsfrom zero-bias voltage to zero-bias voltagefor each of the tubes on each halfof the signal cycle. Second harmonicdistortion is largely cancelled out.6-13 Class B Audio FrequencyPower AmplifiersThe Class B audio-frequency power amplifier(figure 23) operates at a higher platecircuitefficiency than any of the previouslydescribed types of audio power amplifiers.Full-signal plate-circuit efficiencies of 60 toB+ DRIVER -BIAS(CROUND FORZERO BIASOPERATINGCONDITION)8+ MOO.Figure 23CLASS B AUDIO FREQUENCYPOWER AMPLIFIER70 per cent are readily obtainable with thetube types at present available for this typeof work. Since the plate circuit efficiency ishigher, smaller tubes of lower plate diss_i~ationmay be used in a Class B power ampl!f1erof a given power output than can be used inany other conventional type of audio amplifier.An additional factor in favor of the Class Baudio amplifier is the fact that the power inputto the stage is relatively low under nosignalconditions. It is for these reasons thatthis type of amplifier has largely supersededother types in the generation of audio-frequencylevels from perhaps 100 watts on up to levelsof approximately 150,000 watts as required forlarge short-wave broadcast stations.Disadvantages af There are attendant disClass B Amplifier advantageous features to theOperation operation of a power amplifierof this type; but allthese disadvantages can be overcome by properdesign of the circuits associated with thepower amplifier stage. These disadvantagesare: (1) The Class B audio amplifier requiresdriving power in its grid circuit; this disadvantagecan be overcome by the use of anoversize power stage preceding the Class Bstage with a step-down transformer betweenthe driver stage and the Class-8 grids. Degenerativefeedback is sometimes employedto reduce the plate impedance of the driverstage and thus to improve the voltage regulationunder the varying load presented by theClass B grids. (2) The Class B stage requiresa constant value of average grid bias to besupplied in spite of the fact that the grid currenton the stage is zero over most of thecycle but rises to values as high as one-thirdof the peak plate current on the peak of theexciting voltage cycle. Special regulated biassupplies have been used for this application,orB batteries can be used. However, a number
124 Vacuum Tube Amplifiers<strong>THE</strong> R AD I 0of tubes especially designed for Class B audioamplifiers have been developed which requirezero average grid bias for their operation. The811A, 838, 805, 809, HY-5514, and TZ-40 areexamples of this type of tube. All these socalled"zero-bias" tubes have rated operatingconditions up to moderate plate voltageswherein they can be operated without gridbias. As the plate voltage is increased toto their maximum ratings, however, a smallamount of grid bias, such as could be obtainedfrom several 4 ~-volt C batteries, is required.(3), A Class B audio-frequency power amplifieror modulator requires a source of platesupply voltage having reasonably good regulation.This requirement led to the developmentof the swinging choke. The swinging choke isessentially a conventional filter choke inwhich the core air gap has been reduced. Thisreduction in the air gap allows the choke tohave a much greater value of inductance withlow current values such as are encounteredwith no signal or small signal being appliedto the Class B stage. With a higher value ofcurrent such as would be taken by a Class Bstage with full signal applied the inductanceof the choke drops to a much lower value.With a swinging choke of this type, havingadequate current rating, as the input inductorin the filter system for a rectifier power supply,the regulation will be improved to a pointwhich is normally adequate for a power supplyfor a Class B amplifier or modulator stage.Cal cui at ion of OperatingConditions of Class BPower AmplifiersThe following procedurecan be used forthe calculation of theoperating conditionsof Class B power amplifiers when they are tooperate into a resistive load such as the typeof load presented by a Class C power amplifier.This procedure will be found quite satisfactoryfor the application of vacuum tubes asClass B modulators when it is desired tooperate the tubes under conditions which arenot specified in the tube operating characteristicspublished by the tube manufacturer. Thesame procedure can be used with equal effectivenessfor the calculation of the operatingconditions of beam tetrodes as Class AB 2amplifiers or modulators when the restingplate current on the tubes (no signal condition)is less than 25 or 30 per cent of themaximum-signal plate current.1- With the average plate characteristicsof the tube as published by the manufacturerbefore you, select a point onthe EP = Eg (diode bend) line at abouttwice the plate current you expect thetubes to kick to under modulation. Ifbeam tetrode tubes are concerned, selecta point at about the same amount of platecurrent mentioned above, just to theright of the region where the Ib linetakes a sharp curve downward. This willbe the first trial point, and the platevoltage at the point chosen should benot more than about 20 per cent of thed-e voltage applied to the tubes if goodplate-circuit efficiency is desired.2- Note down the value of ipmax and epmin atthis point.3- Subtract the value of epmin from the d-eplate voltage on the tubes.4- Substitute the values obtained in thefollowing equations:ipmax (Ebb - epmin)P 0= ~=Power output2 from 2 tubes(Ebb- e . )RL = 4 . pmmlpmax= Plate-to-plate load for 2 tubesFull signal efficiency (Np) =78.5 1--- epmin )(EbbEffects of Speech All the above equations areClipping true for sine-wave operatingconditions of the tubes concerned.However, if a speech clipper is beingused in the speech amplifier, or if it is desiredto calculate the operating conditions on thebasis of the fact that the ratio of peak powerto average power in a speech wave is approximately4-to-1 as contrasted to the ratio of2-to-1 in a sine wave- in other words, whennon-sinusoidal waves such as plain speech orspeech that has passed through a clipper areconcerned, we are no longer concerned withaverage power output of the modulator as faras its capability of modulating a Class-C amplifieris concerned; we are concerned with itspeak-power-output capability.Under these conditions we call upon other,more general relationships. The first of theseis: It requires a peak power output equal tothe Class-C stage input to modulate that inputfully.The second one is: The average power outputrequired of the modulator is equal to theshape factor of the modulating wave multipliedby the input to the Class-C stage. Theshape factor of unclipped speech is approximately0. 25. The shape factor of a sine waveis 0. 5. The shape factor of a speech wave that
<strong>HANDBOOK</strong>Class B Parameters 125Figure 24Typical Class B a-f amplifierload line. The load line hasbeen drawn on the overagecharacteristics of a type 811tube...:;:;080 0.!::!-0 600a:
126 Vacuum Tube Amplifiers<strong>THE</strong> R AD I 0above by the plate-to-plate load impedance forthe modulator tubes determined in (1) above.The ratio determined in this way is the secondary-co-primaryimpedance ratio. (4) Takethe square root of this ratio to determine thesecondary-to-primary turns ratio. If the turnsratio is greater than one the use of a step-uptransformer is required. If the turns ratio asdetermined in this way is less than one a stepdowntransformer is called for.If the procedure shown in figure · 25 hasbeen used to calculate the operating conditionsfor the modulator tubes, the transformer ratiocalculation can be checked in the followingmanner: Divide the plate voltage on the modulatedamplifier by the total voltage swing onthe modulator tubes: 2 (Ebb - emi 0). This ratioshould be quae close numerically to the transformerturns ratio as previously determined.The reason for this condition is that the ratiobetween the total primary voltage and the d-eplate supply voltage on the modulated stageis equal to the turns ratio of the transformer,since a peak secondary voltage equal co theplate voltage on the modulated stage is requiredto modulate this stage 100 per cent.Use of Clipper SpeechAmplifier with TetrodeModulator TubesWhen a clipper speechamplifier is used inconjunction with a ClassB modulator stage, theplate current on that stage will kick to ahigher value with modulation(due to the greateraverage power output and input) but the platedissipation on the tubes will ordinarily beless than with sine-wave modulation. However,when tetrode tubes are used as modulators,the screen dissipation will be much greaterthan with sine-wave modulation. Care mustbe taken to insure that the screen dissipationrating on the modulator tubes is not exceededunder full modulation conditions witha clipper speech amplifier. The screen dissipationis equal to screen voltage times screencurrent.Practical Aspects afClass B ModulatorsAs stated previously, aClass B audio amplifierrequires the driving stageto supply well-regulated audio power to thegrid circuit of the Class B stage. Since theperformance of a Class Bmodulator may easilybe impaired by an improperly designed driverstage, it is well to study the problems incurredin the design of the driver stage.The grid circuit of a Class B modulator maybe compared to a variable resistance whichdecreases in value as the exciting grid voltageis increased. This variable resistance appearsacross the secondary terminals of thedriver transformer so that the driver stage iscalled upon to deliver power to a varying load,For best operation of the Class B stage, thegrid excitation voltage should not drop as thepower taken by the grid circuit increases.These opposing conditions call for a high orderof voltage regulation in the driver stageplate circuit. In order to enhance the voltageregulation of this circuit, the driver tubes musthave low plate resistance, the driver transformermust have as large a step-down ratioas possible, and the d-e resistance of bothprimary and secondary windings of the drivertransformer should be low.The driver transformer should reflect intothe plate circuit of the driver tubes a load ofsuch value that the required driving power isjust developed with full excitation applied tothe driver grid circuit. If this is done, thedriver transformer will have as high a stepdownratio as is consistent with the maximumdrive requirements of the Class B stage. Ifthe step-down ratio of the driver transformer istoo large, the driver plate load will be sohigh that the power required to drive the ClassB stage to full output cannot be developed.If the step-down ratio is too small the regulationof the driver stage will be impaired.Driver StogeCalculationsThe parameters for the driverstage may be calculated fromthe plate characteristic curve, asample of which is shown in figure 24. Therequired positive grid voltage ( e g-max) for the811A tubes used in the sample calculation isfound at point X, the intersection of the loadline and the peak plate current as found on they-axis. This is + 80 volts. If a vertical lineis dropped from point X to intersect the dottedgrid current curves, it will be found that thegrid current for a single 8llA at this value ofgrid voltage is 100 milliamperes (point Y).The peak grid driving power is therefore80 x 0.100 = 8 watts. The approximate averagedriving power is 4 watts. This is an approximatefigure because the grid impedance is notconstant over the entire audio cycle.A pair of 2A3 tubes will be used as drivers,operating Class A, with the maximum excitationto the drivers occuring just below thepoint of grid current flow in the 2A3 tubes.The driver plate voltage is 300 volts, and thegrid bias is -62 volts. The peak power developedin the primary winding of the drivertransformer is:p.EgPeak Power (Pp) ~ 2RL ((watts)Rp + RLwhere p. is the amplification factor of thedriver tubes (4.2 for 2A3). E 8 is the peak gridswing of the driver stage (62 volts). RP is the);z
<strong>HANDBOOK</strong>Cathode Follower Amplifier 127plate resistance of one driver tube (800 ohms).RL is Yz the plate-to-plate load of the driverstage, and Pp is 8 watts.Solving the above equation for RL, weobtain a value of 14,500 ohms load, plate-toplatefor the 2A3 driver tubes.The peak primary voltage is:f1Egepri ~ 2RL X---- ~ 493 voltsRp + RLand the turns ratio of the driver transformer(primary to Yz secondary) is:epri 493--~ -~ 6.15:1eg(max) 80Plate Circuit One of the commonest causes ofImpedanceMatchingdistortion in a Class B modulatoris incorrect load impedancein the plate circuit. The purposeof the Class B modulation transformer is totake the power developed by the modulator(which has a certain operating impedance) andtransform it to the operating impedance imposedby the modulated amplifier stage.If the transformer in question has the samenumber of turns on the primary winding as ithas on the secondary winding, the turns ratiois 1:1, and the impedance ratio is also 1: I. Ifa 10, 000 ohm resistor is placed across thesecondary terminals of the transformer, a re·fleeted load of 10, 000 ohms would appearacross the primary terminals. If the resistoris changed to one of 2376 ohms, the reflectedprimary impedance would also be 2376 ohms.If the transformer has twice as many turnson the secondary as on the primary, the turnsratio is 2: I. The impedance ratio is the squareof the turns ratio, or 4: I. If a 10, 000 ohmresistor is now placed across the secondarywinding, a reflected load of 2,500 ohms willappear across the primary winding.Effects af Plate It can be seen from theCircuit Mis-match above paragraphs that theClass B modulator plateload is entirely dependent upon the loadplaced upon the secondary terminals of theClass B modulation transformer. If the second·ary load is incorrect, certain changes willtake place in the operation of the Class Bmodulator stage.When the modulator load impedance is toolow, the efficiency of the Class B stageis reduced and the plate dissipation of thetubes is increased. Peak plate current ofthe modulator stage is increased, and satura·tion of the modulation transformer core mayresult. "Talk-back" of the modulation trans·former may result if the plate load impedanceof the modulator stage is too low.When the modulator load impedance is toohigh, the maximum power capability of thestage is reduced. An attempt to increase theoutput by increasing grid excitation to thestage will result in peak-clipping of the audiowave. In addition, high peak voltages may bebuilt up in the plate circuit that may damagethe modulation transformer.6-14 Cathode-Foil owerPower AmplifiersThe cathode-follower is essentially a poweroutput stage in which the exciting signal isapplied between grid and ground. The plate ismaintained at ground potential with respect toinput and output signals, and the output signalis taken between cathode and ground.Types of Cathode-Follower AmplifiersFigure 26 illustrates fourtypes of cathode-followerpower amplifiers in com·mon usage and figure 27 shows the outputimpedance (R 0), and stage gain (A) of bothtriode and pentode (or tetrode) cathode-followerstages. It will be seen by inspection of theequations that the stage voltage gain is alwaysless than one, that the output impedance ofthe stage is much less than the same stageoperated as a conventional cathode-returnamplifier. The output impedance for conventionaltubes will be somewhere between100 and 1000 ohms, depending primarily onthe transconductance of the tube.This reduction in gain and output impedancefor the cathode-follower comes aboutsince the stage operates as though it has 100per cent degenerative feedback applied betweenits output and input circuit. Even though thevoltage gain of the stage is reduced to a valueless than one by the action of the degenerativefeedback, the power gain of the stage (if it isoperating Class A) is not reduced. Althoughmore voltage is required to excite a cathode·follower amplifier than appears across the loadcircuit, since the cathode "follows" alongwith the grid, the relative grid-to-cathode volt·age is essentially theventional amplifier.same as in a con·Use of Cathode- Although the cathode-folFollower Amplifiers lower gives no voltagegain, it is an effectivepower amplifier where it is desired to feed alow-impedance load, or where it is desired tofeed a load of varying impedance with a signalhaving good regulation. This latter capability
1 2 8 Vacuum Tube Amplifiers<strong>THE</strong> R AD I 0TRIODE:JJcFlJJJ+1A = -..,"'J..l'--"R'-"L __ _RL(JJ+1) +RpRo(CA.THOOE) = .UR+,(RK1+RK2) R~RKt + RK2.+ RL 1PENTODE: Ra(CATHODEF 1""""G;;"A = GM ReQFigure 27Equivalent factors lor pentocle (or tetrode)cathocle-follower power amplifiers.@Zo OF CABLEFigure 26CATHODE- FOLLOWER OUTPUTCIRCUITS FOR AUDIO ORVIDEO AMPLIFIERSmakes the cathode follower particularly effectiveas a driver for the grids of a Class Bmodulator stage.The circuit of figure 26A is the type of amplifier,either single-ended or push-pull, whichmay be used as a driver for a Class B modulatoror which may be used for other applicationssuch as feeding a loudspeaker where unusuallygood damping of the speaker is desired.If the d-e resistance of the primary ofthe transformer T 2 is approximately the correctvalue for the cathode bias resistor for the am-+B+Bplifier tube, the components Rk and Ck neednot_ be used. Figure 268 shows an arrangementwhich_ may be used to feed directly a value ofload Impedance which is equal to or higherthan the cathode impedance of the amplifiertube. The ~ al u e of Cc must be quite high,somewhat higher than would be used in a conventionalcircuit, if the frequency response ofthe circuit when operating into a low-impedanceload is to be preserved.Figures 26C and 26D show cathode-followercircuits for use with tetrode or pentode tubes.Figure 26C is a circuit similar to that shownin 26A and essentially the same commentsapply in regard to the components Rk and Ckand the primary resistance of the transformerT 2· Notice also that the screen of the tube ismaintained at the same signal potential as thecathode by means of coupling capacitor Cd·This capacitance should be large enough sothat at the lowest frequency it is desired topass through the stage its reactance will below with respect to the dynamic screen-tocathoderesistance in parallel with Rd T 2 inthis stage as well as in the circuit of figure26A should have the proper turns (or impedance)ratio to give the desired step-down orstep-up from the cathode circuit to the load.Figure 26D is an arrangement frequently usedin video systems for feeding a coaxial cable ofrelatively low impedance from a vacuum-tubeamplifier. A pentode or tetrode tube with acathode impedsnce as a cathode follower(1/Gm) approximately the same as the cableimpedance should be chosen. The 6AG7 and6AC7 have cathode impedances of the sameorder as the surge impedances of certain typesof low-capacitance coaxial cable. An arrangementsuch as 26D is also usable for feedingcoaxial cable with audio or r-f energy whereit is desired to transmit the output signalover moderate distances. The resistor Rk isadded to the circuit as shown if the cathodeimpedance of the tube used is lower than the
<strong>HANDBOOK</strong>Feedback Amplifiers 129characteristic impedance of the cable. If theoutput impedance of the stage is higher thanthe cable impedance a resistance of appropriatevalue is sometimes placed in parallelwith the input end of the cable. The valuesof Cd and Rd should be chosen with the sameconsiderations in mind as mentioned in thediscussion of the circuit of figure 26C above.The Cathode-Follower The cathode followerin R·F Stages may conveniently beused as a method of couplingr-f or i-f energy between two units separateda considerable distance. In such anapplication a coaxial cable should be used tocarry the r-f or i-f energy. One such applicationwould be for carrying the output of a v-f-oto a transmitter located a considerable distancefrom the operating position. Anotherapplication would be where it is desired tofeed a single-sideband demodulator, an FMadaptor, or another accessory with intermediatefrequency signal from a communicationsreceiver. A tube such as a 6CB6 connectedin a manner such as is shown in figure26D would be adequate for the i-f amplifiercoupler, while a 6L6 or a 6AG7 could be usedin the output stage of a v-f-o as a cathodefollower to feed the coaxial line which carriesthe v-f-o signal from the control unit to thetransmitter proper.6-15 Feedback Amp I ifiersIt is possible to modify the characteristicsof an amplifier by feeding back a portion ofthe output to the input. All components, cir·cuits and tubes included between the pointwhere the feedback is taken off and the pointwhere the feedback energy is inserted aresaid to be included within the feedback loop.An amplifier containing a feedback loop issaid to be a feedback amplifier. One stage orany number of stages may be included withinthe feedback loop. However, the difficulty ofobtaining proper operation of a feedback am·plifier increases with the bandwidth of theamplifier, and with the number of stages andcircuit elements included within the feedbackloop.Gain and Phase-shiftin Feedback AmplifiersThe gain and phaseshift of any amplifierare functions of frequency.For any amplifier containing a feedbackloop to be completely stable the gain ofsuch an amplifier, as measured from the inputback to the point where the feedback circuitconnects to the input, must be less than oneVOLTAGE AMPLIFICATION WITH FEEDBACK= _A_1-ABA= GAIN IN ABSENCE OF FEEDBACK8 = FRACTION OF OUTPUT VOLTAGE FED BACK8 IS NEGATIVE FOR NEGATIVE FEEDBACKFEEDBACK IN DECIBELS= 20 LOG (1 -A 8)= 20 LOG ~:~ ;:~6: ~~~~% ~~;Ho~~:~~~~:CKDISTORTION WITH FEEDBACK:: DISTORT I(~ WI~H~Ir FEEDBACKWHERE>R 0 =OUTPUT IMPEDANCE OF AMPLIFIER WITH FEEDBACKRN:: OUTPUT IMPEDANCE OF AMPLIFIER WITHOUT FEEDBACKRL =LOAD IMPEDANCE INTO WHICH AMPLIFIER OPERATI:":SFigure 28FEEDBACK AMPLIFIER RELATIONSHIPSat the frequency where the feedback voltageis in phase with the input voltage of the amplifier.If the gain is equal to or more thanone at the frequency where the feedback voltageis in phase with the input the amplifierwill oscillate. This fact imposes a limitationupon the amount of feedback which may beemployed in an amplifier which is to remainstable. If the reader is desirous of designingamplifiers in which a large amount of feedbackis to be employed he is referred to abook on the subject by H. W. Bode.•Types ofFeedbackFeedback may be either negativeor positive, and the feedback voltagemay be proportional either tooutput voltage or output current. The mostcommonly used type of feedback with a-f orvideo amplifiers is negative feedback proportionalto output voltage. Figure 28 givesthe general operating conditions for feedbackamplifiers. Note that the reduction in distortionis proportional to the reduction in gain ofthe amplifier, also that the reduction in theoutput impedance of the amplifier is somewhatgreater than the reduction in the gain by anamount which is a function of the ratio of theH. W. Bode, "Network Analysis and Feedback AmplifierDesign," D. Van Nostrand Co., 250 Fourth Ave.,New York 3, N.Y.
130 Vacuum Tube Amplifiers<strong>THE</strong> R AD I 008 FEEDBACK : 2.0 LOG [ Rc + RA J~MVZ Ro) J:: zo LOG ( Rz + RA(v'if;AGE G,t.,IN OF Vz)JIIE3Figure 29 illustrates a very simple and effectiveapplication of negative voltage feedbackto an output pentode or tetrode amplifierstage. The reduction in hum and distortionmay amount to 15 to 20 db. The reduction inthe effective plate impedance of the stage willbe by a factor of 20 to 100 dependent upon theoperating conditions. The circuit is commonlyused in commercial equipment with tubes suchas the 6SJ7 for V 1 and the 6V6 or 6L6 for V 2 .6-16 Vacuum-Tube VoltmetersGAIN OF BOTH STAGES = ( GMV1 (~)J X (GMVZ Ro)R8 +R,.,RA = :: ~ ::Ra= _R_2 __GMVz RoRo = REFLECTED LOAD IMPEDANCE ON VzRz:: fEEDBACK RESISTOR (USUALLY ABOUT ~00 K)OUTPUT IMPEDANCE :: RN Rz(Rz+R...,(GMVzRo))X (1+ ~~)RN = PLATE IMPEDANCE OF Vzfigure 29SHUNT FEEDBACK CIRCUITFOR PENTODES OR TETRODESThis circuit requires only the addition ofone resistor, R2, to the norma/ circuit forsuch an application, The plate Impedanceand distortion Introduced by the outputstage are materially reduced.output imp e dan c e of the amplifier withoutfeedback to the load impedance. The reductionin noise and hum in those stages includedwithin the feedback loop is proportional to thereduction in gain. However, due to the reduc·tion in gain of the output section of the amplifiersomewhat increased gain is required ofthe stages preceding the stages included with·in the feedback loop. Therefore the noise andhum output of the entire amplifier may or maynot be reduced dependent upon the relativecontributions of the first part and the latterpart of the amplifier to hum and noise. If mostof the noise and hum is coming from the stagesincluded within the feedback loop the undesiredsignals will be reduced in the outputfrom the complete amplifier. It is most frequentlytrue in conventional amplifiers thatthe hum and distortion come from the latterstages, hence these will be reduced by feedback,but thermal agitation and microphonicnoise come from the first stage and will notbe reduced but may be increased by feedbackunless the feedback loop includes the firststage of the amplifier.The vacuum-tube voltmeter may be consideredto be a vacuum-tube detector in which therectified d-e current is used as an indicationof the magnitude of the applied alternatingvoltage. The vacuum tube voltmeter (v.t.v.m.)consumes little or no power and it may becalibrated at 60 cycles and used at audio orradio frequencies with little change in thecalibration.Basic D-C VacuumTube VoltmeterA simple v.t.v.m. isshown in figure 30.The plate load may bea mechanical device, such as a relay or ameter, or the output voltage may be developedacross a resistor and used for various controlpurposes. The tube is biased by Ec anda fixed value of plate current flows, causinga fixed voltage drop across the plate loadresistor, Rp. When a positive d-e voltage isapplied to the input terminals it cancels partof the negative grid bias, making the gridmore positive with respect to the cathode.This grid voltage change permits a greateramount of plate current to flow, and developsa greater voltage drop across the plate loadresistor. A negative input voltage would decreasethe plate current and decrease thevoltage drop across Rp. The varying voltagedrop across Rp may be employed as a controlvoltage for relays or other devices. When it isdesired to measure various voltages, a voltageFigure 30SIMPLE VACUUM TUBEVOLTMETER
<strong>HANDBOOK</strong>Vacuum Tube Voltmeters 131ZERO-ADJUSTFigure 31D-C VACUUM TUBE VOLTMETERrange switch (figure 31) may precede the v.t.v.m. The voltage to be measured is applied tovoltage divider, R1, R2, R3, by means of the"voltage range" switch. Resistor R4 is usedto protect the meter from excessive inputvoltage to the v.r.v.m. In the plate circuit ofthe tube an additional battery and a variableresistor ("zero adjustment") are used tobalance out the meter reading of the normalplate current of the tube. The zero adjustmentpotentiometer can be so adjusted that themeter M reads zero current with no input voltageto the v.t.v.m. When a d-e input voltageis applied to the circuit, current flows throughthe meter, and the meter reading is proportionalto the applied d-e voltage.The Bridge-type Another important useV.T.V.M.of a d-e amplifier is toshow the exact point ofbalance between two d-e voltages. This isdone by means of a bridge circuit with twod-e amplifiers serving as two legs of thebridge (figure 32). With no input signal, andwith matched triodes, no current will be readon meter M, since the IR drops across R1 andR2 are identical. When a signal is applied toone rube, the IR drops in the plate circuitsbecome unbalanced, and meter M indicatesthe unbalance. In the same way, two d-e voltagesmay be compared if they are applied tothe two input circuits. When the voltages areequal, the bridge is balanced and no currentflows through the meter. If one voltage changes,the bridge becomes unbalanced and indicationof this will be noted by a reading of the meter.A Modern VTVM For the purpose of analysis,the operation of a modernv.r.v.m. will be described. The Heathkit V-7Ais a fit instrument for such a description, sinceit is able to measure positive or negative d-epotentials, a-c r-m-s values, peak-to-peakvalues, and resistance. The circuit of thisunit is shown in figure 33. A sensitive 200 d-e'-------iilllllt------'+Figure 32BRIDGE-TYPE VACUUM TUBEVOLTMETERmicroammeter is placed in the cathode circuitof a 12AU7 twin triode. The zero adjust controlsets up a balance between the two sections ofthe triode such that with zero input voltageapplied to the first grid, the voltage drop acrosseach portion of the zero adjust control is thesame. Under this condition of balance themeter will read zero. When a voltage is appliedto the first grid, the balance in the cathodecircuits is upset and the meter indicates thedegree of unbalance. The relationship betweenthe applied voltage on the first grid and themeter current is linear and therefore the metercan be calibrated with a linear scale. Sincethe tube is limited in the amount of currentit can draw, the meter movement is electronicallyprotected.The maximum test voltage applied to the12AU7 rube is about 3 volts. Higher appliedvoltages are reduced by a voltage dividerwhich has a total resistance of about 10megohms. An additional resistance of 1-megohmis located in the d-e test prod, there by permittingmeasurements to be made in high impedancecircuits with minimum disturbance.The rectifier portion of the v.t.v.m. is shownin figure 34. When a-c measurements are desired,a 6AL5 double diode is used as a fullwave rectifier to provide a d-e voltage proportionaltothe applied a-c voltage. This d-evoltage is applied through the voltage dividerstring to the 12AU7 tube causing the meter toindicate in the manner previously described.The a-c voltage scales of the meter are calibratedin both RMS and peak-to-peak values.In the 1.5, 5, 15, 50, and 150 volt positionsof the range switch, the full a-c voltage beingmeasured is applied to the input of the 6AL5full wave rectifier. On the 500 and 1500 voltpositions of the range switch, a divider networkreduces the applied voltage in order tolimit the voltage input to the 6AL5 to a saferecommended level.
132<strong>THE</strong> R AD I 0,-- --- - - - - - - - - --- -- - - - -- ------_;-- ---- -- - -----z0_,i'1(/)o•Ir--0L_~·~~~~~~~ I•II~ 't-VMivVVVIMVV'f'N'v-+-WM~r>'0N'fv"NifNIHEATHKIT PEAK-TO-PEAK VTVMMODEL V-7AFigure 33
<strong>HANDBOOK</strong>Vacuum Tube Voltmeters 1 33TOVTVMFigure 34FULL-WAVE RECTIFIERFOR V.T.V.M.The a-c calibrate control (figure 33) is usedto obtain the proper meter deflection for theapplied a-c voltage. Vacuum tubes develop acontact potential between tube elements. Suchcontact potential developed in the diode wouldcause a slight voltage to be present at alltimes. This voltage is cancelled out by properapplication of a bucking voltage. The amountof bucking voltage is controlled by the a-cbalance control. This eliminates zero shiftof the meter when switching from a-c to d-ereadings.For resistance measurements, a 1.5 voltbattery is connected through a string of multi·pliers and the external resistance to be meas·ured, thus forming a voltage divider acrossthe battery, and a resultant portion of thebattery voltage is applied to the 12AU7 twinFigure 35R-F PROBE SUITABLEFOR USE IN IKC-100 MCRANGEtriode. The meter scale is calibrated in resistance(ohms) for this function.Test Probes Auxiliary test probes maybe used with the v.t.v.m.to extend the operating range, or to measureradio frequencies with high accuracy. Shownin figure 35 is a radio frequency probe whichprovides linear response to over 100 mega·cycles. A crystal diode is used as a rectifier,and d-e isolation is provided by a .005 uufdcapacitor. The components of the detector aremounted within a shield at the end of a lengthof coaxial line, which terminates in the d-einput jack of the v.t.v.m. The readings obtainedare RMS, and should be multiplied by1.414 to convert to peak readings.
CHAPTER SEVENHigh fidelityTechniquesThe art and science of the reproduction ofsound has steadily advanced, following themajor audio developments of the last decade.Public acceptance of home music reproductionon a "high fidelity" basis probably dates fromthe summer of 1948 when the Columbia L-Pmicrogroove recording techniques were introduced.The term high fidelity refers to the reproductionof sound in which the different distortionsof the electronic system are held belowlimits which are audible to the majority oflisteners. The actual determination, therefore,of the degree of fidelity of a music system islargely psychological as it is dependent uponthe ear and temperament of the listener. By andlarge, a rough area of agreement exists as towhat boundaries establish a "hi-fi" system. Toenumerate these boundaries it is first necessaryto examine sound itself.7-1 The Nature of SoundExperiments with a simple tuning fork inthe seventeenth century led to the discoverythat sound consists of a series of condensationsand rarefactions of the air brought about bymovement of air molecules. The vibrations ofthe prongs of the fork are communicated tothe surrounding air, which in turn transmitsthe agitation to the ear drums, with the resultthat we hear a sound. The vibrating fork producesa sound of extreme regularity, and thisregularity is the essence of music, as opposed tonoise which has no such regularity.134As shown in figure 1, the sound wave ofthe fork has frequency, period, and pitch. Thefrequency is a measure of the number of vibrationsper second of the sound. A fork tunedto produce 261 vibrations per second is tunedto the musical note of middle-C. It is of interestto note that any object vibrating, moving,or alternating 261 times per second will producea sound having the pitch of middle-C.The pitch of a sound is that property which isdetermined by the frequency of vibration ofthe source, and not by the source itself. Thusan electric dynamo producing 261 c.p.s. willhave a hum-pitch of middle-C, as will a siren,a gasoline engine, or other object having thesame period of oscillation.iII))) ) ) )))))) )))))~TUNIN
Nature of Sound 135FREQUENCY (cYCLES PER SECONO)NOTE c D E F G A B C'EQUAL-TEMPERED 2.61.6 293.7 32.9.6 349.2 392.0 440.0 493.9 523.2SCALEFigure 2<strong>THE</strong> EQUAL-TEMPERED SCALE CONTAINS TWELVE INTERVALS, EACH OFWHICH IS 1.06 TIMES <strong>THE</strong> FREQUENCY OF <strong>THE</strong> NEXT LOWEST. <strong>THE</strong> HALFTONE INTERVALS INCLUDE <strong>THE</strong>ABOVE NOTES PLUS FIVE ADDITIONAL NOTES: 277.2, 311.1, 370, 415.3,466.2 REPRESENTED BY <strong>THE</strong> BLACKKEYS OF <strong>THE</strong> PIANO.The MusicalScaleThe musical scale is composedof notes or sounds of variousfrequencies that bear a pleasingaural relationship to one another. Certain combinationsof notes are harmonious to the earif their frequencies can be expressed by thesimple ratios of 1:2,2:3, 3:4, and 4:5. Notesdiffering by a ratio of 1 :2 are said to be separatedby an octave.The frequency interval represented by anoctave is divided into smaller intervals, formingthe musical scales. Many types of scaleshave. been proposed and used, but the scale ofthe piano has dominated western music for thelast hundred or so years. Adapted by J. S.Bach, the equal-tempered scale (figure 2) hastwelve notes, each differing from the next bythe ratio 1: 1.06. The reference frequency, orAmerican Standard Pitch is A, or 440.0 cycles.Harmonics andOvertonesThe complex sounds producedby a violin or a windinstrument bear little resemblanceto the simple sound wave of the tuningfork. A note of a clarinet, for example (whenviewed on an oscilloscope) resembles figure 3.Vocal sounds are even more complex than this.In 1805 Joseph Fourier advanced his monumentaltheorem that made possible a mathematicalanalysis of all musical sounds by showingthat even the most complex sounds aremade up of fundamental vibrations plus harmonics,or overtones. The tonal qualities ofany musical note may be expressed in terms ofthe amplitude and phase relationship betweenthe overtones of the note.To produce overtones, the sound source mustbe vibrating in a complex manner, such as isshown in figure 3. The resulting vibration isa combination of simple vibrations, producinga rich tone having fundamental, the octavetone, and the higher overtones. Any sound -Figure 3<strong>THE</strong> COMPLEX SOUND OF A MUSICALINSTRUMENT IS A COMBINATION OFSIMPLE SINE-WAVE SOUNDS, CALLEDHARMONICS. <strong>THE</strong> SOUND OF LOWESTFREQUENCY IS TERMED <strong>THE</strong> FUNDAMENTAL. <strong>THE</strong> COMPLEX VIBRATIONOF A CLARINET REED PRODUCES ASOUND SUCH AS SHOWN ABOVE.no matter how complex - can be analyzedinto pure tones, and can be reproduced by agroup of sources of pure tones. The numberand degree of the various harmonics of a toneand their phase relationship determine thequality of the tone.For reproduction of the highest quality, theseovertones must be faithfully reproduced. A musicalnote of 523 cycles may be rich in twentiethorder overtones. To reproduce the originalquality of the note, the audio system mustbe capable of passing overtone frequencies ofthe order of 11,000 cycles. Notes of higherfundamental frequency demand that the audiosystem be capable of good reproduction up tothe maximum response limit of the humanear, in the region of 15,000 cycles.ReproductionLimitationsMany factors enter into theproblem of high quality audioreproduction. Most importantof these factors influence the overall design ofthe music system. These are:!-Restricted frequency range.2-Nonlinear distortions.3-Transient distortion.4-Nonlinear frequency response.5-Phase distortion.6-Noise, "wow", and "flutter".A restricted frequency range of reproductionwill tend to make the music sound "tinny"and unrealistic. The fundamental frequencyrange covered by the various musical instrumentsand the human voice lies between 15cycles and 9,000 cycles. Overtones of the instrumentsand the voice extend the upperaudible limit of the music range to 15,000cycles or so. In order to fully reproduce themusical tones falling within this range of frequenciesthe music system must be capable offlawlessly reproducing all frequencies withinthe range without discrimination.
136 High Fidelity Techniques<strong>THE</strong> <strong>RADIO</strong>BASIC LIMITS FOR HI(;H FIDELITY AND"GOOD QUALITY'' REPRODUCTIONLIMITTYPE OFDISTORTION HI
<strong>HANDBOOK</strong>High Fidelity Amplifier 137CONSTANT AMPL!TUD/ /CRFORSEC:,g~~~C.Ywt--------~-~,0 Ij: ICONSTANTVELOCITY50 100 200 500 1000 2000 5000FREQUENCY (CPs)CONSTANT VELOCITY RECORDIN~50 100 200 500 1000 2000 5000FREQUENCY (CPS)CONSTANT AMPLITUDE BELOW CROSSOVERFREQUENCY, CONSTANT VELOCITY ABOVECROSSOVER FREQUENCY®w0::>I-w0::>1--.Ja.::; 0a.1-::>-50w -10>~-1
138 High Fidelity Techniques<strong>THE</strong> <strong>RADIO</strong>Figure 6NEW CRYSTAL "TRANSDUCER"CARTRIDGE PROVIDES HIGHFIDELITY OUTPUT AT RELATIVELYHIGH LEVELorder of one-half volt or so. Inexpensive crystalunits used in 78 r.p.m. record changers andac-dc phonographs may have as much as twoor three volts peak output. The frequency responseoi a typical high quality crystal pickupis shown in figure 7.The variable reluctance pickup is shown infigure 8. The reluctance of the air gap in amagnetic circuit is changed by the movementof the phonograph needle, creating a variablevoltage in a small coil coupled to the magneticlines of force of the circuit. The output impedanceof the reluctance cartridge is of theorder of a few hundred ohms, and the outputis approximately 10 millivolts.For optimum performance, an equalized preamplifierstage is usually employed with thereluctance pickup. The circuit of a suitable unitis shown in figure 9. Equalization is providedby Rs, R., and C3, with a low frequency crossoverat about 500 cycles. Total equalization is15 db. High frequency response may be limitedby reducing the value of R, to 5,000- 15,000ohms.The standard pickup stylus for 78 r.p.m.records has a tip radius of .0025 inch, whereasthe microgroove (33 113 and 45 r.p.m.) stylushas a tip radius of .001 inch. Many pickups,rll:::::: 141ZH==+t -1-12.0 50 100 2.00 500 1000 2000 5000 10000FREQUENCY (cps)Figure 1FREQUENCY RESPONSE OF HIGHQUALITY CRYSTAL PHONOGRAPHCARTRIDGE. !ELECTROVOICE 56-DSPOWER POINT TRANSDUCER)Figure 8"RELUCTANCE" CARTRIDGE ISSTANDARD PICK-UP FORMUSIC SYSTEM.Low stylus pressure of four grams insuresminimum record wear. Dual stylus is usedhaving two needle tip diameters for longplaying and 78 R.P.M. recordings.therefore, are designed to have interchangeablecartridges or needles to accomodate the differentgroove widths.7-3 The High Fidelity AmplifierA block diagram of a typical high fidelitysystem is shown in figure 10. A preamplifieris used to boost the output level of the phonographpickup, and to permit adjustment of inputselection, volume, record compensation,and tone control. The preamplifier may bemounted directly at the phonograph turntableposition, permitting the larger power amplifierto be placed in an out of the way position.The power amplifier is designed to operatefrom an input signal of a volt or so derivedfrom the preamplifier, and to build this signalto the desired power level with a minimumamount of distortion. Maximum power outputlevels of ten to twenty watts are common forhome music systems.The power supply provides the ,smoothed,d-e voltages necessary for operation of the preamplifierand power amplifier, and also the,OUTPUTB+ 100 V.Figure 9PREAMPLIFIER SUITABLE FOR USE WITHLOW LEVEL RELUCTANCE CARTRIDGE.
<strong>HANDBOOK</strong>High Fidelity Amplifier 139I PHONOGRAPH :-=t_L __ _j I -, ~.~PREAMPLIFIER..__ AMPLIFIERr--·-·~L'":__j~115 v. '\...Figure 10BLOCK DIAGRAM OF HIGH FIDELITYMUSIC SYSTEM.R3EQUIVALENT CIRCUITSATTENUATER1R1IN.f IN.rOUT.R~ C~ R~ C1our.R> R3= -=TREBLE BOOSTAND ATTENUATIONINPUTl_BOOSt-OUTPUTATTENUATErIN.} ouT IN.TFigure 11SIMPLE R-C CIRCUITS MAY BEUSED FOR BASS AND TREBLEBOOST OR ATTENUATION.1 }ouTl Ifilament voltages (usually a-c) for the heatersof the various amplifier tubes.The loudspeaker is a device which couplesthe electrical energy of the high fidelity systemto the human ear and usually limits theoverall fidelity of the complete system. Greatadvances in speaker design have been made inthe past years, permitting the loudspeaker towVlz0Q_~ -10f--t---+a:-~OL___L _ __j_L_L _ _t_ __ _J__ _J_ ___ L__10 20 50 100 200 500FREQUENCY (cps)Figure 12FREQUENCY RESPONSE CURVES FOR<strong>THE</strong> BASS AND TREBLE BOOSTAND ATTENUATION CIRCUITSOF FIGURE 11.hold its own in the race for true fideliry.Speaker efficiency runs from about 10% forcone units to nearly 40% for high frequencytweeters. The frequency response of any speakeris a function of the design and constructionof the speaker enclosure or cabinet that mountsthe reproducer.ToneCompensatonEqualizer networks are employedin high fidelity equipmentto ll-tailor the responsecurve of the system to obtain the correctoverall frequency response, 2) -to compensatefor inherent faults in the program material,3 ) -or merely to satisfy the hearing preferenceof the listener. The usual compensation networksare combinations of RC and RL networksthat provide a gradual attenuation overa given frequency range. The basic RC networkssuitable for equlizer service are shownin figure 11. Shunt capacitance is employedfor high frequency attenuation, and seriescapacitance is used for low frequency attenuation.A combination of these simple a-cvoltage dividers may be used to provide almostany response, as shown in figure 12. It iscommon practice to place equalizers betweentwo vacuum tubes in the low level stages ofthe preamplifier, as shown in figure 13. Bassand treble boost and attenuation of the order.04r.--....--lf-- OUTPUTFigure 13BASS AND TREBLE LEVELCONTROLS, ASEMPLOYED IN <strong>THE</strong>HEATHKIT WA-P2PREAMPLIFIER.3.3KB+220 KBASS CONTROLTREBLE CONTROL
140 High Fidelity Techniques <strong>THE</strong> <strong>RADIO</strong>2N 190 2N 190 -12. v 2N19018K18Kt--"{\/1/'----t"---jh:-- 0 U T PUT5BASSTREBLEFigure 14TRANSISTORIZED HIGH-FIDELITY PREAMPLIFIER FOR USE WITHRELUCTANCE PHONOGRAPH CARTRIDGE.+II 0+100-..... ./"PAl N LEVEL~ +9 0:J +-8 0uJ> +7 0uJ_j +6 0~ +5 0"Vl +4 0zuJ +3 0f."" ~1'-v,/~~~~~N~lMtT OF~ +2 00 +I 0 "'-....1 1/z_b, _/:::J 00Vl -1 0 !"-=:2.0 50 100 2.00 500 1 00 2000 5000 1000020000FREQUENCY (CPS)Figure 15<strong>THE</strong> "FLETCHER-MUNSON" CURVEILLUSTRATING <strong>THE</strong> INTENSITYRESPONSE OF <strong>THE</strong> HUMAN EAR.of 15 db may be obtained from such a circuit.A simple transistorized preamplifier usingthis type of equalizing network is shown infigure 14.IN PUT >----10.2R 1 :E------1'!'!'--'500K 100K R2.IMOUTPUTLoudnessCompensotonThe minimum threshold ofhearing and the maximumthreshold of pain vary greatlywith the frequency of the sound as shownin the Fletcher-Munson curves of figure 15.To maintain a reasonable constant tonal balanceas the intensity of the sound is changedit is necessary to employ extra bass and trebleboost as the program level is decreased. Asimple variable loudness control is shown infigure 16 which may be substituted for theordinary volume control used in most audioequipment.The PowerAmplifierThe power amplifier stage ofthe music system must supplydriving power for the loudspeaker.Commercially available loudspeakersare low impedance devices which present a~+10r---,--,---,----,--,---,--J,_---,--,~ + 5·t---t-' +---+------t----t---t- ---t~-----t----r---JZ o I L':'::c L"o. ~j.----, _bV') _ 5 __j______J_ L LsPEAKER --t----t- '1 ~~ ~ ,~ /"-:' 11"+:i! alL : RJESPONSJE~-1 ·~--~--~--~--~--~--~--L---~uJ:.::
<strong>HANDBOOK</strong>High Fidelity Amplifier 141FEEDBACK RESISTORFigure 18TYPICAL TRIODEAMPLIFIER WITHFEEDBACK LOOP.~~---,~O~K----~----~~K~------._--~~i~i:MA:~~F~ ~~~~~*=MATCHED PAIR RESISTORS1 15 V, 'Vvarying load of two to nearly one hundredohms to the output stage (figure 17). It isnecessary to employ a high quality outputtransformer to match the loudspeaker load tothe relatively high impedance plate circuit ofthe power amplifier stage. In general, pushpullamplifiers are employed for the outputstage since they have even harmonic cancellingproperties and permit better low frequencyresponse of the output transformer since thereis no d -c core saturation effect present.To further reduce the harmonic distortionand intermodulation inherent in the amplifiersystem a negative feedback loop is placedaround one or more stages of the unit. Frequencyresponse is thereby improved, and theoutput impedance of the amplifier is sharplyreduced, providing a very low source impedancefor the loudspeaker.Shown in figure 18 is a basic push-pulltriode amplifier, using inverse feedback aroundthe power output and driver stage. A simpletriode inverter is used to provide 180-degreephase reversal to drive the grid circuit of thepower amplifier stage. Maximum undistortedpower output of this amplifier is about 8 watts.A modification of the basic triode amplifieris the popular Williamson circuit (figure 19)developed in England in 1947. This circuitrapidly became the "standard of comparison"in a few short years. Pentode power tubes areconnected as triodes for the output stage, andnegative feedback is taken from the secondaryof the output transformer to the cathode ofthe input stage. Only the most linear portionof the tube characteristic curve is used. Althoughthat portion has been extended byhigher than normal plate supply voltage, itFEEDBACK RESISTOR5 K-15K807~a+400 v.AT 140 MA.Figure 19U. S. VERSION OF BRITISH "WILLIAMSON" AMPLIFIER PROVIDES 10 WATTS POWEROUTPUT AT LESS THAN 2% INTERMODULATION DISTORTION. 6SN7 STAGE USESDIRECT COUPLING.
142 High Fidelity Techniques <strong>THE</strong> <strong>RADIO</strong>807/5881 TO FEEDBACKCIRCUITFROM6SN7C.TPHASEINVERTERI ~0"""'~~._--~~~~~~0.2.&807/5881NOTE: PIN CONNECTIONS AREFOR 807 ruBES+400V.Figure 20"ULTRA-LINEAR" CONFIGURATION OF WILLIAMSON AMPLIFIER DOUBLES POWER OUTPUT, AND REDUCES IM LEVEL. SCREEN TAPS ON OUTPUT TRANSFORMER PERMIT"SEMI-TETRODE" OPERATION.is only a fraction of the curve normally usedin amplifiers. Thus a comparatively low outputpower level is obtained with tubes capable ofmuch more efficient operation under lessstringent requirements. With 400 volts appliedto the output stage, a power output of 10watts may be obained wtih less than 2% intermodulationdistortion.A recent variation of the Williamson circuitinvolves the use of a tapped output transformer.The screen grids of the push-pull amplifierstage are connected to the primary taps,allowing operating efficiency to approach thatof the true pentode. Power output in excessof 25 watts at less than 2% intermodulation dis-Figure 21"BABY HI-FI" AMPLIFIER IS DWARFEDBY 12-INCH SPEAKER ENCLOSUREThis miniature music system is capable of excellentperformance in the small home orapartment. Preamplifier, bass and treble controls,and volume control are all incorporatedin the unit. Amplifier provides 4 watts outputat 4% IM distortion.tortion may be obtained with this circuit(figure 20) .7-4 Amplifier ConstructionWiringTechniquesAssembly and layout of highfidelity audio amplifiers followsthe general technique describedfor other forms of electronic equipment.Extra care, however, must be taken to insurethat the hum level of the amplifier is extremelylow. A good hi-fi system has excellent responsein the 60 cycle region, and even aminute quantity of induced a-c voltage will bedisagreeably audible in the loudspeaker. Spuriouseddy currents produced in the chassis bythe power transformer are usually responsiblefor input stage hum.To insure the lowest hum level, the powertransformer should be of the "upright" typeinstead of the "half-shell" type which canwuple minute voltages from the windings toa steel chassis. In addition, part of the windingsof the half-shell type project below the chassiswhere they are exposed to the input wiring ofthe amplifier. The core of the power transformershould be placed at right angles to thecore of a nearby audio transformer to reducespurious coupling between the two units to aminimum.It is common practice in amplifier designto employ a ground bus return system for allaudio tubes. All grounds are returned to asingle heavy bus wire, which in turn isgrounded at one point to the metal chassis.This ground point is usually at the input jackof the amplifier. When this system is used,a-c chassis currents are not coupled into theamplifying stages. This type of construction isillustrated in the amplifiers described later inthis chapter.
<strong>HANDBOOK</strong>Amplifier Construction 14312AU7.05 220.0110 KBASETREBLEVOLUMECONTROLL-C~O~N~T~R~O~L ___ c~O~N~T~R~O~L~------~+~7~5 s~v.~.-+~~~~+205V.4;gi w~1. ALL RESISTORS 0.5 WATTUNLESS O<strong>THE</strong>RWISE SPECIFIED2. ALL CAPACITOR VALUES IN MFUNLESS O<strong>THE</strong>RWISE SPECIFIED3. RESISTORS MARKED *AREMATCHED PAIRSFigure 22SCHEMATIC, "BABY HI-FI" AMPLIFIERT,-260-0-260 volts at 90 ma., 6.3 volts at 4.0 CH,-1.5 henty at 200 ma. Chicago Standatd C-2327.amp., upright mounting. Chicago-Standatd C,A-B-C-30-20-10 p.fd. 350 volt. Malloty Fp-330.7PC-8420.T:-10 K, CT. to 8, 16 ohms. Peerless (A/tee)NOTE-Feedback loop returns to 8 ohm tap on TzS-510F.when 8 ohm speaker is used.Care should be taken to reduce the capacitanceto the chassis of high impedance circuits,or the high frequency response of the unit willsuffer. Shielded "bath-tub" type capacitorsshould not be used for interstage coupling capacitors.Tubular paper capacitors are satisfactory.These should be spaced well away fromthe chassis.It is a poor idea to employ the chassis as acommon filament return, especially for lowlevel audio stages. The filament center-tap ofthe power transformer should be grounded,and twisted filament wires run to each tubesocket. High impedance audio components andwiring should be kept clear of the filamentlines, which may even be shielded in the vicinityof the input stage. In some instances, thefilament center tap may be taken from the armof a low resistance, wirewound potentiometerplaced across the filament pins of the inputtube socket. The arm of this potentiometer isgrounded, and the setting of the control is adjustedfor minimum speaker hum.7-5 The 11 Baby Hi Fi 11A definite need exists for a compact, highfidelity audio amplifier suitable for use in thesmall home or apartment. Listening tests haveshown that an average power level of less thanone watt in a high efficiency speaker will providea comfortable listening level for a smallroom, and levels in excess of two or three wattsare uncomfortably loud to the ear. The "BabyHi-Fi" amplifier has been designed for usein the small home, and will provide excellentquality at a level high enough to rattle thewindows.Designed around the new Electro-Voice miniatureceramic cartridge, the amplifier will provideover 4 watts power, measured at the secondaryof the output transformer. At this level,the distortion figure is below 1%, and the IMfigure is 4%. At normal listening levels, the IMis much lower, as shown in figure 24.The AmplifierCircuitThe schematic of the amplifieris shown in figure 22.Bass and treble boost controlsare incorporated in the circuit, as is thevolume control. A dual purpose 12AU7 doubletriode serves as a voltage amplifier with cathodedegeneration. A simple voltage dividernetwork is used in the grid circuit to preventamplifier overloading when the ceramic cartridgeis used. The required input signal formaximum output is of the order of 0.3 volts.The output level of the Electro-Voice cartridgeis approximately twice this, as shown in figure7. The use of the high-level cartridge elimin-
144 High Fidelity Techniques<strong>THE</strong> <strong>RADIO</strong>Figure 23UNDER-CHASSISVIEW OF"BABY HI-FI"Low level audio stages are atupper left, with componentsmounted between socket pinsand potentiometer controls.6X5 socket is at lower centerof photo with filterchoke CH, at right. Feedbackresistor Rt is at leftof rectifier socket.ates the necessity of high gain amplifiers requiredwhen low level magnetic pickup headsare used. Problems of hum and distortion introducedby these extra stages are therebyeliminated, greatly simplifying the amplifier.The second section of the 12AU7 is used forbass and treble boost. Simple R-C networksare placed in the grid circuit permitting gainboost of over 12 db at the extremities of theresponse range of the amplifier.A second 12AU7 is employed as a directcoupled "hot-cathode" phase inverter, capacitivelycoupled to two 6AQ5 pentode connectedz0j:00::!!a:1- "' z1••1•/5/./·--- - ---I-"v32'vl_/1""vIIII2 3 •EQUIVALENT SINE WAVE WATTSIIFigure 24INTERMODULATION CURVE FOR"BABY HI-FI" AS MEASURED ONHEATHKIT INTERMODULATIONANALYZER.I/output tubes. The feedback loop is run fromthe secondary of the output transformer to thecathode of the input section of the phase inverter.The power supply of the "Baby Hi-Fi" consistsof a 6X5-GT rectifier and a capacitor inputfilter. A second R-C filter section is usedto smooth the d-e voltage applied to the12AU7 tubes. A cathode-type rectifier is usedin preference to the usual filament type to preventvoltage surges during the warm-up periodof the other cathode-type tubes.AmplifierConstructionThe complete amplifier isbuilt upon a small "amplifierfoundation" chassis and covermeasuring 5"x7"x6" (Bud CA-1754). Heightof the amplifier including dust cover is 6".The power transformer (T,) and output transformer(T,) are placed in the rear corners ofthe chassis, with the •6X5-GT rectifier socketplaced between them. The small filter choke( CH, ) is mounted to the wall of the chassisand may be seen in the under-chassis photographof figure 23. The four audio tubes areplaced in a row across the front of the chassis.Viewed from the front, the 12AU7 tubes areto the left, and the 6AQ5 tubes are to the right.The three section filter capaoitor ( CA, B, C)is a chassis mounting unit, and is placed betweenthe rectifier tube and the four audiotubes. Since the chassis is painted, it is importantthat good grounding points be made ateach tube socket. The paint is cleared away
<strong>HANDBOOK</strong>118 a b y H' 1-F·//1 145Figure 25TYPICALINTERMODULATIONTEST OF AUDIOAMPLIFIER.Audio tones of two frequenciesare applied to inputof amplifier under test,and amplitude of 1 'sum" or11dilference" frequency ismeasured, providing relativeinter-modulation figure.beneath the socket bolt heads, and lock nutsare used beneath the socket retaining nuts toinsure a good ground connection. All groundleads of the first 12AU7 tube are returned tothe socket, whereas all grounds for the rest ofthe circuit are returned to a ground lug offilter capacitor C.Since the input level to the amplifier is ofthe order of one-half volt, the problem ofchassis ground currents and hum is not soprevalent, as is the case with a high gain inputstage.Phonograph-type coax i a 1 receptacles aremounted on the rear apron of the chassis, servingas the input and output connections. Thefour panel controls (bass boost, treble boost,volume, and a-c on) are spaced equidistantacross the front of the chassis.AmplifierWiringThe filament wiring should bedone first. The center-tap of thefilament winding is grounded toa lug of the 6X5-GT socket ring, and the 6.3volt leads from the transformer are attachedto pins 2 and 7 of the same socket. A twistedpair of wires run from the rectifier socket tothe right-hand 6AQ5 socket (figure 23). Thefilament leads then proceed to the next 6AQ5socket and then to the two 12AU7 sockets inturn.The 12AU7 preamplifier stage is wired next.A two terminal phenolic tie-point strip ismounted to the rear of the chassis, holding the12K decoupling resistor and the positive leadof the 10 p,fd., 450-volt filter capacitor. AllB-plus leads are run to this point. Most of thecomponents of the bass and treble boost systemmay be mounted between the tube socket terminalsand the terminals of the two potentiometers.The feedback resistor R1 is mountedbetween the terminal of the coaxial outputconnector and a phenolic tie-point strip placedbeneath an adjacent socket bolt.When the wiring has been completed andchecked, the amplifier should be turned on,and the various voltages compared with thevalues given on the schematic. It is importantthat the polarity of the feedback loop is correct.The easiest way to reverse the feedback polarityFigure 2620-WATT "WILLIAMSON-TYPEAMPLIFIER PROVIDES ULTIMATE INLISTENING PLEASURE FOR <strong>THE</strong>"GOLDEN EAR."Amplifier chassis (left) employs two low levelstages driving push-pull 807 tubes in so-called11 U/tra-linear11 circuit. Power supply is at right.
146 High Fidelity TechniquesINPUTPUT470NOTES'1- A~L. RESISTORS 1-WATT UNLESSO<strong>THE</strong>RWISE SPEC I Fl EO2.- RESISTORS MARKED * AREMATCHED PAl RSBUS5- ALL CAPACITOR VALUES GIVEN INMFD, UNLESS O<strong>THE</strong>RWISE NOTEDe- ~=~~~o:~~~Mrk~ STANCOR A-8072.1 4 3 2.3- VOLTAc;E MEASUREMENTS MADEWITH 1000 OHMS/VOLT VOLTMETER7- PUNCHED AND DRILLED CHASSIS,STANCOR WMB4- JACKS J1 AND J2. ARE INSULATEDFROM CHASSISFigure 27SCHEMATIC OF 20-WATT MUSIC SYSTEM AMPLIFIER.is to cross-connect the two plate leads of the6AQ5 tubes. If the feedback polarization isincorr~t, the amplifier will oscillate at a supersonicfrequency and the reproduced signal willsound fuzzy to the ear. The correct connectionmay be determined with the aid of an oscilloscope,as the oscillation will be easily found.The builder might experiment with differentvalues of feedback resistor R., especially if aspeaker of different impedance is employed.Increasing the value of R. will decrease thedegree of feedback. For an 8-ohm speaker, R,should be decreased in value to maintain thesame amount of feedback.This amplifier was used in conjunction witha General Electric S-1201A 12-inch speakermounted in an Electro-Voice KD6 Aristocratspeaker enclosure which was constructed froma kit. The reproduction was extremely smooth,with good balance of bass and treble.7-6 A High Quality25 Watt AmplifierThis amplifier is recommended for the musiclover who desires the utmost in high fidelity.Capable of delivering 25 watts output at lessthan 1.5% intermodulation distortion, the amplifierwill provide flawless reproduction athigher than normal listening levels. It is truethat other designs have been advocated providinggreater power at a lower IM distortionlevel. However, the improvement in reproductionof such a system is not noticeable - evento the trained ear - over the results obtainedwith this low priced amplifier. A full 20 wattsof audio power are obtained over the frequencyrange of 20 to 50,000 cycles at less than 0.1%harmonic distortion. At the average listeninglevel of 1 watt, the frequency response is plusor minus 1 decibel at 100,000 cycles, assuringtrue high fidelity response over the entire audiblerange. Preamplification and tone compensationare not included in this unit, as thesefunctions are accomplished in the auxiliarydriving amplifier. If a variable reluctance cartridgeis employed, the preamplifiers of figure9 or figure 14 may be used with this unit.The Heathkit W A-P2 preamplifier is alsorecommended for general use when several inputcircuits are to be employed.The AmplifierCircuitThe schematic of this amplifieris shown in figure 27.It is a "Williamson-type"amplifier using the so-called "Ultra-linear"screen-tap circuit on the output stage. A dualtriode 6SN7-GT serves as a voltage amplifierand direct coupled phase inverter. Maximumsignal input to this stage is approximately 0.8volt, r.m.s. for rated output of the amplifier.The feedback loop from the speaker voice coilis returned to the cathode circuit of the inputstage, placing all amplifying stages within theloop.A second 6SN7-GT is used as a push-pull
Figure 28UNDER-CHASSIS VIEWOF 20-WATT AMPLIFIER.807 SOCKETS ARE ATCENTER, WITH CATHODEBALANCINGPOTENTIOMETER ATLOWER RIGHT.Not!t ground bus, starting atcenter, top filter capacitor, andrunning in loop around under·side of chassis. Bus is groundedto chassis at input plug (upperleft). Shielded leads run to volumecontrol (center).driver stage for the high level output amplifier.A plate potential of 430 volts is applied to thisstage to insure ample grid drive to the outputstage. Large value coupling capacitors are usedbetween all stages to prevent signal degenerationat the lower audio frequencies. Two 807beam-power tubes are employed in the finalamplifier stage. Cathode bias is applied to thesetubes, and the current of the tubes may beequalized by potentiometer R, in the biascircuit.Degenerative feedback is taken from the secondaryof the output transformer and appliedto the input of the amplifier. The amount offeedback is controlled by the value of feedbackresistor Ra. A small capacitor is placedacross R, to reduce any tendency towards highfrequency parasitic oscillation sometimes encounteredwhen long leads are used to connectthe amplifier to the loudspeaker.An external power supply is used with theamplifier, delivering 440 volts at a current of1 7 5 milliamperes, and 6.3 volts at 5 amperes.AmplifierConstructionThe amplifier is built upon asteel chassis measuring 9" x7"x2". Punched and drilledchassis for both the amplifier and power supplymay be obtained as standard parts, as specifiedin figure 27, eliminating the necessity ofconsiderable metal work. Placement of themajor components may be seen in the top andunder-chassis photographs. The two cathodecurrent jacks, J, and J, are mounted in oversizeholes and are insulated from the chassiswith tibre shoulder washers placed on the jackstem, and flat fibre washers beneath the retainingnut. The can-type filter capacitors are insulatedfrom the chassis by fibre mountingboards.A common ground bus is used in this amplifier,and all "grounds" are returned to thebus, rather than to the chassis. The bus isgrounded at the input jack of the amplifierand runs around the underside of the chassis,ending at the ground terminals of the highvoltage filter capacitors. Tie-point terminalstrips are used to support the smaller components,and direct point-to-point wiring isused.Matched pairs of resistors should be usedin the balanced audio circuits, and these resistorsare marked ( *) on the schematic. If possible,measure a quantity of resistors in the storewith an ohmmeter, and pick out the pairs ofresistors that are most evenly matched. Theexact resistance value is not critical within 10%as long as the two resistors are close in value.Care must be taken when these resistors aresoldered in the circuit, as the heat of the solderingiron may cause the resistance value tovary. It is wise to grasp the resistor lead witha long-nose pliers, holding the lead betweenthe point where the iron is applied and thebody of the resistor. The pliers will act as a"heat sink", protecting the resistor from excessiveheat.The filament leads are made up of a twistedpair of wires running between the varioussockets. Keep the twisted leads clear of the inputjack to insure minimum hum pickup. Theplate leads from the 807 caps pass through%-inch rubber grommets mounted in the chassisto phenolic tie points mounted beneath thechassis. The plate leads of output transformerT, attach to these tie points. Be sure you ob-
148z0i=..:.J::J00::;a:w ,_z5High Fidelity Techniques4'~2,004 8 12 16 20 24EQUIVALENT SINE WAVE WATTSFigure 29INTERMODULATION CURVEOF 20-WATT AMPLIFIER.28 32serve the color code for these leads, as givenin figure 2 7, as the proper polarization of theleads is required for proper feedback operation.When the amplifier wiring is completed, allconnections should be checked for accidentalgrounds or transpositions. Make sure the meterjacks are insulated from the chassis.The PowerSupplyA companion power supply forthe high fidelity amplifier isshown in figure 30. A cathodetype5V4-G rectifier is used to limit the warmupsurge voltages usually encountered in sucha supply. Filament and plate voltages for theHeathkit W A-P2 preamplifier may be derivedfrom receptacle SO-l. If the preamplifier is notused, or if another type of preamplifier is tobe employed, the center-tap of the 6.3 voltfilament winding of power transformer T,(green/yellow lead) should be grounded tothe chassis of the power supply.Wiring of this unit is straightforward, andno unusual precautions need be taken. It iswise to attach the amplifier to the supply beforeit is turned to limit warm-up voltageexcursions.AmplifierOperationThe amplifier is attached to thepower supply by a short lengthof 4-wire cable. Make sure tahtthe filament leads (pins 1 and 4, p,) aremade of sufficiently heavy wire to insure thatthe filaments of the amplifier receive a full6.3 volts. The amplifier should be turned onand all voltages checked against figure 27. Aspeaker or suitable resistive load should be attachedto output terminal strip p,, and a 0-100d-e milliammeter plugged into jacks J, and ;,.Balance potentiometer R, is now adjusted untilthe two readings are the same. Each measurementwill be very close to 60 milliampereswhen the currents are balanced. The amplifieris now ready for operation, and has a fidelitycurve similar to that shown in figure 29.NOTE: IF HEATHKIT PREAMPLIFIERIS NOT USED, CENTER-TAP WIRE(GREEN-YELLOW) OF a.3 FILAMENTWINDING OF T1 MUST BE GROUNDED.SOCKET FORHEATH WA-P2PREAMPL.IFIERT1- 400-0-400 VOLTS, 2.00 MA .. ~VOLTS,3 AMP., 8.3 VOLTS, 5. AMP.CHICAGO-STANDARD PC-8412CH 1- 4.5 HENRY AT 2.00 MA.CHICAGO-STANDARD C-1411Figure 30POWER SUPPLY FOR 20-WATT AMPLIFIER.
CHAPTER EIGHTRadio frequencyVacuum Tube AmplifiersTUNED RF VACUUM TUBE AMPLIFIERSTuned r-f voltage amplifiers are used in receiversfor the amplification of the incomingr-f signal and for the amplification of intermediatefrequency signals after the incomingfrequency has been converted to the intermediatefrequency by the mixer stage. Signal frequencystages are normally called tuned r-famplifiers and intermediate-frequency stagesare called i-f amplifiers. Both tuned r-f andi-f amplifiers are operated Class A and normallyoperate at signal levels from a fractionof a microvolt to amplitudes as high as 10 to50 volts at the plate of the last i-f stage in areceiver.8-1 Grid CircuitConsiderationsSince the full amplification of a receiver followsthe first tuned circuit, the operating conditionsexisting in that circuit and in its couplingto the antenna on one side and to thegrid of the first amplifier stage on the otherare of greatest importance in determining thesignal-to-noise ratio of the receiver on weaksignals.First TunedCircuitIt is obvious that the highestratio of signal-to-noise be impressedon the grid of the firstr-f amplifier tube. Attaining the optimum ratiois a complex problem since noise will be generatedin the antenna due to its equivalentradiation resistance (this noise is in additionto any noise of atmospheric origin) and in the14 9first tuned circuit due to its equivalent coupledresistance at resonance. The noise voltagegenerated due to antenna radiation resistanceand to equivalent tuned circuit resistanceis similar to that generated in a resistor dueto thermal agitation and is expressed by thefollowing equation:E 02= 4kTRMWhere: En = r-m-s value of noise voltage overthe interval ~fk = Boltzman's constant= 1.374X 10" 22 joule per °K.T = Absolute temperature °K.R = Resistive component of impedanceacross which thermal noiseis developed.~f = Frequency band across whichvoltage is measured.In the above equation ~f is essentially thefrequency band passed by the intermediate frequencyamplifier of the receiver under consideration.This equation can be greatly simplifiedfor the conditions normally encounteredin communications work. If we assume the followingconditions: T = 300° K or 27° C or80.5° F, room temperature; ~f = 8000 cycles(the average pass band of a communicationsreceiver or speech amplifier), the equation reducesto: Er.m.s. = 0.0115 y'Rffiicrovolts. Accordingly,the thermal-agitation voltage appearingin the center of half-wave antenna (assumin·geffective temperature to be 300° K)having a radiation resistance of 73 ohms is
150 R-F Vacuum Tube Amplifiers <strong>THE</strong> R AD I 0approximately 0.096 microvolts. Also, the thermalagitation voltage appearing across a 500,-000-ohm grid resistor in the first stage of aspeech amplifier is approximately 8 microvoltsunder the conditions cited above. Further, thevoltage due to the r m a 1 agitation being impressedon the grid of the first r-f stage in areceiver by a first tuned circuit whose resonantresistance is 50,000 ohms is approximately2.5 microvolts. Suffice to say, however, thatthe value of thermal agitation voltage appearingacross the first tuned circuit when the antennais properly coupled to this circuit willbe very much less than this value.It is common practice to match the impedanceof the antenna transmission line to theinput impedance of the grid of the first r-f amplifierstage in a receiver. This is the conditionof antenna coupling which gives maximumgain in the receiver. However, when u-h-f tubessuch as acorns and miniatures are used at frequenciessomewhat less than their maximumcapabilities, a significant improvement in signal-to-noiseratio can be attained by increasingthe coupling between the antenna and firsttuned circuit to a value greater than that whichgives greatest signal amplitude out of the re·ceiver. In other words, in the 10, 6, and 2 me·ter bands it is possible to attain somewhat improvedsignal-to-noise ratio by increasing antennacoupling to the point where the gain ofthe receiver is slightly reduced.It is always possible, in addition, to obtainimproved signal-to-noise ratio in a v-h-f receiverthrough the use of tubes which haveimproved input impedance characteristics atthe frequency in question over conventionaltypes.Noise Factor The limiting condition for sensitivityin any receiver is thethermal noise generated in the antenna and inthe first tuned circuit. However, with propercoupling between the antenna and the grid ofthe tube, through the first tuned circuit, thenoise contribution of the first tuned circuitcan be made quite small. Unfortunately, though,the major noise contribution in a properly designedreceiver is that of the first tube. Thenoise contribution due to electron flow anddue to losses in the tube can be lumped intoan equivalent value of resistance which, ifplaced in the grid circuit of a perfect tube havingthe same gain but no noise would give thesame noise voltage output in the plate load.The equivalent noise resistance of tubes suchas the 6SK7, 6SG7, etc., runs from 5000 to10,000 ohms. Very high Gm tubes such as the6AC7 and 6AK5 have equivalent noise resistancesas low as 700 to 1500 ohms. The lowerthe value of equivalent noise resistance, thelower will be the notse output under a fixedset of conditions.The equivalent noise resistance of a tubemust not be confused with the actual inputloading resistance of a tube. For highest signal-to-noiseratio in an amplifier the inputloading resistance should be as high as possibleso that the amount of voltage that can bedeveloped from grid to ground by the antennaenergy will be as high as possible. The equivalentnoise resistance should be as low aspossible so that the noise generated by thisresistance will be lower than that attributableto the antenna and first tuned circuit, and thelosses in the first tuned circuit should be aslow as possible.The absolute sensitivity of receivers hasbeen designated in recent years in governmentand commercial work by an arbitraty dimensionlessnumber known as "noise factor" or N.The noise factor is the ratio of noise outputof a "perfect" receiver having a given amountof gain with a dummy antenna matched to itsinput, to the noise output of the receiver underm~asurement having the same amount of gainwith the dummy antenna matched to its input.Although a perfect receiver is not a physicallyrealizable thing, the noise factor of a receiverunder. measurement can be determined by calculauonfrom the amount of additional noise(from a temperature-limited diode or other calibratednoise generator) required to increasethe noise power output of a receiver by a predeterminedamount.Tube InputLoadingAs has been mentioned in a previousparagraph, greatest gainin a receiver is obtained whenthe antenna is matched, through the r-f cou·pling transformer, to the input resistance ofthe r-f tube. However, the higher the ratio oftube input resistance to equivalent noise resistanceof the tube the higher will be the signal-to-noiseratio of the stage-and of course,the better will be the noise factor of the over·all receiver. The input resistance of a tubeis very high at frequencies in the broadcastband and gradually decreases as the frequencyincreases. Tube input resistance on conven·tional tube types begins to become an import·ant factor at frequencies of about 25 Me. andabove. At frequencies above about 100 Me. theuse of conventional tube types becomes im·practicable since the input resistance of thetube has become so much lower than the equivalentnoise resistance that it is impossibleto attain reasonable signal-to-noise ratio onany but very strong signals. Hence, specialv-h-f tube types such as the 6AK5, 6AG5, and6CB6 must be used.The lowering of the effective input resist·
<strong>HANDBOOK</strong>R-F Amplifiers 151ance of a vacuum tube at higher frequenciesis brought about by a number of factors. Thefirst, and most obvious, is the fact that thedielectric loss in the internal insulators, andin the base and press of the tube increaseswith frequency. The second factor is due tothe fact that a finite time is required for anelectron to move from the space charge in thevicinity of the cathode, pass between the gridwires, and travel on to the plate. The fact thatthe electrostatic effect of the grid on the movingelectron acts over an appreciable portionof a cycle at these high frequencies causes acurrent flow in the grid circuit which appearsto the input circuit feeding the grid as a resistance.The decrease in input resistance ofa tube due to electron transit time varies asthe square of the frequency. The undesirableeffects of transit time can be reduced in certaincases by the use of higher plate voltages.Transit time varies inversely as the squareroot of the applied plate voltage.Cathode lead inductance is an additionalcause of reduced input resistance at high frequencies.This effect has been reduced in certaintubes such as the 6SH7 and the 6AK5 byproviding two cathode leads on the tube base.One cathode lead should be connected to theinput circuit of the tube and the other leadshould be connected to the by-pass capacitorfor the plate return of the tube.The reader is referred to the Radiation LaboratorySeries, Volume 23: "Microwave Receivers"(McGraw-Hill, publishers) for additionalinformation on noise factor and input loadingof vacuum tubes.8-2 Plate-CircuitConsiderationsNoise is generated in a vacuum tube by thefact that the current flow within the tube is nota smooth flow but rather is made up of the continuousarrival of particles (electrons) at avery high rate. This shot effect is a source ofnoise in the tube, but its effect is referredback to the grid circuit of the tube since it isincluded in the equivalent noise resistancediscussed in the preceding paragraphs.Plate CircuitCouplingFor the purpose of this section,it will be considered that thefunction of the plate load circuitof a tuned vacuum-tube amplifier is to deliverenergy to the next stage with the greatestefficiency over the required band of frequencies.Figure 1 shows three methods of interstagecoupling for tuned r-f voltage amplifiers.In figure lA omega (w) is 2rr times the resonantfrequency of the circuit in the plate of@AMPLIFICATION AT RESONANCE(APPROX.)=GMWLQ@AMPLIFICATION AT RESONANCE (APPROX.l=GMWMQ©AMPLIFICATION AT RESONANCE(APPROil,Gt.tK ~~~P~sQpQsWHERE: 1. PRI. AND SEC. RESONANT AT SAME FREQUENCY2. K IS COEFFICIENT OF COUPLINGIF PRI. AND SEC. Q ARE APPROXIMATELY <strong>THE</strong> SAME;~~~~~RB~~~;~~~~y = 1. 2 KMAXIMUM AMPLITUDE OCCURS AT CRITICAL COUPLING1WHENK= - --VQPQSFigure 1Gain equations for pentacle r-f amplifierstages operating into a tunec/ loadthe amplifier tube, and L and Q are the inductanceand Q of the inductor L. In figure lB thenotation is the same and M is the mutual inductancebetween the primary coil and the secondarycoil. In figure lC the notation is againthe same and k is the coefficient of couplingbetween the two tuned circuits. As the coefficientof coupling between the circuits isincreased the bandwidth becomes greater butthe response over the band becomes progressivelymore double-humped. The response overthe band is the most flat when the Q' s of primaryand secondary are approximately the sameand the value of each Q is equal to 1. 75/ k.
152 R-F Vacuum Tube Amplifiers <strong>THE</strong> R AD I 0Variable-Mu Tubesin R- F StagesIt is common practice tocontrol the gain of a successionof r-f or i·f am·plifier stages by varying the average bias ontheir control grids. However, as the bias israised above the operating value on a conventionalsharp-cutoff tube the tube becomes in·creasingly non-linear in operation as cutoff ofplate current is approached. The effect of suchnon-linearity is to cause cross modulation be·tween strong signals which appear on the gridof the tube. When a tube operating in such amanner is in one of the first stages of a receivera number of signals are appearing on itsgrid simultaneously and cross modulation betweenthem will take place. The result of thiseffect is to produce a large number of spurioussignals in the output of the receiver-in mostcases these signals will carry the modulationof both the carriers which have been crossmodulated to produce the spurious signal.The undesirable effect of cross modulationcan be eliminated in most cases and greatlyreduced in the balance through the use of avariable·mu tube in all stages which have a·v·cvoltage or other large negative bias applied totheir grids. The variable-mu tube has a char·acteristic which causes the cutoff of plate cur·rent to be grad u a 1 with an increase in gridbias, and the reduction in plate current is ac·companied by a decrease in the effective am·plification factor of the tube. Variable·mu tubesordinarily have somewhat reduced Gm as com·pared to a sharp-cutoff tube of the same group.Hence the sharp-cutoff tube will perform bestin stages to which a·v-c voltage is not applied.<strong>RADIO</strong>-FREQUENCY POWER AMPLIFIERSAll modern transmitters in the medium·fre·quency range and an increasing percentage ofthose in the v·h·f and u·h·f ranges consist ofa comparatively low-level source of radio·fre·quency energy which is multiplied in frequencyand successively amplified to the desired powerlevel. Microwave transmitters are still predom·inately of the self-excited oscillator type, butwhen it is possible to use r·f amplifiers ins·h·f transmitters the flexibility of their ap·plication will be increased. The following por·tion of this chapter will be devoted, however,to the method of operation and calculation ofoperating characteristics of r·f power ampli·fiers for operation in the range of approximate·ly 3.5 to 500 Me.8-3 Class C R-FPower AmplifiersThe majority of r-f power amplifiers fall intothe Class C category since such stages canbe made to give the best plate circuit efficiencyof any present type of vacuum-tube ampli·fier. Hence, the cost of tubes for such a stageand the cost of the power to supply that stageis least for any given power output. Neverthe·less, the Class C amplifier gives less powergain than either a Class A or Class B amplifierunder similar conditions since the grid ofa Class C stage must be driven highly positiveover the portion of the cycle of the excit·ing wave when the plate voltage on the ampli·fier is low, and must be at a large negativepotential over a large portion of the cycle sothat no plate current will flow except whenplate voltage is very low. This, in fact, is thefundamental reason why the plate circuit efficiencyof a Class C amplifier stage can bemade high-plate current is cut off at all timesexcept when the plate-to-cathode voltage dropacross the tube is at its lowest value. ClassC amplifiers almost invariably operate into atuned tank circuit as a load, and as a resultare used as amplifiers of a single frequencyor of a comparatively narrow band of frequencies.Relationships inClass C StageFigure 2 shows the relationshipsbetween the variousvoltages and currents overone c y c 1 e of the exciting grid voltage for aClass C amplifier stage. The notation given infigure 2 and in the discussion to follow is thesame as given at the first of Chapter Six un·der "Symbols for Vacuum· Tube Parameters."The various manufacturers of vacuum tubespublish booklets listing in adequate detail alternativeClass C operating conditions for thetubes which they manufacture. In addition,operating condition sheets for any particulartype of vacuum tube are available for the ask·ing from the different vacuum-tube manufac·turers. It is, nevertheless, often desirable todetermine optimum operating conditions for atube under a particular set of circumstances.To assist in such calculations the followingparagraphs are devoted to a method of calcu·lacing Class C operating conditions which ismoderately simple and yet sufficiently accu·rate for all practical purposes.
<strong>HANDBOOK</strong>Class C R-F Amplifiers 1 53tiona! grid voltage-plate current operatingcurves, the calculation is considerably sim·plified if the alternative "constant-currentcurve" of the tube in question is used. This istrue since the operating line of a Class C amplifieris a straight line on a set of constant·current curves. A set of constant-current curveson the 250TH tube with a sample load linedrawn thereon is shown in figure 5.In calculating and predicting the operationof a vacuum tube as a Class C radio-frequencyamplifier, the considerations which determinethe operating conditions are plate efficiency,power output required, maximum allowableplate and grid dissipation, maximum allowableplate voltage and maximum allowable platecurrent. The values chosen for these factorswill depend both upon the demands of a particularapplication and upon the tube chosen.The plate and grid currents of a Class Camplifier tube are periodic pulses, the dura·tions of which are always less than 180 de·grees. For this reason the average grid cur·rent, average plate current, power output, driv·ing power, etc., cannot be directly calculatedbut must be determined by a Fourier analysisfrom points selected at proper intervals alongthe line of operation as plotted upon the constant-currentcharacteristics. This may be doneeither analytically or graphically. While theFourier analysis has the advantage of accuracy,it also has the disadvantage of beingtedious and involved.The approximate analysis which followshas proved to be sufficiently accurate for mostapplications. This type of analysis also hasthe advantage of giving the desired informa·tion at the first trial. The system is direct ingiving the desired information since the importantfactors, power output, plate efficiency,and plate voltage are arbitrarily selected atthe beginning.Figure 2Instantaneous electrode ancl tank circuitvoltages ancl currents for a Class C r-f poweramplifierCalculation af ClassC Amp I ifier OperatingCharacteristicsAlthough Class C op·erating conditions canbe determined with theaid of the more conven-Method afCalculationThe first step in the method tobe described is to determine thepower which must be deliveredby the Class C amplifier. In making this determinationit is well to remember that ordinarilyfrom 5 to 10 per cent of the power deliveredby the amplifier rube or tubes will be lost inwell-designed tank and coupling circuits atfrequencies below 20 Me. Above 20 Me. thetank and circuit losses are ordinarily somewhatabove 10 per cent.The plate power input necessary to producethe desired output is determined by the plateefficiency: Pin = Po u /N p.For most applications 1t is desirable to oper·ate at the highest practicable efficiency. High·efficiency operation usually requires less ex·pensive tubes and power supplies, and the
154 R-F Vacuum Tube Amplifiers <strong>THE</strong> R AD I 07. 0J0j0lLl _10 •. 0~a:vvv0./.vv3. 0 1.> 1.6 1.&1.00 7. 0i=
<strong>HANDBOOK</strong>Constant Current Calculations155FINAL POINTEIMAC 250THCONSTANT CURRENTCHARACTERISTICS••0a:(!)Ecc=-240.... ....LOAD LINE-PLATE VOLTAGE-VOLTS....FIGURESActive portion of the operating loaclline for an Eimac 2SOTH Class C r-f power amplifier,showing first trial point ancl the final operating point7. Calculate Epm from: Epm =Ebb- epmin•B. Calculate the ratio Ipm/h from:Ipm--=2 Np Ebb9. From the ratio of Ipm/Ib calculated instep 8 determine the ratio ipmax/Ib fromfigure 3.IO. Calculate a new value for ipmax fromthe ratio found in step 9.ip max = (ratio from step 9) h11. Read egmp and igmax from the constantcurrentcharacteristics for the values ofepmin and ip max determined in steps 6and 10.I2. Calculate the cosine of one·half theangle of plate current flow from:13. Calculate the grid bias voltage from:Ecc =~OS {)pfor triodes.II- cos ep----xI- cos epXEpm(-- egmp)ll[_ E;b JEc2]- egmp cos e- --P.ufor tetrodes, where p. 12is the grid-screenamplification factor, and Ec 2 is the d·cscreen voltage.I4. Calculate the peak fundamental grid ex·citation voltage from:Egm = egmp- EccI5. Calculate the ratio E 8m/Ecc for the val-
156 R-F Vacuum Tube Amplifiers <strong>THE</strong> R AD I 0ues of Ecc and Egm found in steps 13and 14.16. Read igma xllc from figure 4 for the ratioEgm/Ecc found in step 15.17. Calculate the ave rage grid current fromthe ratio found in step 16, and the valueof igmax found in step 11:igmaxlc::: --------Ratio from step 1618. Calculate approximate grid driving powerfrom;P d = 0.9 Egmlc19. Calculate grid dissipation from:Pg::: Pd + EcclcP g must not exceed the maximum ratedgrid dissipation for the tube selected.Sample A typical example of a Class CCalculation amplifier calculation is shownin the example below. Referenceis made to figures 3, 4 and 5 in the calculation.1. Desired power output-BOO watts.2. Desired plate voltage-3500 volts.Desired plate efficiency-SO per cent(Np::: 0.80)Pin = 800/0.80 ::: 1000 watts3. P P = 1000 - 800 = 200 wattsUse 250TH; max. Pp = 250w; ll = 37.4. Ib = 1000/3500 = 0.285 ampere (285 rna.)Max. Ib for 250TH is 350 rna.5. Approximate ipmax = 0.285 X 4.5= 1. 28 ampere12. cos e p = 2.32 (1.73 - 1. 57) = o. 37(8p::: 08.3°)13. Ecc::: X1 - 0.373240[' 0.37 (-- 240)37=- 240 volts- 3500]3714. Egm = 240 - (-240) = 480 volts gridswing15. Egm/Ecc::: 480/ - 240::: - 216. igmax/Ic = 5.75 (from figure 4)17. lc = 0.430/5.75:::0.075 amp. (75 rna.grid current)18. P d = 0.9X480X0.075 = 32.5 wattsdriving power19. P 8= 32.5- (-240X0.75)::: 14.5 wattsgrid dissipationMax. P g for 250TH is 40 wattsThe power output of any type of r-f amplifieris equal to:IpmEpm/2::: PoIpm can be determined, of course, from theratio determined in step 8 above (in this typeof calculation) by multiplying this ratio timeslb.It is frequently of importance to know thevalue of load impedance into which a ClassC amplifier operating under a certain set ofconditions should operate. This is simply R L =Epm/Ipm· In the case of the operating conditionsjust determined for a 250TH amplifierstage the value of load impedance is:Epm 3240RL =--= --= 6600 ohmsIpm .4956. epmin = 260 volts (see figure 5 firsttrial point)7. Epm::: 3500 - 260 = 3240 volts8. Ipm/Ib = 2X0.80X3500/3240:::5600/3240 ::: 1. 739. ipmax/Ib = 4.1 (from figure 3)10. ipmax::: 0.285X4.1::: 1.1711. egmp::: 240 voltsigmax = 0.430 amperes(Both above from final point on figure 5)Q of AmplifierTank CircuitIn order to obtain good platetank circuit tuning and lowradiation of harmonics froman amplifier it is necessary that the plate tankcircuit have the correct Q. Charts giving compromisevalues of Q for Class C amplifiersare given in the chapter, Generation of R-FEnergy. However, the amount of inductancerequired for a specified tank circuit Q underspecified operating conditions can be calcu·lated from the following expression:
<strong>HANDBOOK</strong>RLwL=-Qw = 2 TT X operating frequencyL = Tank inductanceRL = Required tube load impedanceQ = Effective tank circuit QA tank circuit Q of 12 to 20 is recommendedfor all normal conditions. However, if a balancedpush-pull amplifier is employed the tankreceives two impulses per cycle and the circuitQ may be 1 ower e d somewhat from theabove values.Quick Method ofCalculating Amp I ifierPlate EfficiencyThe plate circuit efficiencyof a Class B orClass C r-f amplifiercan be determined fromthe following facts. The plate circuit efficiencyof such an amplifier is equal to the product oftwo factors, F,, which is equal to the ratio ofEpm to. Ebb (F, = Epm/Ebb) and F , 2which isproportional to the one-half angle of plate currentflow, 8 p. A graph of F 2against both ()and cos () p is given in figure 6. Either () P 0~cos () p may be used to determine F • 2Cos ()may. be dete_rmined either from the procedurgpreviOusly gtven. for making Class C amplifiercomputatiOns or It may be determined from thefollowing expression:Example afMethodP. Ecc +Ebbcos ()p = - ------/1 Egm- EpmIt is desired to know the one-halfangle of plate current flow andthe plate circuit efficiency foran 812 tube operating under the following con·ditions which have been assumed from inspectionof the data and curves given in the RCATransmitting Tube Handbook HB-3:I. Ebb= llOO voltsEcc = -40 voltsp. = 29Egm = 120 voltsEpm = 1000 volts- 29 X 40 + I 1003. cos ()p = =29 X 120 - 100060-- = 0.02524804. F 2 = 0. 79 (by reference to figure 6)Class B R-F Amplifiers 1571.0.980.960.940.920.900.880.88F20.840.820.800.78•0.70.740.720.70- ~~"'...-"' 1'\\L\.l\.~('\,.~1\o m m ~ ~ ~ ~ ro ~ ~ -•m lBI I"'·.~ELEcTRICAL' DEGREES I I IJ I I I I I I It-0 OtU 0.0. O.IN 0.7 .. Oe4lO.~O.:MZO.t7" O.OO+O.H
158 R-F Vacuum Tube Amplifiers <strong>THE</strong> R A 0 I 01.0N'u .9b"' wa:wa.:::;;4:zwa:(.)"' a:0"' w•76sa:3wa.~ .2w...4:.Ja.I-' \\:IvvII--~. ·~IKTrrc;;: Ec = + 100 I!c2. Ec.=+!.o1//.'/vf.--A- f.-1!v !--""Ec2:: +400 v.Ec~= a v.I Ec = o&
<strong>HANDBOOK</strong>Linear Amplifier Parameters 159plate current of 21 milliamperes willproduce this figute. Referring to figure7, a grid bias of -45 volts is approximate!y correct.2. A practical Class B linear r-f amplifierruns at an efficiency of about 66% at fulloutput, the efficiency dropping to about33% with an unmodulated exciting signal.In the case of single-sideband suppressedcarrier excitation, a no-excitationcondition is substituted for the unmodulatedexcitation case, and the linearamplifier runs at the resting or quiescentinput of 42 watts with no excitingsignal. The peak allowable power inputto the 813 is:Input Peak Power (Wp) =(wat~s)Plate Dissipation X 100-------=( 100 - % plate efficiency)125- x 100 = 379 watts333. The maximum signal plate current is:WP 379ipmax =- = --= 0.189 ampereEP 20004. The plate current flow of the linear amplifieris 1809, and the plate currentpulses have a peak of 3.14 ti!llfs themaximum signal current:3.14 x 0.189 = 0.595 ampere5. Referring to figure 7, a current of 0.605ampere (Point A) will flow at a positivegrid potential of 60 volts and a minimumP!ate potential of 420 volts. The grid isb1ased at -45 volts, so a peak r-f gridvoltage of 60 + 45 volts = 105 volts is required.6. The gri.d driving power required for theClass B linear stage may be found by theaid of figute 8. It is one-quarter the productof the peak grid current times thepeak grid voltage:0.02 X 105P g = ---- = 0.53 watt47. The single tone power output of the 813stage is:Pp = 78.5 (Ep- epmin) X lpPp = 78.5 (2000- 420)x .189 = 235 watts10080eo -"40p0 ~0__\.Ecr=+roov.C I:::: +SO v._""~+eov.Ec1·+ 1 ~t-" L~Ecr:;:+zo .~~2"+400 v.c3=ov.100 200 .300 400PLATE VOLTS EPFigure 8E 9 , VS. Ep CHARACTERISTICS OF 813TUBE8. The plate load resistance is:Ep- epmin 1580RL = --=0.5ipm ax 0.5 X .189= 6000 ohms9. If a loaded plate tank circuit Q of 12 isdesired, the reactance of the plate tankcapacitor at the resonant frequencyshould be:RL 6000Reactance {ohms)=- =--= 500 ohmsQ 1210. For an operating frequency of 4.0 Me.,the effective resonant capacity is:106c------- = 80 !1/lfd.6.28 X 4.0 X 50011. The inductance required to resonate at4.0 Me. with this value of capacity is:Grid CircuitConsiderations500L=---- = 19.9 microhenries6.28 X 4.0I. The maximum positive gridpotential is 60 volts, andthe peak r·f grid voltage is105 volts. Required driving power is 0.53 watt.The equivalent grid resistance of this stage is:
160 R-F Vacuum Tube Amplifiers <strong>THE</strong> R AD I 0(eg) 2 105 2Rg= = =2XPg 2X0.5310,400 ohms2. As in the case of the Class B audio amplifierthe grid resistance of the linearamplifier varies from infinity to a lowvalue when maximum grid current isdrawn. To decrease the effect of thisresistance excursion, a swamping resistorshould be placed across the grid tankcircuit. The value of the resistor shouldbe dropped until a shortage of drivingpower begins to be noticed. For this example,a resistor of 3,000 ohms is used.The grid circuit load for no grid currentis now 3,000 ohms instead of infinity,and drops to 2 400 ohms when maximumgrid current is drawn.3. A circuit Q of 15 is chosen for the gridtank. The capacitive reactance requiredis:2400Xc = --= 160 ohms154. At 4.0 ~!c. the effective capacity is:C= = 248 pp.fd.6.28 X 4 X 1545. The inductive reactance required ro re sonatethe grid circuit at 4.0 Me. is:160L = = 6.4 microhenries6.28X4.06. By substituting the loaded grid resistancefigure in the formula in the firstparagraph, the grid driving power is nowfound to be approximately 2.3 watts.Screen CircuitConsiderationsBy reference to the platecharacteristic curve of the813 tube, it can be seen thatat a minimum plate potential of 500 volts, arida maximum plate current of 0.6 ampere, thescreen current will be approximately 30 milliamperes,dropping to one or two milliamperesin the quiescent state. It is necessary to usea well-regulated screen sup p 1 y to hold thescreen voltage at the correct potential overthis range of current excursion. The use of anelectronic regulated screen supply is recommended.8-5 Special R-F PowerAmplifier CircuitsThe r-f power amplifier discussions of Sections8-4 and 8-5 have been based on the assump,tionthat a conventional grounded-cathodeor cathode-return type of amplifier was in question.It is possible, however, as in the case ofa-f and low-level r-f amplifiers to use circuitsin which electrodes other than the cathode arereturned to ground insofar as the signal potentialis concerned. Both the plate-return orcathode-follower amplifier and the grid-returnor grounded-grid amplifier are effective in certaincircuit applications as tuned r-f poweramplifiers.Disadvantages ofGrounded-CathodeAmplifiersAn undesirable aspect ofthe operation of cathodereturnr-f power amplifiersusing triode tubes is thatsuch amplifiers must be neutralized. Principlesand methods of neutralizing r-f power amplifiersare discussed in the chapter Generationof R-F Energy. As the frequency of operationof an amplifier is increased the stage becomesmore and more difficult to neutralizedue to inductance in the grid and plate leadsof the tubes and in the leads to the neutralizingcapacitors. In other words the bandwidthof neutralization decreases as the frequencyis increased. In addition the very presence ofthe neutralizing capacitors adds additionalundesirable capacitive loading to the grid andplate tank circuits of the tube or tubes. Tolook at the problem in another way, an amplifierthat may be perfectly neutralized at a frequencyof 30 Me. may be completely out ofneutralization at a frequency of 120 Me. Therefore,if there are circuits in both the grid andplate circuits which offer appreciable impedanceat this high frequency it is quite possiblethat the stage may develop a "parasiticoscillation" in the vicinity of 120 Me.Grounded-Grid This condition of restricted-R- F Amp I ifiers range neutralization of r-fpower amplifiers can be greatlyalleviated through the use of a c~- ~ '.return or grounded-grid r-f stage. The groundedgridamplifier has the following advantages:1. The output capacitance of a stage is reducedto approximately one-half the valuewhich would be obtained if the same tubeor tubes were operated as a conventionalneutralized amplifier.2. The tendency toward parasitic oscillationsin such a stage is greatly reduced sincethe shielding effect of the control grid be-
<strong>HANDBOOK</strong>Grounded Grid Amplifier1 6 ltween the filament and the plate is effectiveover a broad range of frequencies.3. The feedback capacitance within the stageis the plate-to-cathode capacitance whichis ordinarily very much less than the gridto-platecapacitance. Hence neutralizationis ordinarily not required. If neutralizationis required the neutralizing capacitors arevery small in value and are cross connectedbetween plates and cathodes in apush-pull stage, or between the oppositeend of a split plate tank and the cathodein a single-ended stage.The disadvantages of a grounded-grid amplifierare:I. A large amount of excitation energy is required.However, only the normal amountof energy is lost in the grid circuit of theamplifier tube; all additional energy overthis amount is delivered to the load circuitas useful output.2. The cathode of a grounded-grid amplifierstage is "hot" to r.f. This means that thecathode must be fed through a suitable impedancefrom the filament supply, or thesecondary of the filament transformer mustbe of the low-capacitance type and adequatelyinsulated for the r-f voltage whichwill be present.3. A grounded-grid r-f amplifier cannot beplate modulated 100 per cent unless theoutput of the exciting stage is modulatedalso. Approximately 70 per cent modulationof the exciter stage as the final stageis being modulated 100 per cent is recommended.However, the grounded-grid r-famplifier is quite satisfactory as a ClassB linear r-f amplifier for single sidebandor conventional amplitude modulated wavesor as an amplifier for a straight c-w orFM signal.Figure 9 shows a simplified representationof a grounded-grid triode r-f power amplifierstage. The relationships between input andout put power and the peak fundamental componentsof electrode voltages and currents aregiven below the drawing. The calculation ofthe complete operating conditions for agrounded-grid amplifier stage is somewhat morecomplex than that for a conventional amplifierbecause the input circuit of the tube is inseries with the output circuit as far as theload is concerned. The primary result of thiseffect is, as stated before, that considerablymore power is required from the driver stage.The normal power gain for a g-g stage is from3 to 15 depending upon the grid circuit conditionschosen for the output stage. The higherthe grid bias and grid swing required on the~COADPOWER OUTPUT TO LOAD : (Ec;M + E:M) ! PM OR ~ + ~POWER DELIVERED BY OUTPUT TUBE =POWER FROM DRIVER TO L.OAD: Ey.M2lPMTOTAL POWER DELIVERED BY DRIVER:[pM2. J PMEGM (l:M+ lGt.dPOWER ABSORBED BY OUTPUT TUBE GRID AND BIAS SUPPLYlK"' (APPROXIMATELY) = ~~.-."'E!'"'iL.e-of-c ~Figure 9EGMZ lpM + 0 9 E
162 R-F Vacuum Tube Amplifiers <strong>THE</strong> R AD I 02. Pin = 850/0.85 = 1000 watts3. P P = 1000 - 850 = 150 wattsType 304TL chosen; max. P P = 300watts, p. = 12.4. lb = 1000/2700 = 0.370 ampere(370 rna.)5. Approximate ipmax = 4.9 X 0.370 = 1.81ampere6. ep min= 140 volts (from 304TL constant-currentcurves)7. Ep m = 2700 - 140 = 2560 volts8. Ipm/h = 2 X 0.85 X 2700/2560 = 1.799. ipmax/Ib = 4.65 (from figure 3)10. ipmax = 4.65 X 0.370 = 1.72 amperes11. e gmp = 140 voltsigmax = 0.480 amperes12. Cos 8P = 2.32 (1.79-1.57) = 0.51ep = 59°113. Ecc = ---X1-0.512560 2700ro.s1 (--140)--JL 12 12= -385 volts14. Egm = 140-(-385) = 525 volts16. igmax/Ic = approx. 8.25 (extrapolatedfrom figure 4)17. lc = 0.480/8.25 = 0.058 (58 rna. d-egrid current)18. P d = 0.9 X 525 X 0.058 = 27.5 watts19. P g = 27.5 -( -385 X 0.058) = 5.2 wattsMax. P g for 304TL is 50 wattsWe can check the operating plate efficiencyof the stage by the method described in Section8-4 as follows:F 1 = Ep m/Eb b = 2560/2700 = 0. 95F 2 for 8P of 59° (from figure 6) = 0.90Np = F, X F 2 = 0.95 X 0.90 = Approx.0.85 (85 per cent plate efficiency)Now, to determine the operating conditionsas a grounded-grid amplifier we must also knowthe peak value of the fundamental componentsof p 1 ate current. This is simp I y equal toOp m/1 b) I b, or:Ipm = 1.79 X 0.370 = 0.660 amperes (from 4and 8 above)The total average power required of thedr~ver (from _fi~ure 9) is equal to Egmlpm/2(since the gnd 1s grounded and the grid swingappears also as cathode swing) plus P d whichis 27.5 watts from 18 above. The total is:525 X 0.660Total drive== 172.5 watts2plus 27.5 watts or 200 wattsTherefore the total power output of the stageis equal to 850 watts (contributed by the304TL) plus 172.5 watts (contributed by thedriver) or 1022.5 watts. The cathode drivingimpedance of the 304TL (again referring tofigure 7) is approximately:zk = 525/(0.660 + 0.116) = approximately 675ohms.PI ate- Return or Circuit diagram, electrodepotentials and curCathode-Follower R-FPower Amp I ifier rents, and operatingconditions for a cathode-followerr-f power amplifier are given infigure 10. This circuit can be used, in additionto the grounded-grid circuit just discussed,as an r-f amplifier with a triode tubeand no additional neutralization circuit. However,the circuit will oscillate if the impedancefrom cathode to ground is allowed to becomecapacitive rather than inductive or resistivewith respect to the operating frequency.The circuit is not recommended except forv-h-f or u-h-f work with coaxial lines as tunedcircuits since the peak grid swing requiredon the r-f amplifier stage is approximatelyequal to the plate voltage on the amplifiertube if high-efficiency operation is desired.This means, of course, that the grid tank mustbe able to withstand slightly more peak voltagethan the plate tank. Such a stage may notbe plate modulated unless the driver stage ismodulated the same percentage as the finalamplifier. However, such a stage may be usedas an amplifier or modulated waves (Class Blinear} or as a c-w or FM amplifier.
<strong>HANDBOOK</strong>304TL G-G Amplifier 163POWER OUTPUT TO LOAD =POWER DELIVERED BY OUTPUT TUBE = ~circuit. If a conventional filament transformeris to be used the cathode tank coil may consistof two parallel heavy conductors (to carrythe high filament current) by-passed at boththe ground end and at the tube socket. Thetuning capacitor is then placed between filamentand ground. It is possible in certain casesto use two r-f chokes of special de sign to feedthe filament current to the tubes, with a conventionaltank circuit between filament andground. Coaxial lines also may be used toserve both as cathode tank and filament feedto the tubes for v-h-f and u-h-f work.POWER FROM DRIVER TO LOAD:EPM2.lGMTOTAL POWER FROM DRIVER =_ (EpM+6GMP) lGM- 2(EPM= + €c;Mp) 1.0 lcAPPRQX.2ASSUMINI; lGM ~ 1.! lc8-6 A Grounded-Grid304TL AmplifierPOWER ABSORBED BY OUTPUT TUBE GRID AND BIAS SUPPLY:= APPROX. 0. 9 (Ecc + € GMP) 1 C(EpM+ €GMP)1.e IcFigure 10CATHODE-FOLLOWER R-F POWERAMPLIFIERShowing the relationships between the tuberotentials and currents and the input andoutput power of the stage. The approximategrid impeclance also is given.The design of such an amplifier stage isessentially the same as the design of agrounded-grid amplifier stage as far as thefirst step is concerned. Then, for the secondstep the operating conditions given in figure10 are applied to the data obtained in the firststep. As an example, take the 304TL stagepreviously described. The total power requiredof the driver will be (from figure 10) approxi·mately (2700X0.58Xl.8)/2 or 141 watts. Ofthis 141 watts 27.5 watts (as before) will belost as grid dissipation and bias loss and thebalance of 113.5 watts will appear as output.The to tal output of the stage will then be approximately963 watts.Cathode Tank forG-GorC-FThe cathode tank circuitfor either a grounded-gridPower Amp I ifier or cathode-follower r·fpower amplifier may be aconventional tank circuit if the filament transformerfor the stage is of the low-capacitancehigh-voltage type. Conventional filament transformers,however, will not operate with thehigh values of r-f voltage present in such aThe 304TL tube is capable of operating asan r-f amplifier of the conventional type withthe full kilowatt input permitted amateur stations.The tube is characterized by an enormousreserve of filament emission, resultingfrom the fact that about 130 watts is requiredmerely to light the four filaments. Where theheavy filament drain, plus the low amplificationfactor of about 12, does not impose hardshipin the design of the transmitter, the 304TLis quite satisfactory for amateur service.The 304TL offers an additional feature inthat its plate-to-cathode capacitance is verylow (about 0.6 fLfL fd.) for a tube of its sizeand power handling capabilities. This featurepermits the tube to be operated, without neutralization,as a grounded-grid r-f power am·plifier.Characteristics ofGrounded-GridOperationA I though the excitationpower of the tube is only27.5 watts (which would bethe actual driving powerrequired if the tube were operating as a neutralizedamplifier) the driving power requiredinto the cathode circuit from the exciter isabout 200 watts. The extra 170 watts or so isnot lost, however, since it appears directly inthe output of the amplifier as additional energy.Thus while the 304TL itself will deliverabout 850 watts to the load circuit, the extra170 watts supplied by the driver over and abovethe 27.5 watts required to excite the 304TLappears added to the 850-watt output of the304TL. Thus the total output of the stagewould be about 1020 watts, even though thed-e input to the 304TL was only one kilowatt.Nevertheless, the tube itself operates only atits normal plate-circuit efficiency, the extrapower output coming directly from the addedeJU;itation power taken from the driver.
164 R-F Vacuum Tube Amplifiers <strong>THE</strong> R AD I 0EXCITATION~-BIAS -H.V.+3000V.10.5V. 1 13 A.Figure 11SCHEMATIC OF <strong>THE</strong>304TL G-G AMPLIFIERL 1 -None required for 3.5 Me., since cathode coli L2tunes to 3.5 Me.7 Mc.-20 turns no. 14 enam., 1%., dla. by 3" long.14 Mc.-10 turns no. 8 bare, 1%" dla. by 3" long.L 2 - Two parallel lengths of no. 12 en am. close-woundto fill Notional type XR-10A coil form.25 turns of the two wltes In parallel.L 3 -S-turn link of no. 16 rubber and cotton covered wire.L 4 -3.5 Me.: 18 turns no. 12 enam.; Nat. XR-10A formwound full, 4 t. link7 Me.: 10 turns no. 12 enam.; Nat. XR-10A form, 3 t.link14 Me.: 4 turns no. 8 bare; Nat. XR-10A form, 2 t.linkRFC-800-ma. r-f choke (National R-175)Since 15 to 20 per cent of the output from agrounded-grid r-f stage passes directly throughit from the driver, it is obvious that conven·tiona! plate modulation of the output stageonly will give incomplete modulation. In fact,if the plate voltage is completely removed fromthe g·g (grounded-grid) stage and the plate re ·turn is grounded, the energy from the driverwill pass directly through the tube and appearin the output circuit. So it becomes necessaryto apply modulation to the driver tube simultaneouslyto that which is applied to the plateof the g·g stage. It is normal practice in com·mercia! application to modulate the driverstage about 60 per cent as much as the g·gstage.For straight c·w operation no special con·siderations are involved in the use of a g·gpower amplifier except that it obviously isnecessary that the driver stage be keyed simultaneous!y with the final amplifier, if the in·stallation is such that the final amplifier is tobe keyed. With excitation, keying the combina·tion of the driver and output stage will oper·ate quite normally.The Grounded-GridClass B LinearThe g·g power amplifieris particularly well suitedto use as a class B lin·ear amplifier to build up the output of a lowerpowered SSB transmitter. Two advantages obtainedthrough the use of the g·g stage as aclass B linear amplifier are (I) no swampingis required in the input circuit of the g·g stage,and (2) nearly the full output of the excitertransmitter appears in the output of the g·gstage, being added to the output of the g·gamplifier.Since operating impedances are loweredwhen a stage is operated as a Class B linear,rather high values of tank capacitance are requiredfor a satisfactory operating circuit Q.The load impedance presented to one .304TLwith the operating conditions specified aboveis about 2500 ohms; hence the plate tank circuitcapacitance for an operating Q of 15should be about 180 ohms. This represents anoperating plate tank capacitance of about 240p.p.fd. for 4 Me., and proportionately smallercapacitances for the higher frequency bands.The cathode impedance for one 304TL underthe operating conditions specified is about700 ohms. Thus, for an operating cathode-circuitQ of 10 the cathode tank capacitanceFigure 12REAR OF <strong>THE</strong> GROUNDED-GRIDAMPLIFIER
<strong>HANDBOOK</strong>304 TL G-G Amplifier 165should have a reactance of 70 ohms. Thisvalue of reactance is represented by about 550J111fd. at 4 Me., and proportionately smaller capacitancesfor higher frequencies. Since thepeak cathode voltage is only about 400, quitesmall spacing in the cathode tank capacitorcan be used.Figure 11 illustrates a practical groundedgridamplifier, using a single 304TL tube. Abifilar wound coil is used to feed the filamentvoltage to the tube. Shunting inductors areused to tune the filament circuit to higher frequencies.When tuning the g-g stage, it mustbe remembered that the input tuned circuit iseffectively in series with the output circuit asfar as the output energy is concerned. Thus,it is not possible to apply full excitation powerto the stage unless plate current is also flowing.If full excitation is applied in the absenceof plate voltage, the grid of the output tubewould be damaged from excessive excitation.Construction of the amplifier is illustrated infigures 12 and 13.A 3000-volt plate supply is used whichshould have good regulation. An output filtercondenser of at least 10 microfarads should beemployed for Class B operation. Class B operatingbias is -260 volts, obtained from a wellregulated bias supply.For operation in TV areas, additional leadfiltering is required, as well as completescreening of the amplifier.Figure 13UNDERSIDE OF <strong>THE</strong> GROUNDED-GRIDAMPLIFIER8-7 Class ABl Radio FrequencyPower AmplifiersClass ABl r-f amplifiers operate undersuch conditions of bias and excitation thatgrid current does not flow over any portion ofthe input cycle. This is desirable, sincedistortion caused by grid current loading isabsent, and also because the stage is capableof high power gain. Stage efficiency is about58% when a plate current operating angle of2100 is chosen, as compared to 62% for ClassB operation.The level of static (quiescent) plate currentfor lowest distortion is quite critical forClass ABl tetrode operation. This value isdetermined by the tube characteristics, and isnot greatly affected by the circuit parametersor operating voltages. The maximum d-e platepotential is therefore limited by the staticdissipation of the tube, since the resting platecurrent figure is fixed. The static plate currentof a tetrode tube varies as the 3/2 power ofthe screen voltage. For example, raising thescreen voltage from 300 to 500 volts willdouble the plate current. The optimum staticplate current for minimum distortion is alsodoubled, since the shape of the Eg-Ip curvedoes not change.In actual practice, somewhat lower staticplate current than optimum may be employedwithout raising the distortion appreciably,and values of static plate current of 0.6 to0.8 of optimum may be safely used, dependingupon the amount of nonlinearity that can betolerated.As with the class B linear stage, the minimumplate voltage swing of the class ABlamplifier must be kept above the d-e screenpotential to prevent operation in the nonlinearportion of the characteristic curve. A low valueof screen voltage allows greater r-f platevoltage swing, resulting in improvement inplate efficiency of the tube. A balance betweenplate dissipation, plate efficiency, andplate voltage swing must be achieved for bestlinearity of the amplifier.The S·Curve The perfect linear amplifierdelivers a signalthat is a replica of the input signal. Inspectionof the plate characteristic curve of a typicaltube will disclose the tube linearity underclass A operating conditions (figure 14). Thecurve is usually of exponential shape, andthe signal distortion is held to a small valueby operating the tube well below its maximumoutput, and centering operation over the mostlinear portion of the characteristic curve.
166 R-F Vacuum Tube Amplifiers <strong>THE</strong> R AD I 0-MOSTL..INEARPORTION OF~JR"JlT.-it'rt-- IP ~-·-f t t-+.1-t-.Q\!]"eyT_SIC:.NALll--R.F.EOUTINPUTSl~NAL.)1R.F. E INFigure 14Eg-lp CURVEAmplifier operation is eonfined tomost I inear portion of choracterist iccurve.The relationship between exc1t1ng voltagein a Class AB1 amplifier and the r-£ platecircuit voltage is shown in figure 15. With asmall value of static plate current the lowerportion of the line is curved. Maximum undistortedoutput is limited by the point on theline (A) where the instantaneous plate voltagedown to the screen voltage. This "hook" inthe line is caused by current diverted fromthe plate to the grid and screen elements ofthe tube. The characteristic plot of the usuallinear amplifier takes the shape of an S-curve.The lower portion of the curve is straightenedout by using the proper value of static platecurrent, and the upper portion of the curve isavoided by limiting minimum plate voltageswing to a point substantially above the valueof the screen voltage.Operating Parameters The approximate operatingparameters may befor the Class AB 1Linear Amplifier obtained from the ConstantCurrent curves(Eg-Ep) or the Eg-lp curves of the tube inquestion. An operating load line is firstapproximated. One end of the load line isdetermined by the d-e operating voltage of thetube, and the required static plate current.As a starting point, let the product of the platevoltage and current equal the plate dissipationof the tube. Assuming we have a 4-400Atetrode, this end of the load line will fall onpoint A (figure 16). Plate power dissipationis 360 watts (3000V @ 120 ma). The oppositeend of the load line will fall on a point determinedby the minimum instantaneous plate voltage,and by the maximum instantaneous platecurrent. The minimum plate voltage, for bestlinearity should be considerably higher thanFigure 15LINEARITY CURVE OFTYPICAL TETRODE AMPLIFIERAt point "A" the Instantaneous platevo·ltage is swinging down to the valueof screen voltage. At point "B" it isswinging well below the sereen and isapproaching the point where saturation,or plate eurrent limiting takes plaee.the screen voltage. In this case, the screenvoltage is 500, so the minimum plate voltageexcursion should be limited to 600 volts.Class AB1 operation implies no grid current,therefore the load line cannot cross the Eg=Oline. At the point Ep=600, Eg=O, the maximumplate current is 580 ma (Point "B ").Each point the load line crosses a grid voltageaxis may be taken as a point for constructionof the Eg•lp curve, just as was done infigure 22, chapter 6. A constructed curveshows that the approximate static bias voltageis -74 volts, which checks closely with point Aof figure 16. In actual practice, the bias voltageis set to hold the actual dissipation slightlybe low the maximum figure of the tube.The single tone power output is:Emax-EminXIpmax, or 3000-600 X. 58= 348 watts4 4The plate current-angle efficiency factor forthis class of operation is 0. 73, and the actualplate circuit efficiency is:Np=Emax-EminXO. 73, or 3000-600 X 0. 73 = 58.4%Emax 3000The power input to the stage is therefore£2_x 100Npor, 348 = 595 watts58.4The plate dissipation is: 595-348 = 247 watts.
<strong>HANDBOOK</strong>+50 I+ a5r .0~POINT6sco 1000aooo 3000 400Ul "I'\"" IS .i ~ f' 4r-~ t---I-50-L!-+-I 1\ r--- -...i. LOAD LINEr-- r-----aPOINTr- A-75Ii :"-·'. l ............. I-100AB 1 R.F. Power Amplifiers 167+100+ I 754-400A TUBEE SCR = 500 VOLTS~IIiI-125 t'!:tVALUE OF VAl-UE OFEP MIN.= eoo v., MAX. PLATE-1.50I~:o.58ADISSIPATION(3000V.X0.12A •J~OWAr;s)-175 l1 1 l J-200.iFigure 16OPERATING PARAMETERS FOR TETRODE LINEARAMPLIFIER ARE OBTAINED FROM CONSTANT-CURRENTCURVES.It can be seen that the limiting factor forthis class of operation is the static plate diss i·pari on, which is quite a bit higher than theoperating dissipation level. It is possible, atthe expense of a higher level of distortion, todrop the static plate dissipation and to increasethe screen voltage to obtain greater power out·put. If the screen voltage is set at 800, andthe bias increased sufficiently to drop thestatic plate current to 90 ma, the single toned·c plate current may rise to 300 ma, for apower input of 900 watts. The plate circuitefficiency is 55 .6%, and the power output is500 watts. Static plate dissipation is 2 70 watts.At a screen potential of 500 volts, the maxi·mum screen current is less than 1 rna, and undercertain loading conditions may be negative.When the screen potential is raised to 800 voltsmaximum screen current is 18 ma. The per·formance of the tube depends upon the voltagefields set up within the tube by the cathode,control grid, screen grid, and plate. The quantityof current flowing in the screen circuit is onlyincidental to the fact that the screen is main·tained at a positive potential with respect tothe electron stream surrounding it.The tube will perform as expected so long asthe screen current, in either direction, doesnot create undesirable changes in the screenvoltage, or cause excessive screen dissipation.Good regulation of the screen supply is thereforerequired. Screen dissipation is highlyresponsive to plate loading conditions, and theplate circuit should always be adjusted so asto keep the screen current below the maximumdissipation level as established by the appliedvoltage.G-G Class B LinearTetrode AmplifierCertain tetrode and pentodetubes, such as the 6AG7,837, and 803 perform we IIas grounded grid class B linear amplifiers. Inthis configuration both grids and the suppressorare grounded, and exc it at ion is applied to thecathode circuit of the tube. So connected, thetubes take on characteristics of high-mu triodes.No bias or screen supplies are required forthis type of operation, and reasonably linearFigure 17SIMPLE GROUNDED-GRIDLINEAR AMPLIFIERB+ 300-700 V.
168 R- F Vacuum Tube Amplifiers <strong>THE</strong> R AD I 0VI837 500Vz837V3837500V4803V;803 500~RF~.001 -=-5 KV+1250V.T, ~EACH1'~'1'01115 v rv+2500Figure 183-STAGE KILOWATT LINEARAMPLIFIER FOR 80 OR 40METER OPERATIONAn open frame filament transformer maybe used for Tl. Cathode taps are adjustedfor proper excitation of followingstage.operation can be had with a very minimum ofcircuit components (figure 17). The input im·pedance of the g·g stage falls between 100and 250 ohms, eliminating the necessity ofswamping resistors, even though considerablepower is drawn by the cathode circuit of theg-g stage.Power gain of a g-g stage varies from approximately20 when tubes ·of the 6AG7 typeare used, down to five or six for the 837 and803 tubes. One or more g-g stages may becascaded to provide up to a kilowatt of power,as illustrated in figure 18.The input and output circuits of cascadedg-g stages are in series, and a variation inload impedance of the output stage reflectsback as a proportional change on the inputcircuit. If the first g-g stage is driven by a highimpedance source, such as a tetrode amplifier,any change in gain will automatically be compensatedfor. If the gain of V 4-VS drops, theinput impedance to that stage will rise. Thischange will reflect through V2:-V3 so that theload impedance of Vl rises. Since V1 has ahigh internal impedance the output voltagewill rise when the load impedance rises. Theincreased output voltage will raise the outputvoltage of each g-g stage so that the overalloutput is nearly up to the initial value beforethe drop in gain of V 4-VS.The tank circuits, therefore, of all g-g stagesmust be resonated with low plate voltage andexcitation applied to the tubes. Tuning of onestage will affect the other stages, and the inputand coupling of each stage must be adjustedin turn until the proper power limit isreached.Operating Data for4-400A GroundedGrid LinearAmplifierExperiments have beenconducted by Collins RadioCo. on a grounded gridlinear amplifier stageusing a 4-400A tube for the h. f. region. Theoperating characteristics of the amplifier aresummarized in figure 19. It can be noted thatunusually low screen voltage is used on thetube. The use of lower screen voltage has theadverse effect of increasing the driving power,but at the same time the static plate currentof the stage is decreased and linearity is imimproved.For grounded grid operation of the4-400A, a screen voltage of 300 volts (filamentto screen) gives a reasonable compromise betweenthese factors.OPERATING DATA FOR 4-400A/4-250AGcG. LINEAR AB1 AMPLIFIER(SINGLE TONE)D-C SCREEN VOLTAGE +300 +300D-C PLATE VOLTAGE + 3000 +3500STATIC PLATE CURRENT 60 MA 60MA.D-C GRID BIAS -60 V. -59 v.PEAK CATHODE SWING87 v .. 113 v.MINIMUM PLATE VOLTA
<strong>HANDBOOK</strong> 16 9TOP VIEW OF A 4-250A AMPL FIERPi-network tetrode amplifier may be operated Class AB,, Class 8, or ass C by varying various potentialsapplied to tube. Same general physical and mechanical des gn applies in each case>.
CHAPTER NINEThe OscilloscopeThe cathode-ray oscilloscope (also calledoscillograph) is an instrument which permitsvisual examination of various electrical phenomenaof interest to the electronic engineer.Instantaneous changes in voltage, current andphase are observable if they take place slowlyenough for the eye to follow, or if they areperiodic for a long enough time so that the eyecan obtain an impression from the screen ofthe cathode-ray tube. In addition, the cathoderayoscilloscope may be used to study anyvariable (within the limits of its frequencyresponse characteristic) which can be convertedinto electrical potentials. This conversionis made possible by the use of some typeof transducer, such as a vibration pickup unit,pressure pickup unit, photoelectric cell, microphone,or a variable impedance. The useof such a transducer makes i:he oscilloscopea valuable tool in fields other than electronics.the rectptent of signals from two sources: thevertical and horizontal amplifiers. The operationof the cathode-ray tube itself has beencovered in Chapter 4; the auxiliary circuitspertaining to the cathode-ray tube will becovered here.The VerticalAmp I ifierThe incoming signal which isto be examined is applied tothe terminals marked VerticalInput and Ground. The Vertical Input terminalis connected through capacitor C 1(figure 2) sothat the a-c component of the input signal appearsacross the vertical amplifier gain controlpotentiometer, R 1 • Thus the magnitude ofthe incoming signal may be controlled to providethe desired deflection on the screen ofVERTICALINPUT9-1 A Typical Cathode-RayOscilloscopeFor the purpose of analysis, the operationof a simple oscilloscope will be described.The Du Mont type 274-A unit is a fit instrumentfor such a description. The block diagramof the 27 4-A is shown in figure 1. The electronbeam of the cathode-ray tube can be movedvertically or horizontally, or the vertical andhorizontal movements may be combined to producecomposite patterns on the tube screen.As shown in figure 1, the cathode-ray tube isGND.O::),Figure 1BLOCK DIAGRAM, TYPE 274-ACATHODE-RAY OSCILLOSCOPE170
Time Base Generator 171c.;;~,·~~-'--~----------INTENSITY MOl),Ca0.1..1.6O+Figure 2TYPICAL AMPLIFIER SCHEMATIC'="120Kthe cathode-ray tube. Also, as shown in figure1, S 2 has been incorporated to by-pass thevertical amplifier and capacitively couple theinput signal directly to the vertical deflectionplate if so desired.In figure 2, V 1is a 6AC7 pentode tube whichis used as the vertical amplifier. As the signalvariations appear on the grid of V, variationsin the plate current of V 1 will take place.Thus signal variations will appear in oppositephase and greatly amplified across the plateresistor, R 7• Capacitor C 2 has been added a·cross R 2 in the cathode circuit of V 1 to flattenthe frequency response of the amplifier at thehigh frequencies. This capacitor because ofits low value has very little effect at low in·put frequencies, but operates more effectivelyas the frequency of the signal increases. Theamplified signal delivered by V 1 is now ap·plied through the second half of switch S 2 andcapacitor C 4 to the free vertical deflectionplate of the cathode-ray tube (figure 3).The HorizontalAmplifierThe circuit of the horizontalamplifier and the circuit ofthe vertical amplifier, de·scribed in the above paragraph, are similar. Aswitch in the input circuit makes provision forthe input from the Horizontal Input terminalsto be capacitively coupled to the grid of thehorizontal amplifier or to the free horizontaldeflection plate thus by-passing the amplifier,or for the output of the sweep generator to becapacitively coupled to the amplifier, as shownin figure 1.The Time BaseGeneratorInvestigation of e I e c t ric alwave forms by the use of acathode-ray tube frequentlyrequires that some means be readily availableto determine the variation in these wave formswith respect to time. When such a time baseis required, the patterns presentedon the cath•ode-ray tube screen show the variation in am·plitude of the input signal with respect toFigure 3SCHEMATIC OF CATHODE-RAY TUBECIRCUITSA SBP lA cathocle-ray tube Is usee/ in thisinstrument. As shown, the necessary potentialsfor operating this tube are obtaineclfrom a voltage clivlcler macle up of resistorsR21 through R26 inclusive. The intensity ofthe beam is aclfustecl by moving the contacton R21• This ac/fusts the potential on thecathocle more or less negative with respectto the gric/ which is operatecl at the full neg•atlve vo/tage-1200 volts. Focusing to theclesirecl sharpness is accompllshecl by acl·fusting the contact on R23 to provicle thecorrect potential for anocle no. J. /ntercle·penclency between the focus one/ the. inten•s/ty controls is inherent In all electrostatl·cally focusec/ cathocle-ray tubes. In short,there Is an optimum seHing of the focus con•trol for every seHing of the intensity con•trol. The secane/ anoc/e of the SBP J A is oper•atecl at grounc/ potential in this Instrument.AI so one of each pair of c/eflect/on plates isoperatec/ at grounc/ potential.The cathocle is operatec/ at a high negativepotential (approximately 7200 volts) so thatthe total overall accelerating voltage of thistube is regarc/ec/ as 7200 volts since the sec•one/ anocle Is operatec/ at grouncl potential.The vertical one/ horizontal positioning controlswhich are connectec/ to their respectivec/eflection plates are capable of-;IJpplylngeither a positive or negative c/.c potentialto the cleflection plates. This permits thespot to be posltionecl at any cleslrecl placeon the ttntire screen.time. Such an arrangement is made possibleby the inclusion in the oscilloscope of a TimeBase-Generator. The function of this genera·tor is to move the spot across the screen at aconstant rate from left to right between twoselected points, to return the spot almost instantaneouslyto its original position, and to
172 The Oscilloscope<strong>THE</strong> R AD I 0ABCFigure 4SAWTOOTH WAVE FORMrepeat this procedure at a specified rate. Thisaction is accomplished by the voltage outputfrom the time base (sweep) generator. Therate at which this voltage repeats the cycleof sweeping the spot across the screen is referredto as the sweep frequency. The sweepvoltage necessary to produce the motion describedabove must be of a sawtooth waveform,such as that shown in figure 4.The sweep occurs as the voltage variesfrom A to B, and the return trace as the voltagevaries from B to C. If A-B is a straightline, the sweep generated by this voltage willbe linear. It should be realized that the sawtoothsweep signal is only used to plot variationsin the vertical axis signal with respectto time. Specialized studies have made necessarythe use of sweep signals of variousshapes which are introduced from an externalsource through the Horizontal Input terminals.The Sawtooth The sawtooth voltage necesGenerator sary to obtain the linear timebase is generated by the circuitof figure 5, which operates as follows:A type 884 gas triode (V 3 ) is used for thesweep generator tube. This tube contains aninert gas which ionizes when the voltage betweenthe cathode and the plate reaches a certainvalue. The ionizing voltage depends uponthe bias voltage of the tube, which is determinedby the voltage divider resistors R, 2-RwWith a specific negative bias applied to the884 tube, the tube will ionize (or fire) at aspecific plate voltage.Capacitors C, 0 -C 14 are selectively connectedin parallel with the 884 tube. Resistor R 11limits the peak current drain of the gas triode.The plate voltage on this tube is obtainedthrough resistors R 28 , R 27and R 11• The voltageapplied to the plate of the 884 tube cannot reachthe power supply voltage because of the chargingeffect this voltage has upon the capacitorwhich is connected across the tube. This capacitorcharges until the plate voltage becomeshigh enough to ionize the gas in the tube. Atthis time, the 884 tube starts to conduct andthe capacitor discharges through the tube untilits voltage falls to the extinction potential ofthe tube. When the tube stops conducting, thecapacitor voltage builds up until the tube firesagain. As this action continues, it results inthe sawtooth wave form of figure 4 appearingat the junction of R 11and R 27•Synchronization Provision has been made sothe sweep generator may besynchronized from the vertical amplifier orfrom an external source. The switch S 1shownin figure 5 is mounted on the front panel to beeasily accessible to the operator.If no synchronizing voltage is applied, thedischarge tube will begin to conduct when theplate potential reaches the value of E 1 (FiringPotential). When this breakdown takes placeand the tube begins to conduct, the capacitoris discharged rapidly through the tube, and theplate voltage decreases until it reaches theextinction potential Ex. At this point conductionceases, and the plate potential rises slowlyas the capacitor begins to charge throughR 27 and R 28 • The plate potential will againreach a point of conduction and the circuitwill start a new cycle. The rapidity of theplate voltage rise is dependent upon the circuitconstants R 2 , R 28 , and the capacitor selected,SWEEPGENERATOROUTPUTR21470KEY.TERNALSYNCR285MFINE FREQ.~,;__ _______J CONTROLFigure 5SCHEMATIC OF SWEEP GENERATOR
<strong>HANDBOOK</strong> The Oscilloscope 173EP vs EgEP+STATIC CONTROLCHARACTERISTIC'!
INTENSITY~~o-----1[~----~ll~_~o~~----------------------------------------------------------------------------------~---------------------.l-470KO. IV65BP1A-1::rI'D0VIn180KIM4,7 My POSFOCUSIM0VIn0"0I'DEXT.SYNC.S1SYNC.SELECTOR390K 500K 12.0K82.0 K0.1------....IOKlOOKIMSlt;NALe>--J#r-XIOKXPOS.4.7 t.A-1:::cmFigure 8CIRCUIT OF TYPE 274-A CATHODE-RAY OSCILLOSCOPE0
<strong>HANDBOOK</strong>Display of Waveforms 175lt~~-0IIII~~c~Z~1=Figure 9PROJECTION DRAWING OF A SINEWAVEAPPLIED TO <strong>THE</strong> VERTICAL AXIS AND ASAWTOOTH WAVE OF <strong>THE</strong> SAME FR~QUENCY APPLIED SIMULTANEOUSLY ON<strong>THE</strong> HORIZONTAL AXISelectrodes of the cathode-ray tube, and forcertain positioning controls.The positive low voltage supply consistsof full-wave rectifier (V 5 ), the output of whichis filtered by a capacitor input filter (20-20 p.fd.and 8 H). It furnishes approximately 400 volts.The high voltage power supply employs a halfwave rectifier tube, V 4 • The output of this rectifieris filtered by a resistance-capacitor filterconsisting of 0.5-0.5 p.fd. and .18M. Avoltage divider network attached from the outputof this filter obtains the proper operatingpotentials for the various electrodes of thecathode-ray tube. The complete schematic ofthe Du Mont 274-A Oscilloscope is shown infigure 8.9-2 Display of WaveformsTogether with a working knowledge of thecontrols of the oscilloscope, an understandingof how the patterns are traced on the screenmust be obtained for a thorough knowledge ofoscilloscope operation. With this in mind acareful analysis of two fundamental waveformpatterns is discussed under the f o 11 owingheadings:a. Patterns plotted against time (using thesweep generator for horizontal deflection).b. Lissajous Figures (using a sine wave forhorizontal deflection).Patterns PlottedAgainst TimeA sine wave is typical ofsuch a pattern and is convenientfor this study. Thisr~sEcl:-t,sEcjrvsJ·~-·1S&·4--- .----~- .~--.:TIME---...1;::: "Figure 10II1 I I7'1:.• IIIIIJII.35 SEC,: ______ j_PROJECTION DRAWING SHOWING <strong>THE</strong> RESULTANT PATTERN WHEN <strong>THE</strong> FR~QUENCYOF <strong>THE</strong> SAWTOOTH IS ONE-HALFOF THAT EMPLOYED IN FIGURE 9wave is amplified by the vertical amplifierand impressed on the vertical (Y-axis) deflectionplates of the cathode-ray tube. Simultaneouslythe sawtooth wave from the time basegenerator is amplified and impressed on thehorizontal (X-axis) deflection plates.The electron beam moves in accordancewith the resultant of the sine and sawtoothsignals. The effect is shown in figure 9 wherethe sine and sawtooth waves are graphicallyrepresented on time and voltage axes. Pointson the two waves that occur simultaneouslyare numbered similarly. For example, point 2on the sine wave and point 2 on the sawtoothwave occur at the same instant. Therefore theposition of the beam at instant 2 is the resultantof the voltages on the horizontal and verticaldeflection plates at instant 2. Referringto figure 9, by projecting lines from the twopoint 2 positions, the position of the electronbeam at instant 2 can be located. If projectionswere drawn from every other instantaneousposition of each wave to intersect on thecircle representing the tube screen, the intersectionsof similarly timed projections wouldtrace out a sine wave.In summation, figure 9 illustrates the principlesinvolved in producing a sine wave traceon the screen of a cathode-ray tube. Each intersectionof similarly timed projections representsthe position of the electron beam actingunder the influence of the varying voltagewaveforms on each pair of deflection plates.Figure 10 shows the effect on the pattern ofdecreasing the frequency of the s a w too t h
176 The Oscilloscope<strong>THE</strong> R AD I 0BCFigure 12METHOD OF CALCULATING FREQUENCYRATIO OF LISSAJOUS FIGURESFigure 11PROJECTION DRAWING SHOWING <strong>THE</strong> RESULTANT LISSAJOUS PATTERN WHEN ASINE WAVE APPLIED TO <strong>THE</strong> HORIZONTAL AXIS IS THREE TIMES THAT AP-PLIED TO <strong>THE</strong> VERTICAL AXISwave. Any recurrent waveform plotted againsttime can be displayed and analyzed by thesame procedure as used in these examples.The sine wave problem just illustrated istypical of the method by which any waveformcan be displayed on the screen of the cathode·ray tube. Such waveforms as square wave,sawtooth wave, and many more irregular recur·rent waveforms can be observed by the samemethod explained in the preceding paragraphs.Obtaining a LissajousPattern on the screenOscilloscope Settings1. The horizontal am·plifier should be dis con·nected from the sweeposcillator. The signalto be examined should be conn e c t e d to thehorizontal amplifier of the oscilloscope.2. An audio oscillator signal should be con·nected to the vertical amplifier of the oscilloscope.3. By adjusting the frequency of the audiooscillator a stationary pattern should be ob·tained on the screen of the oscilloscope. It isnot necessary to stop the pattern, but merelyto slow it up enough to count the loops at theside of the pattern.4. Count the number of loops which intersectan imaginary vertical line AB and the numberof loops which intersect the imaginary horizontalline BC as in figure 12. The ratio ofthe number of loops which intersect AB is to9-3 Lissajous FiguresAnother fundamental pattern is the Lissajousfigure, named after the 19th century Frenchscientist. This type of pattern is of particularuse in determining the frequency ratio betweentwo sine wave signals. If one of these signalsis known, the other can be easily calculatedfrom the pattern made by the two signals uponthe screen of the cathode-ray tube. Commonpractice is to connect the known signal to thehorizontal channel and the unknown signal tothe vertical channel.The presentation of Lissajous figures canbe analyzed by the same method as previouslyused for sine wave presentation. A simple ex·ample is shown in figure 11. The frequencyratio of the signal on the horizontal axis to thesignal on the vertical axis is 3 to 1. If theknown signal on the horizontal axis is 60 cy·des per second, the signal on the verticalaxis is 20 cycles.0 8CD RATIO,,, ®RATIO 2.:1@RATIO 5:~Figure 13O<strong>THE</strong>R LISSAJOUS PATTERNS
<strong>HANDBOOK</strong> Lissajous Figures 177-t-- -t$t- -$ -~PHASE DIFFERENCE=o• PHASE DIFFERENCE=4!t• PHASE DIFFERENCE=9o• PHASE DIFFERENCE=135°-~$*PHASE OIFFERENcE=zz~o PHASE DIFFERENCE 270° PHASE DIFFERENCE 315•Figure 14LISSAJOUS PATTERNS OBTAINED FROM <strong>THE</strong> MAJOR PHASE DIFFERENCE ANGLESthe number of loops which intersect BC as thefrequency of the horizontal signal is to thefrequency of the vertical signal.Figure 13 shows other examples of Lissajousfigures. In each case the frequency ratioshown is the frequency ratio of the signal onthe horizontal axis to that on the verticalaxis.Phase Differ·ence PatternsComing under the heading ofLissajous figures is the methodused to determine the phasedifference between signals of the same frequency.The patterns in v o 1 v e d take on theform of ellipses with different degrees of ec·centricity. ,The following steps should be taken to obtaina phase-difference pattern:1. With no signal input to the oscilloscope,the spot should be centered on the screenof the tube.2. Connect one signal to the vertical ampli·fier of the oscilloscope, and the othersignal to the horizontal amplifier.3. Connect a common ground between thetwo frequencies under investigation andthe oscilloscope.4. Adjust the vertical amplifier gain so asto give about 3 inches of deflection on a5 inch tube, and adjust the calibratedscale of the oscilloscope so that the verticalaxis of the scale coincides preciselywith the vertical deflection of the spot.5. Remove the signal from the vertical amplifier,being careful not to change thesetting of the vertical gain control.6. Increase the gain of the horizontal amplifierto give a deflection exactly thesame as that to which the vertical amplifiercontrol is adjusted (3 inches). Reconnectthe signal to the vertical ampli·fier.The resulting pattern will give an accuratepicture of the exact phase difference betweenthe two waves. If these two patterns are exactlythe same frequency but different in phaseand maintain that difference, the pattern onthe screen will remain stationary. If, 'however,one of these frequencies is drifting slightly,the pattern will drift slowly through 360 °. Thephase angles of 0°, 45°, 90°, 135°, 180°,225 °, 270 °, 315 ° are shown in figure 14.Each of the eight patterns in figure 14 canbe analyzed separately by the previously usedTIME-~ _/~"Figure 15PROJECTION DRAWING SHOWING <strong>THE</strong> RESULTANT PHASE DIFFERENCE PATTERNOF TWO SINE WAVES 45° OUT OF PHASE
178 The Oscilloscope <strong>THE</strong> R AD I 0Figure 16EXAMPLES SHOWING <strong>THE</strong> USE OF <strong>THE</strong> FORMULA FOR DETERMINATION OF PHASEDIFFERENCEprojecthn method. Figure 15 shows two sinewaves which differ in phase being projectedon to the screen of the cathode-ray tube. Thesesignals represent a phase difference of 45 °.It is extremely important: (1) that the spothas been centered on the screen of the cathoderaytube, (2) that both the horizontal and verticalamplifiers have been adjusted to giveexactly the same gain, and (3) that the calibratedscale be originally set to coincide withthe displacement of the signal along the verticalaxis. If the amplifiers of the oscilloscopeare not used for conveying the signal to thedeflection plates of the cathode-ray tube, thecoarse frequency switch should be set to horizontalinput direct and the vertic a 1 inputswitch to direct and the outputs of the twosignals must be adjusted to result in exactlythe same vertical deflection as horizontal deflection.Once this deflection has been set byeither the oscillator output controls or the amplifiergain controls in the oscillograph, itshould not be changed for the duration of themeasurement.Determination ofthe Phase AngleThe relation commonly usedin determining the ph as eangle between signals is:Y interceptSine e = ----y maximumC]TRAPEZOIDALFigure 17MODULATION PATTERNFigure 18MODULATED CARRIER WAVE PATTERNFigure 19PROJECTION DRAWING SHOWING TRAPEZOIDAL PATTERN
<strong>HANDBOOK</strong>Trapezoidal Pattern 179MODULATEDCARRIERfl. F. POWER AMPLIFIERTIME---CROFigure 20PROJECTION DRAWING SHOWING MODU·LATED CARRIER WAVE PATTERNB+NOTE~ IF R.F, PICKUP IS INSUFFICIENT,A TUNED CIRCUIT MAY BE USEDAT <strong>THE</strong> OSCILLOSCOPE AS SHOWN.where e = phase angle between signalsY intercept = point where ellipse crosses ver·tical axis measured in tenths ofinches. (Calibrations on thecalibrated screen)Y maximum = highest vertical point on ellipsein tenths of inchesSeveral examples of the use of the formula aregiven in figure 16. In each case the Y interceptand Y maximum are indicated togetherwith the sine of the angle and the angle itself.For the operator to observe these various patternswith a single signal source such as thetest signal, there are many types of phaseshifters which can be used. Circuits can beobtained from a number of radio text books.The procedure is to connect the original signalto the horizontal channel of the oscilloscopeand the signal which has passed throughthe phase shifter to the vertical channel ofthe oscilloscope, and follow the procedure setforth in this discussion to observe the variousphase shift patterns.9-4 Monitoring TransmitterPerformance with the OscilloscopeThe oscilloscope may be used as an aid forthe proper operation of a radiotelephone trans·mitter, and may be used as an indicator of theoverall performance of the transmitter outputsignal, and as a modulation monitor.Waveforms There are two types of patternsthat can serve as indicators, thetrapezoidal pattern (figure 17) and the modu-Figure 21MONITORING CIRCUIT FOR TRAPEZOI·DAL MODULATION PATTERNlated wave pattern (figure 18). The trapezoidalpattern is presented on the screen by impressinga modulated carrier wave signal on the ver·tical deflection plates and the sign a I thatmodulates the carrier wave signal (the modulatingsignal) on the horizontal de fIe c t ionplates. The trapezoidal pattern can be ana·lyzed by the method used previously in analyzingwaveforms. Figure 19 shows how the sig·nals cause the electron beam to trace out thepattern.The modulated wave pattern is accomplishedby presenting a modulated carrier wave on thevertical deflection plates and by using thetime-base generator for horizontal deflection.The modulated wave pattern also can be usedfor analyzing waveforms. Figure 20 shows howthe two signals cause the electron beam totrace out the pattern.The TrapezoidalPatternThe oscilloscope connectionsfor obtaining a trapezoidalpattern are shown infigure 21. A portion of the audio output of thetransmitter modulator is applied to the horizontalinput of the oscilloscope. The verticalamplifier of the oscilloscope is disconnected,and a small amount of modulated r·f energy iscoupled directly to the vertical de fIe c t ionplates of the oscilloscope. A small pickuploop, loosely coupled to the final amp I if i e rtank circuit and connected to the vertical de·
1 8 0 The Oscilloscope <strong>THE</strong> R AD I 0
<strong>HANDBOOK</strong> Receiver Alignment 181CARRIER WAVE PATTERNFigure 26 Figure 27 Figure 28(LESS THAN 100% MODULATION) (100% MODULATION) {OVER MODULATION)The ResonanceCurve on theScreenTo present a resonance curveon the screen, a frequencymodulatedsignal source mustbe available. This signalsource is a signal generator whose output isthe fundamental i-f frequency which is frequency-modulated5 to 10 kc. each side of thefundamental frequency. A signal generator ofthis type generally takes the form of an ordinarysignal generator with a rotating motordriven tuned circuit capacitor, called a wob·8KC 4KC4KC IKCFigure 29FREQUENCY RESPONSE CURVE OF <strong>THE</strong>1-F OF A LOW PRICED RECEIVERbulator, or its electronic equivalent, a react·ance tube.The method of presenting a resonance curveon the screen is to connect the vertical chan·nel of the oscilloscope across the detectorload of the receiver as shown in the detectorsof figure 31 (between point A and ground) andthe time-base generator output to the horizon·tal channel. In this way the d·c voltage acrossthe detector load varies with the frequencieswhich are passed by the i·f system. Thus, ifthe time-base generator is set at the frequencyof rotation of the motor driven capacitor, orthe reactance tube, a pattern resembling fig·ure 32, a double resonance curve, appears onthe screen.Figure 32 is explained by considering figure33. In half a rotation of the motor drivencapacitor the frequency increases from 445kc. to 465 kc., more than covering the rangeof frequencies passed by the i·f system.Therefore, a full resonance curve is presentedon the screen during this half cycle of rota·tion since only half a cycle of the voltage pro·ducing horizontal deflection has transpired.In the second half of the rotation the motorTRIODE DETECTORDIODE DETECTORFigure 30FREQUENCY RESPONSE OFHIGH-FIDELITY 1-F SYSTEMFigure 31CONNECTION OF <strong>THE</strong> OSCILLOSCOPEACROSS <strong>THE</strong> DETECTOR LOAD
182 The Oscilloscope<strong>THE</strong> R AD I 0Figure 32DOUBLE RESONANCE CURVEFigure 33DOUBLE RESONANCE ACHIEVED BYCOMPLETE ROTATION OF <strong>THE</strong> MOTORDRIVEN CAPACITORcurve is observed as it sweeps the spot acrossthe screen from left to right; and it is observedagain as the sine wave sweeps the spot backagain from right to left. Under these conditionsthe two response curves are superimposedon each other and the high frequencyresponses of both curves are at one end andthe low frequency response of both curves isat the other end. The i-f trimmer capacitorsare adjusted to produce a r e s pons e curvewhich is symmetrical on each side of the fundamentalfrequency.When using sawtooth sweep, the two responsecurves can also be superimposed. Ifthe sawtooth signal is generated at exactlytwice the frequency of rotation of the motordriven capacitor, the two resonance curveswill be superimposed (figure 34) if the i-ftransformers are properly tuned. If the twocurves do not coincide the i-f trimmer capacitorsshould be adjusted. At the point of coincidencethe tuning is correct. It should bepointed out that rarely do the two curves agreeperfectly. As a result, optimum adjustment ismade by making the peaks coincide. This latterprocedure is the one generally used in i-fadjustment. When the two curves coincide, itis evident that the i-f system responds equallyto signals higher and lower than the fundamentali-f frequency.9-6 Single Sideband ApplicationsFigure 34SUPER-POSITION OF RESONANCE CURVESMeasurement of power output and distortionare of particular importance in SSB transmitter.adjustment. These measurements are related tothe extent that distortion rises rapidly whenthe power amplifier is overloaded. The useablepower output of a SSB transmitter is often definedas the maximum peak envelope powerdriven capacitor takes the frequency of thesignal in the reverse order through the rangeof frequencies passed by the i-f system. Inthis interval the time-base generator sawtoothwaveform completes its cycle, drawing theelectron beam further across the screen andthen returning it to the starting point. Subsequentcycles of the motor driven capacitor andthe sawtooth voltage merely retrace the samepattern. Since the signal being viewed is appliedthrough the vertical amplifier, the sweepcan be synchronized internally.Some signal generators, particularly thoseemploying a reactance tube, provide a sweepoutput in the form of a sine wave which issynchronized to the frequency with which thereactance tube is swinging the fundamentalfrequency through its limits, usually 60 cyclesper second. If such a signal is used for horizontaldeflection, it is already synchronized.Since this signal is a sine wave, the response®Figure 3SSINGLE TONE PRESENTATIONOscilloscope trace of SSB signalmodulated by single tone (A).Incomplete carrier supression orspurious products will showmodulated envelope of (B). Theratio of supression is:S = 20 log A+BA-B
<strong>HANDBOOK</strong>S.S.B. Applications 183GERMANIUM2.5 MHR-F SSB INPUT DIODE RFC.67 ~~~i~'oTRAGE (cil-.-1~.....-'00"0'-.-.....---
184 The Oscilloscoper-------------------------~e2OUTPUTSIGNALLEVELFigure 38EFFECT OF INADEQUATERESPONSE OF VERTICALAMPLIFIERINPUT SIGNAL LEVEL0Figure 41ORDINATES ON LINEARITYCURVE FOR 3RD ORDERDISTORTION EQUATIONFigure 39DOUBLE TRACECAUSED BY PHASESHIFTthe envelope detectors in parallel. A perfectlystraight line trace will result when everythingis working properly. One detector is then connectedto the other r-f source through a voltagedivider adjusted so that no appreciable changein the setting of the oscilloscope amplifiercontrols is required. Figure 40 illustrates sometypical linearity traces. Trace A is caused byinadequate static plate current in class A orclass B amplifiers or a mixer stage. To regainlinearity, the grid bias of the stage should bereduced, the screen voltage should be raised,or the signal level should be decreased. TraceB is a result of poor grid circuit regulationwhen grid current is drawn, or a result of non-linear plate characteristics of the amplifiertube at large plate swings. More grid swampingshould be used, or the exciting signal shouldbe reduced. A combination of the effects of Aand Bare shown in Trace C. Trace D illustratesamplifier overloading. The exciting signalshould be reduced.A means of estimating the distortion levelobserved is quite useful. The first and thirdorder distortion components may be derived byan equation that will give the approximatesignal-to-distortion level ratio of a two tonetest signal, operating on a given linearity curve.Figure 41 shows a linearity curve with twoordinates erected at half and full peak inputsignal level. The length of the ordinates nand e2 may be scaled and used in the followingequation:Signal-to-distortion ratio in db= 20 log 8 e 1-e22 et-q0000®TYPICAL LINEARITY TRACES@Figure 40TYPICAL LINEARITYTRACES
CHAPTER TENSpecial VacuumTube CircuitsA whole new concept of vacuum tube applicationshas been developed in recent years.No longer are vacuum tubes chained to thefield of communication. This chapter is devotedto some of the more common circuits encounteredin industrial and military applicationsof the vacuum tube.10-1 Limiting CircuitsThe term limiting refers to the removal orsuppression by electronic means of the ex·tremities of an e I e c t ron i c signal. Circuitswhich perform this function are referred to aslimiters or clippers. Limiters are useful inwave-shaping circuits where it is desirable tosquare off the extremities of the applied signal.A sine wave may be applied to a limitercircuit to produce a rectangular wave. Apeaked wave may be applied to a limiter cir·cuit to eliminate either the positive or nega·ti ve peaks from the output. Limiter circuitsare employed in FM receivers where it is nee·essary to limit the amplitude of the signal ap·plied to the detector. Limiters may be used toreduce automobile ignition noise in short-wavereceivers, or to maintain a high average levelof modulation in a transmitter. They may alsobe used as protective devices to limit inputsignals to special circuits.Diode Limiters The characteristics of adiode tube are such that thetube conducts only when the plate is at a positivepotential with respect to the cathode. Apositive potential may be placed on the cathode,but the tube will not conduct until thevoltage on the plate rises above an equallypositive value. As the plate becomes morepositive with respect to the cathode, the diodeconducts and passes that portion of the wavethat is more positive than the cathode voltage.Diodes may be used as either series or parallellimiters, as shown in figure 1. A diode maybe so biased that only a certain portion of thepositive or negative cycle is removed.Audio PeakLimitingAn audio peak clipper consistingof two diode limiters may be usedto limit the amplitude of an audiosignal to a predetermined value to providea high average level of modulation withoutdanger of overmodulation. An effective limiterfor this service is the series-diode gate clipper.A circuit of this clipper is shown in figure2. The audio signal to be clipped is cou·pled to the clipper through C,. R, and R 2 arethe clipper input and output load resistors.The clipper plates are tied together and areconnected to the clipping level control, R.,through the series resistor, R 3 • R 4 acts as avoltage divider between the high voltage sup·ply and ground. The exact point at which clip-185
186 Special Vacuum Tube Circuits <strong>THE</strong> R AD I 00T _{\_;,.," ·__,__.,I :.~":.----r:::rv, \J 'v-= -=0(\ ~,N___.,i.,. M~'P.,. eouT E--vv=o\Jt©E =VOLTAGE DROPACROSS DIODEeiN -----.-f:\......--.-1-e OUTv v " \}(\ (\ I.,. cv = (\ (\@(\ ~IN -~l.,.-,/WN-'1¥-.,...----- ~U~ iVV=c'r@E =VOLTAGE DROPACROSS DIODE(\~~ __I.,......,M+,._@ ..... E_____ ;;~: \/~-.,. -~)®--I....(\v~~ =--~M~-~~-~ ----0~~~tf®Figure 1VARIOUS DIODE LIMITING CIRCUITSSeries diodes limiting positive and negative peaks are shown in A and B. Parallel diodes limitingpositive and negative peaks are shown in C and D. Parallel diodes limiting above and belowground are shown in E and F. Parallel diode limiters which pass negative and positive peaksare shown in G and H.ping will occur is set by R 4 , which controls thepositive potential applied to the diode plates.Under static conditions, a d-e voltage is obtainedfrom R 4 and applied through R 3to bothplates of the 6AL5 tube. Current flows throughR 4 , R 3 , and divides through the two diodesections of the 6AL 5 and the two load resistors,R, and R 2 • All parts of the clipper circuitare maintained at a positive potential aboveground. The voltage drop between the plateand cathode of each diode is very small comparedto the drop across the 300,000-ohm resistor(R 3 ) in series with the diode plates.The plate and cathode of each diode are thereforemaintained at approximately equal potentialsas long as there is plate current flow.Clipping does not occur until the peak audioinput voltage reaches a value greater than thestatic voltages at the plates of the diode.Assume that R 4 has been set to a point thatwill give 4 volts at the plates of the 6AL5.When the peak audio input voltage is less than4 volts, both halves of the tube conduct at alltimes. As long as the tube conducts, its resistanceis very low compared with the plateresistor R 3 • Whenever a voltage change occursacross input resistor R., the voltage at all ofthe tube elements increases or decreases bythe same amount as the input voltage change,and the voltage drop across R 3 changes by anequal amount. As long as the peak input voltageis less than 4 volts, the 6AL5 acts merelyas a conductor, and the output cathode is permittedto follow all voltage changes at the inputcathode.If, under static conditions, 4 volts appear atthe diode plates, then twice this voltage (8volts) will appear if one of the diode circuits
<strong>HANDBOOK</strong>Clamping Circuits 187B+(\(\vvE-=o=cA£=GRID-CATHODE RESISTANCEWHEN GRID IS DRIVEN POSITIVEFigure 2<strong>THE</strong> SERIES-DIODE GATE CLIPPER FORAUDIO PEAK LiMITINGFigure 3GRID LIMITING CIRCUITis opened, removing its d-e load from the circuit.As long as only one of the diodes continuesto conduct, the voltage at the diodeplates cannot rise above twice the voltage selectedby R 4 • In this example, the voltage cannotrise above 8 volts. Now, if the input audiovoltage applied through C, is increased to anypeak value between zero and plus 4 volts, thefirst cathode of the 6AL 5 will increase in voltageby the same amount to the proper value between4 and 8 volts. The other tube elementswill assume the same potential as the firstcathode. However, the 6AL5 plates cannot increasemore than 4 volts above their original4-volt static level. When the input voltage tothe first cathode of the 6AL5 increases tomore than plus 4 volts, the cathode potentialincreases to more than 8 volts. Since the platecircuit potential remains at 8 volts, the firstdiode section ceases to conduct until the inputvoltage across R, drops below 4 volts.When the input volr.age swings in a negativedirection, it will subtract from the 4-volt dropacross R, and decrease the voltage on the inputcathode by an amount equal to the inputvoltage. The plates and the output cathode willfollow the voltage lf•vel at the input cathodeas long as the input voltage does not swingbelow minus 4 volts. If the input voltage doesnot change more than 4 volts in a negativedirection, the plates of the 6AL5 will also becomenegative. The potential at the outputcathode will follow the input cathode voltageand decrease from its normal value of 4 voltsuntil it reaches zero potential. As the inputcathode voltage decr,~ases to less than zero,I ii> ""-;1 Oo"'0=-rrrrr +tOO-- -- -- ----100-- -- -- -: -: 0~:-the. plates will follow. Howe.ver, the outputcathode, grounded through R 2 , will stop at zeropotential as the plate becomes negative. Conductionthrough the second diode is impossibleunder these conditions. The output cathoderemains at zero potential until the voltage atthe input cathode swings back to zero.The voltage developed across output resistorR 2follows the input voltage variations aslong as the input voltage does not swing to apeak value greater than the static voltage atthe diode plates, determined by R 4 • Effectiveclipping may thus be obtained at any desiredlevel.The square-topped audio waves generatedby this clipper are high in harmonic content,but these higher order harmonics may be greatlyreduced by a low-level speech filter.Grid Limiters A triode grid limiter is shownin figure 3. On positive peaksof the input signal, the triode grid attempts toswing positive, and the grid-cathode resistancedrops to a value on the order of 1000ohms or so. The voltage drop across R (usuallyof the order of 1 megohm) is large comparedto the grid-cathode drop, and the resultinglimiting action removes the top part of thepositive input wave.10-2 Clamping CircuitsA circuit which holds either amplitude extremeof a waveform to a given reference level0 POSITIVE CLAMPING CIRCUIT@ NEGATIVE CLAMPING CIRCUITFigure 4SIMPLE POSITIVE AND NEGATIVE CLAMPING CIRCUITS
188 Special Vacuum Tube Circuits <strong>THE</strong> R AD I 0B+Ct CHARGE PATHC2 DISCHARGE PATHNEGATIVEPLOYED INFigure 5CLAMPING CIRCUIT EMELECTROMAGNETIC SWEEPSYSTEMFigure 7<strong>THE</strong> CHARGE AND DISCHARGE PATHSIN FREE-RUNNING MUL TIVIBRATOR OFFIGURE 6B+is repeated and therefore is "jittery." If aclamping circuit is placed between the sweepamplifier and the deflection element, the startof the sweep can be regulated by adjusting thed-e voltage applied to the clamping tube (fig·ure 5).Figure 6BASIC MUL TIVIBRA TOR CIRCUITof potential is called a clamping circuit or ad-e restorer. Clamping circuits are used afterRC cpupling circuits where the waveformswing is required to be either above or belowthe reference voltage, instead of alternatingon both sides of it (figure 4). Clamping cir·cuits are usually encountered in oscilloscopesweep circuits. If the sweep voltage does notalways start from the same reference point,the trace on the screen does not begin at thesame point on the screen each time the sweep10-3 MultivibratorsThe multivibrator, or relaxation oscillator,is used for the generation of nonsinusoidalwaveforms. The output is rich in harmonics,but the inherent frequency stability is poor.The multivibrator may be stabilized by theintroduction of synchronizing voltages of harmonicor subharmonic frequency.In its simplest form, the multivibrator is asimple two-stage resistance-capacitance coupledamplifier with the output of the secondstage coupled through a capacitor to the gridof the first tube, as shown in figure 6. Sincethe output of the second stage is of the properpolarity to reinforce the input signal appliedto the first tube, oscillations can readily takeplace, started by thermal agitation noise andB+B+ B+®DIRECT-COUPLED CATHODEMULTIVIBRATOR@ ©ELECTRON-COUPLEDMULTIVIBRATORMULTIVIBRATOR WITH SINE-WAVESYNCHRONIZING SIGNAL APPLIEDTO ONE TUBEFigure 8VARIOUS FORMS OF MUL TIVIBRATOR CIRCUITS
<strong>HANDBOOK</strong>Multivibrators189B+B+C1TRIGGER"~INPUTR1PULSEOUTPUTBASIC ECCLES-JORDAN TRIGGERCIRCUIT®ONE -SHOT MULTIVIBRATORFigure 9ECCLES-JORDAN MUL Tl VIBRATOR CIRCUITSmiscellaneous tube noise. Oscillation il' maintainedby the process of building up and dis·charging the store of energy in the grid cou·piing capacitors of the two tubes. The charg·ing and discharging paths are shown in figure7. Various forms of multivibrators are shownin figure 8.The output of a multivibrator may be usedas a source of square waves, as an electronicswitch, or as a means of obtaining frequencydivision. Submultiple frequencies as low asone-tenth of the injected synchronizing fre·quency may easily be obtained.The Eccles-JordanCircuitThe Eccles· Jordan triggercircuit is shown in figure9A. This is not a true multivibrator,but rather a circuit that possessestwo conditions of stable equilibrium. One con·clition is when V 1 is conducting and V 2is cut·off; the other when V 2 is conducting and V 1 iscutoff. The circuit remains in one or the otherof these two stable conditions with no changein operating potentials until some externalaction occurs which causes the nonconductingtube to conduct. The tubes then reverse theirfunctions and remain in the new condition aslong as no plate current flows in the cutofftube. This type of circuit is known as a flipflopcircuit.E'OUT ---1 h~ING{\> 1\1 {\\fi IV lJ~CUTOFFTIMEFigure 10SINGLE-SWING BLOCKING OSCILLATORFigure 9B illustrates a modified Eccles·Jordan circuit which accomplishes a completecycle when triggered with a positive pulse.Such a circuit is called a one-shot multivibra·tor. For initial action, V, is cutoff and V 2isconducting. A large positive pulse applied tothe grid of V 1 causes this tube to conduct, andthe voltage at its plate decreases by virtue ofthe IR drop through R,. Capacitor C 2 is chargedrapidly by this abrupt change in V 1 plate volt·age, and V 2 becomes cutoff while V 1 conducts.This condition exists until C 2 discharges, al·lowing vl to conduct, raising the cathode biasof V 1until it is once again cutoff.A direct, cathode-coupled multivibrator isshown in figure SA. RK is a common cathoderesistor for the two tubes, and coupling takesplace across this resistor. It is impossible fora tube in this circuit to completely cutoff theother tube, and a circuit of this type is calleda free-running multivibrator in which the con·clition of one tube temporarily cuts off theother.+ RF RF RFPULSE PULSE PULSE~ ~ ~Figure 1100 0 I'on'00, I U l.\1----lCUTOFFTIME1----..jCUTOFFTIME'00 11 V VHARTLEY OSCILLATOR USED AS BLOCKINGOSCILLATOR BY PROPER CHOICE OF R 1·C 1
1 90 Special Vacuum Tube Circuits <strong>THE</strong> R AD I 0etN--j ciTVz eouTe1N---1 c~Vz eour+VIR IVIR1@ ® ©POSITIVE COUNTING CIRCUIT NEGATIVE COUNTING CIRCUIT POSITIVE COUNTING Cl RCUIT WITHMETER INDICATIONFigure 12POSITIVE AND NEGATIVE COUNTING CIRCUITS10-4 The Blocking OscillatorA blocking oscillator is any oscillator whichcuts itself off after one or more cycles causedby the accumulation of a negative charge onthe grid capacitor. This negative charge maygradually be drained off through the grid re·sistor of the tube, allowing the circuit to oscillateonce again. The process is repeatedand the tube becomes an intermittent oscilla·tor. The rate of such an occurance is deter·mined by the R-C time constant of the grid circuit.A single-swing blocking oscillator isshown in figure 10, wherein the tube is cutoffbefore the completion of one cycle. The tubeproduces single pulses of energy, the timebetween the pulses being regulated by thedischarge time of the grid R·C network. Theself-pulsing blocking oscillator is shown infigure 11, and is used to produce pulsesof r·f energy, the number of pulses being de·termined by the timing network in the grid cir·cuit of the oscillator. The rate at which thesepulses occur is known as the pulse-repetitionfrequency, or p.r.f.ing units to be counted, and produces a volt·age that is proportional to the frequency ofthe pulses. A counting circuit may be used inconjunction with a blocking oscillator to producea trigger pulse which is a submultiple ofof the frequency of the applied pulse. Eitherpositive or negative pulses may be counted.A positive counting circuit is shown in figure12A, and a negative counting circuit is shownin figure 12B. The positive counter allows acertain amount of current to flow through R 1each time a pulse is applied to C,.The positive pulse charges C, and makesthe plate of V 2 positive with respect to itscathode. V 2 conducts until the exciting pulsep~ssc;s .. C, is then discharged by V,, and thecucult 1s ready to accept another pulse. Theaverage current flowing through R 1increasesas the pulse-repetition frequency increases,and decreases as the p.r.f. decreases.By reversing the diode c on n e c t ion s, asshown in figure 12B, the circuit is made torespond to negative pulses. In this circuit, anincrease in the p.r.f. causes a decrease in theaverage current flowing through R 1 , which isopposite to the effect in the positive counter.10-5 Counting CircuitsA counting circuit, or frequency divider isone which receives uniform pulses, represent·Figure 13STEP-BY-STEP COUNTING CIRCUITFigure 14The step-&y•step counter usee/ to trig9er a&locking oscillator. The &locking oscillatorserves as a frequency clivicler.
<strong>HANDBOOK</strong>R-C Oscillators 1 91+lOOK500'\.o.---------~~ ~.01lP = R 4 = 4 WATT, 110 v. LAMP BULBR1 XC1 =R2 X C2B+Figure 15<strong>THE</strong> WIEN-BRIDGE AUDIO OSCILLATORFigure 16<strong>THE</strong> PHASE-SHIFT OSCILLATORA step-counter is similar to the circuitsdiscussed, except that a capacitor which islarge compared to C, replaces the diode loadresistor. The charge of this condenser is in·creased during the time of each pulse, pro·clueing a step voltage across the output (figure13). A blocking oscillator may be connectedto a step-counter, as shown in figure 14. Theoscillator is triggered into operation when thevoltage across C 2 reaches a point sufficientlypositive to raise the grid of V 3above cutoff.Circuit parameters may be chosen so that acount division up to 1/20 may be obtainedwith reliability.10-6 Resistance-CapacityOscillatorsIn an R·C oscillator, the frequency is de·termined by a resistance capacity network thatprovides regenerative coupling between theoutput and input of a feedback amplifier. Nouse is made of a tank circuit consisting of in·ductance and capacitance to control the fre·quency of oscillation.The Wien-Bridge oscillator employs a Wiennetwork in the R·C feedback circuit and isshown in figure 15. Tube V 1 is the oscillatortube, and tube V 2is an amplifier and phase·inverter tube. Since the feedback voltagethrough C 4 produced by V 2 is in phase with theinput circuit of V, at all frequencies, oscilla·tion is maintained by voltages of any frequen·cy that exist in the circuit. The bridge circuitis used, then, to eliminate feedback voltagesof all frequencies except the single frequencydesired at the output of the oscillator. Thebridge allows a voltage of only one frequencyto be effective in the circuit because of thedegeneration and phase shift provided by thiscircuit. The frequency at which oscillationoccurs is:If"' , when R, X C 1 "'R 2 x C 2277 R 1 C,A lamp L is used as the cathode resistorof V, as a ttermal stabilizer of the oscillatoramplitude. The variation of the resistancewith respect to current of the lamp bulb holdsthe oscillator output voltage at a nearly con·stant amplitude.The phase-shift oscillator shown in figure16 is a single tube oscillator using a threesection phase shift network. Each section ofthe network produces a phase shift in proportionto the frequency of the signal that passesthrough it. For oscillations to be produced,the signal from the plate of the tube must beshifted 180°. Three successive phase shiftsof 60° accomplish this, and the frequency ofoscillation is determined by this phase shift.A high·mu triode or a pentode must be used inthis circuit. In order to increase the frequencyof oscillation, either the resistance or thecapacitance must be decreased.Figure 17<strong>THE</strong> BRIDGE-TYPE PHASE-SHIFTOSCILLATOR
192 Special Vacuum Tube Circuits <strong>THE</strong> R AD I 0r---.-----------------~---8+47 K 1~0 K"VOUT"NOTCH" FREQUENCY 1F=--'- ~2 Tr RC ..JWHERE gC=~C1~IRzCz"NOTCW NETWORK.."J:+t--FREQ. OF OSCILLATIONIII-r-POSITIVEFEEDBACK(LOOP 1)~FREQ. OF OSCILLATION:a~.-------*-------~ .."' ~J:~IFigure 18<strong>THE</strong> NBS BRIDGE-TOSCILLATOR CIRCUIT AS USEDIN <strong>THE</strong> HEATH AG-9 AUDIOGENERATORFigure 19BRIDGE-T FEEDBACKLOOP CIRCUITSOscillation will occur at the nullfrequency of the bridge, at whichfrequency the bridge allowsminimum degeneration in loop 2.A bridge-type phase shift oscillator isshown in figure 17. The bridge is so propor·tioned that at only one frequency is the phaseshift through the bridge 180°. Voltages of otherfrequencies are fed back to the grid of the tubeout of phase with the existing grid signal, andare cancelled by being amplified out of phase.The NBS Bridge-T oscillator developed bythe National Bureau of Standards consists ofa two stage amplifier having two feedbackloops, as shown in figure 18. Loop 1 consistsof a regenerative cathode-to-cathode loop, con·sisting of Lp 1 and C3. The bulb regulates thepositive feedback, and tends to stabilize theoutput of the oscillator, much as in the mannerof the Wien circuit. Loop 2 consists of agrid-cathode degenerative circuit, containingthe bridge-T. Oscillation will occur at thenull frequency of the bridge, at which frequen·cy the bridge allows minimum degenerationin loop 2 (figure 19).10-7 FeedbackFeedback amplifiers have been discussedin Chapter 6, section 15 of this Handbook. Amore general use of feedback is in automaticcontrol and regulating systems. Mechanicalfeedback has been used for many years in suchforms as engine speed governors and steeringservo engines on ships.A simple fee--dback system for temperaturecontrol is shown in figure 20. This is a causeand effect system. The furnace (F) raises theroom temperature (T) to a predetermined valueat which point the sensing thermostat (TH)reduces the fuel flow to the furnace. When theroom temperature drops below the predeterminedvalue the fuel flow is increased by thethermostat control. An interdependent controlsystem is created by this arrangement: theroom temperature depends upon the thermostataction, and the thermostat action depends uponthe room temperature. This sequence of eventsmay be termed a closed loop feedback system.--Figure 20SIMPLE CLOSED LOOPFEEDBACK SYSTEMRoom temperature (T) controlsfuel supply to furnace (F) by feedbockloop through Thermostat(TH) control.IFEEDBACK(ERROR SlbNAL)
<strong>HANDBOOK</strong>Feedback 193INPUT src;NAL/OUTPUT SIGNALg~-*-+~-+~~~~~~r-~-+--~~FEEDBACKS\\~~c;.~A/eo"~ PHASE~ SHIFT
CHAPTER ELEVENElectronicComputersMechanical computing machines were firstproduced in the seventeenth cenmry in Europealthough the simple Chinese abacus (adigital computer) had been in use for centuries.Until the last decade only simple mechanicalcomputers (such as adding and bookkeepingmachines) were in general use.The transformation and transmission of thevolume of information required by moderntechnology requires that machines assume manyof the information processing systems formerlydone by the human mind. Computing machinescan perform routine operations morequickly and more accurately than a human being,processing mathematical and logisticaldata on a production line basis. The computer,however, cannot create, but can only followinstructions. If the instructions are in error,<strong>THE</strong> IBMCOMPUTERAND"MEMORY"The '~704" Computetis used with32,000 "word"memory storageunit for researchprograms. Heartof this auxiliaryunit are small,doughnut-shapediron fettites whichstore informationby means of magnetism.The unitIs the first of Itskind to be Installedwith I B M·'s704 computer.Components of thesystem seen in theforeground are(left) card punchand (right) cardreader. In thecenter of the pictureis the 704'sprocessing unit.
Digital Computers 195NORTH SHORE9BUZZ.ERBATTERYNOTE: ALL BU7TONS HAVE ONE NORMALLY OPEN CONTACT ANDONE NORMALLY CLOSED CONTACT.Figure 2A SEQUENCE COMPUTER.Three correct buttons will sound the buzzer.Figure 1SIMPLE PUZZLES IN LOGIC MAY BESOLVED BY ELECTRIC COMPUTER.<strong>THE</strong> "FARMER AND RIVER"COMPUTER IS SHOWN HERE.the computer will produce a wrong answer.Computers may be divided into two classes:the digital and the analog. The digital computercounts, and its accuracy is limited only bythe number of significant figures provided forin the instrument. The analog computermeasures, and its accuracy is limited by thepercentage errors of the devices used, multipliedby the range of the variables they represent.11-1 Digital ComputersThe digital computer operates in discretesteps. In general, the mathematical operationsare performed by combinations of additions.Thus multiplication is performed by repeatedadditions, and integration is performed bysummation. The digital computer may bethought of as an "on-off" device operatingfrom signals that either exist or do not exist.The common adding machine is a simple computerof this type. The "on-off" or "yes-no"type of situation is well suited to switches, electricalrelays, or to electronic tubes.A simple electrical digital computer may beused to solve the old "farmer and river" problem.The farmer must transport a hen, a bushelof corn, and a fox across a river in a smallboat capable of carrying the farmer pius oneother article. If the farmer takes the fox inthe boat with him, the hen will eat the corn.On the other hand, if he takes the corn, thefox will eat the hen. The circuit for a simplecomputer to solve this problem is shown infigure 1. When the switches are moved from"south shore" to "north shore" in the propersequence the warning buzzer will not sound.An error of choice will sound the buzzer.A second simple "digital computer" is shownis figure 2. The problem is to find the threeproper push buttons that will sound the buzzer.The nine buttons are mounted on a board sothat the wiring cannot be seen.Each switch of these simple computers executesan "on-off' action. When applied to alogical problem "yes-no" may be substitutedfor this term. The computer thus can act outa logical concept concerned with a simplechoice. An electronic switch (tube) may besubstituted for the mechanical switch to increasethe speed of the computer. The earlycomputers, such as the ENIAC (Electronic NumericalIntegrator and Calculator) employedover 18,000 tubes for memory and registeringcircuits capable of "remembering" a 10-digitnumber.11-2 Binary NotationTo simply and reduce the cost of the digitalcomputer it was necessary to modify the systemof operation so that fewer tubes were used perbit of information. The ENIAC-type computerrequires 50 tubes to register a 5-digit number.G) G) G) G) G)® ® ® ® ;~~ ,,,® ~ ® ® ®'•'0 0 0 0 0® ® ® ® ®@ @ @ @ @~--0 0 0 0",I,'@ @ @ @ @® ® ® ~ ,,, ®*@ @ "'Figure 3BINARY NOTATION MAY BE USEDFOR DIGITAL DISPLAY. BINARYBOARD ABOVE INDICATES "73092."@@
196 Electronic Computers<strong>THE</strong> <strong>RADIO</strong>CD ® -~ - ::,,,-~ IDIGIT1 12. 2.3 2.+14 4s 4+10 4+2.7 4+2.+10 0•0+110 0+2.11 8+2+112. 0+413 1+4+1TUBE(S)14 8+4+2.IS 8+ ... +2+1Figure 4BINARY DECIMAL NOTATION. ONLYFOUR TUBES ARE REQUIRED TOREPRESENT DIGITS FROM 1 TO 15.<strong>THE</strong> DIGIT "12" IS INDICATEDABOVE.The tubes (or their indicator lamps) can bearranged in five columns of 10 tubes each.From right to left the columns represent units,tens, hundreds, thousands, etc. The bottom tubein each column represents "zero," the secondtube represents "one," the third tube "two,"and so on. Only one tube in each column isexcited at any given instant. If the number73092 is to be displayed, tube number sevenin the fifth column is excited, tube numberthree in the fourth column, tube number zeroin the third column, etc. as shown in figure 3.A simpler system employs the binary decimalnotation, wherein any number from oneto fifteen can be represented by four tubes.Each of the four tubes has a numerical valuethat is associated with its position in the tubegroup. More than one tube of the group maybe excited at once, as illustrated in figure 4.The values assigned to the tubes in this particulargroup are 1, 2, 4, and 8. Additionaltubes may be added to the group, doubling thenotation of the tube thus: 1, 2, 4, 8, 16, 32,64, 128, 356, etc. Any numerical value lowerthan the highest group number can be displayedby the correct tube combination.A third system employs the binary notationwhich makes use of a bit (binary digit) representinga single morsel of information. Thebinary system has been known for over fortycenturies, and was considered a mystical revelationfor ages since it employed only two sym-DECIMAL NOTATIONBINARY NOTATION0 01 12. 1, 03 1,14 1, o, 0s1, o,,0 1, 1, 07 1,1,10 1, o,o,oI1, 0 1 0 1 I10 1, 0, 1, 0Figure 5BINARY NOTATION SYSTEMREQUIRES ONLY TWO NUMBERS,"0" AND "1."bols for all numbers. Computer service usuallyemploys "zero" and "one" as· these symbols.Decimal notation and binary notation for commonnumbers are shown in figure 5. Thebinary notation represents 4-digit numbers(thousands) with ten bits, and 7 -digit numbers(millions) with 20 bits. Only one electrontube is required to display an informationbit. The savings in components and primarypower drain of a binary-type computer overthe older ENIAC-type computer is obvious.Figure 6 illustrates a computer board showingthe binary indications from one to ten.DigitalComputer UsesThe digital computer is employedin a "yes-no" situation.It may be used forroutine calculations that would ordinarily requireenormous man-hours of time, such aschecking stress estimates in aircraft design, ormilitary logistics, and problems involving themanipulation of large masses of figures.DECIMAL NOTATIONCOMPUTER NOTATION0 •1 • 02. •0 3 • 0 04 •0 5• • 0 •00 0 7• 0 0 08 0• •0 • 010 0• 0 •e=oFF O=oNFigure 6BINARY NOTATION AS REPRESENTEDON COMPUTER BOARD FOR NUMBERSFROM 1 TO 10.
<strong>HANDBOOK</strong>Analog Computers 197~~y LLJ ~eouT- -::~·~'_ ( e, R1 ez Rz) ( R3 -) ..,.eouT- R1+Rz + R1+R2. R1 Rz[R1+Rz]+R3Figure 7SUMMATION OF TWO VOLTAGESBY ELECTRICAL MEANS.+t!IOV.eouT~----------~~-8+EouT11-3 Analog ComputersThe analog computer represents the use ofone physical system as a model for a secondsystem that is usually more difficult to constructor to measure, and that obeys the equationsof the same form. The term analog impliessimilarity of relations or properties betweenthe two systems. The common slide-ruleis a mechanical analog computer. The speedometerin an automobile is a differential analogcomputer, displaying information proportionalto the rate of change of speed of the vehicle.The electronic analog computer employs circuitscontaining resistance, capacitance, and inductancearranged to behave in accordance withanalogous equations. Variables are representedby d-e voltages which may vary with time.Figure 8SUMMATION OF TWO VOLTAGESBY ELECTRONIC MEANS.Thus complicated problems can be solved byd-e amplifiers and potentiometer controls inelectronic circuits performing mathematicalfunctions.Addition andSubtradionIf a linear network is energizedby two voltage sourcesthe voltages may be summedas shown in figure 7. Subtraction of quantitiesmay be accomplished by using negative andpositive voltages. A-c voltages may be employedfor certain additive circuits, and more<strong>THE</strong>HEATHKITELECTRONICANALOGDIGITALCOMPUTER11This electronicslide rule 11 simulatesequations orphysical problemselectronically, substitutingone physicalsystem as amodel lor a secondsystem thatis usually moredifficult or costlyto construct ormeasure, and thatobeys equations olthe same form.
198 Electronic ComputerseouT =EX e1R1J:TL_~~eourFigure 9ELECT~ONIC MULTIPLICATIONMAY BE ACCOMPLISHED BY·CALIBRATED POTENTIOMETERS,WHEN OUTPUT VOLTAGE ISPROPORTIONAL TO <strong>THE</strong> INPUTVOLTAGE MULTIPLIED BY ACONSTANT CR2/R1l.complex circuits employ vacuum tubes, as infigure 8. Synchronous transformers may beused to add expressions of angular rotation,and circuits have been developed for addingtime delays, or pulse counting.Multiplicationand DivisionElectronic multiplication anddivision may be accomplishedwith the use of potentiometerswhere the output voltage is proportional tothe input voltage multiplied by a constantwhich may be altered by changing the physicalarrangement of the potentiometer (figure 9l.Variable autotransformers may also be used toperform multiplication.A simple bridge may be used to obtain anoutput that is the product of two inputs dividedby a third input, as shown in figure 10.Differentiation The time derivative of avoltage can be expressed asa charge on a capacitor by:·-c de1- dt (1)and is shown in figure llA. The charging currentis converted into a voltage by the useof a resistor, R. If the input to the RC circuitis charging at a uniform rate so that the currentthrough C and R is constant, the outputvoltage eo is:OUTPUT"\.. R 1 ~ 2 R3. Figure 10ELECTRONIC MU L Tl PLICATION BYBRIDGE CIRCUIT PROVIDESOUTPUT THAT IS PRODUCT OFTWO INPUTS DIVIDED BY ATHIRD INPUT.c<strong>THE</strong> <strong>RADIO</strong>RCffy---= +~eour e OUT®Figure 11ELECTRONIC DIFFERENTIATIONThe time derivative of a voltage can beexpressed as a charge on a capacitor (A).Operational amplifier (B)back principleemploys feedlor short differentiationtime.deeo=RCdt(2)For highest accuracy, a small RC productshould be used, permitting the maximum possibledifferentiation time. The output of thedifferentiator may be amplified to any suitablelevel.A more accurate differentiating devicemakes use of an operational amplifier. Thisunit is a high gain, negative feedback d-eamplifier (discussed in section 11-4) with theresistance portion of the RC product appearingin the feedback loop of the amplifier (figurellB). A shorter differentiation time may beemployed if the junction point between R andC could be held at a constant potential. Thefeedback amplifier shown inverts the outputsignal and applies it to the RC nerwork, holdingthe junction potential constant.Integration Integration is a process of accumulation,or summation, andrequires a device capable of storing physicalquantities. A capacitor will store an electricalcharge and will give the time integral of acurrent in respect to a voltage:1eo= ci dt ( 3)In most computers, the input signal is inthe form of a voltage, and the input chargingcurrent of the capacitor must be taken througha series resistance as in figure 12. If the integratingtime is short the charging current isapproximately proportional to the input voltage.The charging current may be made atrue measure of the input voltage by the useFigure 12SIMPLE ;:---J1 f3lIN~~~~J~ON ~ fc ~eourMaking use ofcharging currentof capacitor.
<strong>HANDBOOK</strong>Operational Amplifier199~~heo"'., +Figure 13"MILLER FEEDBACK" INTEGRATORSUITABLE FOR COMPUTER USE.11-4 The OperationalAmplifierMathematical operations are performed byusing a high gain d-e amplifier, termed anoperational amplifier. The symbol of this unitis a triangle, with the apex pointing in the directionof operation (figure 15). The gain ofsuch an amplifier is -A, so:eoeo = - Ae., or e. = - A ( 4)Figure 14R-L NETWORK USED FORINTEGRATION PURPOSES.Figure 15OPERATIONAL AMPLIFIER C-AlMathematical operations may be performedby any operational amplifier, usually astable, high-gain d-e amplifier, such asshown in Figure 16.of an operational amplifier wherein the capacitanceportion of the RC product appearsin the feedback loop of the amplifier, holdingthe junction point between R and C at a constantpotential. A simple integrator is shownin figure 13 employing the Miller feedbackprinciple. Integration is also possible with anRL network (figure 14).If -A approaches infinity, e. will be approximatelyzero. In practice this condition isrealized by using amplifiers having open loopgains of 30,000 to 60,000. If eo is set at 100volts, e. will be of the order of a few millivolts.Thus, considering e. equal to zero:-eo - Rr-- oreo=--e•Rr' R,which may be written:Rreo= - Ke, where K =-' R,(6)(5)This amounts to multiplication by a constantcoefficient, since R, and Rr may be fixed invalue. The circuit of a typical operational amplifieris shown in figure 16.AmplifierOperationTwo voltages may be added bythe amplifier, as shown in figure17. Keeping in mind that e. is1M (RF)lOOKeo1~0 K3K(BALANCE)2K2!>0K('AIN)00 K1 KNE-51BIAS -450¥. -2..50 v.1----------------- -- GAl N = -A -Figure 16HIGH GAIN OPERATIONAL AMPLIFIER, SUCH AS USED IN HEATH COMPUTER.
200 Electronic Computers<strong>THE</strong> <strong>RADIO</strong>RFeou_2. ____.!_G ____.fFigure 17TWO VOLTAGES MAY BEADDED BY SUMMATIONAMPLIFIER.essentially at zero (ground) potential:eo==-Rr Rr-e.+--e,R, R,(7)or, eo==- (8)whereAs long as eo does not exceed the inputrange of the amplifier, any number of inputsmay be used:RrRrK,=- andK,=-R, R, (9leo==orRr Rr Rr-et+-e,+--- +-enR, R, Rneo=-Rr+~Rn(10)( 11)Integration is performed by replacing thefeedback resistor Rr with a capacitor Cr, asshown in figure 18. For this circuit (with e.approximately zero) :0- d.1-dt (12)d. deo e• deobut g=C, eo so-= Cr--and -= Cr -' dt dt R, dtThus:anddeo =eo==--R.CrR.c,e• dt(13)(14)e• dt ( 15)Figure 18INTEGRATIONPerformed bySummation Amplifierby replacing teedback resistor with acapacitor.11-5F=f'(tlFigure 19"MASS-SPRINGDAMPER" PROBLEMMAY BE SOLVED BYELECTRICAL ANALOGYWITH SIMPLECOMPUTER.Solving AnalogProblemsBy combining the above operations in variousways, problems of many kinds may besolved For example, consider the mass-springdamperassembly shown in figure 19. The massM is connected to the spring which has anelastic constant K. The viscous damping constantis C. The vertical displacement is y. Thesum of the forces acting on mass M is:d'y dyf (t) =M- + C -+Kydt' dty(16)where f ( t) is the applied force, or forcing/unction.The first step is to set up the analog computercircuit so as to obtain an output voltageproportional to y for a given input voltageproportional to f ( t). Equation ( 16) may berewritten in the form:d'y dyM dr = -ccit-Ky+f (t)( 17)If, in the analog circuit, there is a voltaged'y - dyequal to M dt' it can be converted to dtby passing it through an integrator circuit havingan RC time constant equal to M. This resultingvoltage can be passed through a secondintegrator stage with unit time constant whichwill have an output volage equal to y. Thevoltages representing y, -1: , and f ( t) canthen be summed to give- C ~: - Ky + f (t)which is the right hand side of equation ( 17),and therefore equal to M ~';, . Connecting the
<strong>HANDBOOK</strong>Analog Problems 201OPERATERELAYOPERATERELAYSET VOLTAGE TO Y;Yo ATTIMEt =o (INITIALDISPLACEMENT)1.01.0'"'"'"y--c~; -Kv+f'!tl'lc'" c;; -P(tJTO DISPLAYOS C.\ LLOSCO PE'"f'(t J >------H\1'-----lFigure 20ANALOG SOLUTION FOR "MASS-SPRINGDAMPER" PROBLEM OF FIGURE 19.output of the summing amplifier (A3) to theinput of the first as shown in figure 20 satisfiesthe equation.To obtain a solution to the problem, theinitial displacement and velocity must be specified.This is done by charging the integratingcapacitors to the proper voltages. Three operationalamplifiers and a summing amplifierare required.A second problem that may be solved bythe analog computer is the example of a freelyfalling body. Disregarding air resistance, thebody will fall (due to the action of gravity)with a constant acceleration. The equationdescribing this action is:d'yF = mg = m dt'08 )Integration of equation (18) will give the. dy d. . d .veloCity, or- , an mtegrauon a secon umedtwill give displacement, or y. The block diagramof a suitable computer for this problemis shown in figure 21.If a voltage proportional to g and hence tod'y is introduced into the first amplifier, thedt'dyoutput of that unit will be - dt, or theFigure 21ANALOG COMPUTER FOR"FREELY FALLING BODY"PROBLEM.velocity. That, in turn, will become y, or distance,at the output of the second amplifier.Before the problem can be solved on thecomputer it is necessary to determine the timeof solution desirable and the output amplitudeof the solution. The time of solution is determinedby the RC constant of the integratingamplifiers. If RC is set at unity, computertime is equal to real time. The computer timedesirable is determined by the method of readout.When using an oscilloscope for read-out,a short solution time is desirable. For a recorder,longer solution time is better.Suppose, for example, in the problem of thefalling body, the distance of fall in 2.5 secondsis desired. Using an RC constant of 1 wouldgive a solution time of 2.5 seconds. This wouldbe acceptable for a recorder but is slow for anoscilloscope. A convenient time of solution forthe 'scope would be 25 milliseconds. This is11100 of the real time, so an RC constant of.01 is needed. This can be obtained with Cequal to 0.1 1'-fd, and R equal to 100,000ohms.It is now necessary to choose an input voltagewhich will not overdrive the amplifiers.The value of g is known to be approximately32 ft/ sec./ sec. A check indicates that if weset g equal to 3 2 volts, the voltage representingthe answer will exceed 100 volts. Sincethe linear response of the amplifier is only100 volts, this is undesirable. An input of 16volts, however, should permit satisfactory operationof the amplifiers. Output voltages nearzero should also be avoided. In general, outputvoltage should be about 50 volts or so, withamplifier gains of 20 to 60 being preferable.Thus, for this particular problem the timescalefactor and amplitude-scale factor have
202 Electronic Computers<strong>THE</strong> <strong>RADIO</strong>0.1 0.1eoFigure 22ANALOG SOLUTION FOR"FALLING BODY PROBLEM"OF FIGURE 21.cs=·····.---18 """VOLTS-246 ""'-32-40-46•-5-·-72TIME IN SECONDS-1 2 (2 5} 3""'1\-~----IIIII\I\4 \I,DISTANCE32 INFEET6~ jFigure 23READ-OUT SOLUTION OF "FREELYFALLING BODY" PROBLEM.~r6• (100)been chosen. The problem now looks likefigure 22.To solve the problem, relays A and A' areopened. The solution should now appear onthe oscilloscope as shown in figure 23. Thesolution of the problem leaves the integratingcapacitors charged. It is necessary to removethis charge before the problem can be rerun.This is done by closing relays A and A'.1121 281 44Figure 24LIMITING CIRCUIT TO SIMULATENON-LINEAR FUNCTIONS SUCH ASENCOUNTERED IN HYSTERESIS,BACKLASH, AND FRICTIONPROBLEMS.11-6 Non-linear FunctionsProblems are frequently encountered inwhich non-linear functions must be simulated.Non-linear potentiometers may be used to supplyan unusual voltage source, or diodes maybe used as limiters in those problems in whicha function is defined differently for differentregions of the independent variable. Such afunction might be defined as follows:eo=-K,,e, -K, (19)eo = e, , - K e, K, ( 20)eo = K,, e' K, ( 21)where K, and K, are constants.Various limiting circuits can be used, one ofwhich is shown in figure 24. This is a serieslimiter circuit which is simple and does notrequire special components. Commonly encounteredproblems requiring these or similarlimiting techniques include hysteresis, backlash,and certain types of friction.NOTE: REPLACE C WITH A 1 MEG. RESISTORFOR FUNCTION SETUPRAMP- FUNCTIONGENERATOR820 K -Figure 25SIMPLIFIED DIAGRAM OF FUNCTIONAL GENERATOR TO APPROXIMATE NON-LINEARFUNCTIONS.
<strong>HANDBOOK</strong>+YNon-Linear Functions 203~--------,1II"AND" GATESIIIIPOSITIVEFigure 26TYPICAL NON-LINEAR FUNCTIONWHICH MAY BE SET UP WITHFUNCTION GENERATOR.The FunctionGeneratorA function generator may beused to approximate almostany non-linear function. Thisis done by use of straight line segments whichare combined to approximate curves such asare found in trigonometric functions as wellas in stepped functions. In a typical generatorten line segments are used, five in the plus-xdirection, and five in the minus-x direction.Five 6AL5 double diodes are used. Each linesegment is generated by a modified bridgecircuit (figure 2 5). A ramp function or voltageis fed into one arm of the bridge whilethe opposite arm is connected to a biased diode.The other two arms of the bridge combine toform the output. The voltage appearing acrossone of these arms is fed through a sign-changingamplifier and then summed with the voltageappearing at the opposite arm. If the armof potentiometer P (the slope control) is setin the center, the bridge will be balanced andIBM's new "608," thefirst completely transistorizedcalculatorfor commerdal applications,o p e r a t e swithout the use of asingle vacuum tube.Transistors--tiny germaniumdevices thatperform many of thefunctions of conventionalvacuum tubes-make possible 50%reduction in computer-unitsize and a90% reduction inpower requirementsover a comparableIBM tube-model machine.They aremounted, along withrelated circuitry, onbanks of printed wirIng panels in the608.The machine's internalstorage, or 11 memory,"is made up ofmagnetic cores-minute,doughnut-shapedobjects that can"remembet 11 informationindefinitely, andrecall it for use incalculations in a fewmillionths of acl!lrl\nrl.I_ ___]Figure 27ELECTRONIC PACKAGE INDIGITAL COMPUTER.Stylized diagram of tube package. Lines carryingnegative pulses are marked by a smallcircle at each end. Gates are indicated bya semi-circle with "pins 11 lor each input.the output of the summing amplifier will bezero. If, on the other hand, potentiometer Pis adjusted one way or the other from center,the bridge will be unbalanced and the summingamplifier output will vary linearly withrespect to the input in either a positive or negativey direction, depending upon which sideof center potentiometer P is set.The break voltage, or value of x at which astraight line segment will begin is set bybiasing the diode to the particular voltage levelor value of x desired. The ramp function generatorhas either a positive or negative inputwhich because of the 180 degree phase shiftin the amplifier, gives a minus- or plus-x outputrespectively. A typical function such asshown in figure 26 may be set up with thefunction generator. The initial condition volt-
204 Electronic Computersage is set to the value of X1. The break-voltagecontrol is increased until the output of thesumming amplifier increases abruptly, indicatingthe diode is conducting. The inputvoltage from the initial condition power supplyis set to the value X, The slope control (P)is now set to value Y,. A second function generatormay be used to set points X, and X,,using the break-voltage control and the initialcondition voltage adjustments. Points X, andX, are finally set with a third generator. Thex-output of the function generator system maybe read on an oscilloscope, using the x-outputof the ramp-function generator amplifier asthe horizontal sweep for the oscilloscope.11-7 Digital CircuitryDigital circuits dealing with "and," ··or,and ""not" situations may be excited by electricalpulses representing these logical operations.Sorting and amplifying the pulses can be accomplishedby the use of electronic packages,such as shown in figure 27. Logical operationsmay be accomplished by diode-resistor gatesoperating into an amplifier stage. Negative andpositive output pulses from the amplifier areobtained through diode output gates. The drivingpulses may be obtained from a standardoscillator, operating at or near 1 me.A circuit of a single digital package is shownin figure 28. Other configurations, such as a"flip-flop" may be used. Many such packages<strong>THE</strong> <strong>RADIO</strong>can be connected in series to form operationalcircuits. The input "and" and '"or" gates arebiased to conduction by external voltages. The""and" diode gate transmits a pulse only whenall the input terminals are pulsed positively,and the "or" diode gate transmits a positivepulse applied to any one of its input terminals.The input pulses pass through the gates anddrive the amplifier stage, which delivers anamplified pulse to the positive and negativeoutput gates, and to accompanying memorycircuits.Memory Circuits A memory circuit consistsof some sort of delay linewhich is capable of holding an informationpattern for a period of time. A simple electricaldelay line capable of holding a pulse fora fraction of a microsecond is shown in figure29. The amount of delay is proportional tothe frequency of the input signal as shown inthe graph. A "long" transmission line mayalso be used as a delay line, with the signalbeing removed from the "far" end of the lineafter being delayed an interval equal to thetime of transmission along the line. Lines ofthis type are constructed much in the mannerof a standard coaxial cable, except that theinner conductor is a long, thin coil as illustratedin figure 30. Other memory circuits makeuse of magnetostrictive or piezoelectric effectsto retard the pulse. Information may also bestored in electrostatic storage tubes, upon mag-"AND• GATES+INPUT #1INPUT# 2LIMITINGDIODESINPUT#:3INPUT#4INPUT41:5INPUT'I:6INPUT*- 7INPUT# 8Figure 28TYPICAL DIGITAL PACKAGE SHOWING INPUT AND OUTPUT DIODE GATES ANDPULSE AMPLIFIER.
<strong>HANDBOOK</strong>Digital Circuitry 205INPUTI I IOUTPUT0.50.4t----+--+---+,--+----t-F (MC)Figure 29SIMPLE DELAY LINE COMPOSED OFLUMPED INDUCTANCE ANDCAPACITANCE. GRAPH SHOWS HOWDELAY TIME VARIES WITHFREQUENCY OF INPUT PULSE.netic recording tape, and in ferromagneticcores. Experimental magnetic cores have beendeveloped that will permit construction ofFigure 30UNIFORM DELAY LINE BEARSRESEMBLANCE TO COAXIAL CABLE,EXCEPT FOR COILED INNERCONDUCTOR.memories capable of retaining over 10,000separate bits of information. The memory maybe "read" or "interrogated" by sending pulsesthrough the matrix and observing the resultinginduced voltage in a secondary circuit. Theoutput from the memory circuit may be usedto excite a program control system, permittingthe computer to drive another mechanism inthe proper sequence.
E &. E TECHNI-SHEETTABLE OF ALUMINUM TUBING, ROUND DRAWN, 12 FOOT LENGTHS3S-H 14 TUBING 24S-T3 TUBING61S-T6 TUBINGDIAM. WALL WT.#./FT. DIAM. WALL WT.MT DIAM. WALL WT.#IFT3/16 11 .022 11 .01~ 1/4" .035" .028 3/16" .035" .0181/4" .022" .018 3/8 11 .049" .060 1/4" .035" .0281/4" .035 11 .028 1/2 11 .035 11 .061 1/4" .0 5 8 11 .0413/8" .022" .02.8 518 11 .035" .078 5/16" .035" .0363/8" .035" .060 3/4" .035 11 .094 5/16" .058" .0551/2" .035" .060 3/4" .049" .1 ~0 3/8" .035" .0441/2." .049 11 .083 7/8" .049" .151 3/8" .058" .0681/2" .06 5 II .105 1 II .035" .128 3/8" .065" .0745/8" .0~5" .076 1 II .049" .175 7/16" .035" .0525/8" .049 11 .1 04 1" .065" .228 7/16 11 .065 .0895/8 11 .065 11 .133 1 1/4 11 .049 11 .2.2.1 1/2" .0 2.8" .0493/4" .0~5 11 .092. 1 112" .035" .193 1/2" .035" .0603/4" .049" .128 1 1/2" .049 II .268 1/2" .o58 .0953/4" .058 11 .149 1 1/2" .065 11 .351 1/2" .065" .1 043/4 11 .065" .166 2" .065" .473 518 11 .035" .0767/8" .0~5" .110 5/8" .058" .1217/8 11 .049 11 .150 3/4" .035"' .0927/8" .058 11 .177 3/4" .058" .1487/8" .065" .196 314 11 .08~" .2041 II .035" .126 7/8" .0~5" .1091" .049" .174 7/8" .058" .1751 II .058" .204 1 II .035" .1251" .065 .216 1 II .058" .202.111 .083" .2813S-H14 ALUMINUM TUBING-CAN BE 1 1/4" .058" .256BENT OR FLATTENED READILY. 1 1/4" .083 11 .3572.4S-T3 AI.UMINUM TUBING-HAS HIGHTENSILE STRENGTH AND CAN BE BENT OR1 3/8" .058 11 .282FLATTENED ONLY SLIGHTLY WITHOUT 1/1/2 11 .058" .309CRACKING. GOOD FOR ANTENNA MASTS, 1 1/2" .083 11 .510BOOMS, ELEMENTS AND O<strong>THE</strong>R SECTIONS 2.11 .058" .416REQUIRING STRENGTH AND RIGIDITY. 2.11 .083" .59061S-T6 ALUMINUM TUBING-HAS GREATTENSILE STRENGTH BUT CANNOT BE FLATTENED OR BENT.
CHAPTER TWELVERadio Receiver FundamentalsA conventional reproducing device such asa loudspeaker or a pair of earphones is incapableof receiving_ directly the intelligencecarried by the carrier wave of a radio transmittingstation. It is necessary that an additionaldevice, called a radio receiver, beplaced between the receiving antenna and theloudspeaker or headphones.Radio receivers vary widely in their complexityand basic design, depending upon theintended application and upon economic factors.A simple radio receiver for reception ofradiotelephone signals can consist of an earphone,a silicon or germanium crystal as acarrier rectifier or demodulator, and a lengthof wire as an antenna. However, such a receiveris highly insensitive, and offers nosignificant discrimination between two signalsin the same portion of the spectrum.On the other hand, a dual-diversity receiverdesigned for single-sideband reception andemploying double or triple detection mightoccupy several relay racks and would costmany thousands of dollars. However, conventionalcommunications receivers are intermediatein complexity and performance betweenthe two extremes. This chapter is devoted tothe principles underlying the operation ofsuch conventional communications receivers.20712-1 Detection orDemodulationA detector or demodulator is a device forremoving the modulation (demodulating) ordetecting the intelligence carried by an incomingradio wave.Rod iate lephonyDemodulationFigure 1 illustrates an elementaryform of radiotelephonyreceiver employing adiode detector. Energy from a passing radiowave will induce a voltage in the antenna andcause a radio-frequency current to flow fromantenna to ground through coil L,. The alternatingmagnetic field set up around L 1 linkswith the turns of L 2and causes an r-f currentto flow through the parallel-tuned circuit,L,-C,. When variable capacitor C, is adjustedso that the tuned circuit is resonant at thefrequency of the applied signal, the r-f voltageis maximum. This r-f voltage is applied to thediode detector where it is rectified into a varyingdirect current and passed through the earphones.The variations in this current correspondto the voice modulation placed on thesignal at the transmitter. As the earphonediaphragms vibrate back and forth in accord-
208 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0TRIODE®AUDIO OUTPUTFigure 1ELEMENTARY FORM OF RECEIVERThis is the basis of the "crystal set" type of receiver,although a vacuum cliocle may be usee/ inplace of the crystal cliocle. The tank circuit L2-C1is tuned to the frequency it is clesirecl to receive.The by-pass capacitor across the phones shouldhave a low reactance to the carrier frequency beingreceived, but a high reactance to the moclulationon the received signal.PLATE-TICKLER REGENERATION WITH "THROTTLE'CONDENSER REGENERATION CONTROL.-B +BAUDIO OUTPUTCATHODE-TAP REGENERATION WITH SCREEN VOLTAGEREGENERATION CONTROL.ance with the pulsating current they audiblyreproduce the modulation which was placedupon the carrier wave.The operation of the detector circuit isshown graphically above the detector circuitin figure I. The modulated carrier is shownat A, as it is applied to the antenna. B representsthe same carrier, increased in amplitude,as it appears across the tuned circuit. In Cthe varying d-e output from the detector isseen.Figure 2REGENERATIVE DETECTOR CIRCUITSRegenerative detectors are seldom used at the pre·sent time due to their poor selectivity. However,they do illustrate the sirrplest type of receiverwhich may be used either for racliophone or radiotelegraphreception.RadiotelegraphyReceptionSince a c-w telegraphy sig·nal consists of an unmodulatedcarrier which is interruptedto form dots and dashes, it is apparentthat such a signal would not be made audibleby detection alone. While the keying is a formof modulation, it is composed of such low frequencycomponents that the keying envelopeitself is below the audible range for hand keyingspeeds. Some means must be providedwhereby an audible tone is heard while theunmodulated carrier is being received, the tonestopping immediately when the carrier is interrupted.The most simple means of accomplishingthis is to feed a locally generated carrier ofa slightly different frequency into the samedetector, so that the incoming signal will mixwith it to form an audible beat note. The dif·ference frequency, or heterodyne as the beatnote is known, will of course stop and startin accordance with the incoming c-w radiotelegraphsignal, because the audible heterodynecan exist only when both the incomingand the locally generated carriers are present.The AutodyneDetectorThe local signal which is usedto beat with the desired c·wsignal in the detector may besupplied by a separate low-power oscillatorin the receiver itself, or the detector may bemade to self-oscillate, and thus serve thedual purpose of detector and oscillator. A de·rector which self-oscillates to provide a beatnote is known as an autodyne detector, andthe process of obtaining feedback between thedetector plate and grid is called regeneration.An autodyne detector is most sensitive whenit is barely oscillating, and for this reasona regeneration control is always included inthe circuit to adjust the feedback to the properamount. The regeneration control may be eithera variable capacitor or a variable resistor,as shown in figure 2.With the detector regenerative but not os·cillating, it is also quite sensitive. When thecircuit is adjusted to operate in this manner,modulated signals may be received with con·siderably greater strength than with a nonregenerativedetector.
<strong>HANDBOOK</strong>Superregenerative Detectors20912-2 SuperregenerativeReceiversAr ultra-high frequencies, when ir is desiredto keep weight and cost ar a minimuma special form of rhe regenerative receive~known as rhe superregenerator is often usedfor radi.orelephony reception. The superregeneratoriS essenually a regenerative receiverwHh a means provided to throw the detectorrapidly in and our of oscillation. The frequencyat which the detector is made to go in and outof oscillation varies with the frequency to bereceived, bur is usually between 20,000 and5.00,000 times a second. This superregenerauveaction considerably increases the sensitivityof rhe oscillating detector so rhat theusual "backgrou~d hiss" is greatly amplifiedwhen no s1gnal 1s being received. This hissdiminishes. in proportion ro the strength of therece1ved s1gnal, loud signals eliminating thehiss entirely.QuenchMethodsThere are two systems in commonuse for causing the detector to breakin and our of oscillation rapidly. Inone: a separate interruption-frequency oscillator1s arranged so as to vary the voltage rapidlyon one of rhe detector tube elements (usuallyrhe plate, sometimes the screen) at the highrate. necessary. The interruption-frequencyoscdlaror commonly uses a conventional tick~er-feedback circuit with coils appropriate forHS operating frequency.The s~cond, and simplest, type of superregenerativede.rector circuit is arranged soas ~o p~oduce Hs own interruption frequencyose1llauon, wHhout rhe aid of a separate tube.The detector rube damps (or "quenches") itselfour of signal-frequency oscillation ar a highra~e by vurue of the use of a high value ofgr~d leak and proper size plate-blocking andgnd capacHors, In conjunction wirh an excessof feedback. In this rype of "self-quenched"detector, rhe grid leak is quire often returnedto the P.osiri ve side ofrhe power supply (throughthe co.tl) rather than ro rhe cathode. A representativeself-quenched superregenerarive derectorcircuit is shown in figure 3.Except where ir is impossible ro securesufficient reg.enerarive feedback to permit~uperregeneration, rhe self-quenching circuit~s ro be preferred; ir is simpler, is self-adjusrmgas regards quenching amplitude, and canhave good quenching wave form. To obtainas good results wirh a separately quenchedsul?erregenerator, very careful design is requued.However, separately quenched circuitsare useful when it is possible ro make a certain.t~be. oscil~ate on a very high frequencyb.ur n iS impossible to obtain enough regenerationfor self-quenching action.TO AUDIOAMPLIFIERFigure 3SUPERREGENERATIVE DETECTOR CIRCUITA self-quenched superregenerative detector suchas illustrated above is capable of giving goodsensitivity in the v-h-f range. However, the circuithas the disadvantage that its selectivity is relativelypoor. Also, such a circuit should be precededby an r-f stage to suppress the radiation ofa signal by the oscillating detector.The optimum quenching frequency is a functionof rhe signal frequency. As rhe operatingfrequency goes up, so does rhe optimum quenchingfrequency. When the quench frequency isroo low, maximum sensitivity is nor obtained.When ir is roo high, both sensitivity and selectivitysuffer. In fact, rhe optimum quench frequencyfor an operating frequency below 15 Me.is in rhe audible range. This makes rhe superregeneratorimpracticable for use on rhe lowerfrequencies.The high background noise or hiss whichis heard on a properly designed superregeneratorwhen no signal is being received is notthe quench frequency component; ir is rubeand tuned circuit fluctuation noise, indicatingrhar rhe receiver is extremely sensitive.A moderately strong signal will cause rhebackground noise to disappear completely,because rhe superregeneraror has an inherentand instantaneous automatic volume controlcharacteristic. This same a-v-e characteristicmakes rhe receiver comparatively insensitiveto impulse noise such as ignition pulses-ahighly desirable feature. This characteristicalso resulrs in appreciable distortion of a receivedradiotelephone signal, bur nor enoughro affect rhe intelligibility.The selectivity of a superregenerator israther poor as compared to a superheterodyne,but is surprisingly good for so simple a receiverwhen figured on a percentage basisrather than absolute kc. bandwidth.FM Reception A• superregenerarive receiverwill receive frequency modulatedsignals with resulrs comparing favorablywith amplitude modulation if rhe frequencys~ing of the FM transmitter is sufficientlyhigh. For such reception, rhe receiver is detunedslightly ro either side of resonance.
210 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0I ANTENNAr ----,I R.F. tI AMPLIFIERIIL--- _JAUDIO &2OUTPUT0r- ---,IIAUDIO 1r~p~~~~:J~~~~~1~0IOSCJLLATORII (FORC.W.)IL------'Figure 5ESSENTIAL UNITS OF ASUPERHETERODYNE RECEIVERThe basic portions of the receiver are shownIn solicl blocks. Practicable receivers employthe clotted blocks oncl also usually includesuch oclclitlonol circuits as o noiselimiter, an a-v-e circuit, and a crystal filterin the i-f amplifier.Figure 4<strong>THE</strong> FREMODYNE SUPERREGENERATIVESUPERHETERODYNE DETECTOR FORFREQUENCY MODULATED SIGNALSSuperregenerative receivers radiate a strong,broad, and rough signal. For this reason, it isnecessary in most applications to employ aradio frequency amplifier stage ahead of thedetector, with thorough shielding throughoutthe receiver.The Fremodyne The Haze It in e- FremodyneDetector superregenerative circuit isexpressly designed for receptionof FM signals. This versatile circuitcombines the action of the superregenerativereceiver with the superhetrodyne, convertingFM signals directly into audio signals in onedouble triode tube (figure 4). One section ofthe triode serves as a superregenerative mixer,producing an i-f of 22 Me., an i-f amplifier, anda FM detector. The detector action is accomplishedby slope detection tuning on the sideof the i-f selectivity curve.This circuit greatly reduces the radiatedsignal, characteristic of the superregenerativedetector, yet provides many of the desirablefeatures of the superregenerator. The passbandof the Fremodyne detector is about400 kc.12-3 SuperheterodyneReceiversBecause of its superiority and nearly uni"versa! use in all fields of radio reception, thetheory of operation of the superheterodyneshould be famili~r to every radio student andexperimenter. The following discussion concernssuperheterodynes for amplitud~-modul~tionreception. It is, however, appltcable 1npart to receivers for frequency modulation.Principle ofOperationIn the superheterodyne, the incomingsignal is applied to amixer consisting of a non-linearimpedance such as a vacuum tube or a diode.The signal is mixed with a steady signal generatedlocally in an oscillator stage, with theresult that a signal bearing all the modulationapplied to the original signal but of a frequencyequal to the difference between thelocal oscillator and incoming signal frequenciesappears in the mixer output circuit. Theoutput from the mixer stage is fed into a fixedtunedintermediate-frequency amplifier, whereit is amplified and detected in the usual manner,and passed on to the audio amplifier. Figure5 shows a block diagram of the fundamentalsuperheterodyne arrangement. The basiccomponents are shown in heavy lines, thesimplest superheterodyne consisting simplyof these three units. However, a good communicationsreceiver will comprise all of theelements shown, both heavy and dotted blocks.SuperheterodyneAdvantagesThe advantages of superheterodynereception aredirectly attributable to theuse of the fixed-tuned intermediate,-frequency(i-f) amplifier. Since all signals are convertedto the intermediate frequency, this section ofthe receiver may be designed for optimum selectivityand high amplification. High a~plificationis easily obtained in the intermediatefrequencyamplifier, since it operates at a
<strong>HANDBOOK</strong>The Superhetrodyne 211JHINP TO 0.1 .1JFD.Figure 6TYPICAL 1-F AMPLIFIER STAGErelatively low frequency, where conventionalpentode·type tubes give adequate voltage gain.A typical i-f amplifier is shown in figure 6.From the diagram it may be seen that boththe grid and plate circuits are tuned. The tunedcircuits used for coupling between i-f stagesare known as i-f transformers. These will bemore fully discussed later in this chapter.Choice of Inter- The choice of a frequencymediate Frequency for the i-f amplifier involvesseveral considerations.One of these considerations concernsselectivity; the lower the intermediate frequencythe greater the obtainable selectivity.On the other hand, a rather high intermediatefrequency is desirable from the standpoint ofimage elimination, and also for the receptionof signals from television and FM transmittersand modulated self-controlled oscillators, allof which occupy a rather wide band of frequencies,making a broad selectivity characteristicdesirable. Images are a pecularity common toall superheterodyne receivers, and for thisreason they are given a detailed discussionlater in this chapter.While intermediate frequencies as low as50 kc. are used where extreme selectivity isa requirement, and frequencies of 60 Me. andabove are used in some specialized forms ofreceivers, most present-day communicationssuperheterodynes use intermediate frequenciesaround either 455 kc. or 1600 kc.Home-type broadcast receivers almost alwaysuse an i-f in the vicinity of 455 kc.,while auto receivers usually use a frequencyof about 262 kc. The standard frequency forthe i-f channel of FM receivers is 10.7 Me.Television receivers use an i-f which coversthe band between about 21.5 and 27 Me., althougha new band between 41 and 46 Me. iscoming into more common usage.ArithmeticalSelectivityAside from allowing the use offixed-tuned band-pass amplifierstages, the superheterodyne hasan overwhelming advantage over the tunedradio frequency (t-r-0 type of receiver becauseof what is commonly known as arithmeticalselectivity.This c_an best be illustrated by consideringtwo receivers, one of the t-r-f type and one ofthe superheterodyne type, both attempting toreceive a desired signal at 10,000 kc. andeliminate a strong interfering signal at 10,010kc. In the t-r-f receiver, separating these twosignals in the tuning circuits is practicallyimpossible, since they differ in frequency byonly 0.1 per cent. However, in a superheterodynewith an intermediate frequency of, for example,1000 kc., the desired signal will beconverted to a frequency of 1000 kc. and theinterfering signal will be converted to a freque~cyof 1010 kc., both signals appearing atthe mput of the i-f amplifier. In this case thetwo signals may be separated much more ;eadily,since they differ by 1 per cent, or 10 timesas much as in the first case.The ConverterStageThe converter stage, or mixer,of a superheterodyne receivercan be either one of two types:( 1) it may use a single envelope convertertube, such as a 6K8, 6SA7, or 6BE6, or (2) itmay use two tubes, or two sets of elements inthe same envelope, in an oscillator-mixer arrangement.Figure 7 shows a group of circuitsof both types to illustrate presellt practicewith regard to types of converter stages.Converter tube combinations such as shownin figures 7 A and 7B are relatively simple andinexpensive, and they do an adequate job formost applications. With a converter tube suchas the 6SB7-Y or the 6BA7 quite satisfactoryperformance may be obtained for the receptionof relatively strong signals (as for exampleFM broadcast reception) up to frequencies inexcess of 100 Me. However, the equivalent inputnoise resistance of such tubes is of theorder of 200,000 ohms, which is a rather highvalue indeed. So such tubes are not suited foroperation without an r-f stage in the highfrequencyrange if weak-signal reception isdesired.The 6L 7 mixer circuit shown in figure 7C,and the 6BA7 circuit of figure 7D also arecharacterized by an equivalent inpu~ noise resistanceof several hundred thousand ohms, sothat these also must be preceded by one ormore r-f stages with a fairly high gain perstage if a low noise factor is desired of thecomplete receiver.However, the circuit arrangements shownat figures 7F and 6F are capable of low-noiseoperation within themselves, so that thesecircuits may be fed directly from the antennawithout an r-f stage and still provide a goodnoise factor to the complete receiver. Note
212 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0®~6K8TO I. F.AMP.[[®~6SA 7, 6SB 7·Y,6BE6, 6BA 7TO I. F.[[AMP.TOI.F.AMP.[[TO I.F.[[AMP.2 ..UJJF.+150V.Figure 7TYPICAL FREQUENCY-CONVERTER (MIXER) STAGESThe relative aclvantages of the clifferent circuits are cliscussecl in the textthat both these circuits use control· grid in·jection of both the incoming signal and thelocal-oscillator voltage. Hence, paradoxically,circuits such as these should be preceded byan r·f stage if local- oscillator radiation is tobe held to any reasonable value of field in·tensity.Diode Mixers As the frequency of operation ofa superheterodyne receiver is in·creased above a few hundred megacycles thesignal-to-noise ratio appearing in the platecircuit of the mixer tube when triodes or pen·todes are employed drops to a prohibitivelylow value. At frequencies above the upper·fre·quency limit for conventional mixer stages,mixers of the diode type are most commonlyemployed. The diode may be either a vacuumtubeheater diode of a special u-h-f designsuch as the 9005, or it may be a crystal diodeof the general type of the 1N21 through 1N28series.12-4 Mixer Noiseand ImagesThe effects of mixer noise and images aretroubles common to all superheterodynes. Since
<strong>HANDBOOK</strong>Mixer Characteristics 213both these efiects can largely be obviated bythe same remedy, they will be considered together.Mixer Noise Mixer noise of the shot-effecttype, which is evidenced by ahiss in the audio output of the receiver, iscaused by small irregularities in the plate currentin the mixer stage and will mask weaksignals. Noise of an identical nature is generatedin an amplifier stage, but due to thefact that the conductance in the mixer stageis considerably lower than in an amplifierstage using the same tube, the proportion ofinherent noise present in a mixer usually isconsiderably greater than in an amplifier stageusing a comparable tube.Although this noise cannot be eliminated,its effects can be greatly minimized by placingsufficient signal-frequency amplificationhaving a high signal-to-noise ratio ahead ofthe mixer. This remedy causes the signal outputfrom the mixer to be large in proportion tothe noise generated in the mixer stage. Increasingthe gain after the mixer will be of noadvantage in eliminating mixer noise difficulties;greater selectivity after the mixer willhelp to a certain extent, but cannot be carriedtoo far, since this type of selectivity decreasesthe i-f band-pass and if carried too far willnot pass the sidebands that are an essentialpart of a voice-modulated signal.Triode Mixers A triode having a high transconductanceis the quietestmixer tube, exhibiting somewhat less gain buta better signal-to-noise ratio than a compar·able multi-grid mixer tube. However, below 30Me. it is possible to construct a receiver thatwill get down to the atmospheric noise levelwithout resorting to a triode mixer. The additionaldifficulties experienced in avoidingpulling, undesirable feedback, etc., when usinga triode with control-grid injection tend to makemulti-grid tubes the popular choice for thisapplication on the lower frequencies.On very high frequencies, where set noiserather than atmospheric noise limits the weaksignal response, triode mixers are more widelyused. A 6J6 miniature twin triode with gridsin push-pull and plates in parallel makes anexcellent mixer up to about 600 Me.InjectionVoltageThe amplitude of the injection voltagewill affect the conversion transconductanceof the mixer, and thereforeshould be made optimum if maximum signal-to-noiseratio is desired. If fixed bias isemployed on the injection grid, the optimuminjection voltage is quite critical. If cathodebias is used, the optimum voltage is not socritical; and if grid leak bias is employed, theoptimum injection voltage is not at all criticaljust so it is adequate. Typical optimum injectionvoltages will run from 1 to 10 volts forcontrol grid injection, and 45 volts or so forscreen or suppressor grid injection.Images There always are two signal frequencieswhich will combine with a givenfrequency to produce the same difference frequency.For example: assume a superheterodynewith its oscillator operating on a higherfrequency than the signal, which is commonpractice in present superheterodynes, tuned toreceive a signal at 14,100 kc. Assuming ani-f amplifier frequency of 450 kc., the mixerinput circuit will be tuned to 14,100 kc., andthe oscillator to 14,100 plus 450, or 14,550 kc.Now, a strong signal at the oscillator frequencyplus the intermediate frequency (14, 550plus 450, or 15,000 kc.) will also give a dif·ference frequency of 450 kc. in the mixer outputand will be. heard al_so. Note that the image1s always twtce the mtermediate frequencyaway from the desired signal. Images causerepeat points on the tuning dial.The only way that the image could be eliminatedin this particular case would be to makethe selectivity of the mixer input circuit, andany circuits preceding it, great enough so thatthe 15,000-kc. signal never reaches the mixergrid in sufficient amplitude to produce interference.. For any particular intermediate frequency,image interference troubles become increasinglygreater as the frequency to which thesignal-frequency portion of the receiver istuned is increased. This is due to the fact thatthe percentage difference between the desiredfrequency and the image frequency decreasesas the receiver is tuned to a higher frequency.The ratio of strength between a signal at theimage frequency and a signal at the frequencyto which the receiver is tuned producing equaloutput is known as the image ratio. The higherthis ratio, the better the receiver in regard toimage-interference troubles.With but a single tuned circuit between themixer grid and the antenna, and with 400-500kc. i-f amplifiers, image ratios of 60 db andover are easily obtainable up to frequenciesaround 2000 kc. Above this frequency, greaterselectivity in the mixer grid circuit throughthe use of additional tuned circuits betweenthe mixer and the antenna is necessary if agood image ratio is to be maintained.12-5 R-F StagesSince the necessary tuned circuits betweenthe mixer and the antenna can be combinedwith tubes to form r-f amplifier stages, the
214 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0®GROUNDED-GRID6AB4, t 6J6,6J4, {: 12A T7Figure 8TYPICAL PENTODE R-F AMPLIFIER STAGE®CATHODE -COUPLED6J6+f20V.reduction of the effects of mixer noise and theincreasing of the image ratio can be accom·plished in a single section of the receiver.When incorporated in the receiver, this sectionis known simply as an r-f amplifier; whenit is a separate unit with a separate tuningcontrol it is often known as a preselector.Either one or two stages are commonly usedin the preselector or r-f amplifier. Some pre·selectors use regeneration to obtain stillgreater amplification and selectivity. An r-famplifier or preselector embodying more thantwo stages rarely ever is employed since twostages will ordinarily give adequate gain tooverride mixer noise.©LOW NOISECAS CODE+120 v.Generally speaking, atmosph eric noise in the frequencyrange above 30 Me. is quiteR-F Stages inthe V.H.F Rangelow-so low, in fact, that the noise generatedwithin the receiver itself is greater than thenoise received on the antenna. Hence it is ofthe greatest importance that internally generatednoise be held to a minimum in a receiver.At frequencies much above 300 Me. there isnot too much that can be done at the presentstate of the art in the direction of reducingreceiver noise below that generated in the converterstage. But in the v-h-f range, between30 and 300 Me., the receiver noise factor in awell designed unit is determined by the characteristicsof the first r-f stase.The usual v-h-f receiver, whether for communicationsor for FM or TV reception, usesa miniature pentode for the first r-f amplifierstage. The 6AK5 is the best of presently availabletypes, with the 6CB6 and the 6DC6 closelyapproaching the 6AK5 in performance. Butwhen gain in the first r-f stage is not so important,and the best noise factor must be obtained,the first r-f stage usually uses a triode.Shown in figure 9 are four commonly usedtypes of triode r-f stages for use in the v-h-frange. The circuit at (A) uses few componentsand gives a moderate amount of gain with verylow noise. It is most satisfactory when thefirst r-f stage is to be fed directly from a low-@+250V.+120 v.Figure 9TYPICAL TRIODE V-H-FR-F AMPLIFIER STAGESTriocle r-f stages contribute the least amount ofDQise output for a given signa/ level, hence theirfrequent use in the v·h-f range.impedance coaxial transmission line. Figure9 (B) gives somewhat more gain than (A), butrequires an input matching circuit. The effec·ti ve gain of this circuit is somewhat reducedwhen it is being used to amplify a broad bandof frequencies since the effective Gm of thecathode-coupled dual tube is somewhat less
<strong>HANDBOOK</strong>The Cascode Amplifier215------,IIIIIII11~--~CONVENTIONAL COMMUNI CATIONS 1 1 HIGHLY SELECTIVE ACCESSORY I. F. II RECEIVER jl AMPLIFIER AND DEMODULATOR (Q5'ER)IL---- --------- _jL---------- __ jFigure 10TYPICAL DOUBLE-CONVERSION SUPERHETERODYNE RECEIVERSIllustrated at (A) is the basic circuit of a commercial clouble-conversion superheteroclyne receiver. At (B) isillustrated the application of an accessory sharp i-f channel for obtaining improved selectivity from a con·ventlonal communications receiver through the use of the double-conversion principle.than half the Gm of either of the two tubestaken alone.The CascodeAmplifierThe Cascode r-f amplifier, developedat the MIT RadiationLaboratory during World War II,is a low noise circuit employing a groundedcathode triode driving a grounded grid triode,as shown in figure 9C. The stage gain of sucha circuit is about equal to that of a pentodetube, while the noise figure remains at the lowlevel of a triode tube. Neutralization of thefirst triode tube is usually unnecessary below50 Me. Above this frequency, a definite improvementin the noise figure may be obtainedrhrough the use of neutralization. The neutralizingcoil, LN, should resor.ate at the operatingfrequency with the grid-plate capacity ofthe first triode tube.The 6BQ7 A and 6BZ7 tubes are designed foruse in cascode circuits, and may be used togood advantage in the 144 Me. and 220 Me. amateurbands (figure 9D). For operation at higherfrequencies, the 6AJ4 tube is recommended.Double Conversion As previously mentioned,the use of a higher intermediatefrequency will also improve the imageratio, at the expense of i-f selectivity, byplacing the desired signal and the image fartherapart. To give both good image ratio atthe higher frequencies and good selectivity inthe i-f amplifier, a system known as doubleconversion is sometimes employed. In this system,the incoming signal is first converted toa rather high intermediate frequency, and thenamplified and again converted, this time to amuch lower frequency. The first intermediatefrequency supplies the necessary wide separationbetween the image and the desired signal,while the second one supplies the bulk ofthe i-f selectivity.The double-conversion system, as illustratedin figure 10, is receiving two generaltypes of application at the present time. Thefirst application is for the purpose of attainingextremely good stability in a communicationsreceiver through the use of crystal control ofthe first oscillator. In such an arrangement,as used in several types of Collins receivers,the first oscillator is crystal controlled and isfollowed by a tunable i-f amplifier which thenis followed by a mixer stage and a fixed-tunedi-f amplifier on a much lower frequency.Through such a circuit arrangement the stabilityof the complete receiver is equal to the
216 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0stability of the oscillator which feeds the secondmixer, while the selectivity is determinedby the bandwidth of the second, fixed i-f amplifier.The second common applic"ation of thedouble-conversion principle is for the purposeof obtaining a very high degree of selectivityin the complete communications receiver. Inthis type of application, as illustrated in figure10 (B), a conventional communications receiveris modified in such a manner that itsnormal i-f amplifier (which usually is in the450 to 915 kc. range) instead of being fed toa demodulator and then to the audio system,is alternatively fed to a fixed-tune mixer stageand then into a much lower intermediate frequencyamplifier before the signal is demodulatedand fed to the audio system. The accessoryi-f amplifier system (sometimes called aQ5' er) normally is operated on a frequency of175 kc., 85 kc., or 50 kc.R.f.INPUT 3TOA.V.C. "= +BFigure 11ILLUSTRATING "COMMON POINT"BY-PASSINGTo reduce the detrimental eHects of cathode circuitinductance in v-h-f stages, all by-pass capacitorsshould be returned to the cathode terminalat the socket. Tubes with two cathode leads cangive improved performance if the grid return ismade to one cathocle terminal while the plate andscreen by-pass returns are macle to the cathacleterminal which is connected to the suppressorwithin the tube.12-6 Signal-FrequencyTuned CircuitsThe signal-frequency tuned circuits in highfrequencysuperheterodynes and tuned radiofrequency types of receivers consist of coilsof either the solenoid or universal-wound typesshunted by variable capacitors. It is in thesetuned circuits that the causes of success orfailure of a receiver often lie. The universalwoundtype coils usually are used at frequenciesbelow 2000 kc.; above this frequency thesingle-layer solenoid type of coil is moresatisfactory.Impedanceand QThe two factors of greatest significance in determining the gainper-stageand selectivity, respectively,of a tuned amplifier are tuned-circuitimpedance and tuned-circuit Q. Since the resistanceof modern capacitors is low at ordinaryfrequencies, the resistance usually canbe considered to be concentratE'd in the coil.The resistance to be considered in making Qdeterminations is the r-f resistance, not thed-e resistance of the wire in the coil. The latterordinarily is low enough that it may beneglected. The increase in r-f resistance overd-e resistance primarily is due to skin effectand is influenced by such factors as wire sizeand type, and the proximity of metallic objectsor poor insulators, such as coil forms withhigh losses. Higher values of Q lead to betterselectivity and increased r-f voltage acrossthe tuned circuit. The increase in voltage isdue to an increase in the circuit impedancewith the higher values of Q.Frequently it is possible to secure an increasein impedance in a resonant circuit, andconsequently an increase in gain from an amplifierstage, by increasing the reactancethrough the use of larger coils and smallertuning capacitors (higher L/C ratio).lllput Resistance Another factor which influencesthe operation oftuned circuits is the input resistance of thetubes placed across these circuits. At broadcastfrequencies, the input resistance of mostconventional r-f amplifier tubes is high enoughso that it is not bothersome. But as the frequencyis increased, the input resistance becomeslower and lower, until it ultimatelyreaches a value so low that no amplificationcan be obtained from the r-f stage.The two contributing factors to the decreasein input resistance with increasing frequencyare the transit time required by an electrontraveling between the cathode and grid, andthe inductance of the cathode lead commonto both the plate and grid circuits. As thefrequency becomes higher, the transit timecan become an appreciable portion of the timerequired by an r-f cycle of the signal voltage,and current will actually flow into the grid.The result of this effect is similar to thatwhich would be obtained by placing a resistancebetween the tube's grid and cathode.Superheterodyne Because the oscillator in aTracking superheterodyne operates"offset" from the other frontend circuits, it is necessary to make specialprovisions to allow the oscillator to trackwhen similar tuning capacitor sections are
<strong>HANDBOOK</strong>MIXERPADDING CAPACITORTUNING CAPACITOROSCILLATORSERIES TRACKING CAPACITORTuning Circuits 217ocsITE® ©8~ ~®®Figure 12SERIES TRACKING EMPLOYEDIN <strong>THE</strong> H-F OSCILLATOR OF ASUPERHETERODYNEThe series tracking capacitor permits the use oficlentical gangs in a ganged capacitor, since thetracking capacitor slows clown the rate of frequen·cy change in the oscillator so that a constant clif·ference in frequency between the oscillator anclthe r-f stage (equal to the i-f amplifier frequency)may be maintained.ganged. The usual method of obtaining goodtracking is to operate the oscillator on thehigh-frequency side of the mixer and use aseries tracking capacitor to slow down thetuning rate of the oscillator. The oscillatortuning rate must be slower because it coversa smaller range than does the mixer when bothare expressed as a percentage of frequency.At frequencies above 7000 kc. and with ordi·nary intermediate frequencies, the differencein percentage between the two tuning rangesis so small that it may be disregarded in re·ceivers designed to cover only a small range,such as an amateur band.A mixer and oscillator tuning arrangementin which a series tracking capacitor is provid·ed is shown in figure 12. The value of thetracking capacitor varies considerably withdifferent intermediate frequencies and tuningranges, capacitances as low as .0001 flfd.being used at the lower tuning-range frequen·cies, and values up to .01 flfd. being used atthe higher frequencies.Superheterodyne receivers designed to coveronly a single frequency range, such as thestandard broadcast band, sometimes obtaintracking between the oscillator and the r·f cir·cuits by cutting the variable plates of the oscillatortuning section to a different shapefrom those used to tune the r-f stages.Frequency RangeSelectionThe frequency to which areceiver responds may bevaried by changing the sizeof either the coils or the capacitors in the tuningcircuits, or both. In short-wave receiversFigure 13BANDSPREAD CIRCUITSParallel banclspreacl is illustrated at (A) ancl (B),series banclspreacl at (C), ancl tappecl-coil banclspreaclat (D),a combination of both methods is usually employed,the coils being changed from one bandto another, and variable capacitors being usedto tune the receiver across each band. In practicalreceivers, coils may be changed by oneof two methods: a switch, controllable fromthe panel, may be used to switch coils of differentsizes into the tuning circuits or, alternatively,coils of different sizes may beplugged manually into the receiver, the connectioninto the tuning circuits being made bysuitable plugs on the coils. Where there areseveral plug-in coils for each band, they aresometimes arranged to a single mounting sttip,allowing them all to be plugged in simultan·eously.BandspreadTuningIn receivers using large tuningcapacitors to cover the short·wave spectrum with a minimumof coils, tuning is likely to be quite difficult,owing to the large frequency range covered bya small rotation of the variable capacitors.To alleviate this condition, some method ofslowing down the tuning rate, or bandspreading,must be used.Quantitatively, bandspread is usually desig·nated as being inversely proportional to therange covered. Thus, a large amount of bandspreadindicates that a small frequency rangeis covered by the bandspread control. Conversely,a small amount of bandspread is takento mean that a large frequency range is coveredby the bandspread dial.Types ofBandspreadBandspreading systems are oftwo general types: electrical andmechanical. Mechanical systemsare exemplified by high-ratio dials in whichthe tuning capacitors rotate much more slowly
218 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0than the dial knob. In this system, there isoften a separate scale or pointer c!ither connectedor geared to the dial knob to facilitateaccurate dial readings. However, there is apractical limit to the amount of mechanicalbandspread which can be obtained in a dialand capacitor before the speed-reduction unitand capacitor bearings become prohibitivelyexpensive. Hence, most receivers employ acombination of electrical and mechanical bandspread.In such a system, a moderate reductionin the tuning rate is obtained in the dial,and the rest of the reduction obtained by electricalbandspreading.Stray CircuitCapacitanceIn this book and in other radioliterature, mention is sometimesmade of stray or circuit capacitance.This capacitance is in the usual sensedefined as the capacitance remaining acrossa coil when all the tuning, bandspread, andpadding capacitors across the circuit are attheir minimum capacitance setting.Circuit capacitance can be attributed to twogeneral sources. One source is that due to theinput and output capacitance of the tube whenits cathode is heated. The input capacitancevaries somewhat from the static value whenthe tube is in actual operation. Such factorsas plate load impedance, grid bias, and frequencywill cause a change in input capacitance.However, in all except the extremelyhigh-transconductance tubes, the publishedmeasured input capacitance is reasonably closeto the effective value when the tube is usedwithin its recommended frequency range. Butin the high-transconductance types the effectivecapacitance will vary considerably fromthe published figures as operating conditionsare changed.The second source of circuit capacitance,and that which is more easily controllable, isthat contributed by the minimum capacitanceof the variable capacitors across the circuitand that due to capacitance between the wiringand ground. In well-designed high-frequencyreceivers, every effort is made to keepthis portion of the circuit capacitance at aminimum since a large capacitance reducesthe tuning range available with a given coiland prevents a good L/C ratio, and consequentlya high-impedance tuned circuit, frombeing obtained.A good percentage of stray circuit capacitanceis due also to distributed capacitanceof the coil and capacitance between wiringpoints and chassis.Typical values of circuit capacitance mayrun from 10 to 75 p.p.fd. in high-frequency receivers,the first figure representing concentric-linereceivers with acorn or miniaturetubes and extremely small tuning capacitors,and the latter representing all·wa ve sets withbandswitching, large tuning capacitors, andconventional tubes.12-7 1-F Tuned Circuits!·f amplifiers usually employ bandpass circuitsof some sort. A bandpass circuit is ex~ctly what the name implies-a circuit for passlOga band of frequencies. Bandpass arrangementsca? _be designed for almost any degreeof selectivity, the type used in any particularcase depending upon the ultimate applicationof the amplifier.1-FTransformersIntermediate frequency transformersordinarily consist oftwo or more tuned circuits andsome method of coupling the tuned circuitstogether. Some representative arrangementsare shown in figure 14. The circuit shown atA is the conventional i-f transformer with thecoupling, M, between the tuned circ~its beingprovided by inductive coupling from one coilto the other. As the coupling is increased, theselectivity curve becomes less peaked, and~hen a condition known as critical couplingIS reached, the top of the curve be gins to flattenout. When the coupling is increased stillmore, a dip occurs in the top of the curve.The windings for this type of i-f transformer,as well as most others, nearly always consistof small, flat universal-wound pies mountedeither on a piece of dowel to provide an aircore or on powdered-iron for iron core i-f transformers.The iron-core transformers generallyhave somewhat more gain and better selectivitythan equivalent air·core units.The circuits shown at figure 14-B and C arequite similar. Their only difference is the typeof mutual coupling used, an inductance beingused at B and a capacitance at C. The operationof both circuits is similar. Three resonantcircuits are formed by the components. InB, for example, one resonant circuit is formedby L" C" C 2 and L 2 all in series. The frequencyofthis resonant circuit is just the sameas that of a single one of the coils and capaci·tors, since the coils and capacitors are similarin both sides of the circuit, and the resonantfrequency of the two capacitors and thetwo coils all in series is the same as that ofa single coil and capacitor. The second resonantfrequency of the complete circuit is determinedby the characteristics of each half ofthe circuit containing the mutual coupling device.In B, this second frequency will be lowerthan the first, since the resonant frequency ofL,, C, and the inductance, M, or L,, C 2and Mis lower than that of a single coil and capaci-
<strong>HANDBOOK</strong>1-F Amplifiers 21 9tor, due to the inductance of M being added tothe circuit.The opposite effect takes place at figure14-C, where the common coupling impedanceis a capacitor. Thus, at C the second reson·ant frequency is higher than the first. In eithercase, however, the circuit has two resonant fre·quencies, resulting in a flat-topped selectivitycurve. The width of the top of the curve iscontrolled by the reactance of the mutualcoupling component. As this reactance is in·creased (inductance made greater, capacitancemade smaller), the two resonant frequenciesbecome further apart and the curve is broad·ened.In the circuit of figure 14-D, there is indue·tive coupling between the center coil and eachof the outer coils. The result of this arrangementis that the center coil acts as a sharplytuned coupler between the other two. A signalsomewhat off the resonant frequency of thetransformer will not induce as much currentin the center coil as will a signal of the cor·rect frequency. When a smaller current is inducedin the center coil, it in turn transfersa still smaller current to the output coil. Theeffective coupling between the outer coils in·creases as the resonant frequency is ap·proached, and remains nearly constant over asmall range and then decreases again as theresonant band is passed.Another very satisfactory bandpass arrange·ment, which gives a very straight-sided, flat·topped curve, is the negative-mutual arrange·ment shown at figure 14-E. Energy is trans·£erred between the input and output circuits inthis arrangement by both the negative-mutualcoils, M, and the common capacitive reactance,C. The negative-mutual coils are interwoundon the same form, and connected backward.Transformers usually are made tunable overa small range to permit accurate alignment inthe circuit in which they are employed. Thisis accomplished either by means of a variablecapacitor across a fixed inductance, or bymeans of a fixed capacitor across a variableinductance. The former usually employ eithera mica-compression capacitor (designated"mica tuned"), or a small air dielectric vari·able capacitor (designated "air tuned"). Thosewhich use a fixed capacitor usually employ apowdered iron core on a threaded rod to varythe inductance, and are known as "permea·bility tuned."Shape F actarIt is obvious that to pass modu·lation sidebands and to allowfor slight drifting of the transmitter carrier fre·quency and the receiver local oscillator, thei-f amplifier must pass not a single frequencybut a band of frequencies. The width of thispass band, usually 5 to 8 kc. at maximumFigure 141-F AMPLIFIER COUPLINGARRANGEMENTSTbe interstage caupling arrangements illustrateddx>ve give a beHer shape factor (more straightsided selectivity curve} than would the same num·ber of tuned circuits caupled by means of tubes.width in a good commumcattons receiver, isknown as the pass band, and is arbitrarilytaken as the width between the two frequen·cies at which the response is attenuated 6 db,or is "6 db down." However, it is apparentthat to discriminate against an interfering sig·nal which is stronger than the desired signal,much more than 6 db attenuation is required.The attenuation arbitrarily taken to indicateadequate discrimination against an interferingsignal is 60 db.
220 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0-15 -10 -5 455 +5 +10 +15KC.Figure 16ELECTRICAL EQUIVALENT OFQUARTZ FILTER CRYSTALThe crystal is e~ivalent to a very large value ofinductance in series with small values of capacitanceanrl resistance, with a larger though stillsmall value of capacitance across the whole circuit"(representing holcler capacitance plus straycapacitances).Figure 151-F PASS BAND OF TYPICALCOMMUNICATIONS RECEIVERIt is apparent that it is desirable to havethe bandwidth at 60 db down as narrow aspossible, but it must be done without makingthe pass band ( 6 db points) too narrow for satisfactoryreception of the desired signal. Thefigure of merit used to show the ratio of bandwidthat 6 db down to that at 60 db down isdesignated shape factor. The ideal i-f curve,a rectangle, would have a shape factor of 1.0.The i-f shape factor in typical communicationsreceivers runs from 3.0 to 5.5.The most practicable method of obtaining alow shape factor for a given number of tunedcircuits is to employ them in pairs, as in figure14-A, adjusted to critical coupling (thevalue at which two resonance points just beginto become apparent). If this gives toosharp a "nose" or pass band, then coils oflower Q should be employed, with the couplingmaintained at the critical value. As the Q islowered, closer coupling will be required forcritical coupling.Conversely if the pass band is too broad,coils of higher Q should be employed, thecoupling being maintained at critical. If thepass band is made more narrow by using loosercoupling instead of raising the Q and maintaninigcritical coupling, the shape factor willnot be as good.The pass band will not be much narrowerfor several pairs of identical, critically coupledtuned circuits than for a single pair. However,the shape factor will be greatly improved aseach additional pair is added, up to about 5pairs, beyond which the improvement for eachadditional pair is not significant. Commerciallyavailable communications receivers ofgood quality normally employ 3 or 4 doubletuned transformers with coupling adjusted tocritical or slightly less.The pass band of a typical communicationreceiver having a 455 kc. i-f amplifier is shownin figure 15."MillerEffect"As mentioned previously, the dynamicinput capacitance of a tube variesslightly with bias. As a-v-e voltagenormally is applied to i-f tubes for radiotelephonyreception, the effective grid-cathodecapacitance varies as the signal strengthvaries, which produces the same effect asslight detuning of the i-f transformer. Thiseffect is known as "Miller effect," and canbe minim i zed to the extent that it is nottroublesome either by using a fairly low L/Cratio in the transformers or by incorporatinga small amount of degenerative feedback, thelatter being most easily accomplished by leav·ing part of the cathode resistor unbypassedfor r.f.Crystal Filters The pass band of an intermediatefrequency amplifiermay be made very narrow through the use of apiezoelectric filter crystal employed as aseries resonant circuit in a bridge arrangementknown as a crystal filter. The shape factoris quite poor, as would be expected whenthe selectivity is obtained from the equivalentof a single tuned circuit, but the very narrowpass band obtainable as a result of the extremelyhigh Q of the crystal makes the crystalfilter useful for c-w telegraphy reception.The pass band of a 455 kc. crystal filter maybe made as narrow as 50 cycles, while thenarrowest pass band that can be obtained witha 455 kc. tuned circuit of practicable dimensionsis about 5 kc.The electrical equivalent of a filter crystalis shown in figure 16. For a given frequency,L is very high, C very low, and R (assuming
<strong>HANDBOOK</strong>Crystal Filters 221z,Figure 17EQUIVALENT OF CRYSTALFILTER CIRCUITFor a given voltage out of the generator, the voltagedeveloped across zl depends upon the ratioof the impedance of X to the sum of the impedancesof Z and Z1. Because of the high Q of the crystal,Its impedance changes rapidly with changes infrequency.a good crystal of high Q) is very low. CapacitanceC, represents the shunt capacitance ofthe electrodes, plus the crystal holder andwiring, and is many times the capacitance ofC. This makes the crystal act as a parallelresonant circuit with a frequency only slightlyhigher than that of its f r e que n c y of seriesresonance. For crystal fi 1 t e r use it is theseries resonant characteristic that we are primarilyinterested in.The electrical equivalent of the basic crystalfilter circuit is shown in figure 17. If theimpedance of Z plus Z 1is low compared to theimpedance of the crystal X at resonance, thenthe current flowing through Z 1 , and the voltagedeveloped across it, will be almost in inverseproportion to the impedance of X, which hasa very sharp resonance curve.If the impedance of Z plus Z 1 is made highcompared to the resonant impedance of X, thenthere will be no appreciable drop in voltageacross Z 1 as the frequency departs from theresonant frequency of X until the point isreached where the impedance of X approachesthat of Z plus Z 1 • This has the effect of broadeningout the curve of frequency versus voltagedeveloped across Z., which is another way ofsaying that the selectivity of the crystal filter(but not the crystal proper) has been reduced.In practicable filter circuits the impedancesZ and z, usually are represented by some formof tuned circuit, but the basic principle ofoperation is the same.Practical Filters It is necessary to balanceout the capacitance acrossthe crystal holder (C., in figure 16) to preventbypassing around the crystal undesired signalsoff the crystal resonant frequency. The hal·ancing is done by a phasing circuit whichtakes out-of-phase voltage from a balanced in-Figure 18TYPICAL CRYSTAL FILTER CIRCUITput circuit and passes it to the output side ofthe crystal in proper phase to neutralize thatpassed through the holder capacitance. A representativepractical filter arrangement isshown in figure 18. The balanced input circuitmay be obtained either through the use of asplit-stator capacitor as shown, or by the useof a center-tapped input coil.Varlable-SelectivityFiltersIn the circuit of figure 18, theselectivity is minimum withthe crystal input circuit tunedto resonance, since at resonance the impedanceof the tuned circuit is maximum. As theinput circuit is detuned from resonance, however,the impedance decreases, and the selectivitybecomes greater. In this circuit, the out·put from the crystal filter is tapped down onthe i-f stage grid winding to provide a lowvalue of series impedance in the output circuit.It will be recalled that for maximum selectivity,the total impedance in series with thecrystal (both input and output circuits) mustbe low. If one is made low and the other ismade variable, then the selectivity may bevaried at will from sharp to broad.The circuit shown in figure 19 also achievesvariable selectivity by adding a variable impedancein series with the crystal circuit. Inthis case, the variable impedance is in serieswith the crystal output circuit. The impedanceof the output circuit is varied by varying theQ. As the Q is reduced (by adding resistancein series with the coil), the impedance decreasesand the selectivity becomes greater.The input circuit impedance is made low byusing a non-resonant secondary on the inputtransformer.A variation of the circuit shown at figure19 consists of placing the variable resistanceacross the coil and capacitor, rather than inseries with them. The result of adding the resistoris a reduction of the output impedance,and an increase in selectivity. The circuit behavesoppositely to that of figure 19, however;as the resistance is lowered the selectivitybecomes greater. Still another variation of figure19 is to use the tuning capacitor acrossthe output coil to vary the output impedance.
222 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0+ SELECTIVITYCONTROLFigure 19VARIABLE SELECTIVITYCRYSTAL FILTERThis circuit permits of a greater a>ntrol of selectivitythan cloes the circuit of figure 76, and cloesnot require a split-stator variable capacitor.404550-4 -3 -2 -1 455 +I +Z +3 +4KC.As the output circuit is detuned from resonance,its impedance is lowered, and theselectivity increases. Sometimes a set offixed capacitors and a multipoint switch areused to give step·by·step variation of the outputcircuit tuning, and thus of the crystalfilter selectivity.RejectionNotchAs previously discussed, a filtercrystal has both a resonant(seriesresonant) and an anti-resonant(parallel resonant) frequency, the impedanceof the crystal being quite low at the formerfrequency, and quite high at the latter fre·quency. The anti-resonant frequency is justslightly higher than the resonant frequency,the difference depending upon the effectiveshunt capacitance of the filter crystal andholder. As adjustment of the phasing capacitorcontrols the effective shunt capacitance of thecrystal, it is possible to vary the anti·reso·nant frequency of the crystal slightly withoutunbalancing the circuit sufficiently to let un·desired signals leak through the shunt capac·itance in appreciable amplitude. At the exactanti-resonant frequency of the crystal the at·tenuation is exceedingly high, because of thehigh impedance of the crystal at this frequen·cy. This is cailed the rejection notch, andcan be utilized virtuaily to eliminate theheterodyne image or repeat tuning of c·w sig·nals. The beat frequency osciilator can beso adjusted and the phasing capacitor so ad·justed that the desired beat note is of sucha pitch that the image (the same audio noteon the other side of zero beat) fails in the re·jection notch and is inaudible. The receiverthen is said to be adjusted for single-signaloperation.The rejection notch sometimes can be em·ployed to reduce interference from an undesiredphone signal which is very close infrequency to a desired phone signal. The filteris adjusted to "broad" so as to permit tele·Figure 201-F PASS BAND OF TYPICALCRYSTAL FILTERCOMMUNICATIONS RECEIVERphony reception, and the receiver tuned sothat the carrier frequency of the undesiredsignal falls in the rejection notch. The modu·lation sidebands of the undesired signal stillwiii come through, but the carrier heterodynewiii be effectively eliminated and interferencegreatly reduced.A typical crystal selectivity curve for acommunications receiver is shown in figure 20.Crystal FilterConsiderationsA crystal filter, especiaiiywhen adjusted for single sig·nal reception, greatly reducesinterference and background noise, the latterfeature permitting signals to be copied thatwouldordinarily be too weak to be heard abovethe background hiss. However, when the filteris adjusted for maximum selectivity, the passband is so narrow that the received signalmust have a high order of stability in order tostay within the pass band. Likewise, the localosciliator in the receiver must be highly stable,or constant retuning wiii be required. Anothereffect that wiii be noticed with the filter adjustedtoo "sharp" is a tendency for codecharacters to produce a ringing sound, andhave a hangover or "tails." This effect limitsthe code speed that can be copied satisfac·torily when the filter is adjusted for extremeselectivity.The MechanicalFilterThe Coilins Mechanical Fil·ter (figure 21) is a new conceptin the field of selec·t1v1ty. It is an electro-mechanical bandpassfilter about half the size of a cigarette pack·age. As shown in figure 22, it consists of aninput transducer, a resonant mechanical sec·
<strong>HANDBOOK</strong>Collins Mechanical Filter 223tion comprised of a number of metal discs, andan output transducer.The frequency characteristics of the resonantmechanical section provide the almostrectangular selectivity curves shown in figure23. The input and output transducers serveonly as electrical to mechanical coupling devicesand do not affect the selectivity characteristicswhich are determined by the metaldiscs. An electrical signal applied to the inputterminals is converted into a mechanicalvibration at the input transducer by means ofmagnetostriction. This mechanical vibrationtravels through the resonant mechanical sectionto the output transducer, where it is convertedby magnetostriction to an electricalsignal which appears at the output terminals.In order to provide the most efficient electromechanicalcoupling, a small magnet in themounting above each transducer applies a magneticbias to the nickel transducer core. Theelectrical impulses then add to or subtractfrom this magnetic bias, causing vibration ofthe filter elements that corresponds to theexciting signal. There is no mechanical motionexcept for the imperceptible vibration of themetal discs.Magnetostrictively-driven mechanical filtershave several advantages over electrical equivalents.In the region from 100 kc. to 500 kc.,the mechanical elements are extremely small,and a mechanical filter having better selectivitythan the best of conventional i-f systemsmay be enclosed in a package smaller than onei-f transformer.Since mechanical elements with Q's of 5000or more are readily obtainable, mechanical filtersmay be designed in accordance with thetheory for lossless elements. This permits filtercharacteristics that are unobtainable withelectrical circuits because of the relativelyhigh losses in electrical elements as comparedwith the mechanical elements used in thefilters.Figure 21COLLINS MECHANICAL FILTERSThe Collins Mechanical Filter is anelectro-mechanical bandpass filter whichsurpasses, in one small unit, the se ..lectivity of conventional, space-consumingfilters. At the left is the miniaturizedfilter, less than 2Y.s • long. Type H isnext, ancf two horizontal mounting typesare at right. For exploded view of CollinsMechanical Filter, see figure 46.The frequency characteristics of the mechanicalfilter are permanent, and no adjustment isrequired or is possible. The filter is enclosedin a hermetically sealed case.In order to realize full benefit from the mechanicalfilter's selectivity characteristics,it is necessary to provide shielding betweenthe external input and output circuits, capableof reducing transfer of energy external to theONE SUPPORTINGDISC ATEACH ENDELECTRICAL SIGNAL{INPUT OR OUTPUT)Figure 22MECHANICAL FILTERFUNCTIONAL DIAGRAMELECTRICAL SIGNAL(INPUT OR OUTPUT)Figure 23Selectivity curves of 455-kc. mechanical filterswith nominal 0.8-kc. (doHed line) and 3.1-kc.(solid line) bandwidth at -6 db.
224 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0- --l10KPOT./.
<strong>HANDBOOK</strong> Detector Circuits 225R.f.MIXER-~100M500M.05~Figure 26TYPICAL A-V-C CIRCUIT USING A DOUBLE DIODEAny of the small clual-c/ioc/e tubes may be usee/ in this circuit. Or, if c/esirec/, a c/uo-c/ioc/etrioc/emay be usee/, with the triode acting as the first audio stage. The left-hone/ c/ioc/e servesas the detector, while the right-hone/ side acts as the a-v-e rectifier. The use of separatec/ioc/es for detector one/ a-v·c reduces distortion when receiving an AM signal with a highmodulation percentage.ages on the b·f-o tube is changed, as the lattercircuits usually change the frequency of theb.f.o. at the same time they change the strength,making it necessary to reset the trimmer eachtime the output is adjusted.The b.f.o. usually is provided with a smalltrimmer which is adjustable from the frontpanel to permit adjustment over a range of 5or 10 kc. For single-signal reception the b.f.o.always is adjusted to the high-frequency side,in order to permit placing the heterodyne imagein the rejection notch.In order to reduce the b·f·o signal outputvoltage to a reasonable level which will pre·vent blocking the second detector, the signalvoltage is delivered through a low-capacitance(high-reactance) capacitor having a value of1 to 2 wfd.Care must be taken with the b.f.o. to pre·vent harmonics of the oscillator from beingpicked up at multiples of the b·f·o frequency.The complete b.f.o. together with the couplingcircuits to the second detector, should be thor·oughly shielded to prevent pickup of the har·monies by the input end of the receiver.If b·f·o harmonics still have a tendency togive trouble after complete shielding and iso·lation of the b·f·o circuit has been accom·plished, the passage of these harmonics fromthe b·f·o circuit to the rest of the receiver canbe stopped through the use of a low-pass filterin the lead between the output of the b-f·o cir·cuit and the point on the receiver where theb·f·o signal is to be injected.12-8DetectorsDetector, Audio, andControl CircuitsSecond detectors for use in super·heterodynes are usually of thediode, plate, or infinite-impedance types. Oc·casionall y, grid-leak detectors are used in re·ceivers using one i·f stage or none at all, inwhich case the second detector usually ismade regenerative.Diodes are the most popular second detect·ors because they allow a simple method ofobtaining automatic volume control to be used.Diodes load the tuned circuit to which theyare connected, however, and thus reduce theselectivity slightly. Special i·f transformersare used for the purpose of providing a low·impedance input circuit to the diode detector.Typical circuits for grid-leak, diode, plateand infinite-impedance detectors are shownin figure 25.Automatic Vol· The elements of an automaticume Contro I volume control (a. v. c.) sy s·tern are shown in figure 26.A dual-diode tube is used as a combinationdiode detector and a·v·c rectifier. The left·hand diode operates as a simple rectifier inthe manner described earlier in this chapter.Audio voltage, superimposed on a d·c voltage,appears across the 500,000-ohm potentiometer(the volume control) and the .0001-f.Lfd. capa·citor, and is passed on to the audio amplifier.The right-hand diode receives signal voltagedirectly from the primary of the last i·f ampli·fier, and acts as the a·v·c rectifier. The pul·sating d-e voltage across the 1-megohm a. v.c.diode load resistor is filtered by a 500,000-ohmresistor and a .05-f.Lfd. capacitor, and appliedas bias to the grids of the r·f and i·f amplifiertubes; an increase or decrease in signalstrength will cause a corresponding increaseor decrease in a·v-c bias voltage, and thus thegain of the receiver is automatically adjustedto compensate for changes in signal strength.
226 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0A. C Loading of By disassociating the a. v. c.Second Detector and detecting f u n c t i on sthrough using separatediodes, as shown, most of the ill effects ofa-c shunt loading on the detector diode areavoided. This type of loading causes seriousdistortion, and the additional components re·quired to eliminate it are well worth their cost.Even with the circuit shown, a-c loading canoccur unless a very high (5 megohms, or more)value of grid resistor is used in the followingaudio amplifier stage.RF IF IFA.V.C. inB- F-0- EquippedReceiversIn receivers having a beat·frequency oscillator for thereception of radiotelegraphsignals, the use of a.v.c.can result in a great loss in sensitivity whenthe b.f.o. is switched on. This is because thebeat oscillator output acts exactly like astrong received signal, and causes the a-v-ecircuit to put high bias on the r-f and i-f stages,thus greatly reducing the receiver's sensitivity.Due to the above effect, it is necessary toprovide a method of making the a-v-e circuitinoperative when the b.f.o. is being used. Thesimplest method of eliminating the a-v-e actionis to short the a-v-e line to ground whenthe b.f.o. is turned on. A two-circuit switchmay be used for the dual purpose of turning onthe beat oscillator and shorting out the a.v.c.if desired.®©+B 470Signal StrengthIndicatorsVisual means for determiningwhether or not the receiver isproperly tuned, as well as anindication of the relative signal strength, areboth provided by means of tuning indicators(S meters) of the meter or vacuum-tube type.A d-e milliammeter can be connected in theplate supply circuit of one or more r-f or i-famplifiers, as shown in figure 27 A, so that thechange in plate current, due to the action ofthe a-v-e voltage, will be indicated on the instrument.The d-e instrument MA should havea full-scale reading approximately equal to thetotal plate current taken by the stage or stageswhose plate current passes through the instrument.The value of this current can be estimatedby assuming a plate current on eachstage (with no signal input to the receiver) ofabout 6 ma. However, it will be found to bemore satisfactory to measure the actual platecurrent on the stages with a milliammeter ofperhaps 0-100 ma. full scale before purchasingan instrument for use as an S meter. The50-ohm potentiometer shown in the drawing isused to adjust the meter reading to full scalewith no signal input to the receiver.When an ordinary meter is used in the platecircuit of a stage, for the purpose of indicatingsignal strength, the meter reads backwardsFigure 27SIGNAL-STRENGTH-METER CIRCUITSShown above are four circuits for obtaining a sig·nat-strength reading which is a function of incomingcarrier amplitucle. The circuits ore rJ/scussecllin the accompanying text.with respect to srrength. This is because increaseda-v-e bias on stronger signals causeslower plate current through the meter. For thisreason, special meters which indicate zero atthe right-hand end of the scale are often usedfor signal strength indicators in commercialreceivers using this type of circuit. Alternatively,the meter may be mounted upsidedown, so that the needle moves toward theright with increased strength.The circuit of figure 27B can frequently beused to advantage in a receiver where the cathodeof one of the r-f or i-f amplifier stagesruns directly to ground through the cathodebias resistor instead of running through a cath-
<strong>HANDBOOK</strong>Noise Suppression 227ode-voltage gain control. In this case a 0-1d-e milliammeter in conjunction with a resistorfrom 1000 to 3000 ohms can be used as shownas a signal-strength meter. With this circuitthe meter will read backwards with increasingsignal strength as in the circuit previous! ydiscussed.Figure 27C is the circuit of a forward-readingSmeter as is often used in communicationsreceivers. The instrument is used in an unbalancedbridge circuit with the d-e plate resistanceof one i-f tube as one leg of thebridge and with resistors for the other threelegs. The value of the resistor R must be determinedby trial and error and will be somewherein the vicinity of 50,000 ohms. Sometimesthe screen circuits of the r-f and i-fstages are taken from this point along withthe screen-circuit voltage divider.Electron-ray tubes (sometimes called "magiceyes") can also be used as indicators of relativesignal strength in a circuit similar to thatshown in figure 270. A 6U5/6G5 tube shouldbe used where the a-v-e voltage will be from5 to 20 volts and a type 6E5 tube should beused when the a-v-e voltage will run from 2to 8 volts.Audio Amp I ifiers Audio amp I if i e r s are employedin n e a r I y all radioreceivers. The audio amplifier stage or stagesare usually of the Class A type, although ClassAB push-pull stages are used in some receivers.The purpose of the audio amplifier isto bring the reJatively weak signal from thedetector up to a strength sufficient to operatea pair of headphones or a loud speaker. Eithertriodes, pentodes, or beam tetrodes may beused, the pentodes and beam tetrodes usuallygiving greater output. In some receivers, particularlythose employing grid leak detection,it is possible to operate the headphones directlyfrom the detector, without audio amplification.In such receivers, a single audiostage with a beam tetrode or pentode tube isordinarily used to drive the loud speaker.Most communications receivers, either homeconstructedor factory-made, have a singleendedbeam tetrode (such as a 6L6 or 6V6) orpentode (6F6 or 6K6-GT) in the audio outputstage feeding the loudspeaker. If precautionsare not taken such a stage will actually bringabout a decrease in the effective signal-tonoiseratio of the receiver due to the risinghigh-frequency characteristic of such a stagewhen feeding a loud-speaker. One way of improvingthis condition is to place a mica orpaper capacitor of approximately 0.003 11fd.capacitance across the primary of the outputtransformer. The use of a capacitor in thismanner tends to make the load impedance seenby the plate of the output tube more constantover the audio-frequency range. The speakerand transformer will tend to present a risingi~pedance to the tube as the frequency increases,and the parallel capacitor will tendto make the total impedance more constantsince it will tend to present a decreasing impedancewith increasing audio frequency.A still better way of improving the frequencycharacteristic of the output stage, and at thesame time reducing the harmonic distortion,is to use shunt feedback from the plate of theoutput tube to the plate of a tube such as a6SJ7 acting as an audio amplifier stage aheadof the output stage.12-9 Noise SuppressionThe problem of noise suppression confrontsthe listener who is located in places whereinterference from power lines, electrical appliances,and automobile ignition systems istroublesome. This noise is often of such intensityas to swamp out signals from desiredstations.There are two principal methods for reducingthis noise:(1) A-c llne filters at the source of interference,if the noise is created by anelectrical appliance.(2) Noise-limiting circuits for the reduction,in the receiver itself, of interferenceof the type caused by automobileignition systems.Power LineFiltersMany household appliances, suchas electric mixers, heating pads,vacuum sweepers, refrigerators,oil burners, sewing machines, doorbells, etc.,create an interference of an intermittent nature.The insertion of a line filter near thesource of interference often will effect a completecure. Filters for small appliances canconsist of a 0.1-11fd. capacitor connected across the 110-volt a-c line. Two capacitorsin series across the line, with the midpointconnected to ground, can be used in conjunctionwith ultraviolet ray machines, refrigerators,oil burner furnaces, and other more stubbornoffenders. In severe cases of interference,additional filters in the form of heavydutyr-f choke coils must be connected inseries with the 110-volt a-c line on both sidesof the line right at the interfering appliance.Peak Noise Numerous noise-limiting circuitsLimiters which are beneficial in overcomingkey clicks, automobile ignitioninterference, and similar noise impulseshave become popular. They operate on theprinciple that each individual noise pulse is
228 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0of very short duration, yet of very high amplitude.The popping or clicking type of noisefrom electrical ignition systems may producea signal having a peak value ten to twentytimes as great as the incoming radio signal,but an average power much less than the sig·nal.As the duration of this type of noise peakis short, the receiver can be made inoperativeduring the noise pulse without the human eardetecting the total loss of signal. Some noiselimiters actually punch a hole in the signal,while others merely limit the maximum peaksignal which reaches the headphones or loudspeaker.The noise peak is of such short durationthat it would not be objectionable except forthe fact that it produces an over-loading effecton the receiver, which increases its time con·stant. A sharp voltage peak will give a kickto the diaphragm of the headphones or speak·er, and the momenrum or inertia keeps thediaphragm in motion until the dampening ofthe diaphragm stops it. This movement pro·duces a popping sound which may completelyobliterate the desired signal. If the noise pulsecan be limited to a peak amplitude equal tothat of the desired signal, the resulting inter·ference is practically negligible for moder·atel y low repetition rates, such as ignitionnoise.In addition, the i·f amplifier of the receiverwill also tend to lengthen the duration of thenoise pulses because the relatively high·Q i·fruned circuits will ring or oscillate when ex·cited by a sharp pulse, such as produced byignition noise. The most effective noise limiterwould be placed before the high·Q i·f tunedcircuits. At this point the noise pulse is thesharpest and has not been degraded by pass·age through the i·f transformers. In addition,the pulse is eliminated before it can produceringing effects in the i·f chain.The LambNoise LimiterAn i·f noise limiter is shown infigure 28. This is an adapta·tion of the Lamb noise silencercircuit. The i·f signal is fed into a doublegrid tube, such as a 6L 7, and thence into thei-f chain. A 6AB7 high gain pentode is capa·city coupled to the input of the i-f system.This auxiliary rube amplifies both signal andnoise that is fed to it. It has a minimum ofselectivity ahead of it so that it receives thetrue noise pulse before it is degraded by thei·f strip. A broadly runed i·f transformer is usedto couple the noise amplifier to a 6H6 noiserectifier. The gain of the noise amplifier iscontrolled by a potentiometer in the cathodeof the 6AB7 noise amplifier. This potentiometercontrols the gain of the noise amplifier1ST OET.6AB7ISTI.f,6L76H6Figure 28<strong>THE</strong> LAMB 1-F NOISE SILENCER2ND l,f,stage and in addition sets the bias level onthe 6H6 diode so that the incoming signal willnot be rectified. Only noise peaks louder thanthe signal can overcome the resting bias ofthe 6H6 and cause it to conduct. A noise pulserectified by the 6H6 is applied as a negativevoltage to the control grid of the 6L 7 i·f tube,disabling the tube, and punching a hole in thesignal at the instant of the noise pulse. Byvarying the bias control of the noise limiter,the negative control voltage applied to the6L 7 may be adjusted until it is barely sufficientto overcome the noise impulses appliedto the It 1 control grid without allowing themodulation peaks of the carrier to becomebadly distorted.The BishopNoise LimiterAnother effective i-f noiselimiter is the Bishop limiter.This is a full-wave shunt typediode limiter applied to the primary of the lasti·f transformer of a receiver. The limiter isself-biased and automatically adjusts itselfto the degree of modulation of the receivedsignal. The schematic of this limiter is shownin figure 29. The bias circuit time constant isdetermined by C, and the shunt resistance,which consists of R, and R 2in series. Theplate resistance of the last i-f tube and thecapacity of C 1 determine the charging rate ofthe circuit. The limiter is disabled by openingS, which allows the bias to rise to the valueof the i-f signal.
<strong>HANDBOOK</strong>Noise Limiters 229ter pe~ noise suppression than a standardcommumcations receiver having an i-f bandwidthof perhaps 8 kc. Likewise, when a crystalfilter is used on the "sharp" position ana-f peak" limiter is of little benefit.Audio NoiseLimitersFigure 29<strong>THE</strong> BISHOP 1-F NOISE LIMITERSome of the simplest and mostpractical peak limiters for radio·telephone reception employ oneor two diodes either as shunt or series limitersin the audio system of the receiver. When anoise pulse exceeds a certain predeterminedthreshold value, the limiter diode acts eitheras a short or open circuit, depending uponwhether it is used in a shunt or series circuit.The threshold is made to occur at a level highenough that it will not clip modulation peaksenough to impair voice intelligibility, but lowenough to limit the noise peaks effectively.Because the action of the peak limiter isneeded most on very weak signals, and theseusually are not strong enough to produce propera·v·c action, a threshold setting that is correctfor a strong phone signal is not correctfor optimum limiting on very weak signals. Forthis reason the threshold control often is tiedin with the a-v-e system so as to make theoptimum threshold adjustment automatic insteadof manual.Suppression of impulse noise by means ofan audio peak limiter is best accomplished atthe very front end of the audio system, andfor this reason the function of superheterodynesecond detector and limiter often are combinedin a composite circuit.The amount of limiting that can be obtainedis a function of the audio distortion that canbe tolerated. Because excessive distortionwill reduce the intelligibility as much as willbackground noise, the degree of limiting forwhich the circuit is designed has to be a compromise.Peak noise limiters working at the seconddetector are much more effective when the i-fbandwidth of the receiver is broad, because asharp i-f amplifier will lengthen the pulses bythe time they reach the second detector, makingthe limiter less effective. V-h-f superheterodyneshave an i·f bandwidth considerablywider than the minimum necessary forvoice sidebands (to take care of drift and instability).Therefore, they are capable of bet-PracticalPeak NoiseLimiter CircuitsNoise limiters range all theway from an audio stagerunning at very low screenor plate voltage, to elaborateaffairs employing 5 or more tubes. Ratherthan attempt to show the numerous types, manyof which are quite complex considering theresults obtained, only two very similar typeswill be described. Either is just about as effectiveas the most elaborate limiter that canbe constructed, yet requires the addition ofbut a single diode and a few resistors andcapacitors over what would be employed in agood superheterodyne without a limiter. Bothcircuits, with but minor modifications in resistanceand capacitance values, are incorporatedin one form or another in differenttypes of factory-built communications receivers.Referring to figure 30, the first circuit showsa conventional superheterodyne second detector,a.v.c., and first audio stage with theaddition of one tube element, D 3 ,which maybe either a separate diode or part of a twindiodeas illustrated. Diode D 3 acts as a seriesgate, allowing audio to get to the grid of thea-f tube only so long as the diode is conduct·ing. The diode is biased by a d-e voltage obtainedin the same manner as a-v-e control voltage,the bias being such that pulses of shortduration no longer conduct when the pulsevoltage exceeds the carrier by approximately60 per cent. This also clips voice modulationpeaks, but not enough to impair intelligibility.It is apparent that the series diode clipsonly positive modulation peaks, by limitingupward modulation to about 60 per cent. Negativeor downward peaks are limited automaticallyto 100 per cent in the detector, becauseobviously the rectified voltage out of thediode detector cannot be less than zero. Limitingthe downward peaks to 60 per cent or soinstead of 100 per cent would result in butlittle improvement in noise reduction, and theresults do not justify the additional compon·ents required.It is important that the exact resistancevalues shown be used, for best results, andthat 10 per cent tolerance resistors be usedfor R 3 and R 4 • Also, the rectified carrier voltagedeveloped across C 3 should be at least 5volts for good limiting.The limiter will work well on c-w telegraphyif the amplitude of beat frequency oscillatorinjection is not too high. Variable injection isto be preferred, adjustable from the front panel.
23 0 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0C, -0.1-).Lfd. paperC.,-50-).L).Lfd. micaC,-1 00-).L).Lfd. micaC., Cs -0.01-).Lfd. paperR 1 , R 2 -1 megohm,Y2 wattR 3 , R.-220,000 ohms,Yz wattRs, R,;-1 megohm,Y2 wattR,-2-megohm paten·tiometerFigure 30NOISE LIMITER CIRCUIT, WITH ASSOCIATED A-V-CThis I imiter is of the series type, one/ is seff-acljusting to carrier strength for phone reception. For properoperatio'l several volts shoulcl be clevelopecl across the seconclary of the last i-f transformer (/FT) unclercarrier conditions.If this feature is not provided, the b·f·o injec·tion should be reduced to the lowest value thatwill give a satisfactory beat. When this isdone, effective limiting and a good beat canbe obtained by prope£ adjustment of the r·fand a-f gain controls. It is assumed, of course,that the a. v.c. is cut out of the circuit forc-w telegraphy reception.AlternativeLimiter CircuitThe c i r cui t of figure 31 ismore e ff e c t i v e than thatshown in figure 30 under certainconditions and requires the addition ofonly one more resistor and one more capacitorthan the other circuit. Also, this circuitinvolves a smaller loss in output level thanthe circuit of figure 30. This circuit can beused with equal effectiveness with a combineddiode-triode or diode-pentode tube (6R7,6SR 7, 6Q7, 6SQ7 or similar diode-triodes, or688, 6SF7, or similar diode-pentodes) as diodedetector and first audio stage. However, aseparate diode must be used for the noiselimiter, D 2 • This diode may be one-half of a6H6, 6AL5, 7A6, etc., or it may be a triodeconnected 6]5, 6C4 or similar type.Note that the return for the volume controlmust be made to the cathode of the detectordiode (and not to ground) when a dual tube isused as combined second-detector first-audio.This means that in the circuit shown in figure31 a connection will exist across the pointswhere the "X" is shown on the diagram sincea common cathode lead is brought out of thetube for D, and V,. If desired, of course, asingle dual diode may be used for D, and D 2in this circuit as well as in the circuit of figure30. Switching the limiter in and out withthe switch S brings about no change in volume.In any diode limiter circuit such as the onesshown in these two figures it is important thatthe mid-point of the heater potential for thenoise-limiter diode be as close to groundpotential as possible. This means that thecenter-tap of the heater supply for the tubesshould be grounded wherever possible ratherthan grounding one side of the heater supplyas is often done. Difficulty with hum pickupin the limiter circuit may be encountered whenone side of the heater is grounded due to thehigh values of resistance necessary in thelimiter circuit.The circuit of figure 31 has been used withexcellent success in several home-constructedreceivers, and in the BC-312/BC-342 and BC-348 series of surplus communications receivers.It is also used in certain manufacturedreceivers.An excellent check on the operation of thenoise limiter in any communications receivercan be obtained by listening to the Loran signalsin the 160-meter band. With the limiterout a sharp rasping buzz will be obtainedwhen one of these stations is tuned in. Withthe noise limiter switched into the circuit thebuzz should be greatly reduced and a lowpitchedhum should be heard.The Full-WaveLimiterThe most satisafctory diodenoise limiter is the series fullwavelimiter, shown in figure32. The positive noise peaks are clipped bydiode A, the clipping level of which may beadjusted to clip at any modulation level between25 per cent and 100 per cent. The negaativenoise peaks are clipped by diode B ata fixed level.The TNS Limiter The Twin Noise Squelch,popularized by CQ magazine,is a combination of a diode noise clipperand an audio squelch tube. The squelch cir-
<strong>HANDBOOK</strong>U-H-F Circuits 2 31This circuit is of the self·ad;usting type and gives lessdistortion for a given clegreeof modulation than the morecommon I imiter circuits.R 1 , R 2 -470K, Y, wattR 3 -100K, Y, wattR.., R 5 -1 megohm, J.-'2 watt~-2-megohm potentiometerc,-0.00025 mica (apprax.)C:,-0.01-JLfd. paperC,-0.01-J.Lfd. paperc.-0.01-J.Lfd. paper0 1 , D 2 -6H6, 6AL5, 7A6, ordiode sections of a 658-GTL __________ .J Iv,Figure 31ALTERNATIVE NOISE LIMITER CIRCUITcuir is useful in eliminating the grinding backgroundnoise that is the residual left by thediode clipper. In figure 33, the setting of the470K potentiometer determines the operatinglevel of the squelch action and should be setto eliminate the residual background noise.Because of the low inherent distortion of theTNS, it may be left in the circuit at all times.As with other limiters, the TNS requires ahigh signal level at the second detector formaximum limiting effect.12-10 Special Considerationsin U-H-F Receiver DesignTransmissionLine CircuitsAt increasingly higher frequen·cies, it becomes progressivelymore difficult to obtain a satis·factory amount of selectivity and impedancefrom an ordinary coil and capacitor used as aresonant circuit. On the other hand, quarterwavelength sections of parallel conductors orconcentric transmission line are nor on! y moreefficient but also become of practical dimen·sions.TuningShort LinesTubes and tuning capacitors con·nected to the open end of a trans·mission line provide a capacitancethat makes the resonant length less thana quarter wave-length. The amount of shorten·ing for a specified capacitive reactance isdetermined by the surge impedance of the lineJ.F. STAGEAUDIOFigure 32<strong>THE</strong> FULL-WAVE SERIES AUDIONOISE LIMITERFigure 33<strong>THE</strong> TNS AUDIO NOISE LIMITER
2 32 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0II IICAVITY®CAVITY0~-~~~INETO ANT.II->I-olII TO ANT. I Iu@ ®Figure 34COUPLING AN ANTENNA TO ACOAXIAL·RESONANT CIRCUIT(A) shows the recommenclecl methocl for coup/in!la coaxial line to a coaxial resonant circuit. (BJshows an alternative methocl for use with an openwiretype of antenna feeclline.uHOLE@6itt2ELECTRONBEAMFigure 35METHODS OF EXCITING A RESONANTCAVITYsection. It is given by the equation for resonance:12 rr fC= Z 0tan lin which rr = 3.1416, I is the frequency, C thecapacitance, Z 0the surge impedance of theline, and tan l is the tangent of the electricallength in degrees.The capacitive reactance of the capacitanceacross the end is 1/(2rr 1 C) ohms. For resonance,this must equal the surge impedance ofthe line times the tangent of its electricallength (in degrees, where 90° equals a quarterwave). It will be seen that twice the capacitancewill resonate a line if its surge impedanceis halved; also that a given capacitancehas twice the loading effect when the frequencyis doubled.Coupling IntoLines andCoaxial CircuitsIt is possible to couple intoa parallel-rod line by tappingdirectly on one or bothrods, preferably throughblocking capacitors if any d.c. is present.More commonly, however, a hairpin is inductivelycoupled at the shorting bar end, eitherto the bar or to the two rods, or both. Thisnormally will result in a balanced load. Shoulda loop unbalanced to ground be coupled in,any resulting unbalance reflected into the rodscan be reduced with a simple Faraday screen,made of a few parallel wires placed betweenthe hairpin loop and the rods. These shouldbe soldered at only one end and grounded.An unbalanced tap on a coaxial resonantcircuit can be made directly on the inner conductorat the point where it is properly matched(figure 34). For low impedances, such as a concentricline feeder, a small one-half turn loopcan be inserted through a hole in the outerconductor of the coaxial circuit, being in effecta half of the hairpin type recommendedfor coupling balanced feeders to coaxial resonantlines. The size of the loop and closenessto the inner conductor determines theimpedance matching and loading. Such loopscoupled in near the shorting disc do not alterthe tuning appreciably, if not overcoupled.ResonantCavitiesA cavity is a closed resonantchamber made of metal. It isknown also as a rhumbatron. Thecavity, having both inductance and capacitance,supersedes coil-capacitor and capacitance-loadedtransmission-line tuned circuitsat extremely high frequencies where conventionalL and C components, of even the mostrefined design, prove impractical because ofthe tiny electrical and physical dimensionsthey must have. Microwave cavities have highQ factors and are superior to conventionaltuned circuits. They may be employed in themanner of an absorption wavemeter or as thetuned circuit in other r-f test instruments, andin microwave transmitters and receivers.Resonant cavities usually are closed on allsides and all of their walls are made of electricalconductor. However, in some forms,small openings are present for the purpose ofexcitation. C~vities have been produced inseveral shape~ including the plain sphere,
<strong>HANDBOOK</strong>U-H-F Circuits 233cTUNINGSLUGS0®Figure 36TUNING METHODS FOR CYLINDRICALRESONANT CAVITIESdimpled sphere, sphere with reentrant conesof various sorts, cylinder, prism (includingcube), ellipsoid, ellipsoid-hyperboloid, doughnut-shape,and various reentrant types. In appearance,they resemble in their simpler formsmetal boxes or cans.The cavity actually is a linear circuit, butone which is superior to a conventional coaxialresonator in the s-h-f range. The cavityresonates in much the same manner as does abarrel or a closed room with reflecting walls.Because electromagnetic energy, and theassociated electrostatic energy, oscillates toand fro inside them in one mode or another,resonant cavities resemble wave guides. Themode of operation in a cavity is affected bythe manner in which micro-wave energy is injected.A cavity will resonate to a large numberof frequencies, each being associated witha particular mode or standing-wave pattern.The lowest mode (lowest frequency of operation)of a cavity resonator normally is the oneused.The resonant frequency of a cavity may bevaried, if desired, by means of movable plungersor plugs, as shown in figure 36A, or amovable metal disc (see figure 36B). A cavitythat is too small for a given wavelength willnot oscillate.The resonant frequencies of simple spherical,cylindrical, and cubical cavities may becalculated simply for one particular mode.Wavelength and cavity dimensions (in centimeters)are related by the following simpleresonance formulae:For Cylinder Ar = 2.6 x radius" Cube .\, = 2.83 x half of 1 sideSphere .\, = 2. 28 x radiusButterflyCircuitUnlike the cavity resonator, whichin its conventional form is a devicewhich can tune over a relativelynarrow band, the butterfly circuit is a tunableresonator which permits coverage of a fairly®@Figure 37<strong>THE</strong> BUTTERFLY RESONANT CIRCUITShown at (A) is the physical appearance of thebutterfly circuit as usecl in the v-h-f ancl loweru-h-f range. (B) shows an electrical representa·tion of the circuit.wide u-h-f band. The butterfly circuit is verysimilar to a conventional coil-variable capacitorcombination, except that both inductanceand capacitance are provided by what appearsto be a variable capacitor alone. The Q of thisdevice is somewhat less than that of a concentric-linetuned circuit but is entirely adequatefor numerous applications.Figure 37 A shows construction of a singlebutterfly section. The butterfly-shaped rotor,from which the device derives its name, turnsin relation to the unconventional stator. Thetwo groups of stator "fins" or sectors are ineffect joined together by a semi-circular metalband, integral with the sectors, which providesthe circuit inductance. When the rotor is setto fill the loop opening (the position in whichit is shown in figure 37 A), the circuit inductanceand capacitance are reduced to minimum.When the rotor occupies the position indicatedby the dotted lines, the inductance and capacitanceare at maximum. The tuning range ofpractical butterfly circuits is in the ratio of1.5:1 to 3.5:1.Direct circuit connections may be made topoints A and B. If balanced operation is desired,either point C or D will provide theelectrical mid-point. Coupling may be effectedby means of a small singleturn loop placednear point E or F. The butterfly thus permitscontinuous variation of both capacitance andinductance, as indicated by the equivalentcircuit in figure 37B, while at the same timeeliminating all pigtails and wiping contacts.Several butterfly sections may be stackedin parallel in the same way that variable capacitorsare built up. In stacking these sections,the effect of adding inductances in parallelis to lower the total circuit inductance,while the addition of stators end rotors raisesthe total capacitance, as well as the ratio ofmaximum to minimum capacitance.
234 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0Butterfly circuits have been applied specificallyto oscillators for transmitters, super·heterodyne receivers, and heterodyne frequen·cy meters in the 100-1000-Mc. frequency range.ReceiverCircuitsThe types of resonant circuits de·scribed in the previous paragraphshave largely replaced conventionalcoil-capacitor circuits in the range above 100Me. Tuned short lines and butterfly circuitsare used in the range from about 100 Me. toperhaps 3500 Me., and above about 3500 Me.resonant cavities are used almost exclusively.The resonant cavity is also quite generallyemployed in the 2000-Mc. to 3500-Mc. range.In a properly designed receiver, thermalagitation in the first tuned circuit is amplifiedby subsequent tubes and predominates in theoutput. For good signal-to-set-noise ratio,therefore, one must strive for a high· gain lownoiser·f stage. Hiss can be held down by givingcareful attention to this point. A mixerhas about 0.3 of the gain of an r-f tube of thesame type; so it is advisable to precede amixer by an efficient r-f stage. It is also ofsome value to have good r-f selectivity beforethe first detector in ordelr to reduce noisesproduced by beating noise at one frequencyagainst noise at another, to produce noise atthe intermediate frequency in a superheterodyne.The frequency limit of a tube is reachedwhen the shortest possible external connec·tions are used as the tuned circuit, except forabnormal types of oscillation. Wires or sizeablecomponents are often best considered assections of transmission lines rather than assimple resistances, capacitances, or induct·ances.So long as small triodes and pentodes willoperate normally, they are generally preferredas v-h·f tubes over other receiving methodsthat have been devised. However, the inputcapacitance, input conductance, and transittime of these tubes limit the upper frequencyat which they may be operated. The input resistance,which drops to a low value at veryshort wave-lengths, limits the stage gain andbroadens the tuning.Y·H-FTubesThe first tube in a v-h-f receiver ismost important in raising the signalabove the noise generated in successivestages, for which reason small v·h-f typesare definitely preferred.Tubes employing the conventional grid-controlledand diode rectifier principles have beenmodernized, through various expedients, foroperation at frequencies as high, in some newtypes, as 4000 Me. Beyond that frequency,electron transit time becomes the limiting factorand new principles must be enlisted. Ingeneral, the improvements embodied in existingtubes have consisted of (1) reducing electrodespacing to cut down electron transittime, (2) reducing electrode areas to decreaseinterelectrode capacitances, and (3) shorteningof electrode leads either by mounting theelectrode assembly close to the tube base orby bringing the leads out directly through theglass envelope at nearby points. Through reductionof lead inductance and interelectrodecapacitances, input and output resonant fre·quencies due to tube construction have beet,increased substantially.Tubes embracing one or more of the featuresjust outlined include the later loctaltypes, high-frequency acorns, button-basetypes, and the lighthouse types. Type 6]4button-base triode will reach 500 Me. Type6F4 acorn triode is recommended for use up to1200 Me. Type 1A3 button-base diode has aresonant frequency of 1000 Me., while type9005 acorn diode resonates at 1500 Me. Lighthousetype 2C40 can be used at frequenciesup to 3500 Me. as an oscillator.CrystalRectifiersMore than two decades havepassed since the crystal (mineral)rectifier enjoyed widespread usein radio receivers. Low-priced tubes completelysupplanted the fragile and relatively insensitivecrystal detector, although it did continuefor a few years as a simple meter rectifierin absorption wavemeters after its demiseas a receiver component.Today, the crystal detector is of new importancein microwave communication. It isbeing employed as a detector and as a mixerin receivers and test instruments used at extremelyhigh radio frequencies. At some ofthe frequencies employed in microwave operations,the crystal rectifier is the only satisfactorydetector or mixer. The chief advantagesof the crystal rectifier are very low capacitance,relative freedom from transit-timedifficulties, and its two-terminal nature. Nobatteries or a·c power supply are required forits operation.The crystal detector consists essentially ofa small piece of silicon or germanium mountedin a base of low-melting-point alloy andcontacted by means of a thin, springy feelerwire known as the cat whisker. This arrangementis shown in figure 38A.The complex physics of crystal rectificationis beyond the scope of this discussion. It issufficient to state that current flows from severalhundred to several thousand times morereadily in one direction through the contact ofcat whisker and crystal than in the oppositedirection. Consequently, an alternating current(including one of microwave frequency) will
<strong>HANDBOOK</strong>Receiver Adjustment 235Figure 381N23 MICROWAVE-TYPECRYSTAL DIODEBRASS BASE CONNECTORA small silicon crystal is attached tothe base connector and a fine "catwhisker"wire is set to the most sensitivespot on the crystal. After adjust·ment the ceramic shell is filled withcompound to hold the contact wire inposition. Crystals of this type are usedto over 3 0, 000 me.be rectified by the crystal detector. The load,through which the rectified currents flow, maybe connected in series or shunt with the crys·tal, although the former connection is mostgenerally employed.The basic arrangement of a modern fixedcrystal detector developed during World War IIfor microwave work, particularly radar, isshown in figure 388. Once the cat whisker ofthis unit is set at the factory to the most sen·sitive spot on the surface of the silicon crys·tal and its pressure is adjusted, a filler compoundis injected through the filling hole tohold the cat whisker permanently in position.12-11 Receiver AdjustmentA simple regenerative receiver requireslittle adjustment other than that necessaryto insure correct tuning and smooth regenerationover some desired range. Receivers ofthe tuned radio-frequency type and superheterodynesrequire precise alignment to obtain thehighest possible degree of selectivity andsensitivity.Good results can be obtained from a receiv·er only when it is properly aligned and adjusted.The most practical technique for mak·ing these adjustments is given below.Instruments A very small number of instru·ments will suffice to check andalign a communications receiver, the most importantof these testing units being a modulatedoscillator and a d-e and a-c voltmeter.The meters are essential in checking the voltageapplied at each circuit point from the powersupply. If the a-c voltmeter is of the oxiderectifiertype, it can be used, in addition, asan output meter when connected across thereceiver output when tuning to a modulatedsignal. If the signal is a steady tone, such asfrom a test oscillator, the output meter willindicate the value of the detected signal. Inthis manner, alignment results may be visuallynoted on the meter.T-R-F ReceiverAiignmentAlignment procedure in a multistaget-r-f receiver is exact·ly the same as aligning asingle stage. If the detector is regenerative,each preceding stage is successively alignedwhile keeping the detector circuit tuned to thetest signal, the latter being a station signalor one locally generated by a test oscillatorloosely coupled to the antenna lead. Duringthese adjustments, the r-f amplifier gain controlis adjusted for maximum sensitivity, assumingthat the r-f amplifier is stable anddoes not oscillate. Often a sensitive receivercan be roughly aligned by tuning for maximumnoise pickup.SuperheterodyneAlignmentAligning a superhet is a detailedtask requiring a greatamount of care and patience.It should never be undertaken without a thoroughunderstanding of the involved job to bedone and then only when there is abundanttime to devote to the operation. There are noshort cuts; every circuit must be adjusted individuallyand accurately if the receiver is togive peak performance. The precision of eachadjustment is dependent upon the accuracywith which the preceding one was made.Superhet alignment requires (1) a good sig·nal generator (modulated oscillator) coveringthe radio and intermediate frequencies andequipped with an attenuator; (2) the necessarysocket wrenches, screwdrivers, or "neutralizingtools" to adjust the various i-f and r·ftrimmer capacitors; and (3) some convenienttype of tuning indicator, such as a copper·oxide or electronic voltmeter.Throughout the alignment process, unlessspecifically stated otherwise, the r·f gain controlmust be set for maximum output, the beatoscillator switched off, and the a.v.c. turnedoff or shorted out. \1/'hen the signal output ofthe receiver is excessive, either the attenuatoror the a-f gain control may be turned down, butnever the r-f gain control.1- F AI ignment After the receiver has beengiven a rigid e 1 e c t ric a 1 andmechanical inspection, and any faults whichmay have been found in wiring or the selec·tion and assembly of parts corrected, the i-famplifier may be aligned as the first step inthe checking operations.With the signal generator set to give a modulatedsignal on the frequency at which the i-f
236 Radio Receiver Fundamentals <strong>THE</strong> R AD I 0amplifier is to operate, dip the "hot" outputlead from the generator to the last i-f stagethrough a small fixed capacitor to the controlgrid. Adjust both trimmer capacitors in thelast i-f transformer (the one between the lasti-f amplifier and the second detector) to resonanceas indicated by maximum deflection ofthe output meter.Each i-f stage is adjusted in the same manner,moving the hot lead, stage by stage, backtoward the front end of the receiver and backingoff the attenuator as the signal strengthincreases in each new position. The last adjustmentwill be made to the first i-f transformer,with the hot signal generator lead connectedto the control grid of the mixer. Occasionallyit is necessary to disconnect themixer grid lead from the coil, grounding itthrough a 1,000- or 5,000-ohm resistor, andcoupling the signal generator through a smallcapacitor to the grid.When the last i-f adjustment has been completed,it is good practice to go back throughthe i-f channel, re-peaking all of the transformers.It is imperative that this recheck bemade in sets which do not include a crystalfilter, and where the simple alignment of thei-f amplifier to the generator is final.1- F with There are several ways of alignCrystal Filter ing an i-f channel which containsa crystal-filter circuit.However, the following method is one whichhas been found to give satisfactory results inevery case: An unmodulated signal generatorcapable of tuning to the frequency of the filtercrystal in the receiver is coupled to the grid ofthe stage which precedes the crystal filter inthe receiver. Then, with the crystal filterswitched in, the signal generator is tunedslowly to find the frequency where the crystalpeaks. The receiver "S" meter may be usedas the indicator, and the sound heard from theloudspeaker will be of assistance in findingthe point. \\"hen the frequency at which thecrystal peaks has been found, all the i-f transformersin the receiver should be touched upto peak at that frequency.B- F-0 Adiustment Adjusting the beat oscillatorona receiver that hasno front panel adjustment is relatively simple.It is only necessary to tune the receiver toresonance with any signal, as indicated bythe tuning indicator, and then turn on the b.f.o.and set its trimmer (or trimmers) to producethe desired beat note. Setting the beat oscillatorin this way will result in the beat notebeing stronger on one "side" of the signalthan on the other, which is what is desiredfor c-w reception. The b.f.o. should not be setto zero beat when the receiver is tuned to1-F SIGNAL IN------r----+1-F SICNAL OUTFigure 39<strong>THE</strong> Q-MUL TIPLIER1-F PLUS Q MULTIPLIERThe loss resistance of a high-Q circuitis neutralizecl by regeneration in asimple feedback amplifier. A highlyselective passband is pracluced whichis coupled to the i-f circuit of thereceiver.resonance with the signal, as this will causean equally strong beat to be obtained on bothsides of resonance.Front-End Alignment of the front end of aAlignment home-constructed receiver is arelatively simple process, consistingof first getting the oscillator to coverthe desired frequency range and then of peakingthe various r-f ci~cuits for maximum gain.However, if the frequency range covered bythe receiver is very wide a fair amount of cutand try will be required to obtain satisfactorytracking between the r-f circuits and the oscillator.Manufactured communications receiversshould always be tuned in accordance withthe instructions given in the maintenance manualfor the receiver.12-12 Receiving AccessoriesThe Q-Multiplier The selectivity of a receivermay be increased by raisingthe Q of the tuned circuits of the i-f strip. Asimple way to accomplish this is to add a controlledamount of positive feedback to a tunedcircuit, thus increasing its Q. This is done inthe 0-multiplier whose basic circuit is shownin flgure 39- The circuit L-Cl-C2 is tuned to
<strong>HANDBOOK</strong>Receiving Accessories 237TO PLATE TERMINALOF FIRST 1-F TUBE THRU2.' OF COAXIAL LINE.ooo L1 ~P~o:5- .Q MULTIPLIER "NULL''i 12AX710K5W.,4,Bfill+g+200-30QV_ -= 6.3V.4~5 KCFR EQU ENC.YL1=GRAYBURNE Ve CHOKE (0.6-6.0 MH)L2.=GRAY8URN£ "LOOPSTICK" COILFigure 40Q-MUL TIPLIER NULL CIRCUITThe addition of a second triode permitsthe Q-Multiplier to be used for nullingout an unwanted hetrodyne.the intermediate frequency, and the loss resistanceof the circuit is neutralized by thepositive feedback circuit composed of C3 andthe vacuum tube. Too great a degree of positivefeedback will cause the circuit to break intooscillation.At the resonant frequency, the impedance ofthe tuned circuit is very high, and when shuntedacross an i-f stage will have little effect uponthe signal. At frequencies removed from resonance,the impedance of the circuit is low, resultingin high attenuation of the i-f signal.The resonant frequency of the Q-multiplier mayby varied by changing the value of one of thecomponents in the tuned circuit.The Q-multiplier may also be used to "null"a signal by employing negative feedback tocontrol the plate resistance of an auxiliaryamplifier stage as shown in figure 40. Since thegrid-cathode phase shift through the Q-multiplieris zero, the plate resistance of a second tubemay be readily controlled by placing it acrossthe Q-multiplier. At resonance, the high negativefeedback drops the plate resistance ofV2, shunting the i-f circuit. Off resonance, thefeedback is reduced and the plate resistanceof V2 rises, reducing the amount of signal attenuationin the i-f strip. A circuit combiningboth the "peak" and "null" features is shownin figure 41.The Product DetectorA version of the commonmixer or converter stageFigure 41SCHEMATIC OF A 455KCQ-MUL TIPLIERCoil L7 is required to tune out thereactance of the coaxial line. It is adjustedfor maximum signal response.Ll may be omitted if the Q-multiplieris connected to the receiver with ashort length of wire, and the i-f transformerwithin the receiver is retuned.may be used as a second detector in a receiverin place of the usual diode detector. The diodeis an envelope detector (section 12-1) and developsa d-e output voltage from a single r-fsignal, and audio "beats" from two or moreinput signals. A product detector (figure 42)requires that a local carrier voltage be presentin order to produce an audio output signal.Figure 42<strong>THE</strong> PRODUCTDETECTORAudio output signal isdeveloped only whenlocal oscillator is on.AUDIO OUTPUT
23 8Radio Receiver Fundamentals20 K470T7BEAT OSCSIGNALBEAT OSC.Slt;NALFigure 43PENTAGRID MIXERUSED AS PRODUCTDETECTORSuch a detector is useful for single sidebandwork, since the inter-modulation distortion ISextremely low.A pentagrid product detector is shown infigure 43. The incoming signal is applied togrid 3 of the mixer tube, and the local oscillatoris injected on grid 1. Grid bias is adjusted foroperation over the linear portion of the tubecharacteristic curve. When grid 1 injectionis removed, the audio output from an unmodulatedsignal applied to grid 3 should be reducedapproximately 30 to 40 db below normal detectionlevel. When the frequency of the localoscillator is synchronized with the incomingcarrier, amplitude modulated signals may bereceived by exalted carrier reception, whereinthe local carrier substitutes for the transmittedcarrier of the a-m signal.Three triodes may be used as a productdetector (figure 44). Triodes V1 and V2 actas cathode followers, delivering the sidebandsignal and the local oscillator signal to agrounded grid triode (V3) which functions asthe mixer stage. A third version of the productdetector is illustrated in figure 45. A twintriode tube is used. Section V 1 functions asa cathode follower amplifier. Section V2 is aFigure 44TRIPLE-TRIODEPRODUCT DETECTORVl and V2 act as cathode followers,delivering sideband signaland local oscillator signal togrounded grid triode mixer (V3 ).100.01r-'-"----t-...-(----AU D I 0 OUT1- F S I G. ---j[ f--t-"1-· ~ :=:-+'--+--, 47 K100 KFigure 45DOUBLE-TRIODEPRODUCT DETECTOR"plate" detector, the cathode of which iscommon with the cathode follower amplifier.The local oscillator signal is injected into thegrid circuit of tube V2.II.Figure 46EXPLODED VIEW OF COLLINSMECHANICAL FILTER
CHAPTER THIRTEENGeneration ofRadio Frequency EnergyA radio communication or broadcast transmitterconsists of a source of radio frequencypower, or carrier; a system for modulating thecarrier whereby voice or telegraph keying orother modulation is superimposed upon it; andan antenna system, including feed line, forradiating the intelligence-carrying radio frequencypower. The power supply employed toconvert primary power to the various voltagesrequired by the r-f and modulator portions ofthe transmitter may also be considered partof the transmitter.Voice modulation usually is accomplishedby varying either the amplitude or the frequencyof the radio frequency carrier in accordancewith the components of intelligence to betransmitted.Radiotelegraph modulation (keying) normallyis accomplished either by interrupting,shifting the frequency of, or superimposing anaudio tone on the radio-frequency carrier inaccordance with the dots and dashes to betransmitted.The complexity of the radio-frequency generatingportion of the transmitter is dependentupon the power, order of stability, and frequencydesired. An oscillator feeding an antennadirect! y is the simplest form of radio-frequencygenerator. A modern high-frequency transmitter,on the other hand, is a very complex generator.Such an equipment usually comprises a verystable crystal-controlled or self-controlledoscillator to stabilize the output frequency, aseries of frequency multipliers, one or moreamplifier stages to increase the power up tothe level which is desired for feeding the antennasystem, and a filter system for keepingthe harmonic energy generated in the transmitterfrom being fed to the antenna system.13-1 Self-ControlledOscillatorsIn Chapter Four, it was explained that theamplifying properties of a tube having threeor more elements give it the ability to generatean alternating current of a frequency determinedby the components associated withit. A vacuum tube operated in such a circuitis called an oscillator, and its function isessentially to convert direct current into radiofrequencyalternating current of a predeterminedfrequency.Oscillators for controlling the frequency ofconventional radio transmitters can be dividedinto two general classes: self-controlled andcrystal-controlled.There are a great many types of self-controlledoscillators, each of which is best suited239
240 Generation of R-F Energy <strong>THE</strong> R AD I 0R@ SHUNT-FED HARTLEY @ SHUNT-FED COLPITTS © TUNED PLATE TUNED GRIDRR-·+B@TUNED-PLATE UNTUNED GRID ® ELECTRON COUPLED ® COLPITTS ELECTRON COUPLED-·+BL2@ CLAPP@ CLAPP ELECTRON COUPLEDFigure 1COMMON TYPES OF SELF-EXCITED OSCILLATORSFixecl capacitor ~a/ues ~re ~ypical, but will vary somewhat with the application.In the Clapp OSCillator CirCUitS (G) ancl (H), capacitors cl ancl c2 should have areactance of ~0 to 10~ ohms at t.he operating frequency of the oscillator. Tuning ofthese two osc1llators IS accompl1shecl by capacitor C. In the circuits of (E) (F) ancl(H), tuning of the tank circuit in the plate of the oscillator tube will have ~elatlvelys':'all effect on the frequency. of oscillation. The plate tank circuit also may, if cles~recl,be tunecl to a harmomc of the oscillation frequency, or a broadly resonantcircuit may be usecl In this circuit position.to a particular application. They can furtherbe subdivided into the classifications of: neg·ative-grid oscillators, electron-orbit oscillators,negative-resistance oscillators, velocitymodulation oscillators, and magnetron oscillators.Negotive-Grid A negative-grid oscillator isOscillators essentially a vacuum-tube am·plifier with a sufficient por·tion of the output energy coupled back into theinput circuit to sustain oscillation. The controlgrid is biased negatively with respect tothe cathode. Common types of negative-gridoscillators are diagrammed in figure 1.The Hartley Illustrated in figure 1 (A) is theoscillator circuit which finds themost general application at the present time;this circuit is commonly called the Hartley.The operation of this oscillator will be describedas an index to the operation of allnegative-grid oscillators; the only real differ-
<strong>HANDBOOK</strong>Oscillators 241ence between the various circuits is the mannerin which energy for excitation is coupledfrom the plate to the grid circuit.When plate voltage is applied to the Hartleyoscillator shown at (A), the sudden flow ofplate current accompanying the application ofplate voltage will cause an electro-magneticfield to be set up in the vicinity of the coil.The building-up of this field will cause a potentialdr6p to appear from turn-to-turn alongthe coil. Due to the inductive coupling betweenthe portion of the coil in which the platecurrent is flowing and the grid portion, a potentialwill be induced in the grid portion.Since the cathode tap is between the gridand plate ends of the coil, the induced gridvoltage acts in such a manner as to increasefurther the plate current to the tube. This actionwill continue for a short period of timedetermined by the inductance and capacitanceof the tuned circuit, until the flywheel effectof the tuned circuit causes this action to cometo a maximum and then to reverse itself. Theplate current then decreases, the magneticfield around the coil also decreasing, until aminimum is reached, when the action startsagain in the original direction and at a greateramplitude than before. The amplitude of theseoscillations, the frequency of which is determinedby the coil-capacitor circuit, will increasein a very short period of time to a limitdetermined by the plate voltage of the oscillatortube.The Colpitts Figure 1 (B) shows a version ofthe Colpitts oscillator. It canbe seen that this is essentially the same circuitas the Hartley except that the ratio of apair of capacitances in series determines theeffective cathode tap, instead of actually usinga tap on the tank coil. Also, the net capacitanceof these two capacitors comprisesthe tank capacitance of the tuned circuit. Thisoscillator circuit is somewhat less susceptibleto parasitic (spurious) oscillations thanthe Hartley.For best operation of the Hartley and Colpittsoscillators, the voltage from grid to cathode,determined by the tap on the coil or thesetting of the two capacitors, normally shouldbe from 1/3 to 1/5 that appearing betweenplate and cathode.The T.P. T.G. The tuned-plate tuned-grid oscillatorillustrated at (C) hasa tank circuit in both the plate and grid circuits.The feedback of energy from the plateto the grid circuits is accomplished by theplate-to-grid inter-electrode capacitance withinthe tube. The necessary phase reversal infeedback voltage is provided by tuning thegrid tank capacitor to the low side of the de-sired frequency and the plate capacitor to thehigh side. A broadly resonant coil may be substitutedfor the grid tank to form the T. N. T.oscillator shown at (D).Electron-CoupledOscillatorsIn any of the oscillator circuitsjust described it ispossible to take energy fromthe oscillator circuit by coupling an externalload to the tank circuit. Since the tank circuitdetermines the frequency of oscillation of thetube, any variations in the conditions of theexternal circuit will be coupled back into thefrequency determining portion of the oscillator.These variations will result in frequency instability.The frequency determining portion of anoscillator may be coupled to the load circuitonly by an electron stream, as illustrated in(E) and (F) of figure 1. When it is consideredthat the screen of the tube acts as the plateto the oscillator circuit, the plate merely actingas a coupler to the load, then the similaritybetween the cathode-grid-screen circuitof these oscillators and the cathode-grid-platecircuits of the corresponding prototype can beseen.The electron-coupled oscillator has goodstability with respect to load and voltage variation.Load variations have a relatively smalleffect on the frequency, since the only couplingbetween the oscillating circuit and theload is through the electron stream flowingthrough the other elements to the plate. Theplate is electrostatically shielded from theoscillating portion by the bypassed screen.The stability of the e.c.o. with respect tovariations in supply voltages is explained asfollows: The frequency will shift in one directionwith an increase in screen voltage, whilean increase in plate voltage will cause it toshift in the other direction. By a proper proportioningof the resistors that comprise thevoltage divider supplying screen voltage, it ispossible to make the frequency of the oscillatorsubstantially independent of supply voltagevariations.The ClappOscillatorA relatively new type of oscillatorcircuit which is capable of givingexcellent frequency stability isillustrated in figure IG. Comparison betweenthe more standard circuits of figure IA throughIF and the Clapp oscillator circuits of figuresIG and lH will immediately show one markeddifference: the tuned circuit which controlsthe operating frequency in the Clapp oscillatoris series resonant, while in all the more standardoscillator circuits the frequency controllingcircuit is parallel resonant. Also, thecapacitors C, and C 2 are relatively large interms of the usual values for a Colpitts oscil-
242 Generation of R-F Energy<strong>THE</strong> R AD I 0lator. In fact, the value of capacitors C 1 and·. C 2will be in the vicinity of 0.001 p.fd. to0.0025 p.fd. for an oscillator which is to beoperated in the 1.8-Mc. band.The Clapp oscillator operates in the followingmanner: at the resonant frequency of theoscillator tuned circuit (L, C) the impedanceof this circuit is at minimum (since it operatesin series resonance) and maximum currentflows through it. Note however, that C,and C 2 also are included within the currentpath for the series resonant circuit, so that atthe frequency of resonance an appreciablevoltage drop appears across these capacitors.The voltage drop appearing across C, is appliedto the grid of the oscillator tube as excitation,while the amplified output of theoscillator tube appears across C 2 as the drivingpower to keep the circuit in oscillation.Capacitors C, and C 2should be made aslarge in value as possible, while still permittingthe circuit to oscillate over the full tuningrange of C. The larger these capacitorsare made, the smaller will be the coupling betweenthe oscillating circuit and the tube, andconsequently the better will be oscillator stabilitywith respect to tube variations. High Gmtubes such as the 6AC7, 6AG7, and 6CB6 willpermit the use of larger values of capacitanceat C, and C 2 than will more conventional tubessuch as the 6SJ7, 6V6, and such types. In generalit may be said that the reactance of capacitorsC, and C 2 should be on the order of40 to 120 ohms at the operating frequency ofthe oscillator-with the lower values of reactancegoing with high·Gm tubes and thehigher values being necessary to permit oscillationwith tubes having Gm in the range of2000 micromhos such as the 6S J7,It will be found that the Clapp oscillatorwill have a tendency to vary in power outputover the frequency range of tuning capacitorC. The output will be greatest where C is atits largest setting, and will tend to fall offwith C at minimum capacitance. In fact, ifcapacitors C, and C 2 have too large a valuethe circuit will stop oscillation near the minimumcapacitance setting of C. Hence it willbe necessary to use a slightly smaller valueof capacitance at C, and C 2 (to provide an increasein the capacitive reactance at thispoint), or else the frequency range of the oscillatormust be restricted by paralleling a fixedcapacitor across C so that its effective capacitanceat minimum setting will be increased toa value which will sustain oscillation.In the triode Clapp oscillator, such as shownat figure 1G, output voltage for excitation ofan amplifier, doubler, or isolation stage normallyis taken from the cathode of the oscillatortube by capacitive coupling to the gridof the next tube. However, where greater iso-lation of succeeding stages from the oscillat·ing circuit is desired, the electron-coupledClapp oscillator diagrammed in figure 1H maybe used. Output then may be taken from theplate circuit of the tube by capacitive couplingwith either a tuned circuit, as shown, or withan r-f choke or a broadly resonant circuit inthe plate return. Alternatively, energy may becoupled from the output circuit L 2 -C 3 by linkcoupling. The considerations with regard toC,, C 2, and the grid tuned circuit are the sameas for the triode oscillator arrangement offigure 1G.Negative Resist- Negative-resistance oscilanceOscillators lators often are used when. . . . unusually high frequencystablltty IS desired, as in a frequency meter.The dynatron of a few years ago and the newertransitron are examples of oscillator circuitswhich make use of the negative resistancecharacteristic between different elements insome multi-grid tubes.In the dynatron, the negative resistance is aconsequence of secondary emission of electronsfrom the plate of a tetrode tube. By aprop_er propor~ioning of the electrode voltage,an Increase 10 screen voltage will cause adecrease in screen current, since the increasedscreen voltage will cause the screen to attracta ~arger number of the secondary electronsemttted by the plate. Since the net screen currentflowing from the screen supply will bedecreased by an increase in screen voltage,it is said that the screen circuit presents anegative resistance.If any type of tuned circuit, or even a resistance-capacitancecircuit, is connected inseries with the screen, the arrangement willoscillate-provided, of course, that the externalcircuit impedance is greater than the negativeresistance. A negative resistance effect similarto the dynatron is obtained in the transitroncircuit, which uses a pentode with the suppressorcoupled to the screen. The negative resistancein this case is obtained from a combinationof secondary emission and inter-electrodecoupling, and is considerably more stablethan that obtained from uncontrolled secondaryemission alone in the dynatron. A representativetransitron oscillator circuit is shown infigure 2.The chief distinction between a conventionalnegative grid oscillator and a negativeresistance oscillator is that in the former thetank circuit must act as a phase inverter inorder to permit the amplification of the tubeto act as a negative resistance, while in thelatter the tube acts as its own phase inverter.Thus a negative resistance oscillator requiresonly an untapped coil and a single capacitor
<strong>HANDBOOK</strong>Oscillators 24 3Figure 2TWO-TERMINAL OSCILLATOR CIRCUITSBoth circuits may be usee/ for an audiooscillator or for frequencies into thev·h-f range simply by placing a tank clr·cult tuned to the proper frequency where/nc/icatecl on the drawing. Recommenc/ec/values for the components are given belowfar both oscillators.c,0 TRANSITRON OSCILLATORTRANSITION OSCILLATORc,-0.01-!Lfd. mica for r.f. 10-!Lfd. elect. for a.f.C,-0.00005-!Lfd. mica for r.f. 0.1-!Lid. paper fora. f.C,-0.003- /Lfd. mica for r.f. 0.5- !Lid. paper fora. f.C..-0.01-!Lid. mica for r.f. 8- !Lid. elect. for a. f.R 1 -220K ~-watt carbonR 2 -1800 ohms ~-watt carbonR 3 -22K 2-watt carbonR.-22K 2-watt carbonB+ 100- 1!!10 V.@ CATHODE COUPLED OSCILLATORCATHODE-COUPLED OSCILLATORC,-0.00005-!Lfd. mica for r.f. 0.1-!Lid. poperfor audioC,-0.003- !Lid. mica lor r.f. 8- !Lid. elect. foraudioR 1 -47K ~-watt carbonR 2 -1 K 1-watt carbonas the frequency determining tank circuit, andis classed as a two terminal oscillator. In fact,the time constant of an R/C circuit may beused as the frequency determining element andsuch an oscillator is rather widely used as atunable audio frequency oscillator.The Frankl inOscillatorThe Franklin oscillator makesuse of two cascaded tubes toobtain the negative-resistanceeffect (figure 3). The tubes may be either apair of triodes, tetrodes, or pentodes, a dualtriode, or a combination of a triode and a multi·grid tube. The chief advantage of this oscil·lator circuit is that the frequency determiningtank only has two terminals, and one side ofthe circuit is grounded.The second tube acts as a phase inverter togive an effect similar to that obtained with thedynatron or transitron, except that the effectivetransconductance is much higher. If the tunedcircuit is omitted or is replaced by a resistor,the circuit becomes a relaxation oscillator ora multivibrator.OscillatorStabilityThe Clapp oscillator has provedto be inherently the mOst stableof all the oscillator circuits dis·cussed above, since minimum coupling be·tween the oscillator tube and its associatedtuned circuit is possible. However, this in·herently good stability is with respect to tubevariations; instability of the tuned circuit withrespect to vibration or temperature will ofcourse have as much effect on the frequency ofoscillation as with any other type of oscillatorcircuit. Solid mechanical construction of thecomponents of the oscillating circuit, alongwith a small negative-coefficient compensatingcapacitor included as an element of the tunedcircuit, usually will afford an adequate degreeof oscillator stability.-B+Figure 3<strong>THE</strong> FRANKLIN OSCILLATOR CIRCUITA separate phase Inverter tube is usee/ inthis oscillator to feecl a portion of the outputback to the Input In the proper phase to sus•fain oscillation. The values of C z one/ C zshoulrl be as small as will permit oscillationsto be sustained over the c/es/rec/ frequencyrange.
244 Generation of R-F Energy<strong>THE</strong> R AD I 0V.F.O. Transmit· When used to control the freterControls quency of a transmitter inwhich there are s t ring en tlimitations on frequency tolerance, several precautionsare taken to ensure that a variablefrequency oscillator will stay on frequency.The oscillator is fed from a voltage regulatedpower supply, uses a well designed and temperaturecompensated tank circuit, is of ruggedmechanical construction to avoid the effectsof shock and vibration, is protected againstexcessive changes in ambient room temperature,and is isolated from feedback or straycoupling from other portions of the transmitterby shielding, filtering of voltage supply leads,and incorporation of one or more buffer-amplifierstages. In a high power transmitter a smallamount of stray coupling from the final amplifierto the oscillator can produce appreciabledegradation of the oscillator stability if bothare on the same frequency. Therefore, the oscillatorusually is operated on a subharmonicof the transmitter output frequency, with oneor more frequency multipliers between the os·cillator and final amplifier.13-2 Quartz CrystalOscillatorsQuartz is a naturally occuring crystal havinga structure such that when plates are cutin certain definite relationships to the crystal·lographic axes, these plates will show thepiezoelectric effect-the plates will be deformedin the influence of an electric field,and, conversely, when such a plate is compressedor deformed in any way a potentialdifference will appear upon its opposite sides.The crystal has mechanical resonance, andwill vibrate at a very high frequency becauseof its stiffness, the natural period of vibrationdepending upon the dimensions, the method ofelectrical excitation, and crystallographicorientation. Because of the piezoelectric properties,it is possible to cut a quartz platewhich, when provided with suitable electrodes,will have the characteristics of a series resonantcircuit with a very high L/C ratio andvery high Q. The Q is several times as highas can be obtained with an inductor-capacitorcombination in conventional physical sizes.The equivalent electrical circuit is shown infigure 4A, the resistance component simplybeing an acknowledgment of the fact that theQ, while high, does not have an infinite value.The shunt capacitance of the electrodes andassociated wiring (crystal holder and socket,plus circuit wiring) is represented by the dottedportion of figure 4B. In a high frequency'~~~·; Li(LARGE)R1(SMALL)"fL1(sTRAYI SHUNT)II_..JLl ~LC2I ~:0 @ ©Figure 4EQUIVALENT ELECTRICAL CIRCUIT OFQUARTZ PLATE IN A HOLDERAt (A) is shown the equivalent series-resonantcircuit of the crystal itself, at (B) isshown how the shunt capacitance of theholder electrodes and associated wiring affectsthe circuit to the combination circuitof (C) which exhibits both series resonanceand parallel resonance (anti-resonance), theseparation in frequency between the twomodes being very small and determined bythe ratio of C 1 to C 2·crystal this will be considerably greater thanthe capacitance component of an equivalentseries L/C circuit, and unless the shunt capacitanceis balanced out in a bridge circuit,the crystal will exhibit both resonant (seriesresonant) and anti-resonant (parallel resonant)frequencies, the latter being slightly higherthan the series resonant frequency and approachingit as C 2 is increased.The series resonance characteristic is employedin crystal filter circuits in receiversand also in certain oscillator circuits whereinthe crystal is used as a selective feedbackelement in such a manner that the phase of thefeedback is correct and the amplitude adequateonly at or very close to the series resonantfrequency of the crystal.While quartz, tourmaline, Rochelle salts,ADP, and EDT crystals all exhibit the piezoelectriceffect, quartz is the material widelyemployed for frequency control.As the cutting and grinding of quartz plateshas progressed to a high state of developmentand these plates may be purchased at priceswhich discourage the cutting and grinding bysimple hand methods for one's own use theprocedure will be only lightly touched 'uponhere.The crystal blank is cut from the raw quartzat a predetermined orientation with respect tothe optical and electrical axes, the orientationdetermining the activity, temperature coefficient,thickness coefficient, and other characteristics.Various orientations or "cuts" havinguseful characteristics are illustrated infigure 5.
<strong>HANDBOOK</strong>Crystal Oscillators 245ZERO TE~.!PERATUR( COEFFICIENTOSCILLATORS AND FILTERSHIGH fREQUENCY AT. BTLOW fREQUENCY CT,OT,ET,FTFigure 5AT+35"15'BT-49°CT-t-36°OT-52'ET"-t66°fT-57'ORIENTATION OF <strong>THE</strong>COMMON CRYSTAL CUTSZERO TE~PERATURECOEFFICIENTo• OSCILLATORSA MT LONGITUDINAL CRYSTALB NT FLEXURE CRYSTALDOUGHNUTZERO TEMPERATURECOEFFICIENTLOW TEMP COEFFICIENTt5'FILTERSThe crystal blank is then rough-ground almostto frequency, the frequency increasingin inverse ratio to the oscillating dimension(usually the thickness). It is then finished toexact frequency either by careful lapping, byetching, or plating. The latter process consistsof finishing it to a frequency slightlyhigher than that desired and then silver platingthe electrodes right on the crystal, the frequencydecreasing as the deposit of silver isincreased. If the crystal is not etched, it mustbe carefully scrubbed and "baked" severaltimes to stabilize it, or otherwise the frequencyand activity of the crystal will change withtime. Irradiation by X-rays recently has beenused in crystal finishing.Unplated crystals usually are mounted inpressure holders, in which two electrodes areheld against the crystal faces under slightpressure. Unplated crystals also are sometimesmounted in an air-gap holder, in whichthere is a very small gap between the crystaland one or both electrodes. By making thisgap variable, the frequency of the crystal maybe altered over narrow limits (about 0.3% forcertain types).The temperature coefficient of frequency forvarious crystal cuts of the "-T" rotated familyis indicated in figure 5. These angles aretypical, but crystals of a certain cut will varyslightly. By controlling the orientation and dimensioning,the turning point (point of zerotemperature coefficient) for a BT cut plate maybe made either lower or higher than the 75 degreesshown. Also, by careful control of axesand dimensions, it is possible to get AT cutcrystals with a very flat temperature-frequencycharacteristic.The first quartz plates used were either Ycut or X cut. The former had a very high temperaturecoefficient which was discontinuous,causing the frequency to jump at certain criticaltemperatures. The X cut had a moderatelybad coefficient, but it was more continuous,and by keeping the crystal in a temperaturecontrolled oven, a high order of stability couldbe obtained. However, the X cut crystal wasconsiderably less active than the Y cut, especiallyin the case of poorly ground plates.For frequencies between 500 kc. and about6 Me., the AT cut crystal now is the mostwidely used. It is active, can be made freefrom spurious responses, and has an excellenttemperature characteristic. However, aboveabout 6 Me. it becomes quite thin, and a difficultproduction job. Between 6 Me. and about12 Me., the BT cut plate is widely used. Italso works well between 500 kc. and 6 Me.,but the AT cut is more desirable when a highorder of stability is desired and no crystaloven is employed.For low frequency operation on the order of100 kc., such as is required in a frequencystandard, the GT cut crystal is recommended,though CT and DT cuts also are widely usedfor applications between 50 and 500 kc. TheCT, DT, and GT cut plates are known as contourcuts, as these plates oscillate along thelong dimension of the plate or bar, and aremuch smaller physically than would be thecase for a regular AT or BT cut crystal forthe same frequency.Crystal HoldersCrystals normally are purchasedready mounted. The
246 Generation of R-F Energy<strong>THE</strong> R AD I 0best .type mount is determined by the type crystaland its application, and usually an optimummounting is furnished with the crystal.However, certain features are desirable in allholders. One of these is exclusion of moistureand prevention of electrode oxidization. Thebest means of accomplishing this is a metalholder, hermetically sealed, with glass insulationand a metal-to-glass bond. However, suchholders are more expensive, and a ceramic orphenolic holder with rubber gasket will servewhere requirements are not too exacting.Temperature Control;Crystal OvensWhere the frequency tolerancerequirements arenot too stringent and theambient temperature does not include extremes,an AT-cut plate, or a BT-cut plate with optimum(mean temperature) turning point, willoften provide adequate stability without resortingto a temperature controlled oven. However,for broadcast stations and other applicationswhere very close tolerances must bemaintained, a thermostatically controlled oven,adjusted for a temperature slightly higher thanthe highest ambient likely to be encountered,must of necessity be employed.Harmonic CutCrystalsJust as a vibrating suing canbe made to vibrate on its harmonics,a quartz crystal willexhibit mechanical resonance (and thereforeelectrical resonance) at harmonics of its fundamentalfrequency. When employed in the usualholder, it is possible to excite the crystalonly on its odd harmonics (overtones).By grinding the crystal especially for harmonicoperation, it is possible to enhance itsoperation as a harmonic resonator. BT and ATcut crystals designed for optimum operationon the 3d, 5th and even the 7th harmonic areavailable. The 5th and 7th harmonic types,especially the latter, require special holderand oscillator circuit precautions for satisfactoryoperation, but the 3d harmonic typeneeds little more consideration than a regularfundamental type. A crystal ground for optimumoperation on a particular harmonic may or maynot be a good oscillator on a different harmonicor on the fundamental. One interestingcharacteristic of a harmonic cut crystal is thatits harmonic frequency is not quite an exactmultiple of its fundamental, though the disparityis very small.The harmonic frequency for which the crystalwas designed is the working frequency. Itis not the fundamental since the crystal itselfactually oscillates on this working frequencywhen it is functioning in the proper manner.When a harmonic-cut crystal is employed, aselective tuned circuit must be employed somewherein the oscillator in order to discrimi-EXCITATIONRFCBASIC "PIERCE" OSCILLATOR+B1oo-1~v. ®+B100-15QV.Figure 6HOT-CATHODE "PIERCE'OSCILLATOR<strong>THE</strong> PIERCE CRYSTAL OSCILLATORCIRCUITShown at (A} is the &as/c Pierce crystal oscillatorcircuit. A capacitance of 10 to 75IJ.!J.frl. normally will &e requlrerJ crt C 1 foroptimum operation. If a plate supply voltagehigher than inrlicaterJ is to &e userJ, RFC 1may &e replcrcerJ &y a 22,000-ohm 2·watt resistor.Shown at (B) Is an alternative arrangementwith the r-f grounrl moverJ to theplate, crnrJ with the catharle floating. Thisalternative circuit hers the arlvcrntcrge thatthe full r-f voltage rleveloperJ across thecrystal may &e userl as excitation to the nextstage, since one s/rle of the crystal IsgrounrlerJ.nate against the fundamental frequency or undesiredharmonics. Otherwise the crystal mightnotalways oscillate on the intended frequency.For this reason the Pierce oscillator, laterdescribed in this chapter, is not suitable foruse with harmonic-cut crystals, because theonly tuned element in this oscillator circuitis the crystal itself.Crystal Current;Heating and FractureFor a given crystal operatingas an anti-resonanttank in a given oscillatorat fixed load impedance and plate andscreen voltages, the r-f current through thecrystal will increase as the shunt capacitanceC 2 of figure 4 is increased, because this effectivelyincreases the step-up ratio of C, to C 2•For a given shunt capacitance, C 2 , the crystalcurrent for a given crystal is directly proportionalto the r-f voltage across C 2 • This voltagemay be measured by means of a vacuumtube voltmeter having a low input capacitance,and such a measurement is a more pertinentone than a reading of r-f current by means ofa thermogalvanometer inserted in series withone of the leads to the crystal holder.The function of a crystal is to provide accuratefrequency control, and unless it is usedin such a manner as to take advantage of itsinherent high stability, there is no point inusing a crystal oscillator. For this reason a
<strong>HANDBOOK</strong>Crystal Oscillators 247crystal oscillator should not be run at highplate input in an attempt to obtain considerablepower directly out of the oscillator, assuch operation will cause the crystal to heat,with resultant frequency drift and possiblefracture.13-3 Crystal OscillatorCircuitsConsiderable confusion exists as to nomenclatureof crystal oscillator circuits, due to atendency to name a circuit after its discoverer.Nearly all the basic crystal oscillator circuitswere either first used or else developed independentlyby G. W. Pierce, but he has not beenso credited in all the literature.Use of the crystal oscillator in master oscillatorcircuits in radio transmitters datesback to about 1924 when the first applicationarticles appeared.The Pierce The circuit of figure 6A is the simOscillator plest crystal oscillator circuit. Itis one of those de v e 1 oped byPierce, and is generally known among amateursas the Pierce oscillator. The crystalsimply replaces the tank circuit in a Colpittsor ultra-audion oscillator. The r-f excitationvoltage available to the next stage is low, beingsomewhat less than that developed acrossthe crystal. Capacitor C, will make more ofthe voltage across the crystal available forexcitation, and sometimes will be found necessaryto ensure oscillation. Its value is small,usually approximately equal to or slightlygreater than the stray capacitance from theplate circuit to ground (including the grid ofthe stage being driven).If the r-f choke has adequate inductance, acrystal (even a harmonic cut crystal) will almostinvariably oscillate on its fundamental.The Pierce oscillator therefore cannot be usedwith harmonic cut crystals.The circuit at (B) is the same as that of(A) except that the plate instead of the cathodeis operated at ground r-f potential. All ofthe r-f voltage developed across the crystalis available for excitation to the next stage,but still is low for reasonable values of crystalcurrent. For best operation a tube with lowheater-cathode capacitance is required. Excitationfor the next stage rna y also be takenfrom the cathode when using this circuit.Tuned. Plate The circuit shown in figure7 A is also one used byCrystal OscillatorPierce, but is more widelyreferred to as the "Miller" oscillator. To avoidconfusion, we shall refer to it as the tunedplatecrystal oscillator. It is essentially anArmstrong or tuned plate-tuned grid oscillatorwith the crystal replacing the usual L-C gridtank. The plate tank must be tuned to a frequencyslightly higher than the anti-resonant(parallel resonant) frequency of the crystal.Whereas the Pierce circuits of figure 6 willoscillate at (or very close to) the anti-resonantfrequency of the crystal, the circuits offigure 7 will oscillate at a frequency a littleabove the anti-resonant frequency of thecrystal.The diagram shown in figure 7 A is the basiccircuit. The most popular version of the tuned·plate oscillator employs a pentode or beamtetrode with cathode bias to prevent excessiveplate dissipation when the circuit is not oscillating.The cathode resistor is optional. Itsomission will reduce both crystal current andoscillator efficiency, resulting in somewhatmore output for a given crystal current. Thetube usually is an audio or video beam pentodeor tetrode, the plate-grid capacitance of suchtubes being sufficient to ensure stable oscilla·tion but not so high as to offer excessive feedbackwith resulting high crystal current. The6AG7 makes an excellent all-around tube forthis type circuit.PentodeHarmonic CrystalOscillator CircuitsThe usual type of crystalcontrolledh-f transmitteroperates, at least part ofthe time, on a frequencywhich is an integral multiple of the operatingfrequency of the controlling crystal. Hence,oscillator circuits which are capable of providingoutput on the crystal frequency if desired,but which also can deliver output energyon harmonics of the crystal frequency havecome into wide use. Four such circuits whichhave found wide application are illustrated infigures 7C, 70, 7E, and 7F.The circuit shown in figure 7C is recommendedfor use with harmonic-cut crystalswhen output is desired on a multiple of theoscillating frequency of the crystal. As anexample, a 25-Mc. harmonic-cut crystal maybe used in this circuit to obtain output on 50Me., or a 48-Mc. harmonic-cut crystal may beused to obtain output on the 144-Mc. amateurband. The circuit is not recommended for usewith the normal type of fundamental-frequencycrystal since more output with fewer variableelements can be obtained with the circuits of70 and 7F.The Pierce-harmonic circuit shown in figure70 is satisfactory for many applicationswhich require very low crystal current, buthas the disadvantage that both sides of thecrystal are above ground potential. The Tri-tetcircuit of figure 7E is widely used and can
248 Generation of R-F Energy <strong>THE</strong> R AD I 0FBASIC TUNED-PLATE OSCILLATORRECOMMENDED TUNED-PLATEOSCILLATOR® ©SPECIAL CIRCUIT FOR USE WITHHARMONIC-CUT CRYSTAL.6AG7 F, 2F, 3F, 4F 6AG7 F. 2F. 3F, 4FPIERCE HARMONIC CIRCUIT "TRITET'' CIRCUIT COLPITTS HARMONIC OSCILLATOR@) ® ®Figure 7COMMONLY USED CRYSTAL OSCILLATOR CIRCUITSShown ot (A) is the basic tunec/-p/ate crystal oscillator with a trioc/e oscillator tube.The plate tank must be tunecl on the /ow-capacitance s/c/e pf resonance to sustainasci/lotion. (B) shows the tunecl-p/ote oscillator as it Is normally usecl, with an a-fpower pentacle to permit high output with relotively /qw crystal current.Schematics (C), (D), (E), ancl (F) illustrote crystal oscillator circuits which can cle-1 iver moderate output energy on harmonics of the oscilloting frequency of the crystal.(C) shows a spec/a/ circuit which will permit use of a harmonic-cut crystal toobtain output energy we// into the v-h-f range. (D) is valuable when extremely lowcrystal current is a requirement, but clel Ivers relatively low output. (E) is commonlyusecl, but is subject to crystal clamage if the cothocle circuit is m/stunecl. (F) isrecommenclec/ as the most generally sotlsfactory from the standpoints of: low crys·tal current regarcl/ess of mls-acltustment, goocl output on harmonic frequencies, onesicle of crystal Is grounclecl, will osci//ote with crystals from 1.5 to 10 Me. withoutadjustment, output tank may be tunecl to the crystal frequency for fundamental outputwithout stopping oscillation or changing frequency.give excellent output with low crystal current.However, the circuit has the disadvantages ofrequiring a cathode coil, of requiring carefulsetting of the variable cathode capacitor toavoid damaging the crystal when changing fre·quency ranges, and of having both sides ofthe crystal above ground potential.The Colpitts harmonic oscillator of figure7F is recommended as being the most gener·ally satisfactory harmonic crystal oscillatorcircuit since it has the following advantages:(1) the circuit will oscillate with crystalsover a very wide frequency range with nochange other than plugging in or switching inthe desired crystal; (2) crystal current is ex·tremely low; (3) one side of the crystal isgrounded, which facilitates crystal-switchingcircuits; (4) the circuit will operate straightthrough without frequency pulling, or it maybe operated with output on the second, third,or fourth harmonic of the crystal frequency.Crystal OscillatorTuningThe tunable circuits of alloscillators i 11 us t ratedshould be tuned for maxi·mum output as indicated by maximum excita·tion to the following stage, except that theoscillator tank of tuned-plate oscillators (fig·ure 7 A and figure 7B) should be backed offslightly towards the low capacitance side from
<strong>HANDBOOK</strong>All-band Crystal Oscillator 24 9+300 v.~I. LI"/5.UH (2j" OFB&W#3015)2. l2;f,6JJH (1" OFB&W#3003)3. FOR 160 METER OPERATION ADO 5 .JJ.J./F. CONDENSERBETWEEN PINS 4 & 8 OF 6AG7. PLATE COIL= 55 .JJH,(zf"oF B&W#3016)4, X= 7 MC. CRYSTAL FOR HARMONIC OPERATIONFigure 8751---F, 2F,3F,20-15-lo 4 F.ALL-BAND 6AG7 CRYSTAL OSCILLATORCAPABLE OF DRIVINGBEAM PENTODE TUBEmaximum output, as the oscillator then is ina more stable condition and sure to start immediatelywhen power is applied. This is especiallyimportant when the oscillator is keyed,as for break-in c-w operation.Crystal Switching It is desirable to keep strayshunt capacitances in thecrystal circuit as low as possible, regardlessof the oscillator circuit. If a selector switchis used, this means that both switch and crystalsockets must be placed close to the oscillatortube socket. This is especially true ofharmonic-cut crystals operating on a comparativelyhigh frequency. In fact, on the highestfrequency crystals it is preferable to use aturret arrangement for switching, as the straycapacitances can be kept lower.Crystal Oscillator When the crystal oscillatorKeyingis keyed, it is necessarythat crystal activity and oscillator-tubetransconductance be moderatelyhigh, and that oscillator loading and crystalshunt capacitance be low. Below 2500 kc. andabove 6 Me. these considerations become especiallyimportant. Keying of the plate voltage(in the negative lead) of a crystal oscillator,with the screen voltage regulated at about150 volts, has been found to give satisfactoryresults.A Versatile 6AG7 The 6AG7 tube may beCrystal Oscillator used in a modified Tri-tetcrystal oscillator, capableof delivering sufficient power on all bandsfrom 160 meters through 10 meters to fullydrive a pentode tube, such as the 807, 2E26or 6146. Such an oscillator is extremely usefulfor portable or mobile work, since it combinesall essential exciter functions in onetube. The circuit of this oscillator is shownin figure 8. For 160, 80 and 40 meter operationthe 6AG7 functions as a tuned-plate oscillator.Fundamental frequency crystals areused on these three bands. For 20, 15 and 10meter operation the 6AG7 functions as a Triteroscillator with a fixed-tuned cathode circuit.The impedance of this cathode circuitdoes not affect operation of the 6AG7 on thelower frequency bands so it is left in the circuitat all times. A 7-Mc. crystal is used forfundamental output on 40 meters, and for harmonicoutput on 20, 15 and 10 meters. Crystalcurrent is extremely low regardless of the outputfrequency of the oscillator. The plate circuitof the 6AG7 is capable of tuning a frequencyrange of 2:1, requiring only two outputcoils: one for 80-40 meter operation, and onefor 20, 15 and 10 meter operation. In somecases it may be necessary to add 5 micromicrofaradsof external feedback capacity betweenthe plate and control grid of the 6AG7 tube tosustain oscillation with sluggish 160 metercrystals.Triode OvertoneOscillatorsThe recent development ofreliable overtone crystalscapable of operation on thethird, fifth, seventh (or higher) overtones hasmade possible v-h·f output from a low frequen·cy crystal by the use of a double triode regenerativeoscillator circuit. Some of the newtwin triode tubes such as the 12AU7, 12AV7and 6]6 are especially satisfactory when usedin this type of circuit. Crystals that are groundfor overtone service may be made to oscillateon odd overtone frequencies other than theone marked on the crystal holder. A 24-Mc.overtone crystal, for example, is a speciallyground 8-Mc. crystal operating on its thirdovertone. In the proper circuit it may be madeto oscillate on 40 Me. (fifth overtone), 56 Me.(seventh overtone), or 72 Me. (ninth overtone).Even the ordinary 8-Mc. crystals not designedfor overtone operation may be made to oscil·late readily on 24 Me. (third overtone) in thesecircuits.A variety of overtone oscillator circuits isshown in figure 9. The oscillator of figure 9Ais attributed to Frank Jones, W6AJF. The firstsection of the 6]6 dual triode comprises a re·generative oscillator, with output on either thethird or fifth overtone of the crystal frequency.The regenerative loop of this oscillator consistsof a condenser bridge made up of C 1 andC 2 , with the ratio C,/C, determining the amountof regenerative feedback in the circuit. With
250 Generation of R-F Energy<strong>THE</strong> R AD I 0+300 v.FOR 7 MC. CRYSTAL+300v L1 =zsr. lf24DNNATtONAL xRsoFORML2=sr. f/18 ON NATIONAL XR50FORM® JONES HARMONIC OSCILLATORP12AU7r-----,---jr---6, 9 F50 RFC 7 •• ~ 1 L:O62 K B-= ':"' .001100 100FOR 8 MC. CRYSTAL+30011. Li=9T.*-3003B&WMINIDUCTORl2 = 4 T. 11-3003 S&W MINIOUCTOR<strong>THE</strong>SE COILS MADE FROM SINGLE SECTION9f:tfff1 1 P,}'oCJc~'8R~ 1 flr6lf.~~'fo~{f.N ro© REGENERATIVE HARMONIC OSCILLATOR@ COLPITTS HARMONIC OSCILLATOR1000L13F+30011.f::eMc.6J605 ;::;}0 =100FOR BMC. CRYSTALl1 =tOT.# 3011 B&W MINIOUCTOR,TAP AT 3T. FROM GRID END@ REGENERATIVE HARMONIC OSCILLATORt 12AT7 (oR 6AB4)I K+300V.© CATHODE FOLLOWER OVERTONE OSCILLATOR+200V,l I = 5 T. # !8, -f" 0. SPACEDl2=1r. HOOKUP WIRE, fH D.® V.H.F. OVERTONE OSCILLATORFigure 9VARIOUS TYPES OF OVERTONE OSCILLATORS USING MINIATURE DOUBLE-TRIODETUBESan 8-Mc. crystal, output from the first sectionof the 6J6 tube may be obtained on either 24~!c. or 40 Me., depending upon the resonantfrequency of the plate circuit inductor, L,. Thesecond half of the 6J6 acts as a frequencymultiplier, its plate circuit, L 2 , tuned to thesixth or ninth harmonic frequency when L, istuned to the third overtone, or to the tenthharmonic frequency when L, is tuned to thefifth overtone.Figure 98 illustrates a Colpitts overtoneoscillator employing a 6J6 tube. This is anoutgrowth of the Colpitts harmonic oscillatorof figure 7F. The regenerative loop in thiscase consists of C,, C 2and RFC between thegrid, cathode and ground of the first sectionof the 6J6. The plate circuit of the first sectionis tuned to the second overtone of thecrystal, and the second section of the 6J6doubles to the fourth harmonic of the crystal.This circuit is useful in obtaining 28-Mc. outputfrom a 7-Mc. crystal and is highly popularin mobile work.The circuit of figure 9C shows a typical regenerativeovertone oscillator employing a12AU7 double triode tube. Feedback is controlledby the number of turns in L 2, and thecoupling between L 2 and L,. Only enough feed-
<strong>HANDBOOK</strong>R-F Amplifiers 251back should be employed to maintain properoscillation of the crystal. Excessive feedbackwill cause the first section of the 12AU7 tooscillate as a self-excited TNT oscillator,independent of the crystal. A variety of thiscircuit is shown in figure 9D, wherein a tappedcoil, L 1, is used in place of the two separatecoils. Operation of the circuit is the same ineither case, regeneration now being controlledby the placement of the tap on L 1•A cathode follower overtone oscillator isshown in figure 9E. The cathode coil, L., ischosen so as to resonate with the crystal andtube capacities just below the third overtonefrequency of the crystal. For example, with an8-Mc. crystal, L, is tuned to 24 Me., L 1resonateswith the circuit capacities to 23.5 Me.,and the harmonic tank circuit of the secondsection of the 12AT7 is tuned either to 48 Me.or 72 Me. If a 24-Mc. overtone crystal is usedin this circuit, L, may be tuned to 72 Me., L 1resonates with the circuit capacities to 70Me., and the harmonic tank circuit, L 2, is tunedto 144 Me. If there is any tendency towardsself-oscillation in the circuit, it may be eliminatedby a small amount of inductive couplingbetween L 2and L,. Placing these coils neareach other, with the winding of L 2correctlypolarized with respect to L, will prevent selfoscillationof the circuit.The use of a 144-Mc. overtone crystal isillustrated in figure 9F. A 6AB4 or one-halfof !l 12AT7 tube may be used, with outputdirectly in the 2-meter amateur band. A slightamount of regeneration is provided by the oneturn link, L, which is loosely coupled to the144-~lc. tuned tank circuit, L 1 in the plate circuitof the oscillator tube. If a 12AT7 tubeand a 110-Mc. crystal are employed, direct outputin the 220-Mc. amateur band may be obtainedfrom the second half of the 12A T7.13-4 Radio FrequencyAmplifiersThe output of the oscillator stage in a transmitter(whether it be self-controlled or crystalcontrolled) must be kept down to a fairly lowlevel to maintain stability and to maintain afactor of safety from fracture of the crystalwhen one is used. The low power output ofthe oscillator is brought up to the desiredpower level by means of radio-frequency amplifiers.The two classes of r-f amplifiers thatfind widest application in radio transmittersare the Class B and Class C types.. The Class B Class B amplifiers are used in aAmp I ifierradio-telegraph transmitter whenmaximum power gain and mini-mum harmonic output is desired in a particularstage. A Class B amplifier operates witltcutoff bias and a comparatively small amountof excitation. Power gains of 20 to 200 or soare obtainable in a well-designed Class Bamplifier. The plate efficiency of a Class Bc-w amplifier will run around 65 per cent.The Class BLinearAnother type of Class B amplifieris the Class B linear stageas employed in radiophone work.This type of amplifier is used to increase thelevel of a modulated carrier wave, and dependsfor its operation upon the linear relationbetween excitation voltage and outputvoltage. Or, to state the fact in another manner,the power output of a Class B linear stagevaries linearly with the square of the excitationvoltage.The Class B linear amplifier is operatedwith cutoff bias and a small value of excitation,the actual value of exciting power beingsuch that the power output under carrier conditionsis one-fourth of the peak power capabilitiesof the stage. Class B linears are verywidely employed in broadcast and commercialinstallations, but are comparatively uncommonin amateur application, since tubes with highplate dissipation are required for moderateoutput. The carrier efficiency of such an amplifierwill vary from approximately 30 percent to 35 per cent.The Class CAmplifierClass C amplifiers are very widelyused in all types of transmitters.Good power gain may beobtained (values of gain from 3 to 20 are common)and the plate circuit efficiency may be,under certain conditions, as high as 85 percent. Class C amplifiers operate with considerablymore than cutoff bias and ordinarily witha large amount of excitation as compared to aClass B amplifier. The bias for a normal ClassC amplifier is such that plate current on thestage flows for approximately 120° of the 360°excitation cycle. Class C amplifiers are usedin transmitters where a fairly large amount ofexcitation power is available and good platecircuit efficiency is desired.Plate ModulatedClass CThe characteristic of a ClassC amplifier which makes itline a r with respect tochanges in plate voltage is that which allowssuch an amplifier to be plate modulated forradiotelephony. Through the use of higher biasthan is required for a c-w Class C amplifierand greater excitation, the linearity of suchan amplifier may be extended from zero platevoltage to twice the normal value. The outputpower of a Class C amplifier, adjusted forplate modulation, varies with the square of the
252 Generation of R-F Energy<strong>THE</strong> R AD I 0plate voltage. This is the same condition thatwould take place if a resistor equal to thevoltage on the amplifier, divided by its platecurrent, were substituted for the amplifier.Therefore, the stage presents a resistive loadto the modulator.Grid ModulatedClass CIf the grid current to a ClassC amplifier is reduced to alow value, and the plate loadingis increased to the point where the platedissipation approaches the rated value, suchan amplifier may be grid modulated for radiotelephony.If the plate voltage is raised toquite a high value and the stage is adjustedcarefully, efficiencies as high as 40 to 43 percent with good modulation capability and comparativelylow distortion may be obtained.Fixed bias is required. This type of operationis termed Class C grid-bias modulation.Grid Excitation Adequate grid ex citationmust be available for ClassB or Class C service. The excitation for aplate-modulated Class C stage must be sufficientto produce a normal value of d-e grid currentwith rated bias voltage. The bias voltagepreferably should be obtained from a combinationof grid leak and fixed C-bias supply.Cutoff bias can be calculated by dividingthe amplification factor of the tube into thed-e plate voltage. This is the value normallyused for Class B amplifiers (fixed bias, nogrid resistor). Class C amplifiers use from 1 ~to 5 times this value, depending upon the availablegrid drive, or excitation, and the desiredplate efficiency. Less grid excitation is neededfor c-w operation, and the values of fixedbias (if greater than cutoff) may be reduced, orthe value of the grid leak resistor can be lowereduntil normal rated d-e grid current flows.The values of grid excitation listed for eachtype of tube may be reduced by as much as50 per cent if only moderate power output andplate efficiency are desired. When consultingthe tube tables, it is well to remember thatthe power lost in the tuned circuits must betaken into consideration when calculating theavailable grid drive. At very high frequencies,the r-f circuit losses may even exceed thepower required for actual grid excitation.Link coupling between stages, particularlyto the final amplifier grid circuit, normally willprovide more grid drive than can be obtainedfrom other coupling systems. The number ofturns in the coupling link, and the location ofthe turns on the coil, can be varied with respectto the tuned circuits to obtain the greatestgrid drive for allowable values of bufferor doubler plate current. Slight readjustmentssometimes can be made after plate voltagehas been applied to the driver tube.Excessive grid current damages tubes byoverheating the grid structure; beyond a certainpoint of grid drive, no increase in poweroutput can be obtained for a given plate voltage.13-5 Neutralization ofR.F. AmplifiersThe plate-to-grid feedback capacitance oftriodes makes it necessary that they be neutralizedfor operation as r-f amplifiers at frequenciesabove about 500 kc. Those screengridtubes, pentodes, and beam tetrodes whichhave a plate-to-grid capacitance of 0.1 pp.fd.or less may be operated as an amplifier withoutneutralization in a well-designed amplifierup to 30 Me.NeutralizingCircuitsThe object qf neutralization isto cancel or neutralize the capacitivefeedback of energy fromplate to grid. There are two general methodsby which this energy feedback may be eliminated:the first, and the most common method,is through the use of a capacitance bridge,and the second method is through rhe use of aparallel reactance of equal and opposite polarityto the grid-to-plate capacitance, to nullifythe effect of this capacitance.Examples of the first method are shown infigure 10. Figure lOA shows a capacity neutralizedstage employing a balanced tank circuit.Phase reversal in the tank circuit is obtainedby grounding the center of the tank coilto radio frequency energy by condenser C.Points A and B are 180 degrees out of phasewith each other, and the correct amount of outof phase energy is coupled through the neutralizingcondenser NC to the grid circuit ofthe tube. The equivalent bridge circuit of thisis shown in figure IIA. It is seen that thebridge is not in balance, since the plate-filamentcapacity of the tube forms one leg of thebridge, and there is no corresponding capacityfrom the neutralizing condenser (point B) toground to obtain a complete balance. In addition,it is mechanically difficult to obtain aperfect electrical balance in the tank coil, andthe potential between point A and ground andpoint B and ground in most cases is unequal.This circuit, therefore, holds neutralizationover a very small operating range and unlesstubes of low interelectrode capacity are usedthe inherent unbalance of the circuit will permitonly approximate neutralization.Split-StatorPlate NeutralizationFigure lOB shows the neutralizationcircuit which ismost widely used in singleendedr-f stages. The use of
<strong>HANDBOOK</strong> Neutralization 253NC~1~~ 1 1 ~~':' C RF'C©Figure 10COMMON NEUTRALIZING CIRCUITS FOR SINGLE-ENDED AMPLIFIERSa split-stator plate capacitor makes the electricalbalance of the circuit substantially independentof the mutual coupling within the coiland also makes the balance independent of theplace where the coil is tapped. With conven·tional tubes this circuit will allow one neutralizationadjustment to be made on, say, 28 Me.,and this adjustment usually will hold sufficientlyclose for operation on all lower frequencybands.Condenser C 1is used to balance out theplate-filament capacity of the tube to allow aperfect neutralizing balance at all frequencies.The equivalent bridge circuit is shown in figurellB. If the plate-filament capacity of thetube is extremely low (lOOTH triode, for example),condenser C 1may be omitted, or maymerely consist of the residual capacity of NCto ground.Grid Neutralization A split grid tank circuitmay also be used for neutralizationof a triode tube as shown in figure10C. Out of phase voltage is developed acrossa balanced grid circuit, and coupled throughNC to the single-ended plate circuit of thetube. The equivalent bridge circuit is shownin figure llC. This circuit is in balance untilthe stage is in operation when the loading effectof the tube upon one-half of the grid circuitthrows the bridge circuit out of balance.The amount of unbalance depends upon thegrid-plate capacity of the tube, and the amountof mutual inductance between the two halvesof the grid coil. If an r-f voltmeter is placedbetween point A and ground, and a secondvoltmeter placed between point B and groundthe loading effect of the tube will be noticeable.When the tube is supplied excitationwith no plate voltage, NC may be adjusteduntil the circuit is in balance. When platevoltage is applied to the stage, the voltagefrom point A to ground will decrease, and thevoltage from point B to ground will increase,both in direct proportion to the amount of circuitunbalance. The use of this circuit is notrecommended above 7 Me., and it should beused below that frequency only with low internalcapacity tubes.Push-PullNeutralizationTwo tubes of the same typecan be connected for push-pulloperation so as to obtain twiceas much output as that of a single tube. Apush-pull amplifier, such as that shown in figure12 also has an advantage in that the cir·cuit can more easily be balanced than a singletuber-f amplifier. The various inter-electrodecapacitances and the neutralizing capacitorsare connected in such a manner that the reactanceson one side of the tuned circuits areexactly equal to those on the opposite side.For this reason, push-pull r-f amplifiers canbe more easily neutralized in very-high-frequencytransmitters; also, they usually remainin perfect neutralization when tuning the am·plifier to different bands.The circuit shown in figure 12 is perhaps
254 Generation of R-F Energy<strong>THE</strong> R AD I 0@ BRIDGE EQUIVALENT OF FIGURE 10-Ac,~,Figure 12STANDARD CROSS-NEUTRALIZEDPUSH-PULL TRIODE AMPLIFIERC1YCP-F@ BRIDGE EQUIVALENT OF FIGURE 10-Bc© BRIDGE EQUIVALENT OF FIGURE 10-CFigure 11EQUIVALENT NEUTRALIZING CIRCUITSthe most commonly used arrangement for apush-pull r-f amplifier stage. The rotor of thegrid capacitor is grounded, and the rotor of the ·plate tank capacitor is by-passed to ground.Shunt or CoilNeutralizationThe feedback of energy fromgrid to plate in an unneutralizedr-f amplifier is a result ofthe grid-to-plate capacitance of the amplifiertube. A neutralization circuit is merely anelectrical arrangement for nullifying the effectof this capacitance. All the previous neutralizationcircuits have made use of a bridge circuitfor balancing out the grid-to-plate energyfeedback by feeding back an equal amount ofenergy of opposite phase.Another method of eliminating the feedbackeffect of this capacitance, and hence of neutralizingthe amplifier stage, is shown in figure13. The grid-to-plate capacitance in thetriode amplifier tube acts as a capacitive re-actance, coupling energy back from the plateto the grid circuit. If this capacitance is paralleledwith an inductance having the samevalue of reactance of opposite sign, the re·actance of one will cancel the reactance ofthe other and a high-impedance tuned circuitfrom grid to plate will result.This neutralization circuit can be used onultra-high frequencies where other neutralizationcircuits are unsatisfactory. This is truebecause the lead length in the neutralizationcircuit is practically negligible. The circuitcan also be used with push-pull r-f amplifiers.In this case, each tube will have its own neutralizinginductor connected from grid to plate.The main advantage of this arrangement isthat it allows the use of single-ended tankcircuits with a single-ended amplifier.The chief disadvantage of the shunt neutralizedarrangement is that the stage must be reneutralizedeach time the stage is retuned toa new frequency sufficiently removed that thegrid and plate tank circuits must be retuned toresonance. However, by the use of plug-incoils it is possible to change to a differentband of operation by changing the neutralizingcoil at the same time that the grid andplate coils are changed.The 0.0001-fLfd. capacitor in series withthe neutralizing coil is merely a blocking capacitorto isolate the plate voltage from thegrid circuit. The coil L will have to have avery large number of turns for the band of operatumin order to be resonant with the comparativelysmall grid-to-plate capacitance. Butsince, in all ordinary cases with tubes operatingon frequencies for which they were designed,the L/C ratio of the tuned circuit willbe very high, the coil can use comparativelysmall wire, although it must be wound on airor very low loss dielectric and must be insulatedfor the sum of the plate r-f voltage andthe grid r-f voltage.
<strong>HANDBOOK</strong>Neutralizing Procedure 255Figure 13COIL NEUTRALIZED AMPLIFIERThis neutralization circuit is very effectivewith triode tubes on any frequency, but isparticularly effective in the v-h-f range. Thecoil L is acl;ustec/ so that it resonates at theoperating frequency with the grid-to-platecapacitance of the tube. Capacitor C may bea very small unit of the low-capacitanceneutralizing type and is used to trim the cir·cuit to resonance at the operating frequency.If some means of varying the incluctance ofthe coil a small amount is available, thetrimmer capacitor is not needed.13-6NeutralizingProcedureAn r·f amplifier is neutralized to preventself-oscillation or regeneration. A neon bulb,a flashlight lamp and loop of wire, or an r·fgalvanometer can be used as a null indicatorfor neutralizing low-power stages. The platevoltage lead is disconnected from the r·f amplifierstage while it is being neutralized.Normal grid drive then is applied to the r·fstage, the neutralizing indicator is coupledto the plate coil, and the plate tuning capac·itor is tuned to resonance. The neutralizingcapacitor (or capacitors) then can be adjusteduntil minimum r.f. is indicated for resonantsettings of both grid and plate tuning capac·itor s. Both neutralizing capacitors are ad·jus ted simultaneously and to approximately thesame value of capacitance when a physicallysymmetrical push-pull stage is being neu·tralized.A final check for neutralization should bemade with a d-e milliammeter connected in thegrid leak or grid-bias circuit. There will beno movement of the meter reading as the platecircuit is tuned through resonance (withoutplate voltage being applied) when the stageis completely neutralized.Plate voltage should be completely removedby actually opening the d-e plate circuit. Ifthere is a d-e return through the plate supply,a small amount of plate current will flow whengrid excitation is applied, even though no pri·mary a·c voltage is being fed to the plate trans·former.A further check on the neutralization of anyr·f amplifier can be made by noting whethermaximum grid current on the stage comes atthe same point of tuning on the plate tuningcapacitor as minimum plate current. This checkis made with plate voltage on the amplifierand with normal antenna coupling. As the platetuning capacitor rs detuned slightly from reso·nance on either side the grid current on thestage should decrease the same amount andwithout any sudden jumps on either side ofresonance. This will be found to be a veryprecise indication of accurate neutralizationin either a triode or beam·tetrode r·f amplifierstage, so long as the stage is feeding a loadwhich presents a resistive impedance at theoperating frequency.Push-pull circuits usually can be more com·pletely neutralized than single-ended circuitsat very high frequencies. In the intermediaterange of from 3 to 15 Me., single-ended cir·cuits will give satisfactory results.Neutralization ofScreen-Grid R- FAmplifiersRadio-frequency amplifiersusing screen-grid tubes canbe operated without any ad·ditional provision for neu·tralization at frequencies up to about 15 Me.,provided adequate shielding has been providedbetween the input and output circuits. Specialv-h-f screen-grid and beam tetrode tubes suchas the 2£26, 807W, and 5516 in the low-powercategory and HK-2578, 4£27/8001, 4-125A,and 4-250A in the medium-power category canfrequently be operated at frequencies as highas 100 Me. without any additional provisionfor neutralization. Tubes such as the 807,2£22, HY-69, and 813 can be operated withgood circuit design at frequencies up to 30Me. without any additional provision for neutralization.The 815 tube has been found torequire neutralization in many cases above20 Me., although the 8298 tube will operatequite stably at 100 Me. without neutralization.None of these tubes, however, has perfectshielding between the grid and the plate, acondition brought about by the inherent in·ductance of the screen leads within the tubeitself. In addition, unless "watertight" shieldingis used between the grid and plate circuitsof the tube a certain amount of external leakagebetween the two circuits is present. Thesedifficulties may not be serious enough to re·quire neutralization of the stage to preventoscillation, but in many instances they showup in terms of key-clicks when the stage inquestion is keyed, or as parasitics when thestage is modulated. Unless the designer of theequipment can carefully check the tetrode
256 Generation of R-F Energy <strong>THE</strong> R AD I 0-C -HV +S
<strong>HANDBOOK</strong>Tetrode Neutralization 257NeutralizingSingle- EndedTetrode StagesA single-ended tetrode r-f amplifierstage may be neutralizedin the same manner asiII us t rated for a push-pullstage in figure 14A, provided a split-statortank capacitor is in use in the plate circuit.However, in the majority of single-ended retroder-f amplifier stages a single-section capacitoris used in the plate tank. Hence, otherneutralization procedures must be employedwhen neutralization is found necessary.The circuit shown in figure 14B is not atrue neutralizing circuit, in that the plate-togridcapacitance is not balanced out. However,the circuit can afford the equivalent effect byisolating the high resonant impedance of thegrid tank circuit from the energy fed back fromplate to grid. When NC and C are adjusted tobear the following ratio to the grid-to-platecapacitance and the total capacitance fromgrid-to-ground in the output tube:both ends of the grid tank circuit will be at thesame voltage with respect to ground as a resultof r-f energy fed back to the grid circuit. Thismeans that the impedance from grid to groundwill be effectively equal to the reactance ofthe grid-to-cathode capacitance in parallelwith the stray grid-to-ground capacitance, sincethe high resonant impedance of the tuned circuitin the grid has been effectively isolatedfrom the feedback path. It is important to notethat the effective grid-to-ground capacitanceof the tube being neutralized includes therated grid-to-cathode or input capacitance ofthe tube, the capacitance of the socket, wiringcapacitances and other strays, but it does notinclude the capacitances associated with thegrid tuning capacitor. Also, if the tube is beingexcited by capacitive coupling from a precedingstage (as in figure 14C), the effectivegrid-to-ground capacitance includes the outputcapacitance of the preceding stage andits associated socket and wiring capacitances.Cancellation ofScreen-LeadInductanceThe provisions discussed inthe previous paragraphs arefor neutralization of the small,though still important at thehigher frequencies, grid-to-plate capacitanceof beam-tetrode tubes. However, in the vicinityof the upper-frequency limit of each tube typethe inductance of the screen lead of the tubebecomes of considerable importance. With atube operating at a frequency where the inductanceof the screen lead is appreciable,the screen will allow a considerable amountof energy leak-through from plate to grid eventhough the socket terminal on the tube is carefullyby-passed to ground. This condition takesplace even though the socket pin is bypassedsince the reactance of the screen leadwill allow a moderate amount of r-f potentialto appear on the screen itself inside the electrodeassembly in the tube. This effect hasbeen reduced to a very low amount in suchtubes astheHytron5516, andtheEimac 4Xl50Aand 4X500A but it is still quite appreciable inmost beam-tetrode tubes.The effect of screen-lead inductance on thestability of a stage can be eliminated at anyparticular frequency by one of two methods.These methods are: (1) Tuning out the screenleadinductance by series resonating the screenlead inductance with a capacitor to ground.This method is illustrated in figure 140 and iscommon! y employed in commercial! y-built equipmentfor operation on a narrow frequency bandin the range above about 75 Me. The othermethod (2) is illustrated in figure 14E andconsists in feeding back additional energyfrom plate to grid by means of a small capacitorconnected between these two elements.Note that this capacitor is connected in sucha manner as to increase the effective grid-toplatecapacitance of the tube. This methodhas been found to be effective with 807 tubesin the range above 50 Me. and with tubes suchas the 4-125A and 4-2SOA in the vicinity oftheir upper frequency limits.Note that both these methods of stabilizinga beam-tetrode v-h-f amplifier stage by cancellationof screen-lead inductance are suitableonly for operation over a relatively narrowband of frequencies in the v-h-f range. At lowerfrequencies both these expedients for reducingthe effects of screen-lead inductancewill tend to increase the tendency toward oscillationof the amplifier stage.NeutralizingProblemsWhen a stage cannot be completelyneutralized, the difficultyusually can be traced to one ormore of the following causes: (1) Filamentleads not by-passed to the common ground ofthat particular stage. (2) Ground lead from therotor cqnnection of the split-stator tuning capacitorto filament open or too long. (3) Neutralizingcapacitors in a field of excessiver.f. from one of the tuning coils. (4) Electromagneticcoupling between grid and platecoils, or between plate and preceding bufferor oscillator circuits. (5) Insufficient shieldingor spacing between stages, or between gridand plate circuits in compact transmitters.(6) Shielding placed too close to plate circuitcoils, causing induced currents in the shields.(7) Parasitic oscillations when plate voltageis applied. The cure for the latter is mainly amatter of cut and try-rearrange the parts,
258 Generation of R-F Energy<strong>THE</strong> R AD I 0GRIDLEAKINTERWOUNO COILS(UNITY COUPLING)f"IL.Figure 15+HVGROUNDED-GRID AMPLIFIERThis type of triode ompl ifier requires noneutralization, but can be usee/ only withtubes having o relatively low plate-to-cathodecapacitancechange the length of grid or plate or neutralizingleads, insert a parasitic choke in the gridlead or leads, or eliminate the grid r-f chokeswhich may be the cause of a low-frequencyparasitic (in conjunction with plate r-f chokes).13-7 Grounded GridAmplifiersCertain triodes have a grid configurationand lead arrangement which results in very lowplate to filament capacitance when the controlgrid is grounded, the grid acting as an effectiveshield much in the manner of the screenin a screen-grid tube.By connecting such a triode in the circuit offigure 15, taking the usual precautions againststray capacitive and inductive coupling betweeninput and output leads and components,a stable power amplifier is realized which requiresno neutralization.At ultra-high frequencies, where it is difficultto obtain satisfactory neutralization withconventional triode circuits (particularly whena wide band of frequencies is to be covered),the grounded-grid arrangement is about the onlypracticable means of employing a triode amplifier.Because of the large amount of degenerationinherent in the circuit, considerably more excitationis required than if the same tube wereemployed in a conventional grounded-cathodecircuit. The additional power required to drivea triode in a grounded-grid amplifier is notlost, however, as it shows up in the output circuitand adds to the power delivered to theload. But nevertheless it means that a largerdriver stage is required for an amplifier ofFigure 16CONVENTIONAL TRIODE FREQUENCYMULTIPLIERSmall triodes such as the 6C4 operate sotis·lactorily as frequency multipliers, and candeliver output well into the v-h-f range. ResistorR normally wilt hove o value in thevicinity of 100,000 ohms.given output, because a moderate amount ofpower is delivered to the amplifier load by thedriver stage of a grounded-grid amplifier.13-8 Frequency MultipliersQuartz crystals and variable-frequency oscillatorsare not ordinarily used for direct controlof the output of high-frequency transmitters.Frequency multipliers are usually employedto multiply the frequency to the desiredvalue. These multipliers operate on exact multiplesof the excitation frequency; a 3.6-Mc.crystal oscillator can be made to control theoutput of a transmitter on 7.2 or 14.4 Me., oron28.8 Me., by means of one or more frequencymultipliers. When used at twice frequency,they are often termed frequency doublers. Asimple doubler circuit is shown in figure 16.It consists of a vacuum tube with its plate circuittuned to twice the frequency of the griddriving circuit. This doubler can be excitedfrom a crystal oscillator or another multiplieror amplifier stage.Doubling is best accomplished by operatingthe tube with high grid bias. The grid circuitis driven approximately to the normal value ofd-e grid current through the r-f choke and gridleakresistor, shown in figure 16. The resistancevalue generally is from two to five timesas high as that used with the same tube forstraight amplification. Consequently, the gridbias is several times as high for the samevalue of grid current.Neutralization is seldom necessary in adoubler circuit, since the plate is tuned totwice the frequency of the grid circuit. Theimpedance of the grid driving circuit is verylow at the doubling frequency, and thus thereis little tendency for self-excited oscillation.
<strong>HANDBOOK</strong>Frequency Multipliers 25 9=, TANK CIRCUIT OUTPUT VOLTAGE1\ (\ /\ /\ /\1(CUTOF'F') H - - - IA ~N(CUTOFF)-- - - - - 11E G\ ,'J L\ M 0 /S U\1\../ \_J I I ID HI I I II I I IX[\yL--·t- -1p_-- -ly--- -II\ ,.___ E:XCITATION\ I VOLTAGE'' QwFigure 18ILLUSTRATING <strong>THE</strong> ACTION OF AFREQUENCY DOUBLERFigure 17FREQUENCY MULTIPLIER CIRCUITSThe output of o triocle v-h-f frequency multi·pi ier often moy be lncreasec/ by neutral /rationof the gricl-to-pl ate capacitance as shown at(A) above. Such o stage a/so may be operateclas a straight amplifier when the occasionclemancls. A pentacle frequency multiplieris shown at (B). Conventional powertetrocles operate satisfactorily as multipliersso long as the output frequency is belowabout 100 Me. Above this frequency spec/a/v-h-f tetrocles must be usee/ to obtain satis·factory output.Frequency doublers require bias of severaltimes cutoff; high·/L tubes therefore are desir·able for this type of service. Tubes whichhave amplification factors from 20 to 200 aresuitable for doubler circuits. Tetrodes andpentodes make excellent doublers. Low·/Ltriodes, having amplification constants of from3 to 10, are not applicable for doubler service.In extreme cases the grid voltage must be ashigh as the plate voltage for efficient doublingaction.Angle of Flowin FrequencyMultipliersThe angle of plate current flowin a frequency multiplier is avery important factor in deter·mining the efficiency. As theangle of flow is decreased for a given valueof grid current, the efficiency increases. Toreduce the angle of flow, higher grid bias isrequired so that the grid excitation voltagewill exceed the cutoff value for a shorter por·tion of the exciting-voltage cycle. For a highorder of efficiency, frequency doublers shouldhave an angle of flow of 90 degrees or less,triplers 60 degrees or less, and quadruplers45 degrees or less. Under these conditionsthe efficiency will be on the same order asthe reciprocal of the harmonic on which thestage operates. In other words the efficiencyof a doubler will be approximately ~ or 50per cent, the efficiency of a tripler will beapproximately ';. or 33 per cent and that of aquadrupler will be about 25 per cent. With goodstage design the efficiency can be somewhatgreater than these values, but as the angleof flow is made greater than these limitingvalues, the efficiency falls off rapidly. Thereason is apparent from a study of figure 18.The pulses ABC, EFG, JKL illustrate 180-degree excitation pulses under Class B opera·tion, the solid straight line indicating cutoffbias. If the bias is increased by N times, tothe value indicated by the dotted straight line,and the excitation increased until the peakr·f voltage with respect to ground is the sameas before, then the excitation frequency canbe cut in half and the effective excitationpulses will have almost the same shape asbefore. The only difference is that every otherpulse is missing; MNO simply shows wherethe missing pulse would go. However, if theQ of the plate tank circuit is high, it will havesufficient flywheel effect to carry over throughthe missing pulse, and the only effect will bethat the plate input and r·f output at optimumloading drop to approximately half. As the in·put frequency is half the output frequency, anefficient frequency doubler is the result.By the same token, a tripler or quadruplercan be analyzed, the tripler skipping two ex·citation pulses and the quadrupler three. Ineach case the excitation pulse ideally shouldbe short enough that it does not exceed 180degrees at the output frequency; otherwise theexcitation actually is bucking the output overa portion of the cycle.In actual practice, it is found uneconomicalto provide sufficient excitation to run a tripleror quadrupler in this fashion. Usually the ex·
260 Generation of R-F Energy<strong>THE</strong> R AD I 0EFigure 19PUSH-PUSH FREQUENCY DOUBLERThe output of a cloubler stage may be materiallyincreased through the use of a push-pushcircuit such as illustratecl above.citation pulses will be at least 90 degrees atthe exciting frequency, with correspondinglylow efficiency, but it is more practicable toaccept the low efficiency and build up the out·put in succeeding amplifier stages. The efficiencycan become quite low before the powergain becomes less than unity.Push-PushMultipliersTwo tubes can be connected inparallel to give twice the outputof a single-tube doubler. If thegrids are driven out of phase instead of inphase, the tubes then no longer work simultaneously,but rather one at a time. The effectis to fill 'in the missing pulses (figure 18).Not only is the output doubled, but severaladvantages accrue which cannot be obtainedby straight parallel operation.Chief among these is the effective neutralizationof the fundamental and all odd harmonics,an advantage when spurious emissionsmust be minimized. Another advantage is thatwhen the available excitation is low and excitationpulses exceed 90 degrees, the outputand efficiency will be greater than for thesame tubes connected in parallel.The same arrangement may be used as aquadrupler, with considerably better efficiencythan for straight parallel operation, becauseseldom is it practicable to supply sufficientexcitation to permit 45 degree excitationpulses. As pointed out above, the push-pusharrangement exhibits better efficiency than asingle ended multiplier when excitation is inadequatefor ideal multiplier operation.A typical push-push doubler is illustratedin figure 19. When high transconductance tubesare employed, it is necessary to employ asplit-stator grid tank capacitor to prevent selfoscillation; with well screened tetrodes orpentodes having medium values of transconductance,a split-coil arrangement with a single-sectioncapacitor may be employed (theFigure 20PUSH-PULL FREQUENCY TRIPLERThe push-pull tripler is advantageous in thev-h-1 range since circuit balance is main·tainecl both in the input ancl output circuits.If the circuit is neutralized it may be usee/either as a straight amplifier or as a tripler.Either triocles or tetrocles may be usee/; dualunittetrocles such as the 815, 832A, one/8298 are particularly effective in the v-h-frange.center tap of the grid coil being by-passed toground).Push-Pull FrequencyTriplersIt is frequently desirablein the case of u-h-f andv-h-f transmitters thatfrequency multiplication stages be balancedwith respect to ground. Further it is just aseasy in most cases to multiply the crystal orv-f-o frequency by powers of three rather thanmultiplying by powers of two as is frequentlydone on lower frequency transmitters. Hencethe use of push-pull triplers has become quiteprevalent in both commercial and amateurv-h-f and u-h-f transmitter designs. Such stagesare balanced with respect to ground and appearin construction and on paper essentially thesame as a push-pull r-f amplifier stage withthe exception that the output tank circuit istuned to three times the frequency of the gridtank circuit. A circuit for a push-pull triplerstage is shown in figure 20.A push-pull tripler stage has the furtheradvantage in amateur work that it can also beused as a conventional push-pull r-f amplifiermerely by changing the grid and plate coilsso that they tune to the same frequency. Thisis of some advantage in the case of operatingthe 50-Mc. band with 50-Mc. excitation, andthen changing the plate coil to tune to 144Me. for operation of the stage as a tripler fromexci$8tion on 48 Me. This circuit arrangementis excellent for operation with push-pull beamtetrodes such as the 6360 and 829B, althougha pair of tubes such as the 2£26, or 5763 couldjust as well be used if proper attention weregiven to the matter of screen-lead inductance.
<strong>HANDBOOK</strong>Tank Circuits 26113-9 Tank CircuitCapacitancesIt is necessary that the proper value of Qbe used in the plate tank circuit of any r-famplifier. The following section has been devotedto a treatment of the subject, and chartsare given to assist the reader in the determinationof the proper L/C ratio to be used in aradio-frequency amplifier stage.A Class C amplifier draws plate current inthe form of very distorted pulses of short duration.Such an amplifier is always operated intoa tuned inductance-capacitance or tank circuitwhich tends to smooth out these pulses,by its storage or tank action, into a sine waveof radio-frequency output. Any wave-form distortionof the carrier frequency results in harmonicinterference in higher-frequency channels.A Class A r-f amplifier would produce a sinewave of radio-frequency output if its excitingwaveform were also a sine wave. However, aClass A amplifier stage converts its d-e inputto r-f output by acting as a variable resistance,and therefore heats considerably. A Class Camplifier when driven hard with short pulsesat the peak of the exciting waveform acts moreas an electronic switch, and therefore can convertits d-e input to r-f output with relativelygood efficiency. Values of plate circuit efficiencyfrom 65 to 85 per cent are common inClass C amplifiers operating under optimumconditions of excitation, grid bias, and loading.Tank Circuit Q As stated before, the tank circuitof a Class C amplifierreceives energy in the form of short pulses ofplate current which flow in the amplifier tube.But the tank circuit must be able to storeenough energy so that it can deliver a currentessentially sine wave in form to the load. Theability of a tank to store energy in this mannermay be designated as the effective Q ofthe tank circuit. The effective circuit Q maybe stated in any of several ways, but essentiallythe Q of a tank circuit is the ratio of theenergy stored to 211 times the energy lost percycle. Further, the energy lost per cycle must,by definition, be equal to the energy delivere"dto the tank circuit by the Class C amplifiertube or tubes.The Q of a tank circuit at resonance is equalto its parallel resonant impedance (the resonantimpedance is resistive at resonance) dividedby the reactance of either the capacitoror the inductor which go to make up thetank. The inductive reactance is equal to thecapacitive reactance, by definition, at resonance.Hence we may state:DYNAMICCHARACTER ISTJCGRID SWINGFigure 21CLASS C AMPLIFIER OPERATIONPlate current pulses are shown at (A), (B),and (C). The dip in the top of the plate currentwaveform will occur when the excitationvoltage is such that the minimum plate voltagedips below the maximum grid voltage.A detailed discussion of the operation ofClass C amplifiers is given in Chapter Seven.where RL is the resonant impedance of thetank and Xc is the reactance of the tank capacitorand XL is the reactance of the tankcoil. This value of resonant impedance, RL,is the load which is presented to the Class Camplifier tube in a single-ended circuit suchas shown in figure 21.The value of load impedance, RL, which theClass C amplifier tube sees may be obtained,looking in the other direction from the tankcoil, from a knowledge of the operating conditionson the Class C tube. This load impedancemay be obtained from the following expression,which is true in the general case ofany Class C amplifier:2NpibEbbwhere the values in the equation have the characteristicslisted in the beginning of Chapter 6.The expression above is academic, sincethe peak value of the fundamental componentof plate voltage swing, EPm• is not ordinarilyknown unless a high-voltage peak a-c voltmeteris available for checking. Also, the decimalvalue of plate circuit efficiency is not ordinarilyknown with any degree of accuracy. However,in a normally operated Class C amplifier
262 Generation of R-F Energy<strong>THE</strong> R AD I 00•7••4'2102NDHARMONIC>RDHARMONICI'N.III[[l_t10 1!. 20TANK CIRCUIT QFigure 22RELATIVE HARMONIC OUTPUTPLOTTED AGAINST TANK CIRCUIT Qthe plate voltage swing will be approximatelyequal to 0.85 to 0.9 times the d-e plate voltageon the stage, and the plate circuit efficiencywill be from 70 to 80 per cent (Np of 0. 7 to0.8), the higher values of efficiency normallybeing associated with the higher values ofplate voltage swing. With these two assumptionsas to the normal Class C amplifier, theexpression for the plate load impedance canbe greatly simplified to the following approximatebut useful expression:Rd.c.R L!::::!---2>0which means simply that the resistance presentedby the tank circuit to the Class C tubeis approximately equal to one-half the d-e loadresistance which the Class C stage presentsto the power supply (and also to the modulatorin case high-level modulation of the stage isto be used) .Combining the above simplified expressionfor the r-f impedance presented by the tank tothe tube, with the expression for tank Q givenin a previous paragraph we have the followingexpression which relates the reactance of thetank capacitor or coil to the d-e input to theClass C stage:Xc = X t""J Rd.c.L----2QThe above expression is the basis of theusual charts giving tank capacitance for thevarious bands in terms of the d-e plate voltageand current to the Class C stage, includingthe charts of figure 23, figure 24 and figure 25.Harmonic Radiotionvs. QThe problem of harmonicradiation from transmittershas long been present, butit has become critical only relatively recentlyalong with the extensive occupation of thev-h-f range. Television signals are particularlysusceptible to interference from other signalsfalling within the pass band of the receiver,so that the TVI problem has received the majoremphasis of all the services in the v-h-f rangewhich are susceptible to interference fromharmonics of signals in the h-f or lower v-h-frange.ucia:10~ \1\\ 1\ 1\Q=l21\ f\\1\\~ ..> __'t'_000IV..""" " ~000\ 'I\~ ~ ~1\ \000\ l"... ~ \~~\~~I!t 10 20 50 100 200 500 1000 2000TOTAL CAPACITANCE ACROSS LC CIRCUIT (Ci)~rTF-B ® +SG +B©-c +B-c tB©Figure 23PLATE-TANK CIRCUIT ARRANGEMENTSShown above In the case of each of the tonk circuit tyl?es is the recommenc/ec/ tank circuit ca·pacitance. (A) is a conventional tetroc/e amplifier, (8} is a coil-neutrallzec/ triocle amplifier,(C) is a grounc/ec/-gric/ triocle amplifier, (D) is a gric/-neutra/izec/ trioc/e amplifier.
<strong>HANDBOOK</strong>Tank Circuits 263II2010000000•l\\ \\'ol~~ \ Q=121\ 1\ 1\"' \ I~\ :\\ \ _\[\.\ \_:\ 1\ \\ \1\ i\ \1\ 1\u0\IX1\1\II Z 3 5 7 10 20 30 50 100 200 500 1000CORRECT VALUES OF TANK CIRCUIT CAPACITANCE (C) FOROPERATING Q OF 12 WITH SINGLE-ENDED SPLIT TANK COILSB+RFC®Figure 24PLATE-TANK CIRCUIT ARRANGEMENTSShown above for each of the tank circuit types is the recommenclecl tank circuit capacitance atthe operating frequency for an operating Q of 12. (A) is a split-stator tank, each section of whichis twice the capacity value reac/ on the graph. (B) is circuit using tappec/ coil for phase reversal.Inspection of figure 22 will show quicklythat the tank circuit of a Class C amplifiershould have.an operating Q of 12 or greaterto afford satisfactory rejection of second harmonicenergy. The curve begins to straightenout above a Q of about 15, so that a considerableincrease in Q must be made before an appreciablereduction in second-harmonic energyis obtained. Above a circuit Q of about 10 anyincrease will not afford appreciable reductionin the third-harmonic energy, so that additionalharmonic filtering circuits external to the amplifierproper must be used if increased attenuationof higher order harmonics is desired.The curves also show that push-pull amplifiersmay be operated at Q values of 6 or so, sincethe second harmonic is cancelled to a largeextent if there is no unbalanced coupling betweenthe output tank circuit and the antennasystem.Capacity Charts forCorrect Tank QFigures 23, 24 and 25 illustratethe correct valueof tank capacity for variouscircuit configurations. A Q value of 12has been chosen as optimum for single endedcircuits, and a value of 6 has been chosen forpush-pull circuits. Figure 23 is used when asingle ended stage is employed, and the capacitancevalues given are for the total capacitanceacross the tank coil. This value indudesthe tube interelectrode capacitance(plate to ground), coil distributed capacitance,wiring capacities, and the value of any lowinductanceplate-to-ground by-pass capacitoras used for reducing harmonic generation, inaddition to the actual "in-use" capacitanceof the plate tuning capacitor. Total circuitstray capacitance may vary from perhaps 5micromicrofarads for a v-h-f stage to 30 micromicrofaradsfor a medium power tetrode h-fstage.When a split plate tank coil is employed inthe stage in question, the graph of figure 24should be used. The capacity read from thegraph is the total capacity across the tankcoil. If the split-stator tuning capacitor isused, each section of the capacitor shouldhave a value of capacity equal to twice thevalue indicated by the graph. As in the caseof figure 23, the values of capacity read onthe graph of figure 24 include all residual circuitcapacities.For push-pull operation, the correct valuesof tank circuit capacity may be determinedwith the aid of figure 25. The capacity valuesobtained from figure 25 are the effective valuesacross the tank circuit, and if a split-statortuning capacitor is used, each section of thecapacitor should have a value of capacity equal to twice the value indicated by the graph.As in the case of figures 23 and 24, the valuesof capacity read on the graph of figure 25 includeall residual circuit capacities.The tank circuit operates in the same mannerwhether the tube feeding it is a pentode,beam tetrode, neutralized triode, groundedgridtriode, whether it is single ended or push-
264Generation of R-FEnergy<strong>THE</strong> R AD I 0UJIUUJUJ\:(~..J..J(l.(l.J0a:00001\\ \r-. \o=s5000 \[\1\00008000llr--;-E1 p f.- ~-\ [\6000 \32000\\ \ i\ r-.·~
<strong>HANDBOOK</strong>L and Pi Networks 265FIGURE 26USUAL BREAKDOWN RATINGS OFCOMMON PLATE SPACINGSAir-gap ininches.030.050.070.100.125.150.170.200.250.350.500.700Peak voltagebreakdown1,0002,0003,0004,0004,5005,2006,0007,5009,00011,00015,00020,000Recommended air-gap for use when no d-evoltage appears across plate tank condenser(when plate circuit is shunt fed, or when theplate tank condenser is insulated fromground).D.C. PLATEPLATEVOLTAGE c.w. MOD.400 .030 .050600 .050 . 070750 .050 .0841000 .070 .1001250 .070 .1441500 .078 .2002000 .100 .2502500 .175 .3753000 .200 .5003500 .250 .600Spacings should be multiplied by 1.5 forsame safety factor when d-e voltage appearsacross pi ate tank condenser.These rules apply to a loaded amplifier orbuffer stage. If either is operated without anr-f load, the peak voltages will be greater andcan exceed the d-e plate supply voltage. Forthis reason no amplifier should be operatedwithout load when anywhere near normal d-eplate voltage is applied.If a plate blocking condenser is used, itmust be rated co withstand the d-e plate voltageplus any audio voltage. This capacitorshould be rated at a d-e working voltage of atleast twice the d-e plate supply in a plate modulatedamplifier, and at least equal to the d-esupply in any other type of r-f amplifier.13-10 L and Pi MatchingNetworksThe L and pi networks often can be put toadvantageous ~se in accomplishing an impedancematch between two differing impedances.Common applications are the matching betweena transmission line and an antenna, or betweenthe plate circuit of a single-ended amplifierstage and an antenna transmission line. Suchnetworks may be used to accomplish a matchRFCLRP = Q2 RA (APPROX)RP C RA Q=~~=~; =~~ =~:XL=Xc+B -=PLATE CURRENTRP= 225 RAFOR OPERATING CIRCUITQ OF 15 Xc = ~;XL=~Figure 27<strong>THE</strong> L NETWORK IMPEDANCETRANSFORMERThe L network is useful with a moderateoperating Q for high values of impedancetransformation, ancl it may be used for appli~cations other than in the plate circuit of atube with relatively low values of operatingQ for moderate impedance transformations.Exact and approximate design equations aregiven .between the plate tank circuit of an amplifierand a transmission line, or they may be usedto match directly from the plate circuit of anamplifier to the line without the requirementfor a tank circuit-provided the network is designedin such a manner that it has sufficientoperating Q for accomplishing harmonic attenuation.The L MatchingNetworkThe L network is of limitedutility in impedance matchingsince its ratio of impedancetransformation is fixed at a value equalto (Q 2 +1). The operating Q may be relativelylow (perhaps 3 to 6) in a matching network betweenthe plate tank circuit of an amplifierand a transmission line; hence impedancetransformation ratios of 10 to 1 and even lowermay be attained. But when the network alsoacts as the plate tank circuit of the amplifierstage, as in figure 27, the operating Q shouldbe at least 12 and preferably 15. An operatingQ of 15 represents an impedance transformationof 225; this value normally will be toohigh even for transforming from the 2000 to10,000 ohm plate impedance of a Class C amplifierstage down to a 50-ohm transmissionline.However, the L network is interesting sinceit forms the basis of design for the pi network.Inspection of figure 27 will show that the Lnetwork in reality must be considered as aparallel-resonant tank circuit in which RArepresents the coupled-in load resistance;only in this case the load resistance is di·reedy coupled into the tank circuit rather thanbeing inductively coupled as in the conven--
266 Generation of R-F Energy<strong>THE</strong> R AD I 0Ro.c.+B(Ebb)Ro.c. = ~ ~b X =-RA I RPC 2 RA(Q2+1 )- RPFigure 28<strong>THE</strong> PI NETWORKThe pi network is valuable for use as an im·pee/once transformer over a w I cl e ratio oftransformation values. The operating Q shoulclbe at least 12 ancl preferably 15 to 20 whenthe circuit is to be usee/ in the plate circuitof a Class C amp/ ifier. Design equationsare given above. The inductor Ltot representsa single incluctance, usually variable,with a value equal to the sum of Lt ancl L2.tiona! arrangement where the load circuit iscoupled to the tank circuit by means of a link.When RA is shorted, L and C comprise a con·ventional parallel-resonant tank circuit, sincefor proper operation L and C must be resonantin order for the network to present a resistiveload to the Class C amplifier.The Pi Network The pi impedance matchingnetwork, illustrated in figure28, is much more general in its applicationthan the L network since it offers greater harmonicattenuation, and since it can be usedto match a relatively wide range of impedanceswhile still maintaining any desired operatingQ. The values of C 1 and L 1 in the pi networkof figure 28 can be thought of as having thesame values of the L network in figure 27 forthe same operating Q, but what is more importantfrom the comparison standpoint these valueswill be the same as in a conventional tankcircuit.The value of the capacitance may be determinedby calculation, with the operating Q andthe load impedance which should be reflectedto the plate of the Class C amplifier as thetwo knowns-or the actual values of the ca-pacitance may be obtained for an operating Qof 12 by reference to figures 23, 24 and 25.The inductive arm in the pi network can bethought of as consisting of two inductancesin series, as illustrated in figure 28. The firstportion of this inductance, L 1 , is that value ofinductance which would resonate with C, atthe operating frequency-the same as in a conventionaltank circuit. However, the actualvalue of inductance in this arm of the pi net·work, L, 0, will be greater than L, for normalvalues of impedance transformation. For hightransformation ratios r_ will be only slightlygreater than L 1; for a transformation ratio of1.0, L, , 0will be twice as great ao; L,. Theamount of inductance which must be added toL 1to restore resonance and maintain circuitQ is obtained through use of the expressionfor X., 2in figure 28.The peak voltage rating of the main tuningcapacitor C 1should be the normal value for aClass C amplifier operating at the plate voltageto be employed. The inductor Ltot may bea plug-in coil which is changed for each bandof operation, or some sort of variable inductormay be used. A continuously variable slider·type of variable inductor, such as used in certainitems of surplus military equipment, maybe used to good advantage if available, or atapped inductor such as used in the ART-13may be employed. However, to maintain goodcircuit Q on the higher frequencies when avariable or tapped coil is used on the lowerfrequencies, the tapped or variable coil shouldbe removed from the circuit and replaced bya smaller coil which has been especially designedfor the higher frequency ranges.The peak voltage rating of the output orloading capacitor, C 2, is determined by thepower level and the impedance to be fed. If a50-ohm coaxial line is to be fed from the pinetwork, receiving-type capacitors will besatisfactory even up to the power level of aplate-modulated kilowatt amplifier. In anyevent, the peak voltage which will be impressedacross the output capacitor is ex·pressed by: Epk 2 = 2 Ra W 0 , where Epk is thepeak voltage across the capacitor, R 0is thevalue of resistive load which the network isfeeding, and W 0is the maximum value of theaverage power output of the stage. The har·monic attenuation of the pi network is quitegood, although an external low-pass filter willbe required to obtain harmonic attenuationvalue upward of 100 db such as normally required.The attenuation to second harmonicenergy will be approximately 40 db for an oper·ating Q of 15 for the pi network; the valueincreases to about 45 db for a 1:1 transforma·tion and falls to about 38 db for an impedancestep-down of 80:1, assuming that the operatingQ is maintained at 15.
<strong>HANDBOOK</strong> Grid Bias 267+B~RFC1:t:CeLQ' IU:;~ RFC2 ~ ? ~C2 ~xoAxl-B.,.PLATE LOAD (OHMS) z~~BlWHERE EB IS PLATE VOL TAG£AND I 8 IS PLATE CURRENT!NA,NPI!IiES.CB- .00025JJF. MICA CAPACITOR RATED AT TWICE <strong>THE</strong> D.C.PLATE VOLTAGE.OUTPUTRFC I -N° 28 ENAMELED, CLOSE-WOUND ON A CERAMIC INSULATORI"OIA., 4"LONG OR NATIONAL R-175ARFC2.- 2j MH, NATIONAL R-IOOEstimated Plateload (ohms) 1,000 1,500 2,000 2,500 3,000 3,500 4,000 4,500 s,ooo 6,ooo• NOTESC1 in p.!.J.f, 3.5 Me 520 360 280 210 laO 155 135 120 110 90 The actual capacitance setting7 260 laO 140 lOS 90 76 6a 60 S6 45 for cl equals the value in this14 130 90 70 52 4S 3a 34 30 2a 23 table minus the published tube21 as 60 47 35 31 25 23 20 19 lS output capacitance. Air gap2a 65 45 35 26 23 19 17 lS 14 11 approx. 10 mils/100 v En.L in p.h, 3.5 Me 4.5 6.S a.s 10.5 12.5 14 lS.S la 20 25 Inductance values are lor a7 2.2 3.2 4.2 5.2 6.2 7 7.a 9 10 12.5 SO-ohm load. For a 70·ohm14 1.1 1.6 2.1 2.6 3.1 3.5 3.9 4.5 s 6.2 load, values are approx. 3%21 0.73 l.Oa 1.3a 1.7 2.05 2.3 2.6 3 3.3 -4.1 higher.2a o.ss o.a l.OS 1.2a l.SS 1.7 1.9S 2.25 2.5 3.1C2 in f..L/.Lf, 3.5 Me 2,400 2,100 1,aoo l,SSO 1,400 1,250 1,100 1,000 900 700 For 50-ohm transmission line.7 1.200 1,060 900 760 700 630 S60 soo 460 350 Air gap for C2 is approx. 114 600 530 -450 3ao 350 320 2aO 250 230 175 mil/100 v E~..21 400 350 300 250 230 210 las 165 lSS 1202a 300 265 225 190 175 160 140 125 115 90C2 in ~J.)J..f, 3.5 Me l,aOO 1,500 1,300 1,100 1,000 900 aoo 720 640 soo For 70-ohm transmission line.7 900 7SO 6SO 560 soo 450 400 360 320 25014 450 370 320 2aO 250 220 200 laO 160 12521 300 250 215 190 170 145 130 120 110 as2a 225 las 160 140 125 110 100 90 ao 6S• Values given are approximations. All components shown in Table I are for a Q of 12. For other values of Q, useQa Ca Qa lt,:-:-:: - and - = -· When the estimated plate load is higher than 5,000 ohms, it is recommended that theQ,, C,, Qh .Lacomponents be selected for a circuit Q between 20 and 30.Table I Components for Pi-Coupled Final AmplifiersComponent Chartfor Pi-NetworksTo simplify design procedure,a pi-network chart,compiled by M. Seybold,W2RYI (reproduced by courtesy of R.C.A. TubeDivision, Harrison, N.J.) is shown in table 1.This chart summarizes the calculations of figure28 for various values of plate load.13-11 Grid BiasRadio-frequency amplifiers require someform of grid bias for proper operation. Practicallyall r-f amplifiers operate in such a mannerthat plate current flows in the form ofshort pulses which have a duration of only afraction of an r-f cycle. To accomplish thiswith a sinusoidal excitation voltage, the operatinggrid bias must be at least sufficient tocut off the plate current. In very high efficiencyClass C amplifiers the operating bias maybe many times the cutoff value. Cutoff bias,it will be recalled, is that value of grid voltagewhich will reduce the plate current tozero at the plate voltage employed. The methodfor cal"culating it has been indicated previously.This theoretical value of cutoff will notreduce the plate current completely to zero,due to the variable-11 tendency or "knee"which is characteristic of all tubes as thecutoff point is approached.Class C Bias Amplitude modulated Class Camplifiers ·should be operatedwith the grid bias adjusted to a value greaterthan twice cutoff at the operating plate volt-
268 Generation of R-F Energy41'''""'"'"~~R1IFigure 29GRID-LEAK BIASThe grid leak on an amplifier or multiplierstage may a/ so be used as the shunt feedimpedance to the grid of the tube when ahigh value of grid leak (greater than perhaps20,000 ohms) is used. When a lower value ofgrid leak is to be employed, an r-f chokeshould be used between the grid of the tubeand the grid leak to reduce r·f losses in thegrid leak resistance.age. This procedure will insure that the tubeis operating at a bias greater than cutoff whenthe plate voltage is doubled on positive modu·lation peaks. C-w telegraph and FM transmitterscan be operated with bias as low ascutoff, if only limited excitation is availableand moderate plate efficiency is satisfactory.In a c-w transmitter, the bias supply or re·sistor should be adjusted to the point whichwill allow normal grid current to flow for theparticular amount of grid driving r-f poweravailable. This form of adjustment will allowmore output from the under-excited r·f ampli·fier than when higher bias is used with correspondinglower values of grid current. In anyevent, the operating bias should be set at aslow a value as will give satisfactory opera·tion, since harmonic generation in a stage in·creases rapidly as the bias is increased.Grid·Leak Bias A resistor can be connectedin the grid circuit of a ClassC amplifier to provide grid-leak bias. This resistor,R, in figure 29, is part of the d·c pathin the grid circuit.The r-f excitation applied to the grid cir·cuit of the tube causes a pulsating direct cur·rent to flow through the bias supply lead, dueto the rectifying action of the grid, and anycurrent flowing through R, produces a voltagedrop across that resistor. The grid of the tubeis positive for a short duration of each r-fcycle! and draws electrons from the filamentor cathode of the tube during that time. Theseelectrons complete the circuit through the d·cgrid return. The voltage drop across the re·sistance in the grid return provides a nega·tive bias for the grid.Grid-leak bias automatically adjusts itselfover fairly wide variations of r·f excitation.The value of grid·leak resistance should besuch that normal values of grid current willflow at the maximum available amount of r-f-BIAS SUPPLY<strong>THE</strong> R AD I 0Figure 30COMBINATION GRID·L EAK ANDFIXED BIASGrid-leak bios often is used in conjunctionwith a fixed minimum value of power supplybias. This arrangement permits the operatingbias to be established by the excitation ener.gy, but in the absence of excitation the elec·trode currents to the tube will be held to safevalues by the fixed-minimum power supplybias. If a relatively low value of grid leakis to be used, on r·f choke should be connectedbetween the grid of the tube and thegrid leak as discussed in figure 29.excitation. Grid-leak bias cannot be used forgrid-modulated or linear amplifiers in whichthe . aver~ge d·c grid current is constantlyvarymg with modulation.Safety Bios Grid-leak bias alone provides noprotection against ex c e s s i v eplate current in case of failure of the sourceof r·f grid excitation. A C-battery or C-biassupply can be connected in series with thegrid leak, as shown in figure 30. This fixedprotective bias will protect the tube in theevent of failure of grid excitation. "Zero-bias"tubes do not require this bias source in addi·tion to the grid leak, since their plate currentwill drop to a safe value when the excitationis removed.Cathode Bios A resistor can be connected inseries with the cathode or cen·ter·tapped filament lead of an amplifier to secureautomatic bias. The plate current flowsthrough this resistor, then back to the cathodeor filament, and the voltage drop across theresistor can be applied to the grid circuit byconnecting the grid bias lead to the groundedor power supply end of the resistor R as shownin figure 31.'The grounded (B·minus) end of the cathoderesistor is negative relative to the cathodeby an amount equal to the voltage drop acrossthe resistor. The value of resistance must beso chosen that the sum of the desired gridand plate current flowing through the resistorwill bias the tube for proper operation.. Thi~ type of bias is used more extensivelyIn. a.udio·frequency than in radio-frequency amphfters.The voltage drop across the resistor
<strong>HANDBOOK</strong>\iJDRIVERI FROM---RFCCFigure 31R-F STAGE WITH CATHODE BIASCathode bias sometimes is advantageous foruse in an r·f stage that operates with a relativelysmall amount of r-f excitation.RProtective Circuits 26 9FROM-ju--ORIVERRFC- +.JII _BATTERY-Figure 32R-F STAGE WITHBATTERY BIASBattery bias is seldom usecl, clue to cleteriora•tion of the cells by the reverse gricl current.However, it may be usee/ in certain specialapplications, or the fixed bias voltage maybe supplied by a bias power supply.must be subtracted from the total plate supplyvoltage when calculating the power input tothe amplifier, and this loss of plate voltagein an r·f amplifier may be excessive. A ClassA audio amplifier is biased only to approximatelyone-half cutoff, whereas an r-f amplifiermay be biased to twice cutoff, or more, andthus the plate supply voltage loss may be alarge percentage of the total available voltagewhen using low or medium f-t tubes.Oftentimes just enough cathode bias is employedin an r-f amplifier to act as safety biasto protect the rubes in case of excitation failure,with the rest of the bias coming from agrid leak.Separate BiasSupplyAn external sup p 1 y often isused for grid bias, as shown infigure 32. Battery bias givesvery good voltage regulation and is satisfactoryfor grid-modulated or linear amplifiers,whtch operate at low grid current. In the caseof. Class C amplifiers which operate with highgnd current, battery bias is not satisfactory.This direct current has a charging effect onthe dry batteries; after a few months of servicethe cells will become unstable, bloated, andnoisy.A separate a-c operated power supply iscommonly used for grid bias. The bleeder resistanceacross the output of the filter can bemade sufficiently low in value that the gridcurrent of the amplifier will not appreciablychange the amount of negative grid-bias voltage.Alternately, a voltage regulated grid-biassupply can be used. This type of bias supplyis used in Class B audio and Class B r-f linearamplifier service where the voltage regulationin the C-bias supply is important. Fora Class C amplifier, regulation is not so important,and an economical design of componentsin the power supply, therefore, can beutilized. In this case, the bias voltage mustbe adjusted with normal grid current flowing,as the grid current will raise the bias con-siderably when it is flowing through the biassupplybleeder resistance.13-12 Protective Circuits forTetrode Transmitting TubesThe tetrode transmitting tube requires threeoperating voltages: grid bias, screen voltage,and plate voltage. The current requirements ofthese three operating voltages are somewhatinterdependent, and a change in potential ofone voltage will affect the current drain of thetetrode in respect to the other two voltages.In particular, if the grid excitation voltage isinterrupted as by keying action, or if the platesupply is momentarily interrupted, the resultingvoltage or current surges in the screen circuitare apt to permanently damage the tube.The Series ScreenSupplyA simple method of obtainingscreen voltage is bymeans of a dropping resistorfromthe high voltage plate supply, as shownin figure 33. Since the current drawn by thescreen is a function of the exciting voltageapplied to the tetrode, the screen voltage willrise to equal the plate voltage under conditionsof no exciting voltage. If the control gridis overdriven, on the other hand, the screencurrent may become excessive. In either case,damage to the screen and its associated componentsmay result. In addition, fluctuationsin the plate loading of the tetrode stage willcause changes in the screen current of thetube. This will result in screen voltage fluctuationsdue to the inherently poor voltageregulation of the screen series dropping resistor.These effects become dangerous to tubelife if the plate voltage is greater than thescreen voltage by a factor of 2 or so.
270 Generation of R-F Energy<strong>THE</strong>R AD I 0E+BFigure 33DROPPING· RESISTOR SCREEN SUPPLYNEGATIVE{OPERATING8/ASCUTSOFF CLAMPTUBE ':'"The Clamp Tube A clamp tube may be addedto the series screen supply,as shown in figure 34. The clamp tube is nor·mally cut off by virtue of the d-e grid bias dropdeveloped across the grid resistor of the tet·rode tube. When excitation is removed fromthe tetrode, no bias appears across the gridresistor, and the clamp tube conducts heavily,dropping the screen voltage to a safe value.When excitation is applied to the tetrode theclamp tube is inoperative, and fluctuations ofthe plate loading of the tetrode tube couldallow the screen voltage to rise to a damagingvalue. Because of this factor, the clamp tubedoes not offer complete protection to the tet·rode.The SeparateScreen SupplyA low voltage screen supplymay be used instead of theseries screen dropping resis·tor. This will protect the screen circuit fromexcessive voltages when the other tetrodeoperating parameters shift. However, the screencan be easily damaged if plate or bias voltageis removed from the tetrode, as the screencurrent will reach high values and the screendissipation will be exceeded. If the screensupply is capable of providing slighdy morescreen voltage than the tetrode requires forproper operation, a series wattage-limiting resistormay be added to the circuit as shownin figure 35. With this resistor in the circuitit is possible to apply excitation to the tet·rode tube with screen voltage present (but inthe absence of plate voltage) and still not damagethe screen of the tube. The value of theresistor should be chosen so that the productof the voltage applied to the screen of thetetrode times the screen current never exceedsthe maximum rated screen dissipation of thetube.13-13 lnterstage Coup I ingEnergy is usually coupled from one circuitof a transmitter into another either by cap acitivecoupling, inductive coupling, or link cou·Figure 34CLAMP-TUBE SCREEN SUPPLYpling. The latter is a special form of inductivecoupling. The choice of a coupling methoddepends upon the purpose for which it is tobe used.CapacitiveCaupl ingCapacitive coupling between anamplifier or doubler circuit and apreceding driver stage is shownin figure 36. The coupling capacitor, C, iso·lates the d-e plate supply from the next gridand provides a low impedance path for the r·fenergy between the tube being driven and thedriver tube. This method of coupling is simpleand economical for low powe& amplifier or exciterstages, but has certain disadvantages,particularly for high frequency stages. Thegrid leads in an amplifier should be as shortas possible, but this is difficult to attain inthe physical arrangement of a high power am·plifier with respect to a capacitively-coupleddriver stage.Disadvantages ofCapacitiveCouplingOne significant disadvantageof capacitive couplingis the difficulty of adjustingthe load on the driver stage.Impedance adjustment can be accomplishedby tapping the coupling lead a part of the waydown on the plate coil of the tuned stage ofthe driver circuit; but often when this is doneSERIES RESISTOR -LOW VOLTAGESCREEN SUPPLYFigure 35A PROTECTIVE WATTAGE-LIMITING RESISTOR FOR USE WITH LOW-VOLTAGESCREEN SUPPLY+ B
<strong>HANDBOOK</strong>lnterstage Coupling 271Figure 36CAPACITIVE INTERSTAGE COUPLINGa parasitic oscillation will take place in thestage being driven.One main disadvantage of capacitive coupl·ing lies in the fact that the grid-to-filamentcapacitance of the driven tube is placed directlyacross the driver tuned circuit. Thiscondition sometimes makes the r-f amplifierdifficult to neutralize, and the increased minimumcircuit capacitance makes it difficult touse a reasonable size coil in the v-h-f range.Difficulties from this source can be partiallyeliminated by using a center-tapped or splitstatortank circuit in the plate of the driverstage, and coupling capacitively to the oppo·site end from the plate. This method placesthe plate-to-filament capacitance of the driveracross one-half of the rank and the grid-to·filament capacitance of the following stageacross the other half. This type of coupling isshown in figure 37.Capacitive coupling can be used to advan·tage in reducing the total number of tuned circuitsin a transmitter so as to conserve spaceand cost. It also can be used to advantage be·tween stages for driving beam tetrode or penrodeamplifier or douhler stages.InductiveCouplingInductive coupling (figure 38) re·suits when two coils are electro·magnetically coupled to one an·other. The degree of coupling is controlled byvarying the mutual inductance of the two coils,which is accomplished by changing the spac·ing or the relationship between the axes ofthe coils.Figure 37BALANCED CAPACITIVE COUPLINGBalancecl capacitive coupling sometimes isuseful when it is clesirable to use a rei ativelylarge inc/uctonce in the interstage tank circuit,or wltere the exciting stage is neutraf ...izecl as shown above.Inductive coupling is used extensively forcoupling r·f amplifiers in radio receivers. However,the mechanical prohlems involved in adjustingthe degree of coupling limit the use·fulness of direct inductive coupling in transmitters.Either the primary or the secondaryor both coils may be tuned.Unity Coupling If the grid tuning capacitor offigure 38 is removed and thecoupling increased to the maximum practicablevalue by interwinding the turns of the two coils,the circuit insofar as r.f. is concerned actslike that of figure 36, in which one tank servesboth as plate tank for the driver and grid tankfor the driven stage. The inter-wound gridwinding serves simply to isolate the d-e platevoltage of the driver from the grid of the drivenstage, and to provide a return for d-e grid cur·rent. This type of coupling, illustrated in figure39, is commonly known as unity coupling.Because of the high mutual inductance, bothprimary and secondary are resonated by theone tuning capacitor.INTE'RWOUNDFigure 38INDUCTIVE INTERSTAGE COUPLINGFigure 39"UNITY" INDUCTIVE COUPLINGDue to the high value of coupling betweenthe two coils, one tuning capacitor tunesboth circuits, This a"angement olten is usefulIn coupling from a s/ngle-enclecl to a pushpullstage.
272 Generation of R-F Energy<strong>THE</strong> R AD I 0F~~3U1eLINK COUPLING liifAT "COLO" ENOS.UPPER ENDS "HOT"e 3LINK COUPLINGAT "COLD" CENTER.ENDS "HOT"Figure 40INTERSTAGE COUPLING BY MEANSOF A "LINK"Link interstage coupling is very commonlyused since the two stages may be separatedby a consiclerable distance, since the amountof a coup/ ing between the two stages may beeasily varied, and since the capacitances ofthe two stages may be isolated to permit useof larger incluctances in the v-h-f range.Link Coup I ing A special form of inductivecoupling which is widely employedin radio transmitter circuits is knownas link coupling. A low impedance r-f transmissionline couples the two tuned circuitstogether. Each end of the line is terminatedin one or more turns of wire, or links, woundaround the coils which are being coupled together.These links should be coupled to eachtuned circuit at the point of zero r-f potential,or nodal point. A ground connection to oneside of the link usually is used to reduce harmoniccoupling, or where capacitive couplingbetween two circuits must be minimized. Coaxialline is commonly used to transfer energybetween the two coupling links, although TwinLead may be used where harmonic attenuationis not so important.Typical link coupled circuits are shown infigures 40 and 41. Some of the advantages oflink coupling are the following:(1)(2)(3)(4)(5)(6)It eliminates coupling taps on tuned circuits.It permits the use of series power supplyconnections in both tuned grid and tunedplate circuits, and thereby eliminates theneed of shunt-feed r-f chokes.It allows considerable separation betweentransmitter stages without appreciabler-f losses or stray chassis currents.It reduces capacitive coupling and therebymakes neutralization more easily attainablein r-f amplifiers.It provides semi-automatic impedancematching between plate and grid tunedcircuits, with the result that greater griddrive can be obtained in comparison tocapacitive coupling.It effectively reduces the coupling of harmonicenergy.Figure 41PUSH-PULL LINK COUPLINGThe link-coupling line and links can bemade of no. 18 push-back wire for couplingbetween low-power stages. For coupling betwe.enhigher powered stages the 150-ohmTwtn-Lead transmission line is quite effectiveand has very low loss. Coaxial transmission ismost satisfactory between high powered amplifierstages, and should always be usedwhere harmonic attenuation is important.13-14 Radio-FrequencyChokesRadio-frequency chokes are connected incircuits for the purpose of stopping the passageof r-f energy while still permitting a directcurrent or audio-frequency current to pass.They consist of inductances wound with alarge number of turns, either in the form of asolenoid,. a s.eri~s of solenoids, a single universalpte wmdtng, or a series of pie windings.These inductors are designed to have asmuch inductance and as little distributed orshunt capacitance as possible. The unavoidablesmall amount of distributed capacitanceresonates the inductance, and this frequencynormally should be much lower than the frequencyat which the transmitter or receivercircuit is operating. R-f chokes for operationon several bands must be designed carefullyso that the impedance of the choke will be extremelyhigh (several hundred thousand ohms)in each of the bands.The direct current which flows through ther-f choke largely determines the size of wireto be used in the winding. The inductance ofr-f chokes for the v-h-f range is much lessthan for chokes designed for broadcast andordinary short-wave operation. A very highinductance r-f choke has more distributed capacitancethan a smaller one, with the result
<strong>HANDBOOK</strong>Shunt and Series Feed 2 73-1~RFC! -}PARALLEL PLATE FEED+SG +HVSERIES PLATE FEEDFigure 42ILLUSTRATING PARALLEL ANDSERIES PLATE FEEDParallel plate feecl is desirable fram a safetystandpoint since the tank circuit is at groundpotential with respect to cl.c. However, ahigh-impedance r-f choke is required, anclthe r-f choke must be able to withstand thepeak r-f voltage output of the tube. Seriesplate feed eliminates the requirement for ahigh-performance r-f choke, but requires theuse of a relatively large value of by-passcapacitance at the bottom encl of the tankcircuit, as contrasted to the moderate valueof coupling capacitance which may be usee/at the top of the tank circuit for parallelplate feecl.that it will actually offer less impedance atvery high frequencies.Another consideration, just as important asthe amount of d.c. the winding will carry, isthe r-f voltage which may be placed acrossthe choke without its breaking down. This isa function of insulation, turn spacing, frequency,number and spacing of pies and other factors.Some chokes which are designed to have ahigh impedance over a very wide range of frequencyare, in effect, really two chokes: au-h-f choke in series with a high-frequencychoke. A choke of this type is polarized; thatis, it is important that the correct end of thecombination choke be connected to the "hot"side of the circuit.Shunt andSeries FeedDirect-current grid and p 1 ateconnections are made either byseries or parallel feed systems.Simplified forms of each are shown in figures42 and 43.Series feed can be defined as that in whichthe d-e connection is made to the grid or platecircuits at a point of very low r-f potential.Shunt feed always is made to a point of highr-f voltage and always requires a high impedancer-f choke or a relatively high resistanceto prevent waste of r-f power.-BIASPARALLEL BIAS FEED13-15-BIASFigure 43ILLUSTRATING SERIES ANDPARALLEL BIAS FEEDParallel andPush-Pull Tube CircuitsSERIES BIAS FEEDThe comparative r-f power output from parallelor push-pull operated amplifiers is the sameif proper impedance matching is accomplished,if sufficient grid excitation is available inboth cases, and if the frequency of measurementis considerably lower than the frequencylimit of the tubes.ParallelOperationOperating tubes in par a 11 e I hassome advantages in transmittersdesigned for operation below 10Me., particularly when tetrode or pen rode tubesare to be used. Only one neutralizing capacitoris required for parallel operation of triodetubes, as against two for push-pull. Aboveabout 10 Me., depending upon the tube type,parallel tube operation is not ordinarii y recommendedwith triode tubes. However, paralleloperation of grounded-grid stages and stagesusing low-C beam tetrodes often will give excellentresults well into the v-h-f range.Push-Pull The push-pull connection providesOperation a well-balanced circuit insofar asmiscellaneous capacitances areconcerned; in addition, the circuit can be neutralizedmore completely, especially in highfrequencyamplifiers. The L/C ratio in a pushpullamplifier can be made higher than in aplate-neutralized parallel-tube operated amplifier.Push-pull amplifiers, when perfectlybalanced, have less second-harmonic outputthan parallel or single-tube amplifiers, but inpractice undesired capacitive coupling andcircuit unbalance more or less offset the theoreticalharmonic-reducing advantages of pushpullr·f circuits.
CHAPTER FOURTEENR-f feedbackComparatively high gain is required in singlesideband equipment because the signal isusually generated at levels of one watt or less.To get from this level to a kilowatt requiresabout 30 db of gain. High gain tetrodes maybe used to obtain this increase with a minimumnumber of stages and circuits. Each stage contributessome distortion; therefore, it is goodpractice to keep the number of stages to aminimum. It is generally considered good practiceto operate the low level amplifiers belowtheir maximum power capability in order toconfine most of the distortion to the last twoamplifier stages. R-f feedback can then beutilized to reduce the distortion in the lasttwo stages. This type of feedback is no differentfrom the common audio feedback usedin high fidelity sound systems. A sample ofthe output waveform is applied to the amplifierinput to correct the distortion developedin the amplifier. The same advantages can beobtained at radio frequencies that are obtainedat audio frequencies when feedback is used.14-1 R-F FeedbackCircuitsR-f feedback circuits have been developedby the Collins Radio Co. for use with linearamplifiers. Tests with large receiving and smalltransmitting tubes showed that amplifiers usingthese tubes without feedback developedsignal-to-distortion ratios no better than 30 dbor so. Tests were run employing cathode followercircuits, such as shown in figure lA.Lower distortion was achieved, but at the costof low gain per stage. Since the voltage gainthrough the tube is less than unity, all gainhas to be achieved by voltage step-up in thetank circuits. This gain is limited by the dissipationof the tank coils, since the circuitcapacitance across the coils in a typical transmitteris quite high. In addition, the tuningof such a stage is sharp because of the highQ circuits.The cathode follower performance of thetube can be retained by moving the r-f groundB+~u~q1 OEourfHAS -:-Figure 1SIMILAR CATHODE FOLLOWER CIRCUITS HAVING DIFFERENT R-F GROUND POINTS.274®
R-F Feedback Circuits275B+B+BIASFigure 2SINGLE STAGE AMPLIFIER WITHR-F FEEDBACK CIRCUITB+f-- R-F OUTBIAS B+Figure 4R-F AMPLIFIER WITH FEEDBACKAND IMPEDANCE MATCHINGOUTPUT NETWORK.Tuning and loading ore accomplished by C:and C,,. C1 and L1 are tuned in unison toestablish the correct degree of feedback.BIASFigure 3SINGLE STAGE FEEDBACKAMPLIFIER WITH GROUNDRETURN POINT MODIFIED FORUNBALANCED INPUT ANDOUTPUT CONNECTIONS.point of the circuit from the plate to the cathodeas shown in figure lB. Both ends of theinput circuit are at high r-f potential so inductivecoupling to this type of amplifier isnecessary.Inspection of figure lB shows that by movingthe top end of the input tank down on avoltage divider tap across the plate tank circuit,the feedback can be reduced from 100%,as in the case of the cathode follower circuit,down to any desired value. A typical feedbackcircuit is illustrated in figure 2. This circuitis more practical than those of figure 1, sincethe losses in the input tank are greatly reduced.A feedback level of 12 db may be achievedas a good compromise between distortion andstage gain. The voltage developed across C,will be three times the grid-cathode voltage.Inductive coupling is required for this circuit,as shown in the illustration.The circuit of figure 3 eliminates the needfor inductive coupling by moving the r-fground to the point common to both tankcircuits. The advantages of direct coupling betweenstages far outweigh the disadvantages ofhaving the r-f feedback voltage appear on thecathode of the amplifier tube.In order to match the amplifier to a load,the circuit of figure 4 may be used. The ratioof XL to XC determines the degree of feedback,so it is necessary to tune them in unisonwhen the frequency of operation is changed.Tuning and loading functions are accomplishedby varying C, and Ca. L, may also be varied toadjust the loading.Feedback Around aTwo-Stage AmplifierThe maximum phaseshift obtainable overtwo simple tuned circuitsdoes not exceed 180 degrees, and feedbackaround a two stage amplifier is possible.The basic circuit of a two stage feedbackamplifier is shown in figure 5. This circuitis a conventional two-stage tetrode amplifierexcept that r-f is fed back from the platecircuit of the P A tube to the cathode of thedriver tube. This will reduce the distortionFigure 5BASIC CIRCUIT OF TWO-STAGE AMPLIFIER WITH R-F FEEDBACKFeedback voltage is obtained from o voltage divider across the output circuit andapplied directly to the cathode of the first tube. The input tonk circuit is thusoutside the feedback loop.
276 R-F Feedbackof both tubes as effectively as using individualfeedback loops around each stage, yet willallow a higher level of overall gain. Withonly two tuned circuits in the feedback loop,it is possible to use 12 to 15 db of feedbackand still leave a wide margin for stability. Itis possible to reduce the distortion by nearlyas many db as are used in feedback. This circuithas two advantages that are lacking in thesingle stage feedback amplifier. First, the filamentof the output stage can now be operatedat r-f ground potential. Second, any conventionalpi output network may be used.R- f feedback will correct several types ofdistortion. It will help correct distortion causedby poor power supply regulation, too low gridbias, and limiting on peaks when the platevoltage swing becomes too high.Neutralizationand R-F FeedbackThe purpose of neutralization of an r-f amplifierstage is to balance out effectsof the grid-plate capacitance coupling inthe amplifier. In a conventional amplifier usinga tetrode tube, the effective input capaciryis given by:Input Capacitance= Cn + C.p ( 1 +A cos 8)where: Cn =tube input capacitancecgp =grid-plate capacitanceA = voltage amplification from gridto plate8 = phase angle of loadIn a typical unneutralized tetrode amplifierhaving a stage gain of 33, the input capacitanceof the tube with the plate circuit inresonance is increased by 8 l'l'fd. due to theunneutralized grid-plate capacitance. This isunimportant in amplifiers where the gain (A)remains constant but if the tube gain varies,serious detuning and r-f phase shift may result.A grid or screen modulated r-f amplifier is anexample of the case where the stage gain variesfrom a maximum down to zero. The gainof a tetrode r-f amplifier operating below platecurrent saturation varies with loading so thatif it drives a following stage into grid currentthe loading increases and the gain falls off.The input of the grid circuit is also affectedby the grid-plate capacitance, as shown in thisequation:1Input Resistance =21rf X c.p ( Asin8)This resistance is in shunt with the gridcurrent loading, grid tank circuit losses, anddriving source impedance. When the plate cir-<strong>THE</strong> <strong>RADIO</strong>cuit 1s inductive there is energy transferredfrom the plate to the grid circuit (positivefeedback) which will introduce negative resistancein the grid circuit. When this shuntnegative resistance across the grid circuit islower than the equivalent positive resistanceof the grid loading, circuit losses, and drivingsource impedance, the amplifier will oscillate.When the plate circuit is in resonance(phase angle equal to zero) the input resistancedue to the grid-plate capacitance becomesinfinite. As the plate circuit is tuned to thecapacitive side of resonance, the input resistancebecomes positive and power is actuallytransferred from the grid to the plate circuit.This is the reason that the grid current in anunneutralized tetrode r-f amplifier varies froma low value with the plate circuit tuned on thelow frequency side of resonance to a high valueon the high frequency side of resonance Thegrid current is proportional to the r-f voltageon the grid which is varying under these conditions.In a tetrode class AB1 amplifier, theeffect of grid-plate feedback can be observedby placing a r-f voltmeter across the grid circuitand observing the voltage change as theplate circuit is tuned through resonance.If the amplifier is over-neutralized, the effectsreverse so that with the plate circuittuned to the low frequency side of resonancethe grid voltage is high, and on the high frequencyside of resonance, it is low.AmplifierA useful "rule ofNeutralization Check thumb" method ofchecking neutralizationof an amplifier stage (assuming that itis nearly correct to start with) is to tune bothgrid and plate circuits to resonance. Then, observingthe r-f grid current, tune the plate circuitto the high frequency side of resonance.If the grid current rises, more neutralizationcapacitance is required. Conversely, if the gridcurrent decreases, less capacitance is needed.This indication is very sensitive in a neutralizedtriode amplifier, and correct neutralizationexists when the grid current peaks at thepoint of plate current dip. In tetrode poweramplifiers this indication is less pronounced.Sometimes in a supposedly neutralized tetrodeamplifier, there is practically no change ingrid voltage as the plate circuit is tunedthrough resonance, and in some amplifiers itis unchanged on one side of resonance anddrops slightly on the other side. Another observationsometimes made is a small dip inthe center of a broad peak of grid current.These various effects are probably caused by
<strong>HANDBOOK</strong>R-F Feedback Circuits 277E()UTR-F INFigure 6SINGLE STAGE R-F AMPLIFIERWITH FEEDBACK RATIO OFC,/C, to C"v!C" 1 DETERMINESSTAGE NEUTRALIZATIONR-~ INFigure 7NEUTRALIZED AMPLIFIER ANDINHERENT FEEDBACK CIRCUIT.Neutralization is achieved by varyingthe capacity of Cn.Er")IJTcoupling from the plate to the grid circuitthrough other paths which are not balancedout by the particular neutralizing circuit used.Feedback andNeutralizationof a One-StageR-F AmplifierFigure 6 shows an r-f amplifierwith negative feedback.The voltage developedacross c, due to the voltagedivider action of c, and C.is introduced in series with the voltage developedacross the grid tank circuit and is inphase-opposition to it. The feedback can bemade any value from zero to 100% by properlychoosing the values of C, and C..For reasons stated previously, it is necessaryto neutralize this amplifier, and the relationshipfor neutralization is:c. c.,It is often necessary to add capacitance fromplate to grid to satisfy this relationshipFigure 7 is identical to figure 6 except thatit is redrawn to show the feedback inherent inthis neutralization circuit more clearly. Cn andC replace c, and C., and the main plate tanktuning capacitance is Cs. The circuit of figure7 presents a problem in coupling to the gridcircuit. Inductive coupling is ideal, but theextra tank circuits complicate the tuning of atransmitter which uses several cascaded amplifierswith feedback around each one. Thegrid could be coupled to a high source impedancesuch as a tetrode plate, but the driverthen cannot use feedback because this wouldcause the source impedance to be low. A possiblesolution is to move the circuit groundpoint from the cathode to the bottom end ofthe grid tank circuit. The feedback voltage thenappears between the cathode and ground(figure 8). The input can be capacitivelycoupled, and the plate of the amplifier canbe capacitively coupled to the next stage. Also,cathode type transmitting tubes are availablethat allow the heater to remain at ground po-tential when r-f is impressed upon the cathode.The output voltage available with capacitycoupling, of course, is less than the platecathoder-f voltage developed by the amountof feedback voltage across C..14-2 Feedback andNeutralization of aTwo-Stage R-F AmplifierFeedback around two r-f stages has the advantagethat more of t~he tube gain can berealized and nearly as much distortion reductioncan be obtained using 12 db around twostages as is realized using 12 db around eachof two stages separately. Figure 9 shows abasic circuit of a two stage feedback amplifier.Inductive output coupling is used, althougha pi-network configuration will alsowork well. The small feedback voltage requiredis obtained from the voltage divider C. - C,and is applied to the cathode of the drivertube. C is only a few 1-'1-'fd., so this feedbackvoltage divider may be left fixed for a widefrequency range. If the combined tube gain is160, and 12 db of feedback is desired, the ratioof C, to C is about 40 to 1. This ratio inpractice may be 400 1-'1-'fd. to 2.5 1-'1-'fd., forexample.A complication is introduced into this simplifiedcircuit by the cathode-grid capacitanceFigure 8UNBALANCED INPUT AND OUTPUTCIRCUITS FOR SINGLE-STAGER-F AMPLIFIER WITH FEEDBACK
278 R-F Feedback <strong>THE</strong> <strong>RADIO</strong>V1V2Figure 9TWO-STAGE AMPLIFIER WITH FEEDBACK.Included is a capacitor (C,) for neutra/iring the cathode-grid capacity of the first tube. V, is neutraliredby capacitor c,, and Vz is neutra/ired by the correct ratio of c,;c,.of the first tube which causes an undersiredcoupling to the input grid circuit. It is necessaryto neutralize out this capacitance coupling,as illustrated in figure 9. The relationship forneutralization is:c. c.,c. c.The input circuit may be made unbalancedby making C. five times the capacity of C..This will tend to reduce the voltage acrossthe coil and to minimize the power dissipatedby the coil. For proper balance in this case,C. must be five times the grid-filament capacitanceof the tube.Except for tubes having extremely smallgrid-plate capacitance, it is still necessary toproperly neutralize both tubes. If the ratio ofC. to c, is chosen to be equal to the ratio ofthe grid-plate capacitance to the grid-filamentcapacitance in the second tube (V,), this tubewill be neutralized. Tubes such as a 4X-150Ahave very low grid-plate capacitance and probablywill not need to be neutralized when usedin the first ( V,) stage. If neutralization isnecessary, capacitor C, is added for this purposeand the proper value is given by thefollowing relationship:c.. c., c.c.- c, c.If neither tube requires neutralization, thebottom end of the interstage tank circuit maybe returned to r-f ground. The screen andsuppressor of the first tube should then begrounded to keep the tank output capacitancedirectly across this interstage circuit andto avoid common coupling between the feedbackon the cathode and the interstage circuit.A slight amount of degeneration occurs in thefirst stage since the tube also acts as a groundedgrid amplifier with the screen as the groundedgrid. The JL of the screen is much lower thanthat of the control grid so that this effect maybe unnoticed and would only require slightlymore feedback from the output stage to overcome.Tests ForNeutralizationNeutralizing the circuit offigure 9 balances out couplingbetween the inputtank circuit and the output tank circuit, but itdoes not remove all coupling from the platecircuit to the grid-cathode tube input. Thislatter coupling is degenerative, so applying asignal to the plate circuit will cause a signalto appear between grid and cathode, eventhough the stage is neutralized. A bench testfor neutralization is to apply a signal to theplate of the tube and detect the presence of asignal in the grid coil by inductive couplingto it. No signal will be present when the stageis neutralized. Of course, a signal could be inductivelycoupled to the input and neutralizationaccomplished by adjusting one branch ofthe neutralizing circuit bridge ( C, for example)for minimum signal on the plate circuit.Neutralizing the cathode-grid capacitance ofthe first stage of figure 9 may be accomplishedby applying a signal to the cathode of the tubeand adjusting the bridge balance for minimumsignal on a detector inductively coupled to theinput coil.Tuning a Two-StageFeedback AmplifierTuning the two-stagefeedback amplifier offigure 9 is accomplishedin an unconventional way because theoutput circuit cannot be tuned for maximumoutput signal. This is because the output circuitmust be tuned so the feedback voltageapplied to the cathode is in-phase with theinput signal applied to the first grid. Whenthe feedback voltage is not in-phase, the resultantgrid-cathode voltage increases as shownin figure 10. When the output circuit isproperly tuned, the resultant grid-cathode voltageon the first tube will be at a minimum, andthe voltage on the interstage tuned circuit willalso be at a minimum.
<strong>HANDBOOK</strong>Neutra I ization279VOLTAGE- GRID TO CATHODEVOLTAGEINPUT
280 R-F Feedback <strong>THE</strong> <strong>RADIO</strong>C11~R-FOUTyBIAS B+ BIAS +Figure 13TWO-STAGE AMPLIFIER WITH FEED BACK CIRCUIT.B-When the P A grid circuit is tunedthrough resonance:1-Driver plate current dip2-P A r-f grid voltage peak3-P A grid current peak4-P A power output peakC-When the driver grid circuit is tunedthrough resonance:1-Driver r-f grid voltage peak2-Driver plate current peak3-PA r-f grid current peak4-P A plate current peak5-P A power output peakFour meters may be employed to measurethe most important of these parameters. Themeters should be arranged so that the followingpairs of readings are displayed on meterslocated close together for ease of observationof coincident peaks and dips:1-PA plate current and power output2-PA r-f grid current and PA platecurrent3-P A r-f grid voltage and power output4-Driver plate current and P A r-fgrid voltageThe third pair listed above may not benecessary if the PA plate current dip is pronounced.When this instrumentation is provided,the neutralizing procedure is as follows:1-Remove the r-f feedbackFigure 14FEEDBACK SHORTING DEVICE.2-Neutralize the grid-plate capacitanceof the driver stage3-Neutralize the grid-plate capacitanceof the power amplifier ( P A)stage4-Apply r-f feedback5-Neutralize driver grid-cathode capacitanceThese steps will be explained in more detailin the following paragraphs:Step J, The removal of r-f feedback throughthe feedback circuit must be complete. Theswitch ( s,) shown in the feedback circuit(figure 13) is one satisfactory method. SinceCr. is effectively across the P A plate tank circuitit is desirable to keep it across the circuitwhen feedback is removed to avoid appreciabledetuning of the plate tank circuit. Anothermethod that can be used if properly done isto ground the junction of Cr. and C. Groundingthis common point through a switch orrelay is not good enough because of commoncoupling through the length of the groundinglead. The grounding method shown in figure14 is satisfactory.Step 2. Plate power and excitation are applied.The driver grid tank is resonated by tuningfor a peak in driver r-f grid voltage or driverplate current. The power amplifier grid tankcircuit is then resonated and adjusted for adip in driver plate current. Driver neutralizationis now adjusted until the P A r-f gridvoltage (or P A grid current) peaks at exactlythe point of driver plate current dip. A handyrule for adjusting grid-plate neutralization ofa tube without feedback: with all circuits inresonance, detune the plate circuit to the highfrequency side of resonance: If grid currentto next stage (or power output of the stageunder test) increases, more neutralizing capacitanceis required and vice versa.If the driver tube operates class A so thata plate current dip cannot be observed, a dif-
<strong>HANDBOOK</strong>Neutralization 281RFigure 1 SFEEDBACK NEUTRALIZINGCIRCUIT USINGAUXILIARY RECEIVER.ferent neutralizing procedure is necessary Thiswill be discussed in a subsequent section.Step 3. This is the same as step 2 except itis applied to the power amplifier stage. Adjustthe neutralization of this stage for a peakin power output at the plate current dip.Step 4. Reverse step 1 and apply the r-f feedback.Step 5. Apply plate power and an exciting signalto drive the amplifier to nearly full output.Adjust the feedback neutralization for apeak in amplifier power output at the exactpoint of minimum amplifier plate current.Decrease the feedback ne::tralization capacitanceif the power output rises when the tankcircuit is tuned to the high frequency side ofresonance.The above sequence applies when the neutralizingadjustments are approximately correctto start with. If they are far off, some "cutand-try"adjustment may be necessary. Also,the driver stage may break into oscillation ifthe feedback neutralizing capacitance is notnear the correct setting.It is assumed that a single tone test signalis used for amplifier excitation during theabove steps, and that all tank circuits are atresonance except the one being detuned tomake the observation. There is some interactionbetween the driver neutralization and the feedbackneutralization so if an appreciable changeis made in any adjustment the others shouldbe rechecked. It is important that the grid-plateneutralization be accomplished first when usingthe above procedure, otherwise the feedbackneutralization will be off a little, since it partiallycompensates for that error.NeutralizationTechniquesThe method of neutralizationemploying a sensitive r-f detectorinductively coupled toa tank coil is difficult to apply in some casesbecause of mechanical construction of theequipment, or because of undesired coupling.Another method for observing neutralizationcan be used, which appears to be more accuratein actual practice. A sensitive r-f detectorsuch as a receiver is loosely coupled to thegrid of the stage being neutralized, as shownin figure 15. The coupling capacitance is ofthe order of one or two p,p,fd. It must be smallenough to avoid upsetting the neutralizationwhen it is removed because the total gridgroundcapacitance is one leg of the neutralizingbridge. A signal generator is connected atpoint S and the receiver at point R. If Co isnot properly adjusted the S-meter on the receiverwill either kick up or down as the gridtank circuit is tuned through resonance. C.omay be adjusted for minimum deflection of theS-meter as the grid circuit is tuned throughresonance.The grid-plate capacitance of the tube isthen neutralized by connecting the signal generatorto the plate of the tube and adjustingCu of figure 13 for minimum deflection againas the grid tank is tuned through resonance.The power amplifier stage is neutralized inthe same manner by connecting a receiverloosely to the grid circuit, and attaching asignal generator to the plate of the tube. Ther-f signal can be fed into the amplifier outputterminal if desired.Some precautions are necessary when usingthis neutralization method. First, some drivertubes (the 6CL6, for example) have appreciablymore effective input capacitance whenin operation and conducting plate current thanwhen in standby condition. This increase ininput capacitance may be as great as three orfour p,p,fd, and since this is part of the neutralizingbridge circuit it must be taken intoconsideration. The result of this change ininput capacitance is that the neutralizing adjustmentof such tubes must be made whenthey are conducting normal plate current. Straycoupling must be avoided, and it may provehelpful to remove filament power from thepreceding stage or disable its input circuit insome manner.It should be noted that in each of the aboveadjustments that minimum reaction on thegrid is desired, not minimum voltage.Someresidual voltage is inherent on the grid whenthis neutralizing circuit is used.
CHAPTER FIFTEENAmplitude ModulationIf the output of a c-w transmitter is variedin amplitude at an audio frequency rate insteadof interrupted in accordance with codecharacters, a tone will be heard on a receivertuned to the signal. If the audio signal consistsof a band of audio frequencies comprisingvoice or music intelligence, then thevoice or music which is superimposed on theradio frequency carrier will be heard on thereceiver.When voice, music, video, or other intelligenceis superimposed on a radio frequencycarrier by means of a corresponding variationin the amplitude of the radio frequency outputof a transmitter, amplitude modulation is theresult. Telegraph keying of a c-w transmitteris the simplest form of amplitude modulation,while video modulation in a television transmitterrepresents a highly complex form. Systemsfor modulating the amplitude of a carrierenvelope in accordance with voice, music, orsimilar types of complicated audio waveformsare many and varied, and will be discussedlater on in this chapter.15-1 SidebandsModulation is essentially a form of mtxtngor combining already covered in a previouschapter. To transmit voice at radio frequenciesby means of amplitude modulation, the voice282frequencies are mixed with a radio frequencycarrier so that the voice frequencies are convertedto radio frequency sidebands. Thoughit may be difficult to visualize, tbe amplitudeof the radio frequency carrier does not varyduring conventional amplitude modulation.Even though the amplitude of radio frequencyvoltage representing the compositesignal (resultant of the carrier and sidebands,called the envelope) will vary from zero totwice the unmodulated signal value duringfull modulation, the amplitude of the carriercomponent does not vary. Also, so long asthe amplitude of the modulating voltage doesnot vary, the amplitude of the sidebands willremain constant. For this to be apparent, however,it is necessary to measure the amplitudeof each component with a highly selectivefilter. Otherwise, the measured power or voltagewill be a resultant of two or more of thecomponents, and the amplitude of the resultantwill vary at the modulation rate.If a carrier frequency of 5000 kc. is modulatedby a pure tone of 1000 cycles, or 1 kc.,two sidebands are formed: one at 5001 kc.(the sum frequency) and one at 4999 kc. (thedifference frequency). The frequency of eachsideband is independent of the amplitude ofthe modulating tone, or modulation percentage:the frequency of each sideband is determinedonly by the frequency of the modulatingtone. This assumes, of course, that thetransmitter is not modulated in excess of itslinear capability.
Modulation 283When the modulating signal consists ofmultiple frequencies, as is the case withvoice or music modulation, two sidebands willbe formed by each modulating frequency (oneon each side of the carrier), and the radiatedsignal will consist of a band of frequencies.The band width, or channel taken up in thefrequency spectrum by a conventional doublesidebandamplitude-modulated signal, is equalto twice the highest modulating frequency.For example, if the highest modulating frequencyis 5000 cycles, then the signal (assumingmodulation of complex and varyingwaveform) will occupy a band extending from5000 cycles below the carrier to 5000 cyclesabove the carrier.Frequencies up to at least 2500 cycles, andpreferably 3500 cycles, are necessary for goodspeech intelligibility. If a filter is incorporatedin the audio system to cut out all frequenciesabove approximately 300Q cycles,the band width of a radio-telephone signal canbe limited to 6 kc. without a significant lossin intelligibility. However, if harmonic distortionis introduced subsequent to the filter, aswould happen in the case of an overloadedmodulator or overmodulation of the carrier,new frequencies will be generated and thesignal will occupy a band wider than 6 kc.15-2 Mechanics ofModulationA c-w or unmodulated r-f carrier wave isrepresented in figure IA. An audio frequencysine wave is represented by the curve offigure lB. When the two are combined or"mixed," the carrier is said to be amplitudemodulated, and a resultant similar to IC orID is obtained. It should be noted that undermodulation, each half cycle of r-f voltagediffers slightly from the preceding one andthe following one; therefore at no time duringmodulation is the r-f waveform a pure sinewave. This is simply another way of sayingthat during modulation, the transmitted r-fenergy no longer is confined to a single radiofrequency.It will be noted that the average amplitudeof the peak r-f voltage, or modulation envelope,is the same with or without modulation.This simply means that the modulation issymmetrical (assuming a symmetrical modulatingwave) and that for distortionless modulationthe upward modulation is limited to avalue of twice the unmodulated carrier waveamplitude because the amplitude cannot gobelow zero on downward portions of the modulationcycle. Figure 1D illustrates the maxi-0 nnnnnnnnnnnnnnnnnnnnnnnnn1Av v v~vv~v v vv vvv u VINU ~rrC.W. OR UN MODULATED CARRIER@~SINEWAVE ~AUDIO SIGNAL FROM MODULATOR50% MODULATED CARRIER100% MODULATED CARRIERFigure 1AMPLITUDE MODULATED WAVETop drawing (A) represents on unmodulotedcarrier wave; (B) shows the audio output ofthe modulator. Drawing (C) shows the audiosignal impressed on the carrier wave to theextent of 50 per cent modulation; (D) showsthe carrier with 100 per cent amplitude modu·lotion.mum obtainable distortionless modulation witha sine modulating wave, the r-f voltage at thepeak of the r-f cycle varying from zero totwic~ the unmodulated value, and the r-f powervarytng from zero to four times the unmodulatedvalue (the power varies as the squareof the voltage).While the average r-f voltage of the modulatedwave over a modulation cycle is thesame as for the unmodulated carrier, the averagepower increases with modulation. If theradio frequency power is integrated over theaudio cycle, it will be found with 100 per centsine wave modulation the average r-f powerhas increased 50 per cent. This additionalpower is represented by the sidebands, becauseas previously mentioned, the carrierpower does not vary under modulation. Thuswhen a 10~-watt carrier is modulated 100 pe;cent by a s1ne wave, the total r·f power is ISOwatts; 100 watts in the carrier and 25 wattsin each of the two sidebands.A
2 84 Amplitude Modulation<strong>THE</strong> R AD I 0So long as the relative propor·ModulationPercentage tion of the various sidebandsmaking up voice modulation ismaintained, the signal may be received anddetected without distortion. However, thehigher the average amplitude of the sidebands,the greater the audio signal produced at thereceiver. For this reason it is desirable toincrease the modulation percentage, or degreeof modulation, to the point where maximumpeaks just hit 100 per cent. If the modulationpercentage is increased so that the peaks ex·ceed this value, distortion is introduced, andif carried very far, bad interference to signalson nearby channels will result.@ ®Figure 2GRAPHICAL DETERMINATION OF MODU·LATWN PERCENTAGEThe procedure for determining modulationpercentage from the peak voltage points in·dicated is discussed in the text.ModulationMeasurementThe amount by which a carrieris being modulated may be ex·pressed either as a modulationfactor, varying from zero to 1.0 at maximummodulation, or as a percentage. The percentageof modulation is equal to 100 times themodulation factor. Figure 2A shows a carrierwave modulated by a sine·wave audio tone.A picture such as this might be seen on thescreen of a cathode-ray oscilloscope withsawtooth sweep on the horizontal plates andthe modulated carrier impressed on the verticalplates. The same carrier without modulationwould appear on the oscilloscope screen asfigure 28.The percentage of modulation of the positivepeaks and the percentage of modulationof the negative peaks can be determined separatelyfrom two oscilloscope pictures suchas shown.The modulation factor of the positive peaksmay be determined by the formula:Emax- EcarM= ------------EcarThe factor for negative peaks may be determinedfrom this formula:Ecar- EminM=------EcarIn the above two formulas Em ax is the maximumcarrier amplitude with modulation andEmin is the minimum amplitude; Ecar is thesteady-state amplitude of the carrier with·out modulation. Since the deflection of thespot on a cathode-ray tube is linear with respectto voltage, the relative voltages ofthese various amplitudes may be determinedby measuring the deflections, as viewed onthe screen, with a rule calibrated in inchesor centimeters. The percentage of modulationof the carrier may be had by multiplying themodulation factor thus obtained by 100. Theabove procedure assumes that there is nocarrier shift, or change in average amplitude,with modulation.If the modulating voltage is symmetrical,such as a sine wave, and modulation is accomplishedwithout the introduction of dis·tortion, then the percentage modulation willbe the same for both negative and positivepeaks. However, the distribution and phaserelationships of harmonics in voice and musicwaveforms are such that the percentage modulationof the negative modulation peaks mayexceed the percentage modulation of the positivepeaks, and vice versa. The percentagemodulation when referred to without regardto polarity is an indication of the average ofthe negative and positive peaks.ModulationCapabilityThe modulation capability of atransmitter is the maximum percentageto which that transmittermay be modulated before spurious sidebandsare generated in the output or before the dis·tortion of the modulating waveform becomesobjectionable. The highest modulation cap·ability which any transmitter may have on thenegative peaks is 100 per cent. The maximumpermissible modulation of many transmittersis less than 100 per cent, especially on posi·tive peaks. The modulation capability of atransmitter may be limited by tubes with insufficientfilament emission, by insufficientexcitation or grid bias to a plate-modulatedstage, too light loading of any type of amplifiercarrying modulated r.f., insufficient poweroutput capability in the modulator, or too muchexcitation to a grid-modulated stage or aClass B linear amplifier. In any case, theFCC regulations specify that no transmitterbe modulated in excess of its modulationcapability. Hence, it is desirable to make themodulation capability of a transmitter as nearas possible to 100 per cent so that the carrierpower may be used most effectively.
<strong>HANDBOOK</strong>Modulation Systems 285Speech WaveformDissymmetryThe manner in which thehuman voice is producedby the vocal cords givesrise to a certain dissymmetry in the waveformof voice sounds when they are picked up bya good-quality microphone. This is especiallypronounced in the male voice, and more soon certain voiced sounds than on others. Theresult of this dissymmetry in the waveform isthat the voltage peaks on one side of theaverage value of the wave will be considerablygreater, often two or three times as great,as the voltage excursions on the other sideof the zero axis. The average value of voltageon both sides of the wave is, of course,the same.As a result of this dissymmetry in the malevoice waveform, there is an optimum polarityof the modulating voltage that must be observedif maximum sideband energy is to beobtained without negative peak clipping andgeneration of "splatter" on adjacent channels.A double-pole double-throw "phase revers•ing" switch in the input or output leads of anytransformer in the speech amplifier system willpermit poling the extended peaks in the direc·tion of maximum modulation capability. Theoptimum polarity may be determined easily bylistening on a selective receiver tuned to afrequency 30 to 50 kc. removed from the desiredsignal and adjusting the phase reversingswitch to the position which gives the least"splatter" when the transmitter is modulatedrather heavily. If desired, the switch then maybe replaced with permanent wiring, so long asthe microphone and speech system are not tobe changed.A more conclusive illustration of the lop·sidedness of a speech waveform may be obtainedby observing the modulated waveformof a radiotelephone transmitter on an oscilloscope.A portion of the carrier energy of thetransmitter should be coupled by means of alink direct 1 y to the vertical plates of the'scope, and the horizontal sweep should be asawtooth or similar wave occurring at a rateof approximately 30 to 70 sweeps per second.With the speech signal from the speech am·plifier connected to the transmitter in one po·larity it will be noticed that negative-peakclipping-as indicated by bright "spots" inthe center of the 'scope pattern whenever thecarrier amplitude goes to zero-will occur ata considerably lower level of average modulationthan with the speech signal being fed tothe transmitter in the other polarity. When theinput signal to the transmitter is polarized insuch a manner that the "fingers" of thespeech wave extend in the direction of posi·tive modulation these fingers usually will beclipped in the plate circuit of the modulatorat an acceptable peak modulation level.The use of the proper polarity of the incom·ing speech wave in modulating a transmittercan afford an increase of approximately twoto one in the amount of speech audio powerwhich may be placed upon the carrier for anamplitude-modulated transmitter for the sameamount of sideband splatter. More effectivemethods for increasing t h e amount of audiopower on the carrier of an AM phone transmitterare discussed later in this chapter.Single-SidebandTransmissionBecause the same intelli·gibility is contained in each·of the sidebands associatedwith a modulated carrier, it is not necessaryto transmit sidebands on both sides of thecarrier. Also, because the carrier is simply asingle radio frequency wave of unvarying am·plitude, it is not necessary to transmit thecarrier if some means is provided for insertinga locally generated carrier at the receiver.When the carrier is suppressed but bothupper and lower sidebands are transmitted, itis necessary to insert a locally generatedcarrier at the receiver of exactly the samefrequency and phase as the carrier which wassuppressed. For this reason, suppressedcarrierdouble-sideband systems have littlepractical application.When the carrier is suppressed and only theupper or the lower sideband is transmitted, ahighly intelligible signal may be obtained atthe receiver even though the locally generatedcarrier differs a few cycles from the frequencyof the carrier which was suppressed at thetransmitter. A communications system utilizingbut one group of sidebands with carriersuppressed is known as a single sidebandsystem. Such systems are widely used forcommercial point to point work, and are beingused to an increasing extent in amateur communication.The two chief advantages of thesystem are: (I) an effective power gain ofabout 9 db results from putting all the radiat·ed power in intelligence carrying sidebandfrequencies instead of mostly into radiatedcarrier, and (2) elimination of the selectivefading and distortion that normally occurs ina conventional double-sideband system whenthe carrier fades and the sidebands do not, orthe sidebands fade differently.15-3 Systems of Amp I itudeModulationThere are many different systems and methodsfor amplitude modulating a carrier, butmost may be grouped under three general clas·sifications: ( 1) variable efficiency systemsin which the average input to the stage re-
286 Amplitude Modulation<strong>THE</strong> R AD I 0mains constant with and without modulationand the variations in the efficiency of thestage in accordance with the modulating signalaccomplish the modulation; (2) constantefficiency systems in which the input to thestage is varied by an external source of modulatingenergy to accomplish the modulation;and (3) so-called high-efficiency systems inwhich circuit complexity is increased to obhighplate circuit efficiency in the modulatedstage without the requirement of an externalhigh-level modulator. The various systemsunder each classification have individualcharacteristics which make certain ones bestsuited to particular applications.Variable EfficiencyModulationSince the average inputremains constant in astage employing variableefficiency modulation, and since the averagepower output of the stage increases with modulation,the additional average power outputfrom the stage with modulation must come fromthe plate dissipation of the tubes in the stage.Hence, for the best relation between tube costand power output the tubes employed shouldhave as high a plate dissipation rating perdollar as possible.The plate efficiency in such an amplifier isdoubled when going from the unmodulatedcondition to the peak of the modulation cycle.Hence, the unmodulated efficiency of such anamplifier must always be less than 45 percent, since the maximum peak efficiency obtainablein a conventional amplifier is in thevicinity of 90 per cent. Since the peak efficiencyin certain types of amplifiers will beas low as 60 per cent, the unmodulated efficiencyin such amplifiers will be in the vici·nity of 30 per cent.Assuming a typical amplifier having a peakefficiency of 70 per cent, the following figuresgive an idea of the operation of an idealizedefficiency-modulated stage adjusted for100 per cent sine-wave modulation. It shouldbe kept in mind that the plate voltage is constantat all times, even over the audio cycles.Plate input without modulation ........ 100 wattsOutput without modulation .............. 35 wattsEfficiency without modulation ......... 35%Input on 100% positive modulationpeak (plate current doubles) ........ 200 wattsEfficiency on 100% positive peak .... 70%Output on 100% positive modulationpeak ................................. 140 wattsInput on 100% negative peak ......... .Efficiency on 100% negative peak ... .Output on 100% negative peak ....... .0 watts0%0 wattsAverage input with 1007omodulation .............................. 100 wattsAverage output with 100% modulation(35 watts carrier plus 17.5watts sideband) ...................... 52.5 wattsAverage efficiency with 100%modulation .............................. 52.5%Systems of EfficiencyModulationThere are many systernsof efficiency modulation,but they allhave the general limitation discussed in theprevious paragraph-so long as the carrieramplitude is to remain constant with andwithout modulation, the efficiency at carrierlevel must be not greater than one-half thepeak modulation efficiency if the stage is tobe capable of 100 per cent modulation.The classic example of efficiency modulationis the Class B linear r-f amplifier, to bediscussed below. The other three commonforms of efficiency modulation are controlgridmodulation, screen-grid modulation, andsuppressor-grid modulation. In each case,including that of the Class B linear amplifier,note that the modulation, or the modulatedsignal, is impressed on a control electrodeof the stage.The Class B This is the simplest practiLinear Amplifier cable type amplifier for anamplitude-modulated waveor a single-sideband signal. The system possessesthe disadvantage that excitation, gridbias, and loading must be carefully controlledto preserve the linearity of the stage. Also,the grid circuit of the tube, in the usual applicationwhere grid current is drawn on peaks,presents a widely varying value of load impedanceto the source of excitation. Hence itis necessary to include some sort of swampingresistor to reduce the effect of grid-impedancevariations with modulation. If such aswamping resistance across the grid tank isnot included, or is too high in value, the positivemodulation peaks of the incoming modulatedsignal will tend to be flattened withresultant distortion of the wave being amplified.The dass B linear amplifier has long beenused in broadcast transmitters, but recentlyhas received much more general usage in theh-f range for two significant reasons: (a) theClass B linear is an excellent way of increasingthe power output of a single-sidebandtransmitter, since the plate efficiency withfull signal will be in the vicinity of 70 percent, while with no modulation the input tothe stage drops to a relatively low value; and(b) the Class B linear amplifier operates withrelatively low harmonic output since the gridbias on the stage normally is slightly less
<strong>HANDBOOK</strong>Class B Linear Amplifier 287than the value which will cut off plate currentto the stage in the absence of excitation.Since " Oass B linear amplifier is biasedto extended cutoff with no excitation (thegrid bias at extended cutoff will be approximatelyequal to the plate voltage divided bythe amplification factor for a triode, and willbe approximately equal to the screen voltagedivided by the grid-screen mu factor for atetrode or pentode) the plate current will flowessentially in 180-degree pulses. Due to therelatively large operating angle of plate currentflow the theoretical peak plate efficiencyis limited to 78.5 per cent, with 65 to 70 percent representing a range of efficiency normallyattainable, and the harmonic outputwill be low.The carrier power output from a Class Blinear amplifier of a normal 100 per cent modulatedAM signal will be about one-half therated plate dissipation of the stage, with optimumoperating conditions. The peak outputfrom a Class B linear, which represents themaximum-signal output as a single-sidebandamplifier, or peak output with a 100 per centAM signal, will be about twice the plate dissipationof the tubes in the stage. Thus thecarrier-level input power to a Class B linearshould be about 1.5 times the rated plate dissipationof the stage.The schematic circuit of a Class B linearamplifier is the same as a conventional singleendedor push-pull stage, whether triodes orbeam tetrodes are used. However, a swampingresistor, as mentioned before, must be placedacross the grid tank of the stage if the operatingconditions of the tube are such thatappreciable grid current will be drawn on modulationpeaks. Also, a fixed source of grid biasmust be provided for the stage. A regulatedgrid-bias power supply is the usual source ofnegative bias voltage.Adjustment of a Class With grid bias adjustedB Linear Amplifier to the correct value,and with provision forvarying the excitation voltage to the stageand the loading of the pl~te circuit, a fullymodulated signal is applied to the grid circuitof the stage. Then with an oscilloscope coupledto the output of the stage, excitation andloading are varied until the stage is drawingthe normal plate input and the output waveshapeis a good replica of the input signal.The adjustment procedure normally will requirea succession of approximations, untilthe optimum set of adjustments is attained.Then the modulation being applied to the inputsignal should be removed to check thelinearity. With modulation removed, in thecase of a 100 per cent AM signal, the inputto the stage should remain constant, and thepeak output of the r-f envelope should fall tohalf the value obtained on positive modulationpeaks.Class C.Grid ModulationOne widely used system ofefficiency modulation forcommunications work isClass C control-grid bias modulation. The distortionis slightly higher than for a properlyoperated Class B linear amplifier, but the efficiencyis also higher, and the distortion canbe kept within tolerable limits for communicationswork.Class C grid modulation requires high platevoltage on the modulated stage, if maximumoutput is desired. The plate voltage is normallyrun about 50 per cent higher than formaximum output with plate modulation.The driving power required for operation ofa grid-modulated amplifier under these conditionsis somewhat more than is required foroperation at lower bias and plate voltage, butthe increased power output obtainable overbalancesthe additional excitation requirement.Actually, almost half as much excitationis required as would be needed if the samestage were to be operated as a Class C platemodulatedamplifier. The resistor R acrossthe grid tank of the stage serves as swampingto stabilize the r-f driving voltage. At least50 per cent of the output of the driving stageshould be dissipated in this swamping resistorunder carrier conditions.A comparatively small amount of audio powerwill be required to modulate the amplifier stage100 per cent. An audio amplifier having 20watts output will be sufficient to modulate anamplifier with one kilowatt input. Proportionatelysmaller amounts of audio will be requiredfor lower powered stages. However, theaudio amplifier that is being used as the gridmodulator should, in any case, either employlow plate resistance tubes such as 2A3's,employ degenerative feedback from the outputstage to one of the preceding stages of thespeech amplifier, or be resistance loaded witha resistor across the secondary of the modulationtransformer. This provision of low driv··ing impedance in the grid modulator is to insuregood regulation in the audio driver for the gridmodulated stage. Good regulation of both theaudio and the r-f drivers of a grid-modulatedstage is quite important if distortion-freemodulation approaching 100 per cent is desired,because the grid impedance of the modulatedstage varies widely over the audio cycle.A practical circuit for obtaining grid-biasmodulation is shown in figure 3. The modulatorand bias regulator tube have been combinedin a single 6B4G tube.The regulator-modulator tube operates asa cathode-follower. The average d-e voltage
288Amplitude Modulation<strong>THE</strong> R AD I 0E·per cent. If the antenna coupling is decreasedslightly from the condition just described, andthe excitation is increased to the point wherethe amplifier draws the same input, carrierefficiency of 50 per cent is obtainable withtolerable distortion at 90 per cent modulation.AUDIO INPUTFROM 6SJ7ETC.f1!1V.A.C.2~K fOW.025 47K+BMIDGET CHOKEFigure 3GRID-BIAS MODULATOR CIRCUITon the control grid is controlled by the 70, 000·ohm wire-wound potentiometer and this potentiometeradjusts the average grid bias on themodulated stage. However, a-c signal voltageis also impressed on the control-grid of thetube and since the cathode follows this a-cwave the incoming speech wave is superimposedon the average grid bias, thus effectinggrid-bias modulation of the r-f amplifier stage.An audio voltage swing is required on the gridof the 6B4G of approximately the same peakvalue as will be required as bias-voltageswing on the grid-bias modulated stage. Thisvoltage swing will normally be in the regionfrom 5Q to 200 peak volts. Up to about 100volts peak swing can be obtained from a 6SJ7tube as a conventional speech amplifier stage.The higher voltages may be obtained from atube such as a 6J5 through an audio transformerof 2: 1 or 2~: 1 ratio.With the normal amount of comparativelytight antenna coupling to the modulated stage,a non-modulated carrier efficiency of 40 percent can be obtained with substantially distortion-freemodulation up to practically 100Tuning theGdd-BiasModulated StageThe most satisfactory procedurefor tuning a stagefor grid-bias modulation ofthe Class C type is asfollows. The amplifier should first be neutralized,and any possible tendency toward parasidesunder any condition of operation shouldbe eliminated. Then the antenna should becoupled to the plate circuit, the grid biasshould be run up to the maximum availablevalue, and the plate voltage and excitationshould be applied. The grid bias voltageshould then be reduced until the amplifierdraws the approximate amount of plate currentit is desired to run, and modulation correspondingto about 80 per cent is then applied.If the plate current kicks up when modulationis applied, the grid bias should be reduced;if the plate meter kicks down, increase thegrid bias.When the amount of bias voltage has beenfound (by adjusting the fine control, R2 onthe bias supply) where the plate mete~ remainsconstant with modulation, it is morethan probable that the stage will be drawingeither too much or too little input. The antennacoupling should then be either increasedor decreased (depending on whether the inputwas too little or too much, respectively)until the input is more nearly the correct value.The bias should then be readjusted until theplate meter remains constant with modulationas before. By slight jockeying back and forthof antenna coupling and grid bias, a point canbe reached where the tubes are running atrated plate dissipation, and where the platemilliammeter on the modulated stage remainssubstantially constant with modulation.The linearity of the stage should then bechecked by any of the conventional methods;the trapezoidal pattern method employing acathode-ray oscilloscope is probably the mostsatisfactory. The check with the trapezoidalpattern will allow the determination of theproper amount of gain to employ on the speechamplifier. Too much audio power on the gridof the modulated stage should not be used inthe tuning-up process, as the plate meter willkick erratically and it will be impossible tomake a satisfactory adjustment.Screen-GridModulationC amplifierAmplitude modulation may beaccomplished by varymg thescreen-grid voltage in a Classwhich employs a pentode, beam
<strong>HANDBOOK</strong>Screen Grid Modulation 289tetrode, or other type of screen-grid tube. Themodulation obtained in this way .is not especiallylinear, but screen-grid modulation~oes _offer other advantages and the linearityIs quae adequate for communications work.There are two significant and worthwhileadvantages of screen-grid modulation for communicationswork: ( 1) The excitation requirementsfor an amplifier which is to be modulatedin the screen are not at all critical andgood regulation of the excitation volta~e isnot required. The normal rated grid-circuitoperating conditions specified for Class Cc-w operation are quite adequate for screengridmodulation. (2) The audio modulatingpower requirements for screen-grid modulationare relatively low.A screen-grid modulated r-f amplifier operatesas an efficiency-modulated amplifier, thesame as does a Class B linear amplifier anda grid-modulated stage. Hence, plate circuitloading is relatively critical as in any efficiency-modulatedstage, and must be adjustedto the correct value if normal power outputwith full modulation capability is to be obtained.As in the case of any efficiency-modulatedstage, the operating efficiency at thepeak of the modulation cycle will be between70 and 80 per cent, with efficiency at the carrierlevel (if the stage is operating in the normalmanner with full carrier) about half of thepeak-modulation value.There are two main disadvantages of screengridmodulation, and several factors whichmust be considered if satisfactory operationof the screen-grid modulated stage is to beobtained. The disadvantages are: (1) As mentionedbefore, the linearity of modulation with~espe~t to screen-grid voltage of such a stageIs satisfactory only for communications work,unless carrier-rectified degenerative feed-backis employed around the modulated stage tostraighten the linearity of modulation. ( 2) Theimpedance of the screen grid to the modulatingsignal is non-linear. This means that the modulatingsignal must be obtained from a sourceof quite low impedance if audio distortion ofthe signal appearing at the screen grid is tobe avoided.Screen-GridlmpedanceInstead of being linear with re-spect to modulating voltage, asis the plate circuit of a platemodulatedClass C amplifier, the screen gridpresents approximately a square-law impedanceto the modulating signal over the regionof signal excursion where the screen is positivewith respect to ground. This non-linearitymay be explained in the following manner: Atthe carrier level of a conventional screen·modulated stage the plate-voltage swing ofthe modulated tube is one-half the voltageswing at peak-modulation leveL This conditionmust exist in any type of conventional efficiency-modulated stage if 100 per cent positivemodulation is to be attainable. Since theplate-voltage swing is at half amplitude, andsince the screen voltage is at half its fullmodulationvalue, the screen current is relativelylow. But at the positive modulation peakthe screen voltage is approximately doubled,and the plate-voltage swing also is at twicethe carrier amplitude. Due to the increase inplate-voltage swing with increasing screenvoltage, the screen current increases more thanlinearly with increasing screen voltage.In a test made on an amplifier with an 813tube, the screen current at carrier level wasabout 6 rna. with screen potential of 190 volts;but under conditions which represented a positivemodulation peak the screen current measured25 rna. at a potential of 400 volts. Thusinstead of screen current doubling with twicescreen voltage as would be the case if thescreen presented a resistive impedance, thescreen current became about four times asgreat with twice the screen voltage.Another factor which must be consideredin the design of a screen-modulated stage, iffull modulation is to be obtained, is that thepower output of a screen-grid stage with zeroscreen voltage is still relatively large. Hence,if anything approaching full modulation onnegative peaks is to be obtained, the screenpotential must be made negative with respectto ground on negative modulation peaks. Inthe usual types of beam tetrode tubes thescreen potential must be 20 to 50 volts negativewith respect to ground before cut-off ofoutput is obtained. This condition further complicatesthe problem ofobtaining good linearityin the audio modulating voltage for the screenmodulatedstage, since the screen voltagemust be driven negatively with respect toground over a portion of the cycle. Hence thescreen draws no current over a portion of themodulating cycle, and over the major portionof the cycle when the screen does draw current,it presents approximately a square-lawimpedance.Circuits forScreen-GridModulationLaboratory analysis of a largenumber of circuits for accomplishingscreen modulation hasled to the conclusion that theaudio modulating voltage must be obtainedfrom a low-impedance source if low-distortionmodulation is to be obtained. Figure 4shows a group of sketches of the modulationenvelope obtained with various types of modulatorsand also with insufficient antenna coupling.The result of this laboratory work ledto the conclusion that the cathode-followermodulator of the basic circuit shown in figure
2 90Amplitude Modulation<strong>THE</strong> R AD I 0EENVELOPE OBTAINED WITHINSUffiCIENT ANTENNACOUPLING-c+MOD. +S.G. -!10 V, APPROX,® ©@Figure 4SCREEN-MODULATION CIRCUITSThree common screen modulation circuits are illustrated above. All three circuitsare capable of giving intelligible voice modulation although the waveform distortionin the circuits of (A) and (B) Is likely to be rather severe. The arrangement at (A)is often call eel "clamp tube" screen modulation; by returning the grid leak on theclamp tube to ground the circuit will give control/eel-carrier screen modulation. Thiscircuit has the advantage that it is simple one/ is well suited to use in mobile transmitters.(B) is an arrangement using a transformer coup/eel modulator, and offers noparticular advantages. The arrangement at (C) is capable of giving goocl modulationlinearity clue to the low impedance of the cathode-follower modulator. However, clueto the relatively low heater-cathode ratings on tubes suited for use as the modulator,a separate heater supply for the moclulator tube normally is requirecl. This limi·tation mokes application of the circuit to the mobile transmitter a special problem,since an isolatecl heater supply normally is not available. Shown at (D) as an assist·once in the tuning of a screen·moclulatecl transmitter (or any efficiency·moclulatecltransmitter for that matter) is the type of moclulation envelope which results whenloading to the moclulatecl stage is insufficient.grid stage since it acts as a relatively lowimpedancesource of modulating voltage forthe screen-grid circuit. In addition the cathodefollowermodulator allows the supply voltageboth for the modulator and for the screen gridof the modulated tube to be obtained from thehigh-voltage supply for the plate of the screengridtube or beam tetrode. In the usual casethe plate supply for the cathode follower, andhence for the screen grid of the modulatedtube, may be taken from the bleeder on thehigh-voltage power supply. A tap on the bleedermay be used, or two resistors may be connectedin series to make up the bleeder, with ap-5 is capable of g1vmg good-quality screengridmodulation, and in addition the circuitprovides convenient adjustments for the carrierlevel and the output level on negativemodulation peaks. This latter control, P2 infigure 5, allows the amplifier to be adjustedin such a manner that negative-peak clippingcannot take place, yet the negative modulationpeaks may be adjusted to a level just abovethat at which sideband splatter will occur.The CathodeFollower ModulatorThe cathode follower isideally suited for use asthe modulator for a screen-
<strong>HANDBOOK</strong>Modulation Systems 2 91propriate values such that the voltage appliedto the plate of the cathode follower is ~pp.ropriatefor the tube to be modulated. It IS Importantthat a bypass capacitor be used fromthe plate of the cathode-follower modulatorto ground.The voltage applied to the plate of thecathode follower should be about 100 voltsgreater than the rated screen voltage for thetetrode tube as a c-w Class C amplifier. Hencethe cathode-follower plate voltage should beabout 350 volts for an 815, 2E26, or 8298,about 400 volts for an 807 or 4-125A, about500 volts for an 813, and about 600 volts fora 4-250A or a 4E27. Then potentiometer P1in figure 5 should be adjusted until the carrierlevelscreen voltage on the modulated stageis about one-half the rated screen voltagespecified for the tube as a Class C c-w amplifier.The current taken by the screen of themodulated tube under carrier conditions willbe about one-fourth the normal screen currentfor c-w operation.The only current taken by the cathodefollower itself will be that which will flowthrough the 100,000-ohm resistor between thecathode of the 6L6 modulator and the negativesupply. The current taken from the bleederon the high-voltage supply will be the carrierlevelscreen current of the tube being modulated(which current passes of course throughthe cathode follower) plus that current whichwill pass through the 100,000-ohm resistor.The loading of the modulated stage shouldbe adjusted until the input to the tube is about50 per cent greater than the rated plate dissipationof the tube or tubes in the st~ge. If thecarrier-level screen voltage value Is correctfor linear modulation of the stage, the loadingwill have to be somewhat greater than thatamount of loading which gives maximum outputfrom the stage. The stage may then be modulatedby applying an audio signal to the gridof the cathode-follower modulator, while observingthe modulated envelope on an oscilloscope.If good output is being obtained, a~d t.hemodulation envelope appears as shown In figure4C all is well, except that P2 in figure 5should' be adjusted until negative modulationpeaks, even with excessive modulating signal,do not cause carrier cutoff with its attendantsideband splatter. If the envelope appears asat figure 4D, antenna coupling should be increasedwhile the carrier level is backed downby potentiometer P 1 in figure 5 until a set .ofadjustments is obtained which will give a satisfactorymodulation envelope as shown infigure 4C.Changing BandsAfter a satisfactory set of adjustmentshas been obtained,+250V.22 K2W1~ K POT.P2Figure 5CATHODE-FOLLOWERSCREEN-MODULATION CIRCUITTO SCREEN OF~ODULATED STAGEA cletallecl c/iscusslan of this circuit, whichalso is representee/ in figure 4C, is given inthe accompanying text.-100 v.1t 1s not difficult to readjust the amplifier foroperation on different bands. PotentiometersP1 (carrier level), and P 2 ( neg.ative peak l~vel)may be left fixed after a satisfactory adjustment,with the aid of the scope, has once beenfound. Then when changing bands it is onlynecessary to adjust excitation until the correctvalue of grid current is obtained, and then toadjust antenna coupling until correct platecurrent is obtained. Note that the correct platecurrent for an efficiency-modulated amplifieris only slightly less than the out-of-resonanceplate current of the stage. Hence carrier-levelscreen voltage must be low so that the out_-ofresonanceplate current will not be too h1gh,and relatively heavy antenna coupling must beused so that the operating plate current willbe near the out-of-resonance value, and so thatthe operating input will be slightly greaterthan I. 5 times the rated plate dissipation ofthe tube or tubes in the stage. Since the carrierefficiency of the stage will be only 35 to 40per cent, the tubes will be operating with platedissipation of approximately the rated valuewithout modulation.Speech Clipping in The maximum r-f outputthe Modulated Stage of an efficiency-modulatedstage is limitedby the maximum possible plate voltage swingon positive modulation peaks. In the modulalationcircuit of figure 5 the minimum outputis limited by the minimum vol~age which ~hescreen will reach on a negative modulationpeak, as set by potentiometer P 2 . Hence thescreen-grid-modulated stage, when using themodulator of figure 5, acts effectively as aspeech clipper, provided the modulating signalamplitude is not too much more than that value
292 Amplitude Modulation<strong>THE</strong> R AD I 0which will accomplish full modulation. Withcorrect adjustments of the operating conditionsof the stage it can be made to clip positiveand negative modulation peaks symmetrically.However, the inherent peak clipping ability ofthe stage should not be relied upon as a meansof obtaining a large amount of speech compression,since excessive audio distortion andexcessive screen current on the modulatedstage will result.Characteristics of aTypical ScreenModulated StageAn important characteristicof the screen·modu·lated stage, when usingthe cathode-follower modulator,is that excessive plate voltage on themodulated stage is not required. In fact, fulloutput usually may be obtained with the largertubes at an operating plate voltage from one·half to two-thirds the maximum rated platevoltage for c·w operation. This desirable conditionis the natural result of using a low·impedance source of modulating signal forthe stage.As an example of a typical screen·modu·lated stage, full output of 75 watts of carriermay be obtained from an 813 tube operatingwith a plate potential of only 1250 volts. Noincrease in output from the 813 may be obtainedby increasing the plate voltage, sincethe tube may be operated with full rated platedissipation of 125 watts, with normal plateefficiency for a screen-modulated stage, 37.5per cent, at the 1250-volt potential.The operating conditions of a screen·modu·lated 813 stage are as follows:Plate voltage-1250 voltsPlate current -160 rna.Plate input-200 wattsGrid current-11 rna.Grid bias-110 voltsCarrier screen vol tage-190 voltsCarrier screen current-6 rna.Power output-approx. 75 wattsWith full 100 per cent modulation the platecurrent decreases about 2 rna. and the screencurrent increases about 1 rna.; hence plate,screen, and grid current remain essentiallyconstant with modulation. Referring to figure5, which was the circuit used as modulatorfor the 813, (E 1 ) measured plus 155 volts, ( E2)measured -50 volts, ( E 3 ) measured plus 190volts, ( E 4 ) measured plus 500 volts, and ther.m. s. swing at (E 5 ) for full modulation me as·ured 210 volts, which represents a peak swingof about 296 volts. Due to the high positivevoltage, and the large audio swing, on thecathode of the 6L6 (triode connected) modulatortube, it is important that the heater ofof this tube be fed from a separate filamenttransformer or filament winding. Note also thatthe operating plate-to-cathode voltage on the6L6 modulator tube does not exceed the 360-volt rating of the tube, since the operatingpotential of the cathode is considerably aboveground potential.Suppressor-Grid Still another form of effi·Modulation ciency modulation may beobtained by applying theaudio modulating signal to the suppressor gridof a pentode Class C r·f amplifier. Basically,suppressor-grid modulation operates in thesame general manner as other forms of effi·ciency modulation; carrier plate circuit effi·ciency is about 35 per cent, and antenna couplingmust be rather tight. However, suppressor·grid modulation has one sizeable disadvantage,in addition to the fact that pentode tubes arenot nearly so widely used as beam tetrodeswhich of course do not have the suppressorelement. This disadvantage is that the screen·grid current to a suppressor-grid modulatedamplifier is rather high. The high screen cur·rent is a natural consequence of the rather highnegative bias on the suppressor grid, whichreduces the plate-voltage swing and plate currentwith a resulting increase in the screencurrent.In tuning a suppressor-grid modulated amplifier,the grid bias, grid current, screen voltage,and plate voltage are about the same asfor Class C c·w operation of the stage. Butthe suppressor grid is biased negatively to avalue which reduces the plate-circuit effici·ency to about one-half the maximum obtainablefrom the particular amplifier, with antennacoupling adjusted until the plate input is about1. 5 times the rated plate dissipation of thestage. It is important that the input to thescreen grid be measured to make sure that therated screen dissipation of the tube is notbeing exceeded. Then the audio signal is ap·plied to the suppressor grid. In the normalapplication the audio voltage swing on thesuppressor will be somewhat greater than thenegative bias on the element. Hence sup·pressor-grid current will flow on modulationpeaks, so that the source of audio signal voltagemust have good regulation. Tubes suitablefor suppressor-grid modulation are: 2E22,837, 4E27/8001, 5·125, 804 and 803. A typicalsuppressor-grid modulated amplifier isillustrated in figure 6.15-4 Input ModulationSystemsConstant efficiency variable-input modula·tion systems operate by virtue of the addition
<strong>HANDBOOK</strong>Plate Modulation 293R.F. INPUT ----..--1-CARRIEROUTPUT:.33W.IG=8 MA.....-+-..+--+---+++-l~+------'E22. K6J5 +1500 v."'"'"'€1PEAK+300V. -210V.Figure 6SWING FOR FULLMODULATION= 210 V.AMPLIFIER WITH SUPPRESSOR-GRIDMODULATIONRecommended operating conditions for linearsuppressor-grid modulation of a 4E27/2578/8001 stage ore given on the drawing.of external power to the modulated stage toeffect the modulation. There are two generalclassifications that come under this heading;those systems in which the additional poweris supplied as audio frequency energy from amodulator, usually called plate modulationsystems, and those systems in which the additionalpower to effect modulation is suppliedas direct current from the plate supply.Under the former classification comes Heisingmodulation (probably the oldest type ofmodulation to be applied to a continuous carrier),Class B plate modulation, and seriesmodulation. These types of plate modulationare by far the easiest to get into operation,and they give a very good ratio of power inputto the modulated stage to power output; 65 to80 per cent efficiency is the general rule. Itis for these two important reasons that thesemodulation systems, particularly Class B platemodulation, are at present the most popularfor communications work.Modulation systems coming under the secondclassification are of comparatively recentdevelopment but have been widely applied tobroadcast work. There are quite a few systemsin this class. Two of the more widely usedare the Doherty linear amplifier, and the Terman-Woodyardhigh-efficiency grid-modulatedamplifier. Both systems operate by virtue ofa carrier amplifier and a peak amplifier connectedtogether by electrical quarter-wavelines. They will be described later in thissection.Plate ModulationPlate modulation is the applicationof the audio powerto the plate circuit of an r-f amplifier. The r·famplifier must be operated Class C for thistype of modulation in order to obtain a radiofrequencyoutput which changes in exact accordancewith the variation in plate voltage.The r-f amplifier is 100 per cent modulatedwhen the peak a-c voltage from the modulatoris equal to the d. c. voltage applied to the r- ftube. The positive peaks of audio voltage increasethe instantaneous plate voltage on ther-f tube to twice the d-e value, and the negativepeaks reduce the voltage to zero.The instantaneous plate current to the r-fstage also varies in accordance with the modulatingvoltage. The peak alternating currentin the output of a modulator must be equal tothe d-e plate current of the Class C r-f sta8eat the point of 100 per cent modulation. Thiscombination of change in audio voltage andcurrent can be most easily referred to in termsof audio power in watts.In a sinusoidally modulated wave, the antennacurrent increases approximately 22 percent for 100 per cent modulation with a puretone input; an r-f meter in the antenna circuitindicates this increase in antenna current.The average power of the r-f wave increases50 per cent for 100 per cent modulation, theefficiency remaining constant.This indicates that in a plate-modulatedradiotelephone transmitter, the audio-frequencychannel must supply this additional 50 percent increase in average power for sine-wavemodulation. If the power input to the modulatedstage is 100 watts, for example, theaverage power will increase to 150 watts at100 per cent modulation, and this additional50 watts of power must be supplied by themodulator when plate modulation is used. Theactual antenna power is a constant percentageof the total value of input power.One of the advantages of plate (or power)modulation is the ease with which proper adjustmentscan be made in the transmitter. Also,there is less plate loss in the r-f amplifier fora given value of carrier power than with otherforms of modulation because the plate efficiencyis higher.By properly matching the plate impedanceof the r-f tube to the output of the modulator,the ratio of voltage and current swing to d-evoltage and current is automatically obtained.The modulator should have a peak voltageoutput equal to the average d-e plate voltageon the modulated stage. The modulator shouldalso have a peak power output equal to thed-e plate input power to the modulated stage.The average power output of the modulator willdepend upon the type of waveform. If the am·plifier is being Heising modulated by a ClassA stage, the modulator must have an average
294 Amplitude Modulation<strong>THE</strong> R AD I 0MODU~ATtD C~ASS CR.f'. AMPLIFIERR,f. DRIVE+BFigure 7HEISING PLATE MODULATIONThis type of moc/ulotion was the first formof plate moc/ulotlon. It is sometimes knownas "constant current'' moclulation. Becauseof the eHeetive 1:1 ratio of the couplingehoke, it is impossible to obtain 700 per eentmodulation unless the plate voltage to themoc/ulotec/ stage is dropped slightly by resistorR. The eopoeitor C merely bypassesthe audio around R, so thot the full o-f outputvoltage of the moc/ulotor is Impressedon the Closs C stage.MOD.+BR.F.+BFigure 8CLASS B PLATE MODULATIONThis type of moc/ulotion is the most flexiblein thot the loading adjustment eon be modein o short period of time one/ without el aborotetest equipment alter o ehonge in operotingfrequency of the Class C amplifier hasbeen mac/e.power output capability of one-half the inputto the Class C stage. If the modulator is aClass B audio amplifier, the average powerrequired of it may vary from one· quarter to morethan one-half the Class C input dependingupon the waveform. However, the peak poweroutput of any modulator must be equal to theClass C input to be modulated.HeisingModulationHeising modulation is the oldestsystem of plate modulation, andusually consists of a Class Aaudio amplifier coupled to the r·f amplifier bymeans of a modulation choke coil, as shownin figure 7.The d.c. plate voltage and plate current inthe r·f amplifier must be adjusted to a valuewhich will cause the plate impedance to matchthe output of the modulator, since the modula·tion choke gives a 1-to-1 coupling ratio. Aseries resistor, by-passed for audio frequen·cies by means of a capacitor, must be connect·ed in series with the plate of the r·f amplifierto obtain modulation up to 100 per cent. Thepeak output voltage of a Class A amplifierdoes not reach a value equal to the d-e voltageapplied to the amplifier and, consequently,the d-e plate voltage impressed across ther·f tube must be reduced to a value equal tothe maximum available a-c peak voltage if100% modulation is to be obtained.A higher degree of distortion can be toleratedin low-power emergency phone transmitterswhich use a pentode modulator tube, and theseries resistor and by-pass capacitor areusually omitted in such transmitters.Class BPlate ModulationHigh-level Class B platemodulation is the least ex·pensive method of platemodulation. Figure 8 shows a conventionalClass B plate-modulated Class C amplifier.The statement that the modulator outputpower must be one-half the Class C input for100 per cent modulation is correct only if thewaveform of the modulating power is a sinewave. Where the modulator waveform is unclippedspeech, the average modulator powerfor 100 per cent modulation is considerablyless than one-half the Class C input.Power Relations inSpeech WaveformsIt has been determined experimentallythat the ratioof peak to average powerin a speech waveform is approximately 4 to 1as contrasted to a ratio of 2 to 1 in a sinewave. This is due to the high harmonic contentof such a waveform, and to the fact that
<strong>HANDBOOK</strong>Plate Modulation 2 95this high harmonic content manifests itself bymaking the wave unsymmetrical and causingsharp peaks or "fingers" of high energy contentto appear. Thus for unclipped speech, theaverage modulator plate current, plate dissipation,and power output are approximatelyone-half the sine wave values for a given peakoutput power.Both peak power and average power arenecessarily associated with waveform. Peakpower is just what the name implies; the powerat the peak of a wave. Peak power, althoughof the utmost importance in modulation, is ofno great significance in a-c power work, exceptinsofar as the average power may be determinedfrom the peak value of a known waveform.There is no time element implied in thedefinition of peak power; peak power may beinstantaneous-and for this reason averagepower, which is definitely associated withtime, is the important factor in plate dissipation.It is possible that the peak power of agiven waveform be several times the averagevalue; for a sine wave, the peak power is twicethe average value, and for unclipped speechthe peak power is approximately four timesthe average value. For 100 per cent modulation,the peak (instantaneous) audio powermust equal the Class C input, although theaverage power for this value of peak varieswidely depending upon the modulator waveform,being greater than 50 per cent for speechthat has been clipped and filtered, 50 per centfor a sine wave, and about 25 per cent for typicalunclipped speech tones.Modulation The modulation transformer isTransformer a device for matching the loadColculations impedance of the Class C amplifierto the recommended loadimpedance of the Class B modulator tubes.Modulation transformers intended for communicationswork are usually designed tocarry the Class C plate current through theirsecondary windinss, as shown in figure 8.The manufacturer's ratings should be consultedto insure that the d-e plate currentpassed through the secondary winding doesnot exceed the maximum rating.A detailed discussion of the method ofmaking modulation transformer calculationshas been given in Chapter Six. However, toemphasize the method of making the calcula·tion, an additional example will be given.Suppose we take the case of a Class C amplifieroperating at a plate voltage of 2000with 225 rna. of plate current. This amplifierwould present a load resistance of 2000 dividedby 0.225 amperes or 8888 ohms, The platepower input would be 2000 times 0.225 or 450watts. By reference to Chapter Six we see thata pair of 8ll tubes operating at 1500 platevolts will deliver 225 watts of audio output.The plate-to-plate load resistance for thesetubes under the specified operating conditionsis 18,000 ohms. Hence our problem is to matchthe Class C amplifier load resistance of 8888ohms to the 18,000-ohm load resistance requiredby the modulator tubes.A 200-to-300 watt modulation transformerwill be required for the job. If the taps on thetransformer are given in terms of impedancesit will only be necessary to connect the sec·ondary for 8888 ohms (or a value approximatelyequal to this such as 9000 ohms) and the primaryfor 18,000 ohms. If it is necessary todetermine the proper turns ratio required of thetransformer it can be determined in the followingmanner. The square root of the impedanceratio is equal to the turns ratio, hence:- 18888 = ..; 0.494v TsOoO0.703The transformer must have a turns ratio ofapproximately 1-to-0. 7 step down, total primaryto total secondary. The greater numberof turns always goes with the higher impedance,and vice versa.Plote-and-ScreenModulationWhen only the plate of ascreen-grid tube is modulated,it is impossible to obtainhigh-percentage linear modulation underordinary conditions. The plate current of sucha stage is not linear with plate voltage. However,if the screen is modulated simultaneouslywith the plate, the instantaneous screen voltagedrops in proportion to the drop in the platevoltage, and linear modulation can then be obtained.Four satisfactory circuits for accomplishingcombined plate and screen modulationare shown in figure 9.The screen r-f by-pass capacitor C 2 , shouldnot have a greater value than 0.005 /lfd., preferablynot larger than 0. 001 /lfd. It should belarge enough to bypass effectively all r-f voltagewithout short-circuiting high-frequencyaudio voltages. The plate by-pass capacitorcan be of any value from 0. 002 /lfd. to 0. 005ftld. The screen-dropping resistor, R1, shouldreduce the applied high voltage to the valuespecified for operating the particular tube inthe circuit. Capacitor C 1 is seldom requiredyet some tubes may require this capacitor inorder to keep c2 from attenuating the high frequencies.Different values between .0002 and.002 ftfd. should be tried for best results.Figure 9C shows another method which usesa third winding on the modulation transformer,through which the screen-grid is connected to
2 96 Amplitude Modulation <strong>THE</strong> R A 0 I 0B+ S.G. B+B+S.G. B+Figure 9PLATE MODULATION OF A BEAM TETRODE OR SCREEN-GRID TUBEThese alternative arrangements for plate moc/ulat/on of tetroc/es or pentacles are clis·cussec/ In cletail in the text, The arrangements shown at (B) or (D) are recommenc/ec/for most applications,a low-voltage power supply. The ratio of turnsbetween the two output windings depends uponthe type of screen-grid tube which is beingmodulated. Normally it will be such that thescreen voltage is being modulated 60 per centwhen the plate voltage is receiving 100 percent modulation.If the screen voltage is derived from a droppingresistor (not a divider) that is bypassedfor r.f. but not a.f., it is possible to securequite good modulation by applying modulationonly to the plate. Under these conditions, thescreen tends to modulate itself, the screenvoltage varying over the audio cycle as a resultof the screen impedance increasing withplate voltage, and decreasing with a decreasein plate voltage. This circuit arrangement isillustrated in figure 9B.A similar application of this principle isshown in figure 9D. In this case the screenvoltage is fed directly from a low-voltage supplyof the proper potential through a choke L.A conventional filter choke having an inductancefrom 10 to 20 henries will be satisfactoryfor L.To afford protection of the tube when platevoltage is not applied but screen voltage issupplied from the exciter power supply, whenusing the arrangement of figure 9D, a resistorof 3000 to 10, 000 ohms can be connected inseries with the chokeL. In this case the screensupply voltage should be at least 1Yz_ times as
<strong>HANDBOOK</strong>Cathode Modulation 2 97much as is requtred tor actual screen voltage,and the value of resistor is chosen such thatwith normal screen current the drop throughthe resistor and choke will be such that normalscreen voltage will be applied to the tube.When the plate voltage is removed the screencurrent will increase greatly and the dropthrough resistor R will increase to such avalue that the screen voltage will be loweredto the point where the screen dissipation onthe tube will not be exceeded. However, thesupply voltage and value of resistor R mustbe chosen carefully so that the maximum ratedscreen dissipation cannot be exceeded. Themaximum possible screen dissipation usingthis arrangement is equal to: W = E'/ 4R whereE is the screen supply voltage and R is thecombined resistance of the resistor in figure9D and the d-e resistance of the choke L. Itis wise, when using this arrangement to check,using the above formula, to see that the valueof W obtained is less than the maximum ratedscreen dissipation of the tube or tubes usedin the modulated stage. This same system canof course also be used in figuring the screensupply circuit of a pentode or tetrode amplifierstage where modulation is not to beapplied.The modulation transformer for plate-andscreen-modulation,when utilizing a droppingresistor as shown in figure 9A, is similar tothe type of transformer used for any platemodulated phone. The combined screen andplate current is divided into the plate voltagein order to obtain the Class C amplifier loadimpedance. The peak audio power required toobtain 100 per cent modulation is equal to thed-e power input to the screen, screen resistor,and plate of the modulated r-f stage.15-5 Cathode ModulationCathode modulation offers a workable compromisebetween the good plate efficiency butexpensive modulator of high-level plate modulation,and the poor plate efficiency but inexpensivemodulator of grid modulation. Cathodemodulation consists essentially of an admixtureof the two.The efficiency of the average well-designedplate-modulated transmitter is in the vicinityof 75 to 80 per cent, with a compromise perhapsat 77.5 per cent. On the other hand, theefficiency of a good grid-modulated transmittermay run from 28 to maybe 40 per cent, withthe average falling at about 34 per cent. Nowsince cathode modulation consists of simultaneousgrid and plate modulation, in phasewith each other, we can theoretically obtainany efficiency from about 34 to 77.5 per centfrom our cathode-modulated stage, dependingupon the relative percentages of grid andplate modulation.Since the system is a compromise betweenthe two fundamental modulation arrangements,a value of efficiency approximately half waybetween the two would seem to be the bestcompromise. Experience has proved this to bethe case. A compromise efficiency of about56.5 per cent, roughly half way between thetwo limits, has proved to be optimum. Calculationhas shown that this value of efficiencycan be obtained from a cathode-modulatedamplifier when the audio-frequency modulatingpower is approximately 20 per cent ofthe d-e input to the cathode-modulated stage.An EconomicalSeries CathodeModulatorSeries cathode modulation isideally suited as an economicalmodulating arrangementfor a high-power triode c-wtransmitter. The modulator can be constructedquite compactly and for a minimum componentcost since no power supply is required for it.When it is desired to change over from c-w to'phone, it is only necessary to cut the seriesmodulator into the cathode return circuit of thec-w amplifier stage. The plate voltage for themodulator tubes and for the speech amplifieris taken from the cathode voltage drop of themodulated stage across the modulator unit.Figure 10 shows the circuit of such a modulator,designed to cathode modulate a Class Camplifier using push-pull 810 tubes, runningat a supply voltage of 2500, and with a plateinput of 660 watts. The modulated stage runsat about 50% efficiency, giving a power outputof nearly 350 watts, fully modulated. The voltagedrop across the cathode modulator is 400volts, allowing a net plate to cathode voltageof 2100 volts on the final amplifier. The platecurrent of the 810' s should be about 330 rna.,and the grid current should be approximately40 rna., making the total cathode current of themodulated stage 370 rna. Four parallel 6L6modulator tubes can pass this amount of platecurrent without difficulty. It must be rememberedthat the voltage drop across the cathodemodulator is also the cathode bias of the modulatedstage. In most cases, no extra grid biasis necessary. If a bias supply is used for c-woperation, it may be removed for cathode modulation,as shown in figure 11. With low-mutriodes, some extra grid bias (over and abovethat amount supplied by the cathode modulator)may be needed to achieve proper linearity ofthe modulated stage. In any case, proper operationof a cathode modulated stage should bedetermined by examining the modulated outputwaveform of the St!lge on an oscilloscope.ExcitationThe r-f driver for a cathode-modulatedstage should have about
2 98 Amplitude Modulation<strong>THE</strong>R AD I 06AU66AU66L6 6L6 6L6TO CATHOOEMODULATEDSTAGE~4~.M7~K--.---~:;~-= 10lHJF2.5KlOW+ IOK,IW +·~ ·~ALL R-ESISTORS 0.5 WATT UNLESSO<strong>THE</strong>RWISE NOTED.ALL CAPACITORS IN .lJF UNLESSO<strong>THE</strong>RWISE NOTED.CAUTION -FILAMENTS OF t!JL6 TUBES MUST BE AT OPERATINGTEMPERATURE BEFORE PLATE VOLTAGE IS APPLIEDTO MODULATED AMPLIFIER.Figure 10SERIES CATHODE MODULATOR FOR A HIGH-POWERED TRIODE R-FAMPLIFIERthe same power output capabilities as wouldbe required to drive a c·w amplifier to the sameinput as it is desired to drive the cathode·modulated stage. However, some form of exci·tation control should be available since theamountof excitation power has a direct bearingon the linearity of a cathode-modulated am·plifier stage. If link coupling is used betweenthe driver and the modulated stage, variationin the amount of link coupling will affordiiJIIple excitation variation. If much less than40% plate modulation is employed, the stagebegins to resemble a grid-bias modulatedstage, and the necessity for good r·f regula·tion will apply.Cathode Modulationof TetrodesCathode modulation hasnot proved too satisfac·tory for use with beamtetrode tubes. This is a result of the smallexcitation and grid swing requirements forsuch tubes, plus the fact that some means forholding the screen voltage at the potential ofthe cathode as far as audio is concerned isusually necessary. Because of these factors,cathode modulation is not recommended foruse with tetrode r·f amplifiers.15-6 The Doherty and theTermon-WoodyardModulated AmplifiersThese two amplifiers will be described togethersince they operate upon very similarprinciples. Figure 12 shows a greatly simplifiedschematic diagram of the operation of bothtypes. Both systems operate by virtue of a carriertube (V 1in both figures 12 and 13) whichsupplies the unmodulated carrier, and whoseoutput is reduced to supply negative peaks,and a peak tube (V 2) whose function is tosupply approximately half the positive peakof the modulation cycle and whose additionalfunction is to lower the load impedance on thecarrier tube so that it will be able to supplythe other half of the positive peak of the modulationcycle.The peak tube is enabled to increase theoutput of the carrier tube by virtue of an impedanceinverting line between the plate circuitsof the two tubes. This line is designedto have a characteristic impedance of one-halfthe value of load into which the carrier t1,1beoperates under the carrier conditions. Then aload of one-half the characteristic impedanceof the quarter-wave line is coupled into theoutput. By experience with quarter-wave linesin antenna-matching circuits we know thatsuch a line will vary the impedance at oneend of the line in such a manner that the geometricmean between the two terminal impedanceswill be equal to the characteristic impedanceof the line. thus, if we have a valueof load of one-half the characteristic impedanceof the line at one end, the other end ofthe line will present a value of twice the characteristicimpedance of the lines to the car·rier tube V 1 •This is the situation that exists under thecarrier conditions when the peak tube merelyfloats across the load end of the line and con·tributes no power. Then as a positive peak ofmodulation comes along, the peak tube startsto contribute power to the load until at thepeak of the modulation cycle it is contributingenough power so that the impedance at theload end of the line is equal to R, instead of
<strong>HANDBOOK</strong>Doherty Amplifier 2 99R.F. AMPLIFIERFigure 12DIAGRAMMATIC REPRESENTATION OF<strong>THE</strong> DOHERTY LINEARt.AJC 6AU6 6AU8 4-6L6'SFigure 11CATHODE MODULATOR INSTALLATIONSHOWING PHONE-C.W. TRANSFER SWITCHthe R/2 that is presented under the carrierconditions. This is true because at a positivemodulation peak (since it is delivering fullpower) the peak tube subtracts a negativeresistance of R/2 from the load end of theline.Now, since under the peak condition of modu·lation the load end of the line is terminatedin R ohms instead of R/2, the impedance atthe carrier-tube will be reduced from 2R ohmsto R ohms. This again is due to the impedanceinverting action of the line. Since the load re·sistance on the carrier tube has been reducedto half the carrier value, its output at the peakof the modulation cycle will be doubled. Thuswe have the p.ecessary condition for a 100per cent modulation peak; the amplifier willdeliver four times as much power as it doesunder the carrier conditions.On negative modulation peaks the peak tubedoes not contribute; the output of the carriertube is reduced until on a 100 per cent nega·tive peak its output is zero.The ElectricalQuarter-WaveLineWhile an electrical quarter·wave line (consisting of a pinetwork with the inductanceand capacitance units havinga reactance equal to the characteristic impedanceof the line) does have the desired im·pedance·inverting effect, it also has the undesirableeffect of introducing a 90° phaseshift across such a line. If the shunt elementsare capacitances, the phase shift across theline lags by 90°; if they are inductances, thephase shift leads by 90°. Since there is an un-desirable phase shift of 90° between the platecircuits of the carrier and peak tubes, an equaland opposite phase shift must be introduced inthe exciting voltage to the grid circuits of thetwo tubes so that the resultant output in theplate circuit will be in phase. This additionalphase shift has been indicated in figure 12 anda method of obtaining it has been shown infigure 13.Comparison BetweenLinear ondGrid ModulatorThe difference betweenthe Doherty linear jllll·plifier and the TermanWoodyard grid-modulatedamplifier is the same as the difference betweenany linear and grid-modulated stages. Modulatedr.f.is applied to the grid circuit of the Dohertylinear amplifier with the carrier tube biased tocutoff and the peak tube biased to the pointwhere it draws substantially zero plate currentat the carrier condition.In the Terman·Woodyard grid-modulated amplifierthe carrier tube runs Class C with com·paratively high bias and high plate efficiency,while the peak tube again is biased so that itdraws almost no plate current. Unmodulatedr.f. is applied to the grid circuits of the twotubes and the modulating voltage is insertedin series with the fixed bias voltages. Fromone-half to two-thirds as much audio voltageis required at the grid of the peak tube as isrequired at the grid of the carrier tube.OperatingEfficienciesThe resting carrier efficiency ofthe grid-modulated amplifier mayrun as high as is obtainable inany Class C stage, 80 per cent or better. Theresting carrier efficiency of the linear will beabout as good as is obtainable in any ClassB amplifier, 60 to 70 per cent. The overallefficiency of the bias-modulated amplifier at100 per cent modulation will run about 75 percent; of the linear, about 60 per cent.In figure 13 the plate tank circuits are detunedenough to give an effect equivalent tothe shunt elements of the quarter-wave "line"of figure 12. At resonance, the coils L, andL2 in the grid circuits of the two tubes have
300 Amplitude Modulation<strong>THE</strong> R AD I 0other undesirable features which make theiruse impracticable alongside the more conventionalmodulation systems. Nearly all thesecircuits have been published in the I. R. B.Proceedings and the interested reader can referto them in back copies of that journal.15-7 Speech ClippingFigure 13SIMPLIFIED SCHEMATIC OF A"HIGH EFFICIENCY" AMPLIFIERThe basic system, comprising a "carrier"tube and a "peak" tube interconnecled bylumped-constant quarter-wave lines, is thesame lor either gricl·bias moclulation or loruse as a linear amplifier of a moclulateclwave.each an inductive reactance equal to the capacitivereactance of the capacitor C1. Thuswe have the effect of a pi network consistingof shunt inductances and series capacitance.In the plate circuit we want a phase shift ofthe same magnitude but in the opposite direc·tion; so our series element is the inductanceL3 whose reactance is equal to the characteristicimpedance desired of the network. Thenthe plate tank capacitors of the two tubes c2and C3 are increased an amount past resonance,so that they have a capacitive reactanceeq~al to_ the_ inductive reactance of the coil L3.It ts qutte tmportant that there be no couplingbetween the inductors.Although both these types of amplifiers arehighly efficient and require no high-level audioequipment, they are difficult to adjust-particularlyso on the higher frequencies-and itwould be an extremely difficult problem to d~signa multiband transmitter employing thecircuit. However, the grid-bias modulation systemhas advantages for the high-power transmitterwhich will be operated on a single frequencyband.Other High-Efficiency Many other high-efficienModulation Systems cy modulation systemshave been described sinceabout 1936. The majority of these, howeverhave received little application either by commercialinterests or by amateurs. In most casesthe circuits are difficult to adjust, or they haveSpeech waveforms are characterized by frequentlyrecurring high-intensity peaks of veryshort duration. These peaks will cause overmodulationif the average level of modulationon loud syllables exceeds approximately 30per cent. Careful checking into the nature ofspeech sounds has revealed that these highintensitypeaks are due primarily to the vowelsounds. Further research has revealed that thevowel sounds add little to intelligibility, themajor contribution to intelligibility comingfrom the consonant sounds such as v b k st, and /. Measurements have shown' that 'th~power contained in these consonant soundsmay be down 30 db or more from the energy inthe vowel sounds in the same speech passage.Obviously, then, if we can increase the relativeenergy content of the consonant soundswith respect to the vowel sounds it will bepossible to understand a signal modulated withsuch a wave-form in the presence of a muchhigher level of background noise and interference.Experiment has shown that it is possibleto accomplish this desirable result simplyby cutting off or clipping the high-intensitypeaks and thus building up in a relative mannerthe effective level of the weaker sounds.Such clipping theotetically can be accomplishedsimply by increasing the gain of thespeech amplifier until the average level ofmodulation on loud syllables approaches 90per cent. This is equivalent to increasing thespeech power of the consonant sounds by about10 times or, conversely, we can say that 10 dbof clipping has been applied to the voice wave.However, the clipping when accomplished inthis manner will produce higher order sidebandsknown as "splatter," and the transmittedsignal would occupy a relatively tremendousslice of spectrum. So another method of accomplishingthe desirable effects of clipping mustbe employed.A considerable reduction in the amount ofsplatter caused by a moderate increase in thegain of the speech amplifier can be obtainedby poling the signal from the speech amplifierto the transmitter such that the high-intensitypeaks occur on upward or positive modulation.Overloading on positive modulation peaks producesless splatter than the negative-peakclipping which occurs with overloading on the
\<strong>HANDBOOK</strong>Speech Clipping 301Figure 14SPEECH-WAVEFORM AMPLITUDEMODULATIONShowing the effect of using the prop·er polarity of a speech wave formodulating a transmitter. (A) showsthe effect of proper speech polarityon a transmitter having an upwardmodulation capability of greaterthan 100 per cent. (B) shows the ®effect of using proper speech polar·ity on a transmitter having an up·ware/ modulation capability of only100 per cent. Both these conditionswill give a clean signal withoutob;ectionable splatter. (C) showsthe effect of the use of improperspeech polarity. This condition willcause serious splatter clue to nega·tive·peak clipping in the moclulatecl· @amp/ ifier stage.0negative peaks of modulation. This aspect ofthe problem has been discussed in more detailin the section on Speech Waveform Dissymmetryearlier in this chapter. The effect of feedingthe proper speech polarity from the speech am·plifier is shown in figure 14.A much more desirable and effective methodof obtaining speech clipping is actual! y to em·ploy a clipper circuit in the earlier stages ofthe speech amplifier, and then to filter out theobjectionable distortion components by meansof a sharp low·pass filter having a cut·off frequencyof approximately 3000 cycles. Tests onclipper-filter speech systems have shown that6 db of clipping on voice is just noticeable,12 db of clipping is quite acceptable, andvalues of clipping from 20 to 25 db are tolerableunder s..;ch conditions that a high degreeof clipping is necessary to get through heavyQRM or QRN. A signal with 12 db of clippingdoesn't sound quite natural but it is not unpleasantto listen to and is much more readablethan an unclipped signal in the presenceof strong interference.The use of a clipper-filter in the speech amplifier,to be completely effective, requiresthat phase shift between the clipper-filterstage and the final modulated amplifier be keptto a mm1mum. However, if there is phase shiftafter the clipper-filter the system does notcompletely break down. The presence of phaseshift mere! y requires that the audio gain followingthe clipper-filter be reduced to the pointwhere the cant applied to the clipped speechwaves still cannot cause overmodulation. Thiseffect is illustrated in figures 15 and 16.The cant appearing on the tops of the squarewaves leaving the clipper-filter centers aboutthe clipping level. Hence, as the frequencybeing passed through the system is lowered,the amount by which the peak of the cantedwave exceeds the clipping level is increased.Phase ShiftCorrectionIn a normal transmitter having amoderate amount of phase shiftthe cant applied to the tops ofthe waves will cause overmodulation on frequenciesbelow those for which the gain followingthe clipper-filter has been adjusted unlessremedial steps have been taken. The followingsteps are advised:( 1) Introduce bass suppression into the speechamplifier ahead of the clipper-filter.(2) Improve the low-frequency response characteristicinsofar as it is possible in the
3 02 Amplitude Modulation <strong>THE</strong> R AD I 0®INCOMING SPEECH WAVECLIPPED AND FILTERED SPEECH WAVENEGATIVE CLIPPING LEVELFigure 15ACTION OF A CLIPPER-FILTERON A SPEECH WAVEThe drawing (A) shows the incomIng speech wave before it reachesthe clipper stage. (B) shows theoutput of the clipper-filter, illus·tratlng the manner in which theAVERAGE LEVEL peaks are clipped and then thesharp edges of the clipped waveremoved by the filter. (C) showsthe eHect of phase shift in thestages following the clipper-filter.(C) a/so shows the manner in whichthe transmiHer may be adjusted for100 per cent modulation of' the"canted" peaks of the wave, thesloping top of the wave reachingAVERA VE about 70 per cent modulation.- _fQQ ~P_Q_S!.!.!~ MO...QUh_A..II~___ _!0~ _!()!JTJYE_MO_Q_UJ..~TI,P~__ .!._O~N~GA_!!V~QQ~A_I!O~100 "'b NEGATIVE MODULATIONMODULATED WAVE AFTER UNDERGOING PHASE SHIFTstages following the clipper-filter. Feedingthe plate current to the final amplifierthrough a choke rather than through thesecondary of the modulation transformerwill help materially.Even with the normal amount of improvementwhich can be attained through the steps mentionedabove there will still be an amount ofwave cant which must be compensated in somemanner. This compensation can be done ineither of two ways. The first and simpler wayis as follows:(1) Adjust the speech gain ahead of the clipper-filteruntil with normal talking intothe microphone the distortion being introducedby the clipper-filter circuit is quiteapparent but not objectionable. This amountof distortion will be apparent to the normallistener when 10 to 15 db of clipping istaking place.( 2) Tune a selective communications receiverabout 15 kc. to one side or the other of thefrequency being transmitted. Use a shortantenna or no antenna at all on the receiverso that the transmitter is not blockingthe receiver.(3) Again with the normal talking into themicrophone adjust the gain following theclipper-filter to the point where the sidebandsplatter is being heard, and thenslightly back off the gain after the clipper-filteruntil the splatter disappears.If the phase shift in the transmitter or modulatoris not excessive the adjustment proceduregiven above will allow a clean signal to beradiated regardless of any reasonable voicelevel being fed into the microphone.If a cathode-ray oscilloscope is availablethe modulated envelope of the transmittershould be checked with 30 to 70 cycle sawtoothwaves on the horizontal axis. If the upperhalf of the envelope appears in general thesame as the drawing of figure 15C, all is welland phase-shift is not excessive. However, ifmuch more slope appears on the tops of thewaves than is illustrated in this figure, it willbe well to apply the second step in compensationin order to insure that sideband splattercannot take place and to afford a still higheraverage percentage of modulation. This secondstep consists of the addition of a high-levelsplatter suppressor such as is illustrated infigure 17.
<strong>HANDBOOK</strong>Splatter Suppression 303+ l\- 71 - 7\- -r;: -l'i - 7\- -1'1 - ..,..- -ri - 7'1 - ~-0~~4-~~~4-~-+~+-~-+4-~~_L3000 '\..WAVE+B MOO.FIL. TRANS.---INSULATED(0001 FOR H.V.11!JV.A.C.TOPLATE-MODULATEDCLASS-C AMPLIFIER7500-10000 OHMSLOAD\..r--+B R.F. FINALI 000 '\..WAVEFigure 17HIGH-LEVEL SPLATTER SUPPRESSORThis circuit is effective In reclucing splattercausecl by negative-peak clipping in the moclulateclamplifier stage. The use of a twosectionFilter as shown is recommenclec/, althougheither a single m-clerivecl or a con•stant·k section may be usecl for greater econ·omy. Suitable chokes, a Ion g with recom·menclecl capacitor values, are available fromseveral manufacturers.300 '\..WAVEFigure 16ILLUSTRATING <strong>THE</strong> EFFECT OF PHASESHIFT AND FILTERED WAVES OF DIF-FERENT FREQUENCYSketch (A} shows the eHect of a clipper ancla filter having a cutoff of about 3500 cycleson a wave of 3000 cycles. Note that no har·monics are present in the wave so that phaseshift following the clipper-filter will haveno significant effect on the shape of thewave. (B) ancl (C) show the effect of phaseshift on waves well below the cutoff frequen·cy of the filter. Note that the" cant" p/acec/upon the top of the wave causes the peakvalue to rise higher ancl higher above theclipping I eve/ as the frequency is lo werecl.It is for this reason that bass suppressionbefore the clipper stage is desirable. lm·provecl low-frequency response following theclipper-filter will recluce the phase shift ancltherefore the canting of the wave at the lowervoice frequencies.The use of a high-level splatter suppressorafter a clipper-filter system will afford the re·sult shown in figure 18 since such a devicewill not permit the negative-peak clippingwhich the wave cant caused by audio-systemphase shift can produce. The high-level splat·ter suppressor operates by virtue of the factthat it will not permit the plate voltage on themodulated amplifier to go completely to zeroregardless of the incoming signal amplitude.Hence negative-peak clipping with its attend·ant splatter cannot take place. Such a devicecan, of course, also be used in a transmitterwhich does not incorporate a clipper-filter sys·tem. However, the full increase in averagemodulation level without serious distortion,afforded by the clipper-filter system, will notbe obtained. ·A word of caution should be noted at thistime in the case of tetrode final modulatedamplifier stages which afford screen voltagemodulation by virtue of a tap or a separatewinding on the modulation transformer suchas is shown in figure 9C of this chapter. Ifsuch a system of modulation is in use, thehigh-level splatter suppressor shown in figure17 will not operate satisfactorily since nega·tive·peak clipping in the stage can take placewhen the screen voltage goes too low.Clipper Circuits Two effective low-level clip·per-filter circuits are shownin figures 19 and 20. The circuit of figure 19employs a 6J6 double triode as a clipper, eachhalf of the 6]6 clipping one side of the ;m·pressed waveform. The optimum level at whichthe clipping operation begins is set by thevalue of the cathode resistor. A maximum of12 to 14 db of clipping may be used with thiscircuit, which means that an extra 12 to 14 dbof speech gain must precede the clipper. Fora peak output of 8 volts from the clipper-filter,a peak audio signal of about 40 volts must beimpressed upon the clipper input circuit. The6C4 speech amplifier stage must therefore beconsidered as a part of the clipper circuit as
304 Amplitude Modulation<strong>THE</strong> R AD I 0\ ,'~, SPLATTER-CAUSING...... NEGATIVE OVERMODULATION PEAKCUTOFF BY "HIGH-LEVEL'SPLATTER SUPPRESSOR!'ZERO AXIS100 CIJb NEG. MODULATIONFigure 18ACTION OF HIGH-LEVELSPLATTER SUPPRESSORA high-level splatter suppressormay be used in a transmitter withouta cl ipper•filter to reduce nega•five-peak clipping, or such a unitmay be used following a clipper·filter to allow a higher averagemodulation level by eliminating thenegative-peak clipping which thewave-cant caused by phase shiftmight produce.it compensates for the 12 to 14 db loss of gainincurred in the clipping process. A simple low·pass filter made up of a 20 henry a.c. • d.creplacement type filter choke and two micacondensers follows the 6]6 clipper. This fil·ter is designed for a cutoff frequency of about3500 cycles when operating into a load im·pedance of Yz megohm. The output level of 8volts peak is ample to drive a triode speechamplifier stage, such as a 6C4 or 6J5.A 6AL5 double diode series clipper is em·ployed in the circuit of figure 20, and a com·mercially made low-pass filter is used to givesomewhat better high frequency cutoff characteristics.A double triode is employed as aspeech amplifier ahead of the clipper circuit.The actual performance of either circuit isabout the same.To eliminate higher order products that maybe generated in the stages following the clipper-filter,it is wise to follow the modulatorwith a high-level filter, as shown in figure 21.Clipper Adjustment These c 1 i p per circuitshave two adjustments:Adjust Gain and Adjust Clipping. The Adj.Gain control determines the modulation levelof the transmitter. This control should be setso that over-modulation of the transmitter isimpossible, regardless of the amount of clippingused. Once the Adj. Gain control hasbeen roughly set, the Adj. Clip. control maybe used to set the modulation level to any percentagebelow 100%. As the modulation levelis decreased, more and more clipping is introducedinto the circuit, until a full 12 db ofclipping is used. This me an s that the Adj.Gain control may be advanced some 12 db pastthe point where the clipping action started.Clipping action should start at 85% to 90%modulation when a sine wave is used for circuitadjustment purposes.High-Level Even though we may have cut offFilters all frequencies above 3000 or 3500cycles through the use of a filtersystem such as is shown in the circuits of figures19 and 20, higher frequencies may againb~ int.rodu_ced into the modulated wave bydistortion 10 stages following the speech amplifier.Harmonics of the incoming audio frequenciesmay be generated in the driver stagefor the modulator; they may be generated in theplate circuit of the modulator; or they may begenerated by non-linearity in the modulatedamplifier itself./ADJUST GAIN500 KTO NEXTGRIDPEAK OUTPUT APPROX.8 V. MAX. WITH JZ 08OF CLIPPING.5K 1 1WIKIW+250 v.ALL RESISTORS 0.5 WArr UNLESSO<strong>THE</strong>RWISE MARKED.ALL CAPACITORS IN .J.JF UNLESSO<strong>THE</strong>RWISE NOTED.Figure 19CLIPPER FILTER USING 6J6 DOUBLE TRIODE STAGE
<strong>HANDBOOK</strong>Splatter Suppression30556 KPEAK OUTPUT APPROX.5 V MAX. WITH 12 08OF CLIPPING+300 v.IK,IWALL RESISTORS 0.5 WATT UNLESSO<strong>THE</strong>RWISE MARKED.ALL CAPACITORS IN JJF UNLESSO<strong>THE</strong>RWISE MARKED.Figure 20CLIPPER FILTER USING 6ALS STAGERegardless of the point in the system followingthe speech amplifier where the highaudio frequencies may be generated, these fre·quencies can still cause a broad signal to betransmitted even though all frequencies above3000 or 3500 cycles have been cut off in thespeech amplifier. The effects of distortion inthe audio system following the speech amplifiercan be eliminated quite effectively throughthe use of a post-modulator filter. Such a filtermust be used between the modulator plate cir·cuit and the r·f amplifier which is being modulated.This filter may take three general forms ina normal case of a Class C amplifier plate mod·ulated by a Class B modulator. The best meth·od is to use a high level low-pass filter asshown in figure 21 and discussed previously.Another method which will give excellent re·sults in some cases and poor results in others,dependent upon the characteristics of the mod·ulation transformer, is to "build out" the mod·ulation transformer into a filter section. Thisis accomplished as shown in figure 22 by plac·ing mica capacitors of the correct value acrossthe primary and secondary of the modulationtransformer. The proper values for the capaci-CLASS C AMPLIFIERMODUL.ATORMODULATOR~IIB+ MOD. 8+ R.F.Figure 21ADDITIONAL HIGH-LEVEL LOW-PASS FIL·TER TO FOLLOW MODULATOR WHEN ALOW-LEVEL CLIPPER FILTER IS USEDSuitable choke, alon9 with recommended ca·pacitor values, is a v a i I a b I e lrom severalmanufacturers.Figure 22"BUILDING-OUT" <strong>THE</strong> MODULATIONTRANSFORMERThis expedient uti/ires the leakage react·once of the moc/ul at ion translormer in con•junction with the capacitors shown to makeup a single-section low·pass li/ter. In orderto determine exact values for CI and c2 plusC3, It is necessary to use a measurementsetup such as Is shown In figure 23. How•ever, experiment has shown in the case of anumber of commercially available modulotiontransformers that a value for C 1 of 0.002-f.l-fd.ancl c2 plus c3 af 0.004-J.Lfc/. will give satls·factory results.
306Amplitude Modulation<strong>THE</strong> R AD I 0AUDIOOSCILLATORR1Figure 23OSCILLOSCOPE2 R 3 .C, yg[\.METE+C3TEST SETUP FOR BUILDING-OUTMODULATION TRANSFORMERThrough the use of a test setup such as isshown ancJ the methocJ cJescribecJ in the textit is possible to determine the correct valuesfor a specified filter characteristic In thebuilt-out modulation transformer.tors cl and c2 must, in the ideal case, be de·termined by trial and error. Experiment witha number of modulators has shown, however,that if a 0. 002 l!fd. capacitor is used for C1,and if the sum of C2 and y is made 0. 004 l!fd.(0.002 l!fd. for C2 and 0. 002 for C3) the idealcondition of cutoff above 3000 cycles will beapproached in most cases with the "multiple·match" type of modulation transformer.If it is desired to determine the optimumvalues of the capacitors across the transformerthis can be determined in several ways, all ofwhich require the use of a calibrated audiooscillator. One way is diagrammed in figure 23.The series resistors R1 and R2 should each beequal to l-i the value of the recommended plate·to-plate load resistance for the Class B modulatortubes. Resistor ~ should be equal to thevalue of load resistance which the Class Cmodulated stage will present to the modulator.The meter V can be any type of a·c voltmeter.The indicating instrument on the secondary ofthe transformer can be either a cathode-rayoscilloscope or a high-impedance a·c volt·meter of the vacuum-tube or rectifier type.With a set-up as shown in figure 23 a plot ofoutput voltage against frequency is made, atall times keeping the voltage across V con·stant, using various values of capacitance forcl and c2 plus c3. When the proper values ofcapacitance have been determined which givesubstantially constant output up to about 3000or 3500 cycles and decreasing output at all fre·quencies above, high-voltage mica capacitorscan be substituted if receiving types were usedin the tests and the transformer connected tothe modulator and Class C amplifier.With the transformer reconnected in thetransmitter a check of the modulated-waveoutput of the transmitter should be made usingan audio oscillator as signal generator and anoscilloscope coupled to the transmitter output.With an input signal amplitude fed to the speech0I2f----.o 0 ~J4ID2- 5z -60f= -7..:::Jz -wII- -..:/9II. I10 III12II?jcfIIIv /v /~ ]Io"~ Tlj_(_'?(~~~~~:'tt:t;,~~7f_,R~~~ FORII I I€f1!.RG=sooK... -JUIll100 200 300 .500 700 1000 2000 3000 5000FREQUENCY (CPS)Figure 24BASE ATTENUATION CHARTFrequency attenuation caused by variousvalues of coupling capacitor with a grlcJ re·sistor of 0.5 megohm in the following stage(RG > RL)amplifier of such amplitude that limiting doesnot take place, a substantially clean sine waveshould be obtained on the carrier of the transmitterat all input frequencies up to the cutofffrequency of the filter system in the speechamplifier and of the filter which includes themodulation transformer. Above these cutofffrequencies very little modulation of the carrierwave should be obtained. To obtain a checkon the effectiveness of the "built out" modulationtransformer, the capacitors across theprimary and secondary should be removed forthe test. In most cases a marked deteriorationin the waveform output of the modulator willbe noticed with frequencies in the voice rangefrom 500 to 1500 cycles being fe
<strong>HANDBOOK</strong>Bass Suppression307bility. This means that the speech level maybe increased considerably without overmodulationor overload of the audio system. Inaddition, if speech clipping is used, attenuationof the lower audio frequencies before theclipper will reduce phase shift and canting ofthe clipper output.A simple method of bass suppression is toreduce the size of the interstage coupling capacitorsin a resistance coupled amplifier.Figure 24 shows the frequency characteristicscauseJ by such a suppression circuit. A secondsimple bass suppression circuit is toplace a small a.c. - d.c. type filter choke fromgrid to ground in a speech amplifier stage, asshown in figure 25.Modulated Amplifier The systems describedDistortion in the preceding paragraphswill have no effectin reducing a broad signal caused by nonlinearityin the modulated amplifier. Eventhough the modulating waveform impressed uponthe modulated stage may be distortion free,if the modulated amplifier is non-linear distortionwill be generated in the amplifier. Theonly way in which this type of distortion maybe corrected is by making the modulated amplifiermore linear. Degenerative feedbackwhich includes the modulated amplifier in theloop will help in this regard.Plenty of grid excitation and high grid biaswill go a long way toward making a platemodulatedClass C amplifier linear, althoughsuch operating conditions will make more difficultthe problem of TVI reduction. If this stilldoes not give adequate linearity, the precedingbuffer stage may be modulated 50 per centor so at the same time and in the same phaseas the final amplifier. The use of a grid leakto obtain the majority of the bias for a ClassC stage will improve its linearity.The linearity of a grid-bias modulated r-famplifier can be improved, after proper adjustmentsof excitation, grid bias, and antennacoupling have been made by modulating thestage which excites the grid-modulated amplifier.The preceding driver stage may be gridbiasmodulated or it may be plate modulated.Modulation of the driver stage should be inthe same phase as that of the final modulatedamplifier.15-8 The Bias-ShiftHeising ModulatorThe simple Class A modulator is limited toan efficiency of about 30%, and the tube mustdissipate the full power input during periodsof quiescence. Class 1\B and class B audiosystems have largely taken the place of theold Heising modulator because of this great.01sr~+B L .. "IOHENRY"'MIDGETA C-DC FILTER CHOKE(STANCOR C-/JJJ)Figure 25USE OF PARALLEL INDUCTANCEFOR BASS SUPPRESSIONwaste of power. It is possible, however, tovary the operating bias of the class A modulatorin such a way as to allow class A operationonly when an audio signal is applied tothe grid of the tube. During resting periods,the bias can be shifted to a higher value,dropping the resting plate current and platedissipation of the tube. When voice waveformshaving low average power are ~mployed, theefficiency of the system is comparable to thepopular class B modulator.The characteristic curve for a class A modulatoris shown in figure 26. Normal bias isused, and the operating point is placed in themiddle of the linear portion of the Eg-Ip curve.Maximum plate input is limited by the platedissipation of the tube under quiescent condition.The bias-shift modulator is biasedclose to plate current cut-off under no signalcondition (figure 27). Resting plate current+1P-IP\ __ -----v__J(;RID INPUTSI(;NALFigure 26CHARACTERISTIC GRIDVOLTAGE-PLATE CURRENTCURVE FOR CLASS AHEISING MODULATORi PLATE~OUTPUTI SI(;NAL
3 08 Amplitude Modulation<strong>THE</strong> <strong>RADIO</strong>IEXCUrSION+lPI-EG/1CUT-OFF@ 0BIASPLATE CURRENT81 AS -SH 1FT ~ -[pEXC URSIONCLASS A OPERATINGBIAS LINEI p (MAX SIGNAL)I p (NO Sf GNAL)+EoCLASS CAMPLIFIERQUIESCENTBIAS LINEFigure 27BIAS-SHIFT MODULATOROPERATING CHARACTERISTICSModulator is biased close to platecurrent cufaoff under no signal,condition, B. Upon applicationof audio signal, the bias of thestage is shifted toward the classA operating point, A. Bias-shiftvoltage is obtained from audiosignal.and plate dissipation are therefore quite low.Upon application of an audio signal, the biasof the stage is shifted toward the class Aoperating point, preventing the negative peaksof the applied audio voltage from cutting offthe plate current of the tube. As the audiovoltage increases, the operating bias point isshifted to the right on figure 27 until the classA operating point is reached at maximum ex·citation.The bias-shift voltage may be obtaineddirectly from the exciting signal by rectification,as shown in figure 28. A simple lowpass filter system is used that will pass onlythe syllabic components of speech. Enoughnegative bias is applied to the bias-shift modulatorto cut the resting plate current to thedesired value, and the output of the bias controlrectifier is polarized so as to "buck"the fixed bias voltage. No spurious modulationfrequencies are generated, since themodulator operates class A throughout theaudio cycle.This form of grid pulsing permits the modulatorstage to work with an pverall efficiencyof greater than 50%, comparing favorably withthe class B modulator. The expensive class Bdriver and output transformers are not required,since resistance coupling may be used in theinput circuit of the bias-shift modulator, andFigure 28BLOCK DIAGRAM OFBIAS-SHIFT MODULATORa heavy-duty filter choke will serve as an impedancecoupler for the modulated stage.Series and Parallel The bias-shift systemControl Circuits make take one of severalforms. A "series" controlcircuit is shown in figure 29. Resting bias isapplied to the bias-shift modulator tube throughthe voltage divider R2/R4. The bias controltube is placed across resistor R2. Quiescentbias for the modulator is set by adjusting R2.As the internal resistance of the bias controltube is varied at a syllabic rate the voltagedrop across R2 will vary in unison. The modulatorbias, therefore varies at the same rate.Excitation for the bias control tube is obtainedfrom the audio signal through potentiometerRl which regulates the amplitude of the controlsignal. The audio signal is rectified bythe bias control rectifier, and filtered by networkR3-Cl in the grid circuit of the bias controltube.The "parallel" control system is illustratedin figure 30. Resting bias for the modulator isobtained from the voltage divider R2/R4.Potentiometer R2 adjusts the resting biaslevel, determining the static plate current ofthe modulator. Resistor R3 serves as a biasresistor for the control tube, reducing its platecurrent to a low level. When an audio signalis applied via Rl to the grid of the controltube the internal resistance is lowered, decreasingthe shunt resistance across R2. Thenegative modulator bias is therefore reduced.The bias axis of the modulator is shifted fromthe cut-off region to a point on the linearportion of the operating curve. The amount ofbias-shift is controlled by the setting ofpotentiometer Rl. Capacitor Cl in conjunctionwith bias resistor R3 form a syllabic filter for
<strong>HANDBOOK</strong>Heising Modulator 309B+ TO MODULATEDR-F AMPLl Fl ERSPEECH·~+B+ 10 MODULATEDR-F AMPLIFIERB lAS CONTROLRECTI Fl ERNE
310304-TL+300 v.CH -CHICA~O SiANOARO SR-300 ('I( LOAO)T 1- CHIC'A'O SiANOARO I'C-~40PTz -cHICA'o srANOARo P-1~9'T.1- 6 VOLTS AT 40 AMP5,NOi!- CONN6Ci ONE Flt.AM£Ni PIN O!J 6$N! CONiROLr1111 ro CArHODI (I' IN J).Figure 31500-WATT BIAS SHIFT MODULATORTwo 304-TL triode tubes in bias controlled class A modulator willeffectively modulate one kilowatt input to the class C amplifier stage.Modulator tubes are operated at 2500 volts, and approximately -280volts bias. Resting bias is ad;usted for static plate current of 40milliamperes.
E & E TECHNI-SHEETDECIBEL CONVERSION CHART-- --- --··--- -- -voLTAGEAND 1 - .. ~ - --- r-voL-i-AGEAND-~c-~s ___;_~~ER RATIO___c;URRENT ~i_ D=-=IB=~S POWER RATio_______cuRRENT RATIO.,, ,. . JF ___,:- -- .....___()_:1 I. ,_02_3___ ____•. 0,_2 _______ ~ ____ ~53 4.467 _0.2 . 1.047 ---1----- 1.023 14 25 119 5.012~, ~ ..m -o.4 1.096 1.047 I, 16 39 611 6.310------0.5 1.122 1.059 17 50.119 7.079o.e ,_,4a , 012. IL-.......- 1s 63.o9e 7.94_3 ___---lr---- 0. 7 1.1_7_5 __, 1.084 =-:]~--- - 19 ' -79.433 8.9_1_3 ____~~:: -- -c-~: ::~:: ·EJ' --~+---=1' -~.::0 :::::: --1.0 1.259 1.122 24 251.19 15.849~---------~----------+---------11.2 I 1318 1.148 26 388,11 19.953'·' I ;"' - ' ---;-;;,----- L 34 ----1----2~- 50.1191.4 1 380 1 175 ~B I 630 96 25.119r------1-.-6----------:-==---• 4~±- 1.202 =~~---~- 1.514 1 230 I 32 I 1564 9 39.611_-~-----j----'-"-oo_._o_o _____ --'----3_1_._6_2_3~~~~~~----1-l2. 2 o 1. 288 L 36 ---+l_::3_9_8..c'c.·c.'____-+--6-3_.c:.oc:.9_6c____2.4 8 1.318 I ---38___ 6309.6 i 79.4332.6 1 1.820 1.349 40 __ l~~co __·c:_o__ ~--+-i __,oo.ooo2.8 ~ 42 1.585 X 10 4 125.893.5 46 3.981X10 4 199.533.0 .44 2.512 X 10 4 _l 158.49---------j4.0 48 6.310-X--10-4~----f--2-5_1_.-,-9-----j4. 5 ------t---------Il--·- --5=-oc-------"ir----, o-5--- ---+--+-. ---3-,-6--'.-2-3----------15.0 3.162 1 778 52 1.585 X 10 5 398.11f------5--. 5--------+-----3-. 5~--;.------- 1. 684 54 2. 51 2 X 10 5 -;~-----------T----------1f----6:_ ____ +----3::_.__c9:_::61 1 . 9 9 5 L- 56 3. 98 1 X 10 6 3 0. 967 5.012 2. .. 239 _L- 58 =t 6._310 X 10 6 794.338 6.3>o 2.5>2 60 10 6 I 1ooo.o9 ___ -~~- ..-9 -,-__ - ______ 2.~---~ 70-- ----,-o-7 _3_,__6_ _ _______,2 3r----------t------------j------';--·-'-6"'2=-~~~~:::_], I 00-40 0__ -~--- -±=--;-;-s ------ ~o_o ___ -1 - 0 7-_11 12.589 3.548 !1 9o _____ ~------l-~'623 or---1-2 _____ +--,-5-.-8-4_9 ___ --t---3-.-9-8-,-------J, 1------1-o--o- , o' o 1 , ooo_o_o_o ____ ___,
CHAPTER SIXTEENFrequency Modulation andRadioteletype TransmissionExciter systems for FM and single sidebandtransmission are basically similar in that modificationof the signal in accordance with theintelligence to be transmitted is normally accomplishedat a relatively low level. Then theintelligence-bearing signal is amplified to thedesired power level for ultimate transmission.True, amplifiers for the two types of signalsare basically different; linear amplifiers of theClass, A or Class B type being used for ssbsignals, while Class C or non-linear Class Bamplifiers may be used for FM amplification.But the principle of low-level generation andsubsequent amplification is standard for bothtypes of transmission.16-1 Frequency ModulationThe use of frequency modulation and theallied system of phase modulation has becomeof increasing importance in recent years. Foramateur communication frequency and phasemodulation offer important advantages in thereduction of broadcast and TV interferenceand in the elimination of the costly high-levelmodulation equipment most commonly employedwith amplitude modulation. For broadcast workFM offers an improvement in signal-to-noiseratio for the high field intensities availablein the local-coverage area of FM and TV broadcaststations.In this chapter various points of differencebetween FM and amplitude modulation transmissionand reception will be discussed andthe advantages of FM for certain types of communicationpointed out. Since the distinguishingfeatures of the two types of transmissionlie entirely in the modulating circuits at thetransmitter and in the detector and limiter circuitsin the receiver, these parts of the communicationsystem will receive the major portionof attention.Modulation Modulation is the process of alteringa radio wave in accordancewith the intelligence to be transmitted. Thenature of the intelligence is of little importanceas far as the process of modulation isconcerned; it is the method by which this in·telligence is made to give a distinguishingcharacteristic to the radio wave which willenable the receiver to convert it back into intelligencethat determines the type of modulationbeing used.Figure 1 is a drawing of an r-f carrier amplitudemodulated by a sine-wave audio voltage.After modulation the resultant modulatedr-f wave is seen still to vary about the zeroaxis at a constant rate, but the strength of theindividual r-f cycles is proportional to the amplitudeof the modulation voltage.In figure 2, the carrier of figure 1 is shownfrequency modulated by the same modulatingvoltage. Here it may be seen that modulationvoltage of one polarity causes the carrier frequencyto decrease, as shown by the fact thatthe individual r-f cycles of the carrier atespaced farther apart. A modulating voltage ofthe opposite polarity causes the frequency to312
Frequency Modulation313wClCARRIER;:: SIDE FREQUENCY SIDE FREQUENCYJ0..::;
314 FM Transmission<strong>THE</strong> R AD I 0w:::> "1-:;0..~clulation frequency, but their arrpl ituclevaries greatly as the amount of modulation Ischanged. The ca"ier strength also varies greatlywith frequency modulation. The sicle frequenciesshown represent a case where the clevlation eachsicle of the "carrier" frequency is equal to fivetimes the moclul ati ng frequency. Other amounts ofdeviation with the same moclulat/on frequencywoulcl cause the relative strengths of the varioussiclebancls to change widely.strength of the component at the carrier frequencyvaries widely in FM and it may evendisappear entirely under certain conditions.The variation of strength of the carrier componentis useful in measuring the amount offrequency modulation, and will be discussedin detail later in this chapter.One of the great advantages of FM over AMis the reduction in noise at the receiver whichthe system allows. If the receiver is made responsiveonly to changes in frequency, a considerableincrease in signal-to-noise ratio ismade possible through the use of FM, whenthe signal is of greater strength than the noise.The noise reducing capabilities of FM arisefrom the inability of noise to cause appreciablefrequency modulation of the noise-plus-signalvoltage which is applied to the detector in thereceiver.FM Terms Unlike amplitude modulation, theterm percentage 11KJdulation meanslittle in FM practice, unless the receiver characteristicsare specified. There are, however,three terms, deviation, modulation index, anddeviation ratio, which convey considerableinformation concerning the character of theFMwave.Deviation is the amount of frequency shifteach side of the unmodulated carrier frequencywhich occurs when the transmitter is modulated.Deviation is ordinarily measured in kilocycles,and in a properly operating FM trans-mitter it will be directly proportional to theamplitude of the modulating signal. When asymmetrical modulating signal is applied tothe transmitter, equal deviation each side ofthe resting frequency is obtained during eachcycle of the modulating signal, and the totalfrequency range covered by the FM transmitteris sometimes known as the swing. If, for instance,a transmitter operating on 1000 kc.has its frequency shifted from 1000 kc. to 1010kc., back to 1000 kc., then to 990 kc., andagain back to 1000 kc. during one cycle of themodulating wave, the deviation would be 10kc. and the swing 20 kc.The modulation index of an FM signal isthe ratio of the deviation to the audio modulatingfrequency, when both are expressed inthe same units. Thus, in the example aboveif the signal is varied from 1000 kc. to 1010kc. to 990 kc., and back to 1000 kc. at a rate(frequency) of 2000 times, a second, the modulationindex would be 5, since the deviation(10 kc.) is 5 times the modulating frequency(2000 cycles, or 2 kc.).The relative strengths of the FM carrier andthe various side frequencies depend directlyupon the modulation index, these relativestrengths varying widely as the modulationindex is varied. In the preceding example, forinstance, side frequencies occur on the highside of 1000 kc. at 1002, 1004, 1006, 1008,1010, 1012, etc., and on the low frequencyside at 998, 996, 994, 992, 990, 988, etc. Inproportion to the unmodulated carrier strength(100 per cent), these side frequencies havethe following strengths, as indicated by amodulation index of 5: 1002 and 998-33 percent, 1004 and 996-5 per cent, 1006 and 994-36 per cent, 1008 and 992-39 per cent, 1010and 990-26 per cent, 1012 and 988-13 percent. The carrier strength (1000 kc.) will be18 per cent of its unmodulated value. Changingthe amplitude of the modulating signal willchange the deviation, and thus the modulationindex will be changed, with the result that theside frequencies, while still located in thesame places, will have different strength valuesfrom those given above.The deviation ratio is similar to the modulationindex in that it involves the ratio betweena modulating frequency and deviation.In this case, however, the deviation in questionis the peak frequency shift obtained underfull modulation, and the audio frequency to beconsidered is the maximum audio frequency tobe transmitted. When the maximum audio frequencyto be transmitted is 5000 cycles, forexample, a deviation ratio of 3 would call fora peak deviation of 3 x 5000, or 15 kc. at full.nodulation. The noise-suppression capabilitiesof FM are directly related to the deviationratio. As the deviation ratio is increased,
<strong>HANDBOOK</strong>Narrow Band FM 315the noise suppression becomes better if thesignal is somewhat stronger than the noise.Where the noise approaches the signal instrength, however, low deviation ratios allowcommunication to be maintained in many caseswhere high-deviation-ratio FM and conventionalAM are incapable of giving service.This assumes that a narrow-band FM receiveris in use. For each value of r-f signal-to-noisequio at the receiver, there is a maximum deviationratio which may be used, beyond whichthe output audio signal-to-noise ratio decreases.Up to this critical deviation ratio,however, the noise suppression becomes progressivelybetter as the deviation ratio is increased.For high-fidelity FM broadcasting putposes,a deviation ratio of 5 is ordinarily used, themaximum audio frequency being 15,000 cycles,and the peak deviation at full modulation being75 kc. Since a swing of 150 kc. is coveredby the transmitter, it is obvioll's that widebandFM transmission must necessarily beconfined to the v-h-f range or higher, whereroom for the signals is available.In the case of television sound, the deviationratio is 1.67; the maximum modulationfrequency is 15,000 cycles, and the transmitterdeviation for full modulation is 25 kc.The sound carrier frequency in a standard TVsignal is located exactly 4.5 Me. higher thanthe picture carrier frequency. In the intercarrierTV sound system, which recently hasbecome quite widely used, this constant differencebetween the picture carrier and the soundcarrier is employed within the receiver to obtainan FM sub-carrier at 4.5 Me. This 4.5Me. sub-carrier then is demodulated by the FMdetector to obtain the sound signal whichaccompanies the picture.Narrow-Band Narrow-band FM trans-FM Transmission mission has become standardizedfor use by the mobileservices such as police, fire, and taxicabcommunication, and also on the basis ofa temporary authorization for amateur work inportions of each of the amateur radiotelephonebands. A maximum deviation of 15 kc. hasbeen standardized for the mobile and commercialcommunication services, while a maximumdeviation of 3 kc. is authorized for amateurNBFM communication.Bandwidth Re·qulred by FMAs the above discussion hasindicated, many side frequenciesare set up when a radiofrequencycarrier is frequency modulated; theoretically,in fact, an infinite number of sidefrequencies is formed. Fortunately, however,the amplitudes of those side frequencies fallingoutside the frequency range over whichthe transmitter is swung are so small thatmost of them may be ignored. In FM transmission,when a complex modulating wave(speech or music) is used, still additionalside frequencies resulting from a beating togetherof the various frequency componentsin the modulating wave are formed. This is asituation that does not occur in amplitudemodulation and it might be thought that thelarge number of side frequencies thus formedmight make the frequency spectrum producedby an FM transmitter prohibitively wide. Analysisshows, however, that the additional sidefrequencies are of very small amplitude, and,instead of increasing the bandwidth, modulationby a complex wave actually reduces theeffective bandwidth of the FM wave. This isespecially true when speech modulation isused, since most of the power in voiced soundsis concentrated at low frequencies in the vicinityof 400 cycles.The bandwidth required in an FM receiveris a function of a number of factors, both theoreticaland practical. Basically, the bandwidthrequired is a function of the deviation ratioand the maximum frequency of modulation,although the practical consideration of driftand ease of receiver tuning also must be considered.Shown in figure 5 are the frequencyspectra (carrier and sideband frequencies)associated with the standard FM broadcastsignal, the TV sound signal, and an amateurbandnarrow-band FM signal with full modulationusing the highest permissible modulatingfrequency in each case. It will be seen thatfor low deviation ratios the receiver bandwidrhshould be at least four times the maximumfrequency deviation, but for a deviationratio of 5 the receiver bandwidth need be onlyabout 2. 5 times the maximum frequency deviation.16-2 Direct FM CircuitsFrequency modulation may be obtained eitherby the direct method, in which the frequencyof an oscillator is changed directly by themodulating signal, or by the indirect methodwhich makes use of phase modulation. Phasemodulationcircuits will be discussed in section16-3.A successful frequency modulated transmittermust meet two requirements: (1) Thefrequency deviation must be symmetrical abouta fixed frequency, for symmetrical modulationvoltage. (2) The deviation must be directlyproportional to the amplitude of the modulation,and independent of the modulation frequency.There are several methods of directfrequency modulation which will fulfill these
316 FM Transmission<strong>THE</strong>R AD I 00 FM BROADCASTDEVIATION- 75 KC·MOO. FREQ,-15 KC.MOO. INDEX-~OSCILLATOR INf. 7 5 MC. RANGE® TV SOUND-45KCDEVIATION- 25 KC.MOO. FREQ.- 1~ KC.MOD. I NOEX -1.67+1S0-200V.REGULATEDFigure 6REACTANCE-TUBE MODULATORThis circuit is convenient for direct frequen·cy modulation of an oscillator in the 1. 75-Mc.range. Capacitor C 3 may be only the inputcapacitance of the tube, or a small trimmercapacitor may be incluclecl to permit a varia·tion in the sensitivity of the reactance tube.© AMATEUR NBFMCENTERFREQUENCYDEVIATION- 3 KC.MOD. FREQ.- 3 KC.MOO. INDEX -IFigure 5EFFECT OF FM MODULATION INDEXShowing the side-frequency arrpl itucle ancl distributionfor the three most common moclulotion inc/icesusee/ in FM work. The maximum moclulotingfrequency and maximum deviation are shown ineach case.requirements. Some of these methods will bedescribed in the following paragraphs.Reactance-TubeMadul atorsOne of the most practicalways of obtaining direct fre·quency modulation is throughthe use of a reactance-tube modulator. In thisarrangement the modulator plate-cathode cir·cuit is connected across the oscillator tankcircuit, and made to appear as either a capaci·tive or inductive reactance by exciting themodulator grid with a voltage which eitherleads or lags the oscillator tank voltage by90 degrees. The leading or lagging grid volt·age causes a corresponding leading or laggingplate current, and the plate-cathode circuitappears as a capacitive or inductive reactanceacross the oscillator tank circuit. When thetransconductance of the modulator tube isvaried, by varying one of the element voltages,the magnitude of the reactance across the os-dilator tank is varied. By applying audio modulatingvoltage to one of the elements, thetransconductance, and hence the frequency,may be varied at an audio rate. When properlydesigned and operated, the reactance-tubemodulator gives linear frequency modulation,and is capable of producing large amounts ofdeviation.There are numerous possible configurationsof the reactance-tube modulator circuit. Thedifference in the various arrangements liesprincipally in the type of phase-shifting circuitused to give a grid voltage which is inphase quadrature with the r-f voltage at themodulator plate.Figure 6 is a diagram of one of the mostpopular forms of reactance-tube modulators.The modulator tube, which is usually a pentodesuch as a 6BA6, 6AU6, or 6CL6, has itsplate coupled through a blocking capacitor,C,, to the "hot" side of the oscillator gridcircuit. Another blocking capacitor, C 2, feedsr.f. to the phase shifting network R-C 3in themodulator grid circuit. If the resistance of Ris made large in comparison with the reactanceof C 3 at the oscillator frequency, the currentthrough the R-C 3 combination will benearly in phase with the voltage across thetank circuit, and the voltage across C, willlag the oscillator tank voltage by almost 90degrees. The result of the 90-degree laggingvoltage on the modulator grid is that its platecurrent lags the tank voltage by 90 degrees,and the reactance tube appears as an inductancein shunt with the oscillator inductance,thus raising the oscillator frequency.The phase-shifting capacitor C 3 can consistof the input capacitance of the modulator tubeand stray capacitance between grid and ground.
<strong>HANDBOOK</strong>Reactance Tube 317C3, .001AUDIOIN47KlOOK+I!:.0-250 v.REGULATEDXrl~ LQ-Figure 7ALTERNATIVE REACTANCE-TUBEMODULATORThis circuit is often preferable for use in thelower frequency range, although it may beusee! at 1. 75 Me. ancl above if clesirecl. In theschematic above the reactance tube is shownconnected across the voltage·clivicler capaci·tors of a Clapp oscillator, although the moclulatorcircuit may be usee/ with any commontype of oscillator.However, better control of the operating conditionsof the modulator may be had throughthe use of a variable capacitor as C 3 • ResistanceR will usually have a value of between4700 and 100,000 ohms. Either resistance ortransformer coupling may be used to feed audiovoltage to the modulator grid. When a resistancecoupling is used, it is necessary to shieldthe grid circuit adequately, since the high impedancegrid circuit is prone to pick up strayr-f and low frequency a-c voltage, and causeundesired frequency modulation.An alternative reactance modulator circuitis shown in figure 7. The operating conditionsare generally the same, except that the r-fexcitation voltage to the grid of the reactancetube is obtained effectively through reversingthe R and C 3 of figure 6. In this circuit a smallcapacitance is used to couple r.f. into the gridof the reactance tube, with a relatively smallvalue of resistance from grid to ground. Thiscircuit has the advantage that the grid of thetube is at relatively low impedance with respectto r.f. However, the circuit normally isnot suitable for operation above a few megacyclesdue to the shunting capacitance withinthe tube from grid to ground.Either of the reactance-tube circuits maybe used with any of the common types of oscillators.The reactance modulator of figure6 is shown connected to the high-impedancepoint of a conventional hot-cathode Hartleyoscillator, while that of figure 7 is shown connectedacross the low-impedance capacitorsof a series·tuned Clapp oscillator.There are several possible variations of thebasic reactance-tube modulator circuits shownin figures 6 and 7. The audio input may beapplied to the suppressor grid, rather than thecontrol grid, if desired. Another modificationis to apply the audio to a grid other than thecontrol grid in a mixer or pentagrid convertertube which is used as the modulator. Generally,it will be found that the transconductancevariation per volt of control-element voltagevariation will be greatest when the control(audio) voltage is applied to the control grid.In cases where it is desirable to separate completelythe audio and r·f circuits, however,applying audio voltage to one of the other elementswill often be found advantageous despitethe somewhat lower sensitivity.Adjusting thePhase ShiftOne of the simplest methodsof adjusting the phase shift tothe correct amount is to placea pair of earphones in series with the oscilla·tor cathode-to-ground circuit and adjust thephase-shift network until minimum sound isheard in the phones when frequency modulationis taking place. If an electron-coupled orHartley oscillator is used, this method requiresthat the cathode circuit of the oscillatorbe inductively or capacitively coupled tothe grid circuit, rather than tapped on the gridcoil. The phones should be adequately bypassedfor r.f. of course.Stabilization Due to the presence of the reactance-rubefrequency modulator,the stabilization of an FM oscillator in regardto voltage changes is considerably more involvedthan in the case of a simple self-controlledoscillator for transmitter frequencycontrol. If desired, the oscillator itself may bemade perfectly stable under voltage changes,but the presence of the frequency modulatordestroys the beneficial effect of any suchstabilization. It thus becomes desirable toapply the stabilizing arrangement to the modulatoras well as the oscillator. If the oscillatoritself is stable under voltage changes, itis only necessary to apply voltage-frequencycompensation to the modulator.Reactance· TubeModulatorsTwo simple reactance-tubemodulators that may be appliedto an existing v.f.o. are illustratedin figures 8 and 9. The circuit of figure8 is extremely simple, yet effective. Only twotubes are used exclusive of the voltage regulatortubes which perhaps may be already incorporatedin the v.f.o. A 6AU6 serves as ahigh-gain voltage amplifier stage, and a 6CL6is used as the reactance modulator since itshigh value of transconductance will permit alarge value of lagging current to be drawnunder modulation swing. The unit should be
318 FM Transmission<strong>THE</strong> R AD I 06AU66CL6RFC.NOTE: ALL RESISTORS 0.5 WATT UNLESS2..5 MH O<strong>THE</strong>RWISE NOTEDALL CAPACITORS IN ..UF UNLESSO<strong>THE</strong>RWISE NOTEDADJUST FOR CORRECTVR CURRENTrFigure 8SIMPLE FM REACTANCE-TUBE MODULATORmounted in close proximity to the v.f.o. so thatthe lead from the 6CL6 to the grid circuit ofthe oscillator can be as short as possible. Apractical solution is to mount the reactancemodulator in a small box on the side of thev·f·o cabinet.By incorporating speech clipping in the re·actance modulator unit, a much more effectiveuse is made of a given amount of deviation.When the FM signal is received on an AM re·ceiver by means of slope detection, the useof speech clipping will be noticed by the great·ly increased modulation level of the FM sig·nal, and the attenuation of the center frequencynull of no modulation. In many cases, it isdifficult to tell a speech·clipped FM signalfrom the usual AM signal.A more complex FM reactance modulator in·corporating a speech clipper is shown in fig·ure 9. A 12AX7 double triode speech amplifierprovides enough gain for proper clipper actionwhen a high level crystal microphone is used.A double diode 6AL5 speech clipper is used,the clipping level being set by the potentiome·ter controlling the plate voltage applied to thediode. A 6CL6 serves as the reactance modu·lator.The reactance modulator may best be ad·justed by listening to the signal of the v·f·oexciter at the operating frequency and adjust·ing the gain and clipping controls for the bestmodulation level consistent with minimum side·band splatter. Minimum clipping occurs whenthe Adj. Clip. potentiometer is set for ma11:imumvoltage on the plates of the 6AL5 clipper tube.As with the case of all reactance modulators,a voltage regulated plate supply is required.Linearity Test It is almost a necessity to runa static test on the reactancetubefrequency modulator to determine its line·arity and effectiveness, since small changesin the values of components, and in stray ca·pacitances will almost certainly alter the modu·lator characteristics. A frequency·versus·con·trol·voltage curve should be plotted to ascer·tain that equal increments in control voltage,both in a positive and a negative direction,cause equal changes in frequency. If the curveshows that the modulator has an appreciableamount of non-linearity, changes in bias, electrodevoltages, r-f excitation, and resistance6AL56CL6ADJUST FORCORRECT VR CURRENrFigure 9FM REACTANCE MODULATOR WITH SPEECH CLIPPER
<strong>HANDBOOK</strong>Phase Modulation 319&ooon.TO MODUl-ATORCONTROL ELEMENTwhich the modulator operates, but is depend·ent only upon the phase deviation which isbeing produced and upon the modulation fre·quency. Expressed as an equation:4.!)V.ltllltil4.~v.- + +Figure 10REACTANCE· TUBE LINEARITY CHECKERvalues may be made to obtain a straight-linecharacteristic.Figure 10 shows a method of connectingtwo 4~-volt C batteries and a potentiometerto plot the characteristic of the modulator. Itwill be necessary to use a zero-center voltmeterto measure the grid voltage, or else re·verse the voltmeter leads when changing frompositive to negative grid voltage. When astraight-line characteristic for the modulatoris obtained by the static test method, the ca·pacitances of the various by-pass capacitorsin the circuit must be kept small to retain thischaracteristic when an audio voltage is usedto vary the frequency in place of the d·c volt·age with which the characteristic was plotted.16-3 Phase ModulationBy means of phase modulation (PM) it ispossible to dispense with self-controlled os·cillators and to obtain directly crystal-con·trolled FM. In the final analysis, PM is sim·ply frequency modulation in which the devia·tion is directly proportional to the modulationfrequency. If an audio signal of 1000 cyclescauses a deviation of ~ kc., for example, a2000-cycle modulating signal of the same am·plitude will give a deviation of 1 kc., and soon. To produce an FM signal, it is necessaryto make the deviation independent of the modulationfrequency, and proportional only to themodulating signal. With PM this is done byincluding a frequency correcting network inthe transmitter. The audio correction networkmust have an attenuation that varies directlywith frequency, and this requirement is easilymet by a very simple resistance-capacity net·work.The only disadvantage of PM, as comparedto direct FM such as is obtained through theuse of a reactance-tube modulator, is the factthat very little frequency deviation is pro·duced directly by the phase modulator. Thedeviation produced by a phase modulator isindependent of the actual carrier frequency onF d"' Mp modulating frequencyWhere F d is the frequency deviation one wayfrom the mean value of the carrier, and Mp isthe phase deviation accompanying modulationexpressed in radians (a radian is approximately57 .3°). Thus, to take an example, if the phasedeviation is ~ radian and the modulating fre·quency is 1000 cycles, the frequency deviationapplied to the carrier being passed throughthe phase modulator will be 500 cycles.It is easy to see that an enormous amountof multiplication of the carrier frequency isrequired in order to obtain from a phase modu·lator the frequency deviation of 75 kc. requiredfor commercial FM broadcasting. However, foramateur and commercial narrow-band FM work(NBFM) only a quite reasonable number ofmultiplier stages are required to obtain a deviationratio of approximately one. Actually,phase modulation of approximately one-halfradian on the output of a crystal oscillator inthe SO-meter band will give adequate deviationfor 29-Mc. NBFM radiotelephony. For example;if the crystal frequency is 3700 kc., the de·viation in phase produced is ~ radian, and themodulating frequency is 500 cycles, the devia·tion in the SO-meter band will be 250 cycles.But when the crystal frequency is multipliedon up to 29,600 kc. the frequency deviationwill also be multiplied by S so that the result·ing deviation on the 10-meter band will be 2 kc.either side of the carrier for a total swing incarrier frequency of 4 kc. This amount of deviationis quite adequate for NBFM work.Odd-harmonic distortion is produced whenFM is obtained by the phase-modulation meth·od, and the amount of this distortion that canbe tolerated is the limiting factor in determin·ing the amount of PM that can be used. Sincethe aforementioned frequency-correcting net·work causes the lowest modulating frequencyto have the greatest amplitude, maximum phasemodulation takes place at the lowest modu·lating frequency, and the amount of distortionthat can be tolerated at this frequency deter·mines the maximum deviation that can be ob·tained by the PM method. For high-fidelitybroadcasting, the deviation produced by PM islimited to an amount equal to about one-thirdof the lowest modulating frequency. But forNBFM work the deviation may be as high as0.6 of the modulating frequency before distor·tion becomes objectionable on voice modulation.In other terms .this means that phase de·viations as high as 0.6 radian may be used foramateur and commercial NBFM transmission.
320 FM Transmission<strong>THE</strong> R AD I 0REACTANCETUBEFigure 11REACTANCE-TUBE MODULATION OFCRYSTAL OSCILLATOR STAGEPhase-ModulationCircuits+BA simple reactance modulatornormally used for FMmay also be used for PM byconnecting it to the plate circuit of a crystaloscillator stage as shown in figure 11.Another PM circuit, suitable for operationon 20, 15 and 10 meters with the use of 80meter crystals is shown in figure 12. A doubletriode 12AX7 is used as a combination Piercecrystal oscillator and phase modulator. C,should not be thought of as a neutralizing condenser,but rather as an adjustment for thephase of the r·f voltage acting between thegrid and plate of the 12AX7 phase modulator.C 2 acts as a phase angle and magnitude control,and both these condensers should be adjustedfor maximum phase modulation capabilitiesof the circuit. Resonance of the circuit isestablished by the iron slug of coil L,-L 2 • A6CL6 is used as a doubler to 7 Me. and deliversapproximately 2 watts on this band. Ad·ditional doubler stages may be added after the6CL6 stage to reach the desired band of operation.Still another PM circuit, which is quite widelyused commercially, is shown in figure 13.In this circuit L and C are made resonant at afrequency which is 0. 707 times the operatingfrequency. Hence at the operating frequencythe inductive reactance is twice the capacitivereactance. A cathode follower tube acts as avariable resistance in series with the L andC which go to make up the tank circuit. Theoperating point of the cathode follower shouldbe chosen so that the effective resistance inseries with the tank circuit (made up of theresistance of the cathode-follower tube in parallelwith the cathode bias resistor of the cathodefollower) is equal to the capacitive react·ance of the tank capacitor at the operating frequency.The circuit is capable of about plusor minus Yz radian deviation with tolerable distortion.When a single-frequency modMeasurementof Deviationulating voltage is used with anFM transmitter, the relativeamplitudes of the various sidebands and thecarrier vary widely as the deviation is variedby increasing or decreasing the amount of modulation.Since the relationship between theamplitudes of the various sidebands and carrierto the audio modulating frequency and thedeviation is known, a simple method of measuringthe deviation of a frequency modulatedtransmitter is possible. In making the measurement,the result is given in the form of themodulation index for a certain amount of audioinput. As previously described, the modulationindex is the ratio of the peak frequency deviationto the frequency of the audio modulation.The measurement is made by applying asine-wave audio voltage of known frequencyto the transmitter, and increasing the modulationuntil the amplitude of the carrier componentof the frequency modulated wave reacheszero. The modulation index for zero carriermay then be determined from the table below.As may be seen from the table, the first pointof zero carrier is obtained when the modulationindex has a value of 2.405,-in other words,when the deviation is 2.405 times the modulationfrequency. For example, if a modulationfrequency of 1000 cycles is used, and themodulation is increased until the first carriernull is obtained, the deviation will then be2.405 times the modulation frequency, or 2.405kc. If the modulating frequency happened to be2000 cycles, the deviation at the first nullwould be 4.810 kc. Other carrier nulls will beobtained when the index is 5.52, 8.654, and atincreasing values separated approximately by17. The following is a listing of the modulationindex at successive carrier nulls up to thetenth:Zero carrierpoint no.12345678910Modulationindex2.4055.5208.65411.79214.93118.07121.21224.35327.49430.635The only equipment required for making themeasurements is a calibrated audio oscillatorof good wave form, and a communication receiverequipped with a beat oscillator andcrystal filter. The receiver should be usedwith its crystal filter set for minimum bandwidthto exclude sidebands spaced from thecarrier by the modulation frequency. The un-
<strong>HANDBOOK</strong>FM Reception 32112AX76CL6L 1-36T. #36£.} SPACED i'APART ONf"FORML2-15T. tt36E. PGWOEREO!RON CORL3-37T. #20£. CLOSE-SPACED f' DIA.NOTE: ALL RESISrDRS 0.5 WATT UNLESSO<strong>THE</strong>RWISE NOTED.ALL CAPACITORS INJJF UNLESSO<strong>THE</strong>RWISE NOTEDFigure 12REACTANCE MODULATOR FOR 10, 15 AND 20 METER OPERATIONmodulated carrier is accurately tuned in on thereceiver with the beat oscillator operating.Then modulation from the audio oscillator isapplied to the transmitter, and the modulationis increased until the first carrier null is obtained.This carrier null will correspond to amodulation index of 2.405, as previously mentioned.Successive null points will correspondto the indices listed in the table.A volume indicator in the transmitter audiosystem may be used to measure the audio levelrequired for different amounts of deviation,and the indicator thus calibrated in terms offrequency deviation. If the measurements aremade at the fundamental frequency of the oscillator,it will be necessary to multiply thefrequency deviation by the harmonic upon whichthe transmitter is operating, of course. It willprobably be most convenient to make the determinationat some frequency intermediate betweenthat of the oscillator and that at whichthe transmitter is operating, and then to multiplythe result by the frequency multiplicationbetween that frequency and the transmitteroutput frequency.as used by FM broadcast stations, TV sound,and mobile communications FM.The FM receiver must have, first of all, abandwidth sufficient to pass the range of frequenciesgenerated by the FM transmitter. Andsince the receiver must be a superheterodyneif it is to have good sensitivity at the frequenciesto which FM is restricted, i-f bandwidthis an important factor in its design.The second requirement of the FM receiveris that it incorporate some sort of device forconverting frequency changes into amplitudechanges, in other words, a detector operatingon frequency variations rather than amplitudevariations. The third requirement, and one whichis necessary if the full noise reducing capa-PHASE-MODULATEDOUTPUTAUDIO IN16-4 Reception of FMSignalsA conventional communications receiver maybe used to receive narrow-band FM transmissions,although performance will be much poorerthan can be obtained with an NBFM receiveror adapter. However, a receiver specificallydesigned for FM reception must be used whenit is desired to receive high deviation FM suchFigure 13CATHODE-FOLLOWER PHASEMODULATORThe phase moc/u/ator i/lustratec/ above Isquite satisfactory when the stage is to beoperated on a single frequency or over a norrowrange of frequendes.
322 FM Transmission<strong>THE</strong> R AD I 0Figure 14FM RECEIVER BLOCK DIAGRAMUp to the amp/ /tude I /miter stage, the FMreceiver is similar to an AM receiver, exceptfor a somewhat wider i-f bandwidth. The lim·iter removes any amplitude moclulation, one/the frequency detector following the limiterconverts frequency variations into amplituclevariations.Figure ISSLOPE DETECTION OF FM SIGNALOne side of the response characteristic of atuned circuit or of an i-f amp/ ifier may beused as shown to convert frequency variationsof an incoming signal into amplitudevariations.bilities of the FM system ot transmission aredesired, is a limiting device to eliminate am·plitude variations before they reach the de·tector. A block diagram of the essential partsof an FM receiver is shown in figure 14.The FrequencyDetectorThe simplest device for convertingfrequency variationsto amplitude variations is an"off·tune" resonant circuit, as illustrated infigure 15. With the carrier tuned in at point"A," a certain amount of r-f voltage will bedeveloped across the tuned circuit, and, asthe frequency is varied either side of this frequencyby the modulation, the r-f voltage willincrease and decrease to points "C" and "B"in accordance with the modulation. If the volt·age across the tuned circuit is applied to anordinary detector, the detector output will varyin accordance with the modulation, the ampli·tude of the variation being proportional to thedeviation of the signal, and the rate beingequal to the modulation frequency. It is obviousfrom figure 15 that only a small portion of theresonance curve is usable for linear conversionof frequency variations into amplitude variations,since the linear portion of the curve israther shon. Any frequency variation whichexceeds the linear portion will cause distortionof the recovered audio. It is also obviousby inspection of figure 15 that an AM receiverused in this manner is wide open to signalson the peak of the resonance curve and alsoto signals on the other side of the resonancecurve. Further, no noise limiting action is af·forded by this type of reception. This system,therefore, is not recommended for FM reception,although widely used by amateurs foroccasional NBFM reception.Travis Discriminator Another form of frequen·cy detector or dis crimi·nator, is shown in figure 16. In this arrange·ment two tuned circuits are used, one tunedon each side of the i-f amplifier frequency,and with their resonant frequencies spacedslightly more than the expected transmitterswing. Their outputs are combined in a differ·entia! rectifier so that the voltage across theseries load resistors, R, and R 2, is equal tothe algebraic sum of the individual outputvoltages of each rectifier. When a signal at theFROMI.F".AMP.Figure 16TRAVIS DISCRIMINATORThis type of discriminator makes use of twooff-tuner/ resonant circuits couplec/ to a sin·gle primary winding. The circuit is capableof excellent linearity, but Is difficult to a-lign.At Its "center" Ire•quency the dlscrimi•nato r produces zerooutput voltage. One i t h e r s i de of thisfrequency It givesa voltage of a polar·lty and magnitudewhich depend on thedirection and amountFREQUENCYof frequency shift.Figure 17DISCRIMINATOR VOLTAGE-FREQUENCYCURVE
<strong>HANDBOOK</strong>FM Reception 323Figure 18FOSTER-SEELEY DISCRIMINATORThis discriminator is the most wiclely usee/circuit since it is capable of excellent lin•earity one/ is relatively simple to align whenproper test equipment is available.i-f mid-frequency is received, the voltagesacross the load resistors are equal and opposite,and the sum voltage is zero. As the r-fsignal varies from the mid-frequency, however,these individual voltages become unequal, anda voltage having the polarity of the larger voltageand equal to the difference between thetwo voltages appears across the series resistors,and is applied to the audio amplifier.The relationship between frequency and discriminatoroutput voltage is shown in figure17. The separation of the discriminator peaksand the linearity of the output voltage vs. frequencycurve depend upon the discriminatorfrequency, the Q of the tuned circuits, and thevalue of the diode load resistors. As the intermediate(and discriminator) frequency is increased,the peaks must be separated further tosecure good linearity and output. Within limits,as the diode load resistance or the Q is reduced,the linearity improves, and the separationbetween the peaks must be greater.Foster-SeeleyDiscriminatorThe most widely used form ofdiscriminator is that shown infigure 18. This type of discriminatoryields an output-voltage-versus-frequencycharacteristic similar to that shown in figure19. Here, again, the output voltage is equalto the algebraic sum of the voltages developedacross the load resistors of the two diodes,the resistors being connected in series toground. However, this Foster-Seeley discriminatorrequires only two tuned circuits insteadof the three used in the previous discriminator.The operation of the circuit results from thephase relationships existing in a transformerhaving a tuned secondary. In effect, as a closeexamination of the circuit will reveal, the primarycircuit is in series, for r.f., with eachhalf of the secondary to ground. When the receivedsignal is at the resonant frequency ofthe secondary, the r-f voltage across the secondaryis 90 degrees out of phase with thatacross the primary. Since each diode is connectedacross one half of the secondary wind-Figure 19DISCRIMINATOR VECTOR DIAGRAMA signal at the resonant frequency of thesecondary will cause the secondary voltageto be 90 clegrees out of phase with the primaryvoltage, as shown at A, one/ the resultantvoltages R one/ R' are equal. If the signalfrequency changes, the phase relationshipalso changes, one/ the resultant voltagesare no longer equal, as shown at B. A clifferentialrectifier is usee/ to give an output voltageproportional to the difference betweenR one/ R'.ing and the primary winding in series, the resultantr-f voltages applied to each are equal,and the voltages developed across each diodeload resistor are equal and of opposite polarity.Hence, the net voltage between the top ofthe load resistors and ground is zero. This isshown vectorially in figure 19A where the resultantvoltages R and R' which are appliedto the two diodes are shown to be equal whenthe phase angle between primary and secondaryvoltages is 90 degrees. If, however, thesignal varies from rhe resonant frequency, the90-degree phase relationship no longer existsbetween primary and secondary. The result ofthis effect is shown in figure 198 where thesecondary r-f voltage is no longer 90 degreesout of phase with respect to the primary voltage.The resultant voltages applied to the twodiodes are now no longer equal, and a d-evoltage proportional to the difference betweenthe r-f voltages applied to the two diodes willexist across the series load resistors. As thesignal frequency varies back and forth acrossthe resonant frequency of the discriminator,an a-c voltage of the same frequency as theoriginal modulation, and proportional to thedeviation, is developed and passed on to theaudio amplifier.RatioDetectorOne of the more recent types of FMdetector circuits, called the ratiodetector is diagrammed in figure 20.The input transformer can be designed so thatthe parallel input voltage to the diodes can betaken from a tap on the primary of the trans-
324 FM Transmission<strong>THE</strong>R AD I 0RFCA.V.C OUTPUT33 fOJJF.K +100K+250TODISCRIMINATORA.F. OUTPUTFigure 20RATIO DETECTOR CIRCUITThe parallel voltage to the c/ioc/es in a ratiodetector may be obtained from a tap on theprimary winding of the transformer or froma third winding. Note that one of the clioclesis reversed from the system usee/ with theFoster-Seeley discriminator, ancl that theoutput circuit is completely different. Theratio detector c/oes not hove to be prececleclby a limiter, but is more difficult to align fordistortion-free output than the conventionaldiscriminator.former, or this voltage may be obtained from atertiary winding coupled to the primary. Ther-f choke used must have high impedance atthe intermediate frequency used in the receiver,although this choke is not needed if the transformerhas a tertiary winding.The circuit of the ratio detector appearsvery similar to that of the more conventionaldiscriminator arrangement. However, it will benoted that the two diodes in the ratio detectorare poled so that their d-e output voltages add,as contrasted to the Foster-Seeley circuitwherein the diodes are poled so that the d-eoutput voltages buck each other. At the centerfrequency to which the discriminator transformeris tuned the voltage appearing at thetop of the !-megohm potentiometer will be onehalfthe d-e voltage appearing at the a-v-e outputterminal-since the contribution of eachdiode will be the same. However, as the inputfrequency varies to one side or the other ofthe tuned value (while remaining within thepass band of the i-f amplifier feeding the detector)the relative contributions of the twodiodes will be different. The voltage appearingat the top of the 1-megohm volume control willincrease for frequency deviations in one directionand will decrease for frequency deviationsin the other direction from the mean or tunedvalue of the transformer. The audio output voltageis equal to the ratio of the relative contributionsof the two diodes, hence the nameratio detector.The ratio detector offers several advantagesover the simple discriminator circuit. The circuitdoes not require the use of a limiter precedingthe detector since the circuit is inherentlyinsensitive to amplitude modulation onFigure 21LIMITER CIRCUITOne, or sometimes two, limiter stages nor ..molly precede the discriminator sa that a constantsignal level will be feel to the FM de·tector. This procedure eliminates amplitudevariations in the signal feel to the discrimi·nator, so that it will respond only to frequencychanges.an incoming signal. This factor alone meansthat the r-f and i-f gain ahead of the detectorcan be much less than the conventional discriminatorfor the same overall sensitivity.Further, the circuit provides a-v-e voltage forcontrolling the gain of the preceding r-f andi-f stages. The ratio detector is, however, susceptibleto variations in the amplitude of theincoming signal as is any other detector circuitexcept the discriminator with a limiterpreceding it, so that a-v-e should be used onthe stages preceding the detector.Limiters The limiter of an FM receiver usinga conventional discriminator servesto remove amplitude modulation and pass onto the discriminator a frequency modulatedsignal of constant amplitude; a typical circuitis shown in figure 21. The limiter tube is operatedas an i-f stage with very low plate voltageand with grid leak bias, so that it overloadsquite easily. Up to a certain point theoutput of the limiter will increase with an increasein signal. Above this point, however,the limiter becomes overloaded, and furtherlarge increases in signal will not give any increasein output. To operate successfully, thelimiter must be supplied with a large amountof signal, so that the amplitude of its outputwill not change for rather wide variations inamplitude of the signal. Noise, which causeslittle frequency modulation but much amplitudemodulation of the received signal, is virtuallywiped out in the limiter.The voltage across the grid resistor varieswith the amplitude of the received signal. Forthis reason, conventional amplitude modulatedsignals may be received on the FM receiverby connecting the input of the audio amplifierto the top of this resistor, rather than to thediscriminator output. When properly filtered
<strong>HANDBOOK</strong>NBFM Adapter 325by a simple R-C circuit, the voltage acrossthe grid resistor may also be used as a-v-evoltage for the receiver. When the limiter isoperating properly, a.v.c. is neither necessarynor desirable, however, for FM reception alone.Receiver DesignConsiderationsOne of the m o s t importantfactors in the design of anFM receiver is the frequencyswing which it is intended to handle. It willbe apparent from figure 17 that if the straightportion of the discriminator circuit covers awider range of frequencies than those generatedby the transmitter, the audio output willbe reduced from the maximum value of whichthe receiver is capable.In this respect, the term "modulation percentage"is more applicable to the FM receiverthan it is to the transmitter, since the modulationcapability of the communication systemis limited by the receiver bandwidth and thediscriminator characteristic; full utilization ofthe linear portion of the characteristic amounts,in effect, to 100 per cent modulation. Thismeans that some sort of standard must be agreedupon, for any particular type of communication,to make it unnecessary to vary the transmitterswing to accommodate different receivers.Two considerations influence the receiverbandwidth necessary for any particular type ofcommunication. These are the maximum audiofrequency which the system will handle, andthe deviation ratio which will be employed.For voice communication, the maximum audiofrequency is more or less fixed at 3000 to 4000cycles. In the matter of deviation ratio, however,the amount of noise suppression whichthe FM system will provide is influenced bythe ratio chosen, since the improvement insignal-to-noise ratio which the FM systemshows over amplitude modulation is equivalentto a constant multiplied by the deviationratio. This assumes that the signal is somewhatstronger than the noise at the receiver,however, as the advantages of wideband FMin regard to noise suppression disappear whenthe signal-to-noise ratio approaches unity.On the other hand, a low deviation ratio ismore satisfactory for strictly communicationwork, where readability at low signal-to-noiseratios is more important than additional noisesuppression when the signal is already appreciablystronger than the noise.As mentioned previously, broadcast FM practiceis to use a deviation ratio of 5. When thisratio is applied to a voice-communication system,the total swing becomes 30 to 40 kc. Withlower deviation ratios, such as are most frequentlyused for voice work, the swing becomesproportionally less, until at a deviation ratioof 1 the swing is equal to twice the highestaudio frequency. Actually, however, the re-R.oz=FR~OM~----~%~~--J~~~DISCRIMINATORR=zzoK., C=340.U.J.JF.R= 100 K, C= 750 ..UJ.JF.R= 47K, C= 1600JJ.UF.R= 22 K, C = 3400 .U..UF.TO AUDIO GRIDFigure 2275-MICROSECOND DE-EMPHASISCIRCUITSThe audio signal transmitted by FM one/ TVstations has received high-frequency pre·emphasis, so that a cle-emphasi s c i r c u itshould be in clue/eel between the output of theFM detector one/ the input of the audiosystem.ceiver bandwidth must be slightly greater thanthe expected transmitter swing, since for distortionlessreception the receiver must passthe complete band of energy generated by thetransmitter, and this band will always cover arange somewhat wider than the transmitterswing.Pre-Emphasis Standards in FM broadcastand De-Emphasis and TV sound work call forthe pre-emphasis of all audiomodulating frequencies above about 2000cycles, with a rising slope such as would beproduced by a 75-microsecond RL network.Thus the FM receiver should include a compensatingde-emphasis RC network with a timeconstant of 75 microseconds so that the overallfrequency response from microphone toloudspeaker will approach linearity. The useof pre-emphasis and de-emphasis in this mannerresults in a considerable improvement inthe overall signal-to-noise ratio of an FM system.Appropriate values for the de-emphasisnetwork, for different values of circuit impedanceate given in figure 22.A NBFM 455-kc.Adapter UnitThe unit diagrammed in figure23 is designed to provideNBFM reception when attachedto any communication receiver having a 455-kc.i-f amplifier. Although NBFM can be receivedon an AM receiver by tuning the receiver toone side or the other of the incoming signal,a tremendous improvement in signal-to-noiseratio and in signal to amplitude ratio will beobtained by the use of a true FM detector system.The adapter uses two tubes. A 6AU6 isused as a limiter, and a 6AL5 as a discriminator.The audio level is approximately 10
326 FM TransmissionRadio Teletype6AU6":" 0.139K·2We+ 250V.AT 3 t.tA.AUDIOOUTlOOK---,1oo[Y,K -100 _---?lJUF • VOLTT I -J.W. MILLER 012-CJNOTE: ALL CAPAC/ rDRS IN .JJF UNLESS OTJIERWISE NOTEDAU RESISTORS 0.5 WATT UNLESS O<strong>THE</strong>RWISE NOTEDFigure 23NBFM ADAPTER FOR 455-KC. 1-F SYSTEMMETER~volts peak for the maximum deviation whichcan be handled by a conventional 455·kc. i-fsystem. The unit may be tuned by placing ahigh resistance d-e voltmeter across R, andtuning the trimmers of the i-f transformer formaximum voltage when an unmodulated signalis injected into the i-f strip of the receiver.The voltmeter should next be connected acrossthe audio output terminal of the discriminator.The receiver is now tuned back and forth a·cross the frequency of the incoming signal,and the movement of the voltmeter noted. Whenthe receiver is exactly tuned on the signal thevoltmeter reading should be zero. When the re·ceiver is tuned to one side of center, the volt·meter reading should increase to a maximumvalue and then decrease gradually to zero asthe signal is tuned out of the passband of thereceiver. When the receiver is tuned to theother side of the signal the voltmeter shouldincrease to the same maximum value but inthe opposite direction or polarity, and thenfall to zero as the signal is tuned out of thepassband. It may be necessary to make smalladjustments to C 1 and C 2 to make the volt·meter read zero when the signal is tuned inthe center of the passband.16-5 Radio TeletypeThe teletype machine is an electric type·writer that is stimulated by d.c. pulses origi·nated by the action of a second machine. Thepulses may be transmitted from one machineto another by wire, or by a radio signal. Whenradio transmission is used, the system istermed radio teletype (RTTY).The d.c. pulses that comprise the teletypesignal may be converted into three basic typesof emission suitable for radio transmission.These are: 1- Frequency shift keying (FSK),designated as F1 emission; 2- Make-breakkeying (MBK),designated as AI emission, and;3- Audio frequency shift keying (AFSK),designated as F2 emission.Frequency shift keying is obtained by vary·ing the transmitted frequency of the radiosignal a fixed amount (usually 850 cycles)during the keying process. The shift is ac·complished in discrete intervals designatedmark and space. Both types of intervals conveyinformation to the teletype printer. Make-breakkeying is analogous to simple c-w transmissionin that the radio carrier conveys informationby changing from an off to an on condition.Early RTTY circuits employed MBK equipment,which is rapidly becoming obsolete since itis inferior to the frequency shift system.Audio frequency shift keying employs asteady railio carrier modulated by an audiotone that is shifted in frequency according tothe R TTY pulses. Other forms of informationtransmission may be employed by a RTTYsystem which also encompass the translationof RTTY pulses into r·f signals.TeletypeCodingThe RTTY code consists ofthe 26 letters of the alphabet,the space, the line feed, thecarriage return, the bell, the upper case shift,and the lower case shift; making a total of 32coded groups. Numerals, punctuation, andsymbols may be taken care of in the case shift,since all transmitted letters are capitals.The FSK system normally employs the higherradio frequency as the mark, and the lowerfrequency as the space. This relationshipholds true in the AFSK system also. The loweraudio frequency (mark) is normally 2125 cyclesand the higher audio tone (space) is 2975cycles, giving a frequency difference of 850cycles.The TeletypeSystemA simple FSK teletype systemmay be added to any c-w trans·mitter. The teletype keyboardprints ~he keyed letters on a tape, and at thesame tlme generates the electrical code groupthat describes the letter. The d.c. pulses areimJXessed upon a distributor unit which ar·ranges the typing and spacing pulses in propersequence. The resulting series of impulsesate applied to the transmitter frequency controldevice, which may be a reactance modulator,actuated by a polar relay.The received signal is hetrodyned againsta beat oscillator to provide the two audio toneswhich are limited in amplitude and passedthrough audio filters to separate them. Recti·fication of the tones permits operation of apolar relay which can provide d.c. pulsessuitable for operation of the tele-typewriter.
CHAPTER SEVENTEENSideband TransmissionWhile single-sideband transmtmon (SSB)has attracted significant interest on amateurfrequencies only in the past few years, the principleshave been recognized and put to use invarious commercial applications for many years.Expansion of single-sideband for both commercialand amateur communication has awaitedthe development of economical componentspossessing the required characteristics (such assharp cutoff filters and high stability crystals)demanded by SSB techniques. The availabilityof such components and precision test equipmentnow makes possible the economical testing,adjustment and use of SSB equipment ona wider scale than before. Many of the seeminglyinsurmountable obstacles of past yearsno longer prevent the amateur from achievingthe advantages of SSB for his class of operation.17-1 CommercialApplications of SSBBefore discussion of amateur SSB equipment,it is helpful to review some of the commercialapplications of SSB in an effort to avoid problemsthat are already solved.The first and only large scale use of SSBhas been for multiplexing additional voice circuitson long distance telephone toll wires.Carrier systems came into wide use during the30's, accompanied by the development of highQ toroids and copper oxide ring modulatorsof controlled characteristics.327The problem solved by the carrier systemwas that of translating the 300-30()0 cyclevoice band of frequencies to a higher frequency(for example, 40.3 to 43-0 kc.) for transmissionon the toll wires, and then to reversethe translation process at the receiving terminal.It was possible in some short-haul equipmentto amplitude modulate a 40 kilocyclecarrier with the voice frequencies, in whichcase the resulting signal would occupy a bandof frequencies between 3 7 and 43 kilocycles.Since the transmission properties of wires andcable deteriorate rapidly with increasing frequency,most systems required the bandwidthconservation characteristics of single-sidebandtransmission. In addition, the carrier wave wasgenerally suppressed to reduce the powerhandling capability of the repeater amplifiersand diode modulators. A substantial body ofliterature on the components and circuit techniquesof SSB has been generated by the largeand continuing development effort to produceeconomical carrier telephone systems.The use of SSB for overseas radiotelephonyhas been practiced for several years thoughthe number of such circuits has been numericallysmall. However, the economic value ofsuch circuits has been great enough to warrantelaborate station equipment. It is fromthese stations that the impression has been obtainedthat SSB is too complicated for all buta corps of engineers and technicians to handle.Components such as lattice filters with 40or more crystals have suggested astronomicalexpense.
328 Sideband Transmission®"~SIDEBAND I SIDEBANDCARRIERFREQ,FREQUENCY SPECTRUM WITHCOMPLEX MODULATING. WAVE®CARRIER ENVELOPE WITHCOMPLEX MODULATING WAVEFigure 1REPRESENTATION OF ACONVENTIONAL AM SIGNALMore recently, SSB techniques have beenused to multiplex large numbers of voice channelson a microwave radio band using equipmentprincipally developed for telephone ca;rier fipplications. It should be noted that allproduction equipment employed in these servicesuses the filter method of generating thesingle-sideband signal, though there is a widevariation in the types of filters actually used.The SSB signal is generated at a low frequencyand at a low level, and then translatedand linearly amplified to a high levelat the operating frequency.Considerable development effort has beenexpended on high level phasing type transmitterswherein the problems of linear amplificationare exchanged for the problems ofaccurately controlled phase shifts. Such equipmenthas featured automatic tuning circuits,servo-driven to facilitate frequency changing,but no transmitter of this type has been sufficientlyattractive to warrant appreciable production.17-2 Derivation ofSingle-Sideband SignalsThe single-sideband method of communicationis, essentially, a procedure for obtainingmore efficient use of available frequency spectrumand of available transmitter capability.As a starting point for the discussion of single-sidebandsignals, let us take a conventionalAM signal, such as shown in figure 1,as representing the most common method fortransmitting complex intelligence such as. .voice or musiC.It will be noted in figure 1 that there arethree distinct portions to the signal: the carrier,and the upper and the lower sidebandgroup. These three portions always are presentin a conventional AM signal. Of all these portionsthe carrier is the least necessary andthe most expensive to transmit. It is an actual<strong>THE</strong> <strong>RADIO</strong>fact, and it can be proved mathematically (andphysically with a highly selective receiver)that the carrier of an AM signal remains unchangedin amplitude, whether it is being modulatedor not. Of course the carrier appearsto be modulated when we observe the modulatedsignal on a receiving system or indicatorwhich passes a sufficiently wide band thatthe carrier and the modulation sidebands areviewed at the same time. This apparent changein the amplitude of the carrier with modulationis simply the result of the sidebands beatingwith the carrier. However, if we receivethe signal on a highly selective receiver, and ifwe modulate the carrier with a sine wave of3000 to 5000 cycles, we will readily see thatthe carrier, or either of the sidebands can betuned in separately; the carrier amplitude, asobserved on a signal strength meter, will remainconstant, while the amplitude of the sidebandswill vary in direct proportion to themodulation percentage.Elimination ofthe Carrier andOne SidebandIt is obvious from the previousdiscussion that thecarrier is superfluous so faras the transmission of intelligenceis concerned. It is obviously a convenience,however, since it provides a signalat the receiving end for the sidebands to beatwith and thus to reproduce the original modulatingsignal. It is equally true that the transmissionof both sidebands under ordinary conditionsis superfluous since identically thesame intelligence is contained in both sidebands.Several systems for carrier and sidebandelimination will be discussed in thischapter.Power Advantageof SSB over AMSingle sideband is a veryefficient form of voicecommunication by radio.The amount of radio frequency spectrum occupiedcan be no greater than the frequencyrange of the audio or speech signal transmitted,whereas other forms of radio transmission requirefrom two to several times as much spectrumspace. The r-f power in the transmittedSSB signal is directly proportional to the powerin the original audio signal and no strongcarrier is transmitted. Except for a weak pilotcarrier present in some commercial usage,there is no r-f output when there is no audioinput.The power output rating of a SSB transmitteris given in terms of peak envelopepower (PEP). This may be defined as ther-m-s power at the crest of the modulation
<strong>HANDBOOK</strong>envelope. The peak envelope power of a conventionalamplitude modulated signal at 100%modulation is four times the carrier power.The average power input to a SSB transmitteris therefore a very small fraction of the powerinput to a conventional amplitude modulatedtransmitter of the same power rating.Single sideband is well suited for longrangecommunications because of its spectrumand power economy and because it is less susceptibleto the effects of selective fading andinterference than amplitude modulation.Theprincipal advantages of SSB arise from theelimination of the high-energy carrier andfrom further reduction in sideband power permittedby the improved performance of SSBunder unfavorable propagation conditions.In the presence of narrow band man-madeinterference, the narrower bandwidth of SSBreduces the probability of destructive interference.A statistical study of the distributionof signals on the air versus the signal strengthshows that the probability of successful communicationwill be the same if the SSB poweris equal to one-half the power of one of thetwo a-m sidebands. Thus SSB can give from 0to 9 db improvement under various conditionswhen the total sideband power is equal inSSB and a-m. In general, it may be assumedthat 3 db of the possible 9 db advantage willbe realized on the average contact. In this case,the SSB power required for equivalent performanceis equal to the power in one of thea-m sidebands. For example, this would ratea 1 00-watt SSB and a 400 watt (carrier) a-mtransmitter as having equal performance. Itshould be noted that in this comparison it isassumed that the receiver bandwidth is justsufficient to accept the transmitted intelligencein each case.To help evaluate other methods of comparisonthe following points should be considered.In conventional amplitude modulation twosidebands are transmitted, each having a peakenvelope power equal to Y-4-carrier power. Forexample, a 100-watt a-m signal will have 25-watt peak envelope power in each sideband,or a total of 50 watts. When the receiver detectsthis signal, the voltages of the two sidebandsare added in the detector. Thus the detectoroutput voltage is equivalent to that ofa 1 00-watt SSB signal. This method of comparisonsays that a 100 watt SSB transmitteris just equivalent to a 100-watt a-m transmitter.This assumption is valid only when thereceiver bandwidth used for SSB is the sameas that required for amplitude modulationDerivation 329tbtbd!"' c "' c c ::0::0 "' ...... ... -:J ::::i .J~0 4KC ~AUDIO SPECTRUM®4000KC. 4004KC 3996KC.4000KC ~SSB SPECTRUM(UPPER SIDEBAND)®SSB SPECTRUM(LOWERS IDE BAND)©Figure 2RELATIONSHIP OF AUDIO ANDSSB SPECTRUMSThe single sideband components are the sameas the original audio components except thatthe frequency of each is raised by the frequencyof the carrier. The relative amplitudeol the various components remains the same.(e.g., 6 kilocycles), when there is no noise orinterference other than broadband noise, andif the a-m signal is not degraded by propagation.By using half the bandwidth for SSBreception (e.g., 3 kilocycles) the noise is reduced3 db so the 100 watt SSB signal becomesequivalent to a 200 watt carrier a-msignal. It is also possible for the a-m signalto be degraded another 3 db on the averagedue to narrow band interference and poorpropagation conditions, giving a possible 4to 1 power advantage to the SSB signal.It should be noted that 3 db signal-to-noiseratio is lost when receiving only one sidebandof an a-m signal. The narrower receiving bandwidthreduces the noise by 3 db but the 6 dbadvantage of coherent detection is lost, leavinga net loss of 3 db. Poor propagation will deg!'adethis "one sideband" reception of an a-msignal less than double sideband reception,however. Also under severe narrow band interferenceconditions (e.g., an adjacent strongsignal) the ability to reject all interference onone side of the carrier is a great advantage.The Nature af aSSB SignalThe nature of a singlesideband signal is easilyvisualized by noting thatthe SSB signal components are exactly the sameas the original audio components except thatthe frequency of each is raised by the frequencyof the carrier. The relative amplitude of thevarious components remains the same, however.(The first statement is only true for theupper sideband since the lower sideband frequencycomponents are the difference betweenthe carrier and the original audio signal) .Figure 2A, B, and C shows how the audiospectrum is simply moved up into the radiospectrum to give the upper sideband.Thelower sideband is the same except inverted, asshown in figure 2C. Either sideband may beused. It is apparent that the carrier frequency
330 Sideband Transmission<strong>THE</strong> <strong>RADIO</strong>SIN(;LE TONETWO TONE®Figure 3A SINGLE SINE WAVE TONE INPUTTO A SSB TRANSMITTER RESULTSIN A STEADY SINGLE SINE WAVER-F OUTFIT (Al. TWO AUDIO TONESOF EQUAL AMPLITUDE BEATTOGE<strong>THE</strong>R TO PRODUCE HALF-SINEWAVES AS SHOWN IN (8).of a SSB signal can only be changed by addingor subtracting to the original carrier frequency.This is done by heterodyning, usingconverter or mixer circuits similar to thoseemployed in a superheterodyne receiver.It is noted that a single sine wave tone inputto a SSB transmitter results in a singlesteady sine wave r-f ouput, as shown in figure3A. Since it is difficult to measure the performanceof a linear amplifier with a singletone, it has becoine standltd practice to usetwo tones of equal amplitude for test purposes.The two radio frequencies thus producedbeat together to give the SSB envelopeshown in figure 3B. This figure has the shapeof half sine waves, and from one null to thenext represents one full cycle of the differencefrequency. How this envelope is generated isshown more fully in figures 4A and 4B. f1and f, represent the two tone signals. Whena vector representing the lower frequency tonesignal is used as a reference, the other vectorrotates around it as shown, and this actionFigure 5TWO-TONE SSBENVELOPE WHENONE TONE HASTWICE <strong>THE</strong>AMPLITUDE OF<strong>THE</strong> O<strong>THE</strong>R.Figure 6THREE-TONE SSBENVELOPE WHENEQUAL TONES OFEQUAL FREQUENCYSPACINGSARE USED.generates the SSB envelope When the twovectors are exactly opposite in phase, the outputis zero and this causes the null in the envelope.If one tone has twice the amplitude ofthe other, the envelope shape is shown infigure 5. Figure 6 shows the SSB envelope ofthree equal tones of equal frequency spacingsand at one particular phase relationship. Figure7 A shows the SSB envelope of four equaltones with equal frequency spacings and atone particular phase relationship. The phaserelationships chosen are such that at some instantthe vectors representing the several tonesare all in phase. Figure 7B shows a SSB envelopeof a square wave. A pure square wave requiresinfinite bandwidth, so its SSE enveloperequires infinite amplitude. This emphasizesthe point that the SSE envelope shape is notthe same as the original audio wave shape, andusually bears no similarity to it. This is becausethe percentage difference between theradio frequencies is very small, even thoughone audio tone may be several times the otherin terms of frequency. Speech clipping as used. I IFREQUENCY f"t f"2OFCARRIERf"2 /II,., I®Figure 4VECTOR REPRESENTATION OFTWO-TONE SSB ENVELOPEIA0Figure 7AFOUR TONESSB ENVELOPEwhen equal toneswith equal frequencyspacings are used®Figure 78SSB ENVELOPEOF A SQUAREWAVE.Peak of wave reachesinfinite amplitude.
<strong>HANDBOOK</strong>Derivation 331in amplitude modulation is of no practicalvalue in SSB because the SSB r-f envelopesare so different than the audio envelopes. Aheavily clipped wave approaches a square waveand a square wave gives a SSB envelope withpeaks of infinite amplitude as shown in figure7B.Carrier FrequencyStability RequirementsReception of a SSBs i g n a 1 is accomplishedby simplyheterodyning the carrier down to zero frequency.(The conversion frequency used in thelast heterodyne step is often called the reinsertedcarrier) . If the SSB signal is not heterodyneddown to exactly zero frequency, eachfrequency component of the detected audiosignal will be high or low by the amount ofthis error. An error of 10 to 20 c.p s. for speechsignals is acceptable from an intelligibilitystandpoint, but an error of the order of 50c.p.s. seriously degrades the intelligibility. Anerror of 20 c.p.s. is not acceptable for thetransmission of music, however, because theharmonic relationship of the notes would bedestroyed. For example, the harmonics of 220c.p.s. are 440, 660, 880, etc., but a 10 c.p s.error gives 230, 450, 670, 890, etc., or 210,430, 650, 870, etc., if the original error is onthe other side. This error would destroy theoriginal sound of the tones, and the harmonybetween the tones.Suppression of the carrier is common in amateurSSB work, so the combined frequency stabilitiesof all oscillators in both the transmittingand receiving equipment add together togive the frequency error found in detection.In order to overcome much of the frequencystability problem, it is common commercialpractice to transmit a pilot carrier at a reducedamplitude. This is usually 20 db belowone tone of a two-tone signal, or 26 db belowthe peak envelope power rating of the transmitter.This pilot carrier is filtered out fromthe other signals at the receiver and either amplifiedand used for the reinserted carrier orused to control the frequency of a local oscillator.By this means, the frequency drift ofthe carrier is eliminated as an error in detection.Advantage of SSB On long distance communicationcircuitswith Selective Fadingusing a-m, selectivefading often causes severe distortion and attimes makes the signal unintelligible. Whenone sideband is weaker than the other, distor-~0---:JPUSH-PULLAUDIO INPUSH-PULLAUDIO INFigure 8SHOWING TWO COMMON TYPESOF BALANCED MODULATORSc:::-;uo~~~Notice that a balanced modulator changesthe circuit condition from single ended topush-pull, or vice verso. Choice of circuit dependsupon external circuit conditions sinceboth the (A) and (B) arrangements can givesatisfactory generation of a double-sidebandsuppressed-carrier signal.ticn results; but when the carrier becomesweak and the sidebands are strong, the distortionis extremely severe and the signal maysound like "monkey chatter." This is becausea carrier of at least twice the amplitude ofeither sideband is necessary to demodulate thesignal properly. This can be overcome by usingexalted carrier reception in which the carrieris amplified separately and then reinsertedbefore the signal is demodulated or detected.This is a great help, but the reinserted carriermust be very close to the same phase as theoriginal carrier. For example, if the reinsertedcarrier were 90 degrees from the originalsource, the a-m signal would be converted tophase modulation and the usual a-m detectorwould deliver no output.The phase of the reinserted carrier is of noimportance in SSB reception and by using astrong reinserted carrier, exalted carrier receptionis in effect realized. Selective fading withone sideband simply changes the amplitudeand the frequency response of the system andvery seldom causes the signal to become unintelligible.Thus the receiving techniques usedwith SSB are those which inherently greatlyminimize distortion due to selective fading.
332 Sideband Transmission<strong>THE</strong> <strong>RADIO</strong>BRIDGEMODULATORRINGMODULATORro~OOO)CARRIERVOLTAGE( 000001CARRIERVOLTAGEFigure 9TWO TYPES OF DIODE BALANCEDMODULATORSuch balanced modulator circuits are commonlyused in carrier telephone work and insingle-sideband systems where the carrierfrequency and modulating frequency are relativelyclose together. Vacuum diodes, copperoxiderectifiers, or crystal diodes may beused in the circuits.17-3 Carrier EliminationCircuitsISlOEBANDOUTPUTVarious circuits may be employed to eliminatethe carrier to provide a double sidebandsignal. A selective filter may follow the carrierelimination circuit to produce a single sidebandsignal.Two modulated amplifiers may be connectedwith the carrier inputs 180 o out of phase, andwith the carrier outputs in parallel. The car-rier will be balanced out of the output circuit,leaving only the two sidebands. Such a circuitis called a balanced modulator.Any non-linear element will produce modulation.That is, if two signals are put in, sumand difference frequencies as well as the originalfrequencies appear in the output. Thisphenomenon is objectionable in amplifiers anddesirable in modulators or mixers.In addition to the sum and difference frequencies,other outputs (such as twice onefrequency plus the other) may appear. Allcombinations of all harmonics of each inputfrequency may appear, but in general these areof decreasing amplitude with increasing orderof harmonic. These outputs are usually rejectedby selective circuits following the modulator.All modulators are not alike in themagnitude of these higher order outputs. Balanceddiode rings operating in the square lawregion are fairly good and pentagrid convertersmuch poorer. Excessive carrier level in tubemixers will increase the relative magnitudeof the higher order outputs. Two types oftriode balanced modulators are shown in figure8, and two types of diode modulators in figure9. Balanced modulators employing vacuumtubes may be made to work very easily to apoint. Circuits may be devised wherein bothinput signals may be applied to a high impedancegrid, simplifying isolation and loadingproblems. The most important difficultieswith these vacuum tube modulator circuitsare: ( 1) Balance is not independent of signallevel. ( 2) Balance drifts with time and environment.( 3) The carrier level for low "highorderoutput" is critical, and ( 4) Such circuitshave limited dynamic range.A number of typical circuits are shown infigure 10. Of the group the most satisfactoryperformance is to be had from plate modulatedtriodes.ws.~TPUSH-PULl_AUDIO IN®B+PLATE MODULATED BALANCEDTRIODE MODULATOR®B+ PUSH-PULLAUDIO INBALANCED TRIODE MODULATORWITH Sl NGLE ENDED INPUT CIRCUITSFigure 10BALANCED MODULATORS©8+BALANCED PENTAGRID CONVERTER MODULATOR
<strong>HANDBOOK</strong> Carrier Elimination 333HIGH ZSIDEBANDOUTPUTLOW ZMODULATINGVOLTAGELOWZSIDEBANDOUTPUTCARRIER VOLTAGECARRIER VOLTAGECARRIER VOLTAGE0 ® ©DOUBLE-BALANCED RING MODULATOR SHUNT-QUAD MODULATOR SERIES-QUAD MODULATORFigure 11DIODE RING MODULATORSDiode RingModulatorsModulation in telephone carrierequipment has been verysuccessfully accomplished withcopper-oxide double balanced ring modulators.More recently, germanium diodes have beenapplied to similar circuits. The basic diodering circuits are shown in figure 11. The mostwidely applied is the double balanced ring(A). Both carrier and input are balanced withrespect to the output, which is advantageouswhen the output frequency is not sufficientlydifferent from the inputs to allow ready separationby filters. It should be noted that thecarrier must pass through the balanced inputand output transformers. Care must be takenin adapting this circuit to minimize the carrierpower that will be lost in these elements. Theshunt and series quad circuits are usable whenthe output frequencies are entirely different(i.e.: audio and r.f.). The shunt quad (B)is used with high source and load impedancesand the series quad (C) with low source andload impedances. These two circuits may beadapted to use only two diodes, substituting abalanced transformer for one side of thebridge, as shown in figure 12. It should benoted that these circuits present a half-waveload to the carrier source. In applying any ofthese circuits, r-f chokes and capacitors mustbe employed to control the path of signal andcarrier currents. In the shunt pair, for example,a blocking capacitor is used to prevent the r-fload from shorting the audio input.To a first approximation, the source andload impedances should be an arithmeticalmean of the forward and back resistances ofthe diodes employed. A workable rule ofthumb is that the source and load impedancesbe ten to twenty times the forward resistancefor semi-conductor rings. The high frequencylimit of operation in the case of junction andcopper-oxide diodes may be appreciably extendedby the use of very low source and loadimpedances.Copper-oxide diodes suitable for carrierwork are normally manufactured to order. Theyoffer no particular advantage to the amateur,though their excellent long-term stability isimportant in commercial applications. Rectifiertypes intended to be used as meter rectifiersare not likely to have the balance or highfrequency response desirable in amateur SSBtrans mir:ters.Vacuum diodes such as the 6AL5 may beused as modulators. Balancing the heatercathodecapacity is a major difficulty exceptwhen the 6AL5 is used at low source and loadimpedance levels. In addition, contact potentialsof the order of a few tenths of a volt mayalso disturb low level applications (figure 13).The double diode circuits appear attractive,but in general it is more difficult to balance atransformer at carrier frequency than an additionalpair of diodes. Balancing potentiometersmay be employed, but the actual cause of theunbalance is far more subtile, and cannot beadequately corrected with a single adjustment.A signal produced by any of the above circuitsmay be classified as a double sideband,suppressed-carrier signal.CARRIER VOLTAGE®SERIES-PAIRMODULATORSIDEBANDOUTPUT0SHUNT-PAl RMODULATOR~~' "' D)>---,,-o.-.-A-No~ 0VOLTAGEOUTPUTCARRIER VOL.TAGEFigure 12DOUBLE-DIODE PAIRED MODULATORS
334 Sideband Transmission<strong>THE</strong> <strong>RADIO</strong>A'JDIOIN~PUSH-PULL R.F.CARRIER INB+I470@ SERIES-BALANCED DIODE MODULATORUSING 6AL5 TUBE0-1 0-2 0-30 ~~~--+--+--+-~---~uv\w-• 00~ \-70f----l--f----1r----l---l--~-80>1--l----1r----l'-----l---l--l- 1 -----\--lR.F.CAR·RIER INAUDIOIN6AL5@ RING-DIODE MODULATOR USING 6AL5 TUBEFigure 13VACUUM DIODE MODULATOR CIRCUITS17-4 Generation ofSingle-Sideband SignalsIn general, there are two commonly usedmethods by which a single-sideband signal maybe generated. These systems ar€: ( 1) The FilterMethod, and ( 2) The Phasing Method.The systems may be used singly or in combination,and either method, in theory, maybe used at the operating frequency of thetransmitter or at some other frequency withthe signal at the operating frequency being obtainedthrough the use of frequency changers(mixers).The FilterMethodtelephoneThe filter method for obtaininga SSB signal is the classic methodwhich has been in use by thecompanies for many years both for-o -!1 -4 -3 -2 -1 o +IKILOCYCLES DEVIATIONFigure 15BANDPASS CHARACTERISTIC OFBURNELL S-15000 SINGLESIDEBAND FILTERland-line and radio communications. The modeof operation of the filter method is diagrammedin figure 14, in terms of components andfilters which normally would be available tothe amateur or experimenter. The output ofthe speech amplifier passes through a conventionalspeech filter to limit the frequencyrange of the speech to about 200 to 3000cycles. This signal then is fed to a balancedmodulator along with a 50,000-cycle first carrierfrom a self-excited oscillator. A low-frequencybalanced modulator of this type mostconveniently may be made up of four diodesof the vacuum or crystal type cross connectedin a balanced bridge or ring modulator circuit.Such a modulator passes only the sidebandcomponents resulting from the sum and differencebetween the two signals being fed tothe balanced modulator. The audio signal andthe 50-kc. carrier signal from the oscillatorboth cancel out in the balanced modulator sothat a band of frequencies between 47 and 50kc. and another band of frequencies between50 and 53 kc. appear in the output.The signals from the first balanced modulatorare then fed through the most criticalFigure 14BLOCK DIAGRAM OF FILTER EXCITER EMPLOYING A 50-K.C.SIDEBAND FILTER
<strong>HANDBOOK</strong> Generation of S.S. B. 3356AU6 t 12AU7 6AL5 t 12AU7 6C4SHUNT-DIODE MODULATOR R.F. AMPLIFIERPHASE~JNVERTER.01 0.1 2~0.UUF+3!10V.$0htA.PUSH-PULL R.F. TO BALANCEDMODULATOR FOR CONVERSIONTO 160 METERS12AU750KC. OSCILLATORNOTE: UNLESS O<strong>THE</strong>RWISE SPECIFIED;RESISTORS ARE 0.5 WATT.CAPACITORS IN J.JF.Figure 16OPERATIONAL CIRCUIT FOR SSB EXCITER USING <strong>THE</strong> BURNELL50-KC. SIDEBAND FILTERcomponent in the whole system-the first sidebandfilter. It is the function of this first sidebandfilter to separate the desired 47 to 50 kc.sideband from the unneeded and undesired 50to 53 kc. sideband. Hence this filter must havelow attenuation in the region between 47 and50 kc., a very rapid slope in the vicinity of50 kc., and a very high attenuation to thesideband components falling between 50 and53 kilocycles.Burnell & Co., Inc., of Yonkers, New Yorkproduce such a filter, designated as BurnellS-15,000. The passband of this filter is shownin figure 15.Appearing, then, at the output of the filteris a single sideband of 47 kc. to 50 kc. Thissideband may be passed through a phase inverterto obtain a balanced output, and thenfed to a balanced mixer. A local oscillatoroperating in the range of 1750 kc. to 1950 kc.is used as the conversion oscillator. Additionalconversion stages may now be added to translatethe SSB signal to the desired frequency.Since only linear amplification may be used,it is not possible to use frequency multiplyingstages. Any frequency changing must be doneby the beating-oscillator technique. An operationalcircuit of this type of SSB exciter isshown in figure 16.A second type of filter-exci~er for SSB maybe built around the Collins Mechanical Filter.Such an exciter is diagrammed in figure 17.Voice frequencies in the range of 200-3000cycles are amplified and fed to a low impedancephase-inverter to furnish balanced audio.This audio, together with a suitably chosenr-f signal, is mixed in a ring modulator, madeup of small germanium diodes. Dependingupon the choice of frequency of the r-f oscillator,either the upper or lower sideband may beapplied to the input of the mechanical filter.The carrier, to some extent, has been rejectedby the ring modulator. Additional carrier rejectionis afforded by the excellent passbandFigure 17BLOCK DIAGRAM OF FILTER EXCITER EMPLOYING A 455-KC.MECHANICAL FILTER FOR SIDEBAND SELECTION
336 Sideband Transmission<strong>THE</strong> <strong>RADIO</strong>CH~OCH!aOCH~OCH$0460.~K.C. 10OUTzoI.../1\\FT-241 CHANNEL 49 CRYSTAL c 451.1 KC.FT-241 CHANNEL SO CRYSTAL = 462,9KC.4- CARRIER0 FREQUENCY.Figure 18SIMPLE CRYSTAL LATTICE FILTER\Ij0459 460 461 462 463 464FREQUENCY (K.C.)characteristics of the mechanical filter. Forsimplicity, the mixing and filtering operationusually takes place at a frequency of 455 kilocycles.The single-sideband signal appearingat the output of the mechanical filter may betranslated directly to a higher operating frequency.Suitable tuned circuits must followthe conversion stage to eliminate the signalfrom the conversion oscillator.Wove Filters The heart of a filter-type SSBexciter is the sideband filter.Conventional coils and capacitors may beused to construct a filter based upon standardwave filter techniques. The Q of the filter inductancesmust be high when compared withthe reciprocal of the fractional bandwidth. Ifa bandwidth of 3 kc. is needed at a carrier frequencyof 50 kc., the bandwidth expressed interms of the carrier frequency is 3/50 or 6%.This is expressed in terms of fractional bandwidthas 1 I 16. For satisfactory operation, the;o08 zz00f- 30z 4 0I1Jff
<strong>HANDBOOK</strong>Generation of S.S.B. 337TO POWER AMPLIFIER STAGESOR DIRECTLY TO ANTENNA SYSTEMPHASE DIFFERENCE BETWEEN lf/1 AND $2 = to•PHASE DIFFERENCE BETWEEN 91 AND 62 = 90°Figure 20BLOCK DIAGRAM OF <strong>THE</strong> "PHASING" METHODThe phasing method of obtaining a single-sideband signal is simpler than the filter system in regard tothe number of tubes ond circuits required. The system is also less expensive In regard to the componentsrequired, but is more critical in regard to adjustments for the transmission of a pure single-sideband signal.at center frequencies of 250 kc. and 455 kc.The 2 50 kc. series is specifically intended forsideband selection. The selectivity attained bythese filters is intermediate between good LCfilters at low center frequencies and engineeredquartz crystal filters. A passband of two 250kc. filters is shown in figure 19. In applicationof the mechanical filters some special precautionsare necessary. The driving and pick-upcoils should be carefully resonated to the operatingfrequency. If circuit capacities are unknown,trimmer capacitors should be usedacross the coils. Maladjustment of these tunedcoils will increase insertion loss and the peakto-valleyratio. On high impedance filters (tento twenty thousand ohms) signals greater than2 volts at the input should be avoided. D-eshould be blocked out of the end coils. Whilethe filters are rated for 5 rna. of coil current,they are not rated for d-e plate voltage.The PhasingSystemThere are a number of pointsof view from which the operationof the phasing systemof SSB generation may be described. We maystate that we generate two double-sidebandsuppressed carrier signals, each in its own balancedmodulator, that both the r-f phase andthe audio phase of the two signals differ by90 degrees, and that the outputs of the twobalanced modulators are added with the resultthat one sideband is increased in amplitudeand the other one is cancelled. This, of course,is a true description of the action that takesplace. But it is much easier to consider thephasing system as a method simply of adding(or of subtracting) the desired modulationfrequency and the nominal carrier frequency.The carrier frequency of course is not transmitted,as· is the case with all SSB transmissions,but only the sum or the difference of themodulation band from the nominal carrier istransmitted (figure 20) .The phasing system has the obvious advantagethat all the electrical circuits which giverise to the single sideband can operate in apractical transmitter at the nominal output frequencyof the transmiter. That is to say thatif we desire to produce a single sideband whosenominal carrier frequency is 3.9 Me., thebalanced modulators are fed with a 3-9-Mc.signal and with the audio signal from thephase splitters. It is not necessary to go throughseveral frequency conversions in order to obtaina sideband at the desired output frequency,as in the case with the filter methodof sideband generation.Assuming that we feed a speech signal tothe balanced modulators along with the 3900-kc. carrier ( 3.9 Me.) we will obtain in the outputof the balanced modulators a signal whichis either the sum of the carrier signal and thespeech band, or the difference between the carrierand the speech band. Thus if our speechsignal covers the band from 200 to 3000cycles, we will obtain in the output a band offrequencies from 3900.2 to 3903 kc. (the sumof the two, or the "upper" sideband), or a bandfrom 3897 to 3899.8 kc. (the difference betweenthe two or the "lower" sideband) . Afurther advantage of the phasing system ofsideband generation is the fact that it is a verysimple matter to select either the upper sidebandor the lower sideband for transmission.A simple double-pole double-throw reversingswitch in two of the four audio leads to thebalanced modulators is all that is required.
338 Sideband Transmission<strong>THE</strong> <strong>RADIO</strong>o• 1eo• goo 210•FOUR-PHASE A F.a• teoo goo 210•~FOUR-PHASE A.F.Figure 21TWO CIRCUITS FOR SINGLESIDEBAND GENERATION BY <strong>THE</strong>PHASING METHOD.The circuit ol (A) oilers the advantages olsimplicity in the single-ended input circuitsplus a push-pull output circuit. Circuit (B) requiresdouble-ended input circuits but allowsall the plates to be connected in parallel lorthe output circuit.High-LevelPhasing Vs.Low-Level PhasingThe plate-circuit efficiencyof the four tubesusually used to make upthe two balanced modulatorsof the phasing system may run as highas 50 to 70 per cent, depending upon the operatingangle of plate current flow. Hence it ispossible to operate the double balanced modulatordirectly into the antenna system as theoutput stage of the transmitter.The alternative arrangement is to generatethe SSB signal at a lower level and then toamplify this signal to the level desired bymeans of class A or class B r-f power amplifiers.If the SSB signal is generated at a levelof a few milliwatts it is most common to makethe first stage in the amplifier chain a classA amplifier, then to use one or more class Blinear amplifiers to bring the output up to thedesired level.BalancedModulator CircuitsIllustrated in figure 8 arethe two basic balancedmodulator circuits whichgive good results with a radio frequency carrierand an audio modulating signal. Note thatone push-pull and one single ended tank circuitis required, but that the push-pull circuitmay be placed either in the plate or the gridcircuit. Also, the audio modulating voltage alwaysis fed into the stage in push-pull, andthe tubes normally are operated Class A.When combining two balanced modulatorsto make up a double balanced modulator asused in the generation of an SSB signal by thephasing system, only one plate circuit is requiredfor the two balanced modulators. However,separate grid circuits are required sincethe grid circuits of the two balanced modulatorsoperate at an r-f phase difference of 90degrees. Shown in figure 21 are the two typesof double balanced modulator circuits used forgeneration of an SSB signal. Note that the circuitof figure 21A is derived from the balancedmodulator of figure SA, and similarlyfigure 21B is derived from figure SB.Another circuit that gives excellent performanceand is very easy to adjust is shown infigure 22. The adjustments for carrier balanceare made by adjusting the potentiometer forvoltage balance and then the small variable capacitorfor exact phase balance of the balancedcarrier voltage feeding the diode modulator.12AU7\R-F CARRIER >--··-2.5 VOLTSFigure 22BALANCED MODULATOR FOR USEWITH MECHANICAL FILTER0.1VOLTSSBOUTPUT
<strong>HANDBOOK</strong>Generation of S.S. B. 339+U0\1.EFigure 23LOW-Q R-F PHASE-SHIFT NETWORKThe r-f phase-shift system illustrated aboveis convenient in a case where it is desired tomake small changes in the operating frequencyof the system without the necessityof being precise in the adJustment of twocoupled circuits as used tor r-f phase shiftin the circuit of figure 21.Radio-FrequencyPhasingA single-sideband generatorof the phasing typerequires that the two balancedmodulators be fed with r-f signals havinga 90-degree phase difference. This r-fphase difference may be obtained through theuse of two loosely coupled resonant circuits,such as illustrated in figures 21A and 21B.The r-f signal is coupled directly or inductivelyto one of the tuned circuits, and the couplingbetween the two circuits is varied until, atresonance of both circuits, the r-f voltagesdeveloped across each circuit have the sameamplitude and a 90-degree phase difference.The 90-degree r-f phase difference also maybe obtained through the use of a low-Q phaseshifting network, such as illustrated in figure23; or it may be obtained through the use ofa lumped-constant quarter-wave line. The lowQ phase-shifting system has proved quite practicablefor use in single-sideband systems,particularly on the lower frequencies. In suchan arrangement the two resistances R havethe same value, usually in the range between100 and a few thousand ohms. Capacitor C, inshunt with the input capacitances of the tubesand circuit capacitances, has a reactance atthe operating frequency equal to the value ofthe resistor R. Also, inductor L has a net inductivereactance equal in value at the operatingfrequency to resistance R.The inductance chosen for use at L musttake into account the cancelling effect of theinput capacitance of the tubes and the circuitcapacitance; hence the inductance should be+1~0V•Figure 24DOME AUDIO-PHASE-SHIFT NETWORKThis circuit arrangement is convenient for obtainingthe audio phase shift when it is desiredto use a minimum of circuit components andtube elements.variable and should have a lower value of inductancethan that value of inductance whichwould have the same reactance as resistor R.Inductor L may be considered as being madeup of two values of inductance in parallel; (a)a value of inductance which will resonate atthe operating frequency with the circuit andtube capacitances, and (b) the value of inductancewhich is equal in reactance to the resistanceR. In a network such as shown in figure23, equal and opposite 45-degree phase shiftsare provided by the RL and RC circuits, thusproviding a 90-degree phase difference betweenthe excitation voltages applied to thetwo balanced modulators.Audio-FrequencyPhasingThe audio-frequency phaseshiftingnetworks used ingenerating a single-sidebandsignal by the phasing method usually arebased on those described by Dome in an articlein the December, 1946, Electronics. Arelatively simple network for accomplishingthe 90-degree phase shift over the range from160 to 3500 cycles is illustrated in figure 24.The values of resistance and capacitance mustbe carefully checked to insure minimum deviationfrom a 90-degree phase shift over the 200to 3000 cycle range.Another version of the Dome network isshown in figure 25. This network employsthree 12AU7 tubes and provides balanced outputfor the two balanced modulators. As withthe previous network, values of the resistanceswithin the network must be held to very closetolerances. It is necessary to restrict the speechrange to 300 to 3000 cycles with this network.Audio frequencies outside this range will nothave the necessary phase-shift at the output
340Sideband Transmission<strong>THE</strong> <strong>RADIO</strong>. 01o-j12AU7TO BAL.MOO.tt1TO BAL.MOD.•2o.oo--j.... ...1.6K lOOK lOOKrPUSH-PULL 1215AUDIO 100 JJJJFINPUTTO BAL.MOD.l* 1TO BAL.1.6K 133.3 K rMOO. tt zo-----j...o.oFigure 26PASSIVE AUDIO-PHASE-SHIFTNETWORK, USEFUL OVER RANGEOF 300 TO 3000 CYCLES.+10$V. REI;ULATEOFigure 25A VERSION OF <strong>THE</strong> DOMEAUDIO-PHASE-SHIFTNETWORKof the network and will show up as spuriousemissions on the sideband signal, and alsoin the region of the rejected sideband. A lowpass3 500 cycle speech filter, such as theChicago Transformer Co. LPF-2 should beused ahead of this phase-shift network.A passive audio phase-shift network thatemploys no tubes is shown in figure 26. Thisnetwork has the same type of operating restrictionsas those described above. Additionalinformation concerning phase-shift networkswill be found in Single Sideband Techniquespublished by the Cowan Publishing Corp.,New York, and The Single Sideband Digestpublished by the American Radio RelayLeague. A comprehensive sideband review iscontained in the December, 1956 issue ofProceedings of the l.R.E.Comparison of Filterand Phasing Methodsof SSB GenerationEither the filter or thephasing method ofsingle-sideband generationis theoreticallycapable of a high degree of performance.In general, it may be said that a high degreeof unwanted signal rejection may be attainedwith less expense and circuit complexity withthe filter method. The selective circuits forrejection of unwanted frequencies operate at arelativly low frequency, are designed for thisone frequency and have a relatively high orderof Q. Carrier rejection of the order of 50 db orso may be obtained with a relatively simplefilter and a balanced modulator, and unwantedsideband rejection in the region of 60 db iseconomically possible.The phasing method of SSB generation exchangesthe problems of high-Q circuits andlinear amplification for the problems of accuratelycontrolled phase-shift networks. If thephasing method is employed on the actualtransmitting frequency, change of frequencymust be accompanied by a corresponding rebalanceof the phasing networks. In addition,it is difficult to obtain a phase balance withordinary equipment within 2% over a band ofaudio frequencies. This means that carriersuppression is limited to a maximum of 40 dbor so. However, when a relatively simple SSBtransmitter is needed for spot frequency operation,a phasing unit will perform in a satisfactorymanner.Where a high degree of performance in theSSB exciter is desired, the filter methodand the phasing method may be combined.Through the use of the phasing method in thefirst balanced modulator those undesired sidebandcomponents lying within 1000 cycles ofthe carrier may be given a much higher degreeof rejection than is attainable with the filtermethod alone, with any reasonable amount ofcomplexity in the sideband filter. Then thesideband filter may be used in its normal wayto attain very high attenuation of all undesiredsideband components lying perhaps furtherthan 500 cycles away from the carrier, and torestrict the sideband width on the desired sideof the carrier to the specified frequency limit.17-5 Single SidebandFrequency Conversion SystemsIn many instances the band of sidebandfrequencies generated by a low level SSB transmittermust be heterodyned up to the desiredcarrier frequency. In receivers the circuitswhich perform this function are called convertersor mixers. In sideband work they areusually termed mixers or modulators.Mixer Stages One circuit which can be usedfor this purpose employs arece1vmg-type mixer tube, such as the 6BE6.The output signal from the SSB generator isfed into the # 1 grid and the conversion fre-
<strong>HANDBOOK</strong>rf+l6BE6CONVERSION 2000KC.::;:;: +FREQUENCY ::: TUNE TO SELECT(2.5 v.) ~ 2.000+~~0=22.~0KC.':"' •IH c::_.2000-2.$0=t750 KC.Frequency Conversion 341+100t---;~ SSB OUTPUT2.50 KC. SSB +SIGNAL(0.25V.)Figure 27PENTAGRID MIXER CIRCUIT FORSSB FREQUENCY CONVERSIONquency into the # 3 grid. This is the reverse ofthe usual grid connections, but it offers about10 db improvement in distortion. The platecircuit is tuned to select the desired outputfrequency product. Actually, the output of themixer tube contains all harmonics of the twoinput signals and all possible combinations ofthe sum and difference frequencies of all theharmonics. In order to avoid distortion of theSSB signal, it is fed to the mixer at a lowlevel, such as 0.1 to 0.2 volts. The conversionfrequency is fed in at a level about 20 dbhigher, or about 2 volts. By this means, harmonicsof the incoming SSB signal generatedin the mixer tube will be very low. Usuallythe desired output frequency is either the sumor the difference of the SSB generator carrierfrequency and the conversion frequency. Forexample, using a SSB generator carrier frequencyof 250 kc. and a conversion injectionfrequency of 2000 kc. as shown in figure 27,the output may be tuned to select either 2250kc. or 1750 kc.Not only is it necessary to select the desiredmixing product in the mixer output but alsothe undesired products must be highly attenuatedto avoid having spurious output signalsfrom the transmitter. In general, all spurioussignals that appear within the assigned frequencychannel should be at least 60 db belowthe desired signal, and those appearing outsideof the assigned frequency channel at least80 db below the signal level.When mixing 250 kc. with 2000 kc. as inthe above example, the desired product is the2250 kc. signal, but the 2000 kc. injectionfrequency will appear in the output about 20db stronger than the desired signal. To reduceit to a level 80 db below the desired signalmeans that it must be attenuated 100 db.The principal advantage of using balancedmodulator mixer stages is that the injectionfrequency theoretically does not appear in theoutput. In practice, when a considerable frequencyrange must be tuned by the balancedmodulator and it is not practical to trim theFigure 28TWIN TRIODE MIXER CIRCUIT FORSSB FREQUENCY CONVERSIONpush-pull circuits and the tubes into exactamplitude and phase balance, about 20 dbof injection frequency cancellation is all thatcan be depended upon. With suitable trimmingadjustments the cancellation can be madeas high as 40 db, however, in fixed frequencycircuits.The Twin Triode Mixer The mixer circuitshown in figure 28has about 10 db lower distortion than the conventional6BE6 converter tube. It has a lowervoltage gain of about unity and a lower outputimpedance which loads the first tunedcircuit and reduces its selectivity. In some applicationsthe lower gain is of no consequencebut the lower distortion level is importantenough to warrant its use in high performanceequipment. The signal-to-distortion ratio ofthis mixer is of the order of 70 db comparedto approximately 60 db for a 6BE6 mixerwhen the level of each of two tone signals is0.5 volt. With stronger signals, the 6BE6distortion increases very rapidly, whereas the12AU7 distortion is much better comparatively.6AS6's-BIAS CARRIER + 120 V.IN AT SMA,Figure 29BALANCED MODULATOR CIRCUITFOR SSB FREQUENCY CONVERSION
342 Sideband Transmission <strong>THE</strong> <strong>RADIO</strong>III80Figure 30RESPONSE OF "N" NUMBER OF TUNED CIRCUITS,ASSUMING EACH CIRCUIT Q IS 50
<strong>HANDBOOK</strong>In practical equipment where the injectionfrequency is variable and trimming adjustmentsand tube selection cannot be used, itmay be easier and more economical to obtainthis extra 20 db of attenuation by using anextra tuned circuit in the output than by usinga balanced modulator circuit. A balanced modulatorcircuit of interest is shown in figure29, providing a minimum of 20 db of carrierattenuation with no balancing adjustment.Selective Tuned Circuits The selectivity requirementsof thetuned circuits following a mixer stage oftenbecome quite severe. For example, using aninput signal at 250 kc. and a conversion injectionfrequency of 4000 kc. the desired outputmay be 4250 kc. Passing the 4250 kc.signal and the associated sidebands withoutattenuation and realizing 100 db of attenuationat 4000 kc. (which is only 250 kc. away) isa practical example. Adding the requirementthat this selective circuit must tune from 2250kc. to 4250 kc. further complicates the basicrequirement. The best solution is to cascade anumber of tuned circuits. Since a large numberof such circuits may be required, the mostpractical solution is to use permeability tuning,with the circuits tracked together. An exampleof such circuitry is found in the CollinsKWS-1 sideband transmitter.If an amplifier tube is placed between eachtuned circuit, the overall response will be thesum of one stage multiplied by the numberof stages (assuming identical tuned circuits) .Figure 30 is a chart which may be used todetermine the number of tuned circuits requiredfor a certain degree of attenuation atsome nearby frequency. The Q of the circuitsis assumed to be 50, which is normally realizedin small permeability tuned coils. The numberof tuned circuits with a Q of 50 required forproviding 100 db of attenuation at 4000 kc.while passing 4250 kc. may be found as follows:tl.f is 4250-4000=250 kc.f, is the resonant frequency, 4250 kc.tl.f 250and - = -- = 0.059f, 4250The point on the chart where .059 intersects100 db is between the curves for 6 and 7tuned circuits, so 7 tuned circuits are required.Another point which must be considered inpractice is the tuning and tracking error ofthe circuits. For example, if the circuits wereFrequency Conversion 343actually tuned to 4220 kc. instead of 4250 kc.,tl.f 220 .the Lwould be 4220or 0.0522. Check1ngthe curves shows that 7 circuits would justbarely provide 100 db of attenuation. Thisillustrates the need for very accurate tuningand tracking in circuits having high attenuationproperties.Coupled TunedCircuitsWhen as many as 7 tunedcircuits are required for properattenuation, it is notnecessary to have the gain that 6 isolating amplifiertubes would provide. Several vacuumtubes can be eliminated by using two or threecoupled circuits between the amplifiers. Witha coefficient of coupling between circuits 0.5of critical coupling, the overall response isvery nearly the same as isolated circuits. Thegain through a pair of circuits having 0.5coupling is only eight-tenths that of two criticallycoupled circuits, however. If criticalcoupling is used between two tuned circuits,the nose of the response curve is broadenedand about 6 db is lost on the skirts of eachpair of critically coupled circuits. In somecases it may be necessary to broaden the noseof the response curve to avoid adversely affectingthe frequency response of the desiredpassband. Another tuned circuit may be requiredto make up for the loss of attenuationon the skirts of critically coupled circuits.Frequency ConversionProblemsThe example in theprevious section showsthe difficult selectivityproblem encountered when strong undesiredsignals appear near the desired frequency. Ahigh frequency SSB transmitter may be requiredto operate at any carrier frequency inthe range of 1.75 Me. to 30 Me. The problemis to find a practical and economical means ofheterodyning the generated SSB frequency toany carrier frequency in this range. There aremany modulation products in the output of themixer and a frequency scheme must be foundthat will not have undesired output of appreciableamplitude at or near the desired signal.When tuning across a frequency range someproducts may "cross over" the desired frequency.These undesired crossover frequenciesshould be at least 60 db below the desiredsignal to meet modern standards. The amplitudeof the undesired products depends uponthe particular characteristics of the mixer andthe particular order of the product. In general,most products of the 7th order and higherwill be at least 60 db down. Thus any cross-
344 Sideband Transmission<strong>THE</strong> <strong>RADIO</strong>FRO>J sseCENERATORDELAY BIAS VOLTAGEFROM POWER SUPPLYFigure 31SSB DISTORTION PRODUCTS,SHOWN UP TO NINTH ORDERover frequency lower than the 7th must beavoided since there is no way of attenuatingthem if they appear within the desired passband.The General Electric Ham News, volume11 # 6 of Nov.-Dec., 1956 covers the subjectof spurious products and incorporates a "mixselector"chart that is useful in determiningspurious products for various different mixingschemes.In general, for most applications when theintelligence bearing frequency is lower thanthe conversion frequency, it is desirable thatthe ratio of the two frequencies be between 5to 1 and 10 to 1. This is a compromise betweenavoiding low order harmonics of thissignal input appearing in the output, andminimizing the selectivity requirements of thecircuits following the mixer stage.17-6 Distortion ProductsDue to Nonlinearity ofR-F AmplifiersWhen the SSB envelope of a voice signalis distorted, a great many new frequencies aregenerated. These represent all of the possiblecombinations of the sum and difference frequenciesof all harmonics of the original frequencies.For purposes of test and analysis,two equal amplirude tones are used as theSSB audio source. Since the SSB radio frequencyamplifiers use tank circuits, all distortionproducts are filtered out except thosewhich lie close to the desired frequencies.These are all odd order products; third order,fifth order, etc .. The third order products are2p-q and 2q-p where p and q represent thetwo SSB r-f tone frequencies. The fifth orderproducts are 3p-2q and 3q-2p. These andsome higher order products are shown infigure 31. It should be noted that the frequencyspacings are always equal to the differencefrequency of the two original tones.Thus when a SSB amplifier is badly over-Figure 32BLOCK DIAGRAM OF AUTOMATICLOAD CONTROL (A.L.C.) SYSTEMloaded, these spurious frequencies can extendfar outside the original channel width andcause an unintelligible "splatter" type of interferencein adjacent channels. This is usuallyof far more importance than the distortion ofthe original tones with regard to intelligibilityor fidelity. To avoid interference in anotherchannel, these distortion products should bedown at least 40 db below adjacent channelsignal. Using a two-tone test, the distortion isgiven as the ratio of the amplitude of onetest tone to the amplitude of a third orderproduct. This is called the signal-to-distortionratio ( S/D) and is usually given in decibels.The use of feedback r-f amplifiers make SIDratios of greater than 40 db possible and practical.AutomaticLoad ControlTwo means may be used tokeep the amplitude of thesedistortion products down toacceptable levels. One is to design the amplifierfor excellent linearity over its amplirudeor power range. The other is to employ ameans of limiting the amplitude of the SSBenvelope to the capabilities of the amplifier.An automatic load control ssytem ( ALC) maybe used to accomplish this result. It should benoted that the r.f wave shapes of the SSB signalare always sine waves because the tank circuitsmake them so. It is the change in gainwith signal level in an amplifier that distortsthe SSB envelope and generates unwanted distortionproducts. An ALC system may be usedto limit the input signal to an amplifier toprevent a change in gain level caused by excessiveinput level.The ALC system is adjusted so the poweramplifier is operating near its maximum powercapability and at the same time is protectedfrom being over-driven. In amplitude modulatedsystems it is common to use speech compressorsand speech clipping systems to performthis function. These methods are not
<strong>HANDBOOK</strong>Distortion Products 3453 DB2. DB 1 A.LC1LEVELt =- =- ~ ~ ~~-=----_- ~-= -t-~ --~I I Ir-r---- --,--,----' I~II 2ooaDB SIGNAL LEVEL INPUTFigure 33PERFORMANCE CURVE OFA.L.C. CIRCUITequally useful in SSB. The reason for this isthat the SSB envelope is different from theaudio envelope and the SSB peaks do notnecessarily correspond with the audio peaksas explained earlier in this chapter. For thisreason a "compressor" of some sort locatedbetween the SSB generator and the power amplifieris most effective because it is controlledby SSB envelope peaks rather than audio peaks.Such a "SSB signal compressor" and the meansof obtaining its control voltage comprises asatisfactory ALC system.The ALC Circuit A block diagram of an ALCcircuit is shown in figure32. The compressor or gain control part ofthis circuit uses one or two stages of remotecutoff tubes such as 6BA6, operating very similarlyto the intermediate frequency stages ofa receiver having automatic volume control.The grid bias voltage which controls thegain of the tubes is obtained from a voltagedetector circuit connected to the power amplifiertube plate circuit. A large delay bias isused so that no gain reduction takes place untilthe signal is nearly up to the full power capabilityof the amplifier. At this signal level,the rectified output overcomes the delay biasand the gain of the preamplifier is reducedrapidly with increasing signal so that there isvery little rise in output power above thethreshold of gain control.When a signal peak arrives that would normallyoverload the power amplifier, it is desireablethat the gain of the ALC amplifier bereduced in a few milliseconds to a value whereoverloading of the power amplifier is overcome.After the signal peak passes, the gainshould return to the normal value in aboutone-tenth second. These attack and releasetimes are commonly used for voice communications.For this type of work, a dynamic rangeof at least 10 db is desirable. Input peaks ashigh as 20 db above the threshold of compressionshould not cause loss of control althoughsome increase in distortion in the upperrange of compression can be tolerated becausepeaks in this range are infrequent. Anotherlimitation is that the preceding SSBgenerator must be capable of passing signalsabove full power output by the amount ofcompression desired. Since the signal levelthrough the SSB generator should be maintainedwithin a limited range, it is unlikelythat more than 12 db ALC action will beuseful. If the input signal varies more thanthis, a speech compressor should be used tolimit the range of the signal fed into the SSBgenerator.Figure 33 shows the effectiveness of theALC in limiting the output signal to the capabilitiesof the power amplifier. An adjustmentof the delay bias will place the threshold ofcompression at the desired power output. Figure34 shows a simplified schematic of an ALCsystem. This ALC uses two variable gain am-6BA6Figure 34SIMPLIFIED SCHEMATIC OF AUTOMATICLOAD CONTROL AMPLIFIER. OPERATINGPOINT OF ALCCIRCUIT MAY BESET BY VARYINGBLOCKING BIAS ONCATHODE OF 6X4SIGNAL RECTIFIER+I+TOP-APLATEls10K+2.!0V.2.2KALC':"COMPRESSIONINDICATORZEROADJ. "='
346 Sideband Transmissionso~-;cE~A-FSOURCElFisure 35SSB JR. MODULATOR CIRCUITR-F and A-F sources are applied in seriesto balanced modulator.plifier stages and the maximum overall gain isabout 20 db. A meter is incorporated which iscalibrated in db of compression. This is usefulin adjusting the gain for the desiredamount of load control. A capacity voltagedivider is used to step down the r-f voltageat the plate of the amplifier tube to about 50volts for the ALC rectifier. The output of theALC rectifier passes through R-C networksto obtain the desired attack and release times<strong>THE</strong> <strong>RADIO</strong>and through r-f filter capacitors. The 3.3Kresistor and 0.1 /Lfd. capacitor across the rectifieroutput stabilizes the gain around the ALCloop to prevent "motor-boating."17-7 Sideband ExcitersSome of the most popular sideband excitersin use today are variations of the simple phasingcircuit introduced in the November, 1950issue of General Electric Ham News. Called theSSB, ]r., this simple exciter is the basis formany of· the phasing transmitters now in use.Employing only three tubes, the SSB, ]r. is aclassic example of sideband generation reducedto its simplest form.The SSB, Jr. This phasing exciter employsaudio and r-f phasing circuitsto produce a SSB signal at one spot frequency.The circuit of one of the balanced modulatorstages is shown in figure 35. The audio signaland r-f source are applied in series to two germaniumdiodes serving as balanced modulatorsPHASE SHIFT NETWORKT 12AU712AU712AT76AG7+300 v. -10.5V.C+,B-C2.A,B,C,O=EACH SECTION 2.0JJF, 450 V. ELECTROLYTIC G1,2.,3,4= 1N52. GERMANIUM DIODE OR EQUIVALENTC1= 2430 lJJJFD (.002.JJFD MICA ±5"=' WITH 170-780 UUFO TRIMMER) L 1, L2.= 33T. N° 2.1 E. WIRE CLOSE WOUND ON MILLEN N• ~904eC8=4860JJJJF0.(.0043JJFDMICA ±-5• WITH 170-780.U1JFOTRIMMER) IRON CORE ADJUSTABLE SLUG COIL FORM. LINK OF 6C9= 12.1!1 .U.UFD.(.OOI .UFO MICA :1:5% WITH 50-380JJ1JFDTRIMMER) TURNS OF HOOKUP WIRE WOUND ON OPEN END.C10=607.~.U.UFD (500.UUFD MICA ±5'Jb WITH 9-180.UJJFDTRIMMER)L3=16T. N° 19 E. WIRE SPACED TO FILL MILLEN N° 69046Ct8=350JJJJFD 600V. MIC.A ± 10'M:I (2.$0.U.UFO AND 100JJJJFO PARALLEL) COIL FORM. TAP AT 8 TURNS. LINK OF 1 TURN AT CENTER.R7,R10=133,3000HMS 1 112WATT ±:1% l4=SAMEAS l1 EXCEPTNOLINKUSED.R8,R9= 100,000 OHMS, 1/2 WATT ± 1'Yo L5= 2.8 T. OFN°19 E. WIRE. LINK ON END TO MATCH LOAD.T1 =STANCO!/ A-S3C TRANSFORMER.(4 TURN LINK MATCHES 72. OHM LOAD)T2,T3= UTC R-38A TRANSFORMER.51= DPDT TOGGLE SWITCH *=MOUNTING END OF COILS.Figure 36SCHEMATIC, SSB, JR.
<strong>HANDBOOK</strong>having a push-pull output circuit tuned to ther-f "carrier" frequency. The modulator drivesa linear amplifier directly at the output frequency.The complete circuit of the exciteris shown in figure 36.The first tube, a 12AU7, is a twin-triodeserving as a speech amplifier and a crystaloscillator. The second tube is a 12AT7, actingas a twin channel audio amplifier followingthe phase-shift audio network. The linearamplifier stage is a 6AG 7, capable of a peakpower output of 5 watts.Sideband switching is accomplished by thereversal of audio polarity in one of the audiochannels (switch s,), and provision is madefor equalization of gain in the audio channels(R,). This adjustment is necessary in orderto achieve normal sideband cancellation, whichmay be of the order of 35 db or better. Phaseshiftnetwork adjustment may be achieved byadjusting potentiometer R,. Stable modulatorbalance is achieved by the balance potentiometersR" and Rn in conjunction with thegermanium diodes.The SSE, Jr. is designed for spot frequencyoperation. Note that when changing frequencyL,, L,, L,, L,, and L should be readjusted,since these circuits constitute the tuning adjustmentsof the rig. The principal effect ofmistuning L,, L,, and L, will be lower output.The principal effect of mistuning L,, however,will be degraded sideband suppression.Power requirements of the SSE, Jr. are 300volts at 60 rna., and -10.5 volts at 1 rna.Under load the total plate current will rise toabout 80 rna. at full level with a single toneinput. With speech input, the total currentwill rise from the resting value of 60 rna. toabout 70 rna., depending upon the voice waveform.The "Ten-A"ExciterThe Model 1 OcA phasing exciterproduced by CentralElectronics, Inc. is an advancedversion of the SSE, Jr. incorporatingextra features such as VFO control, voice operation,and multi-band operation. A simplifiedschematic of the Model 1 OA is shown infigure 3 7. The 12AX7 two stage speech amplifierexcites a transformer coupled V2-12BH7low impedance driver stage and a voice operated(VOX) relay system employing a 12AX7and a 6AL5. A transformer coupled 12AT7follows the audio phasing network, providingtwo audio channels having a 90-degree phasedifference. A simple 90-degree r-f phase shiftnetwork in the plate circuit of the 9 Me. crys-S.B. Exciters 347tal oscillator stage works into the matched,balanced modulator consisting of four 1N48diodes.The resulting 9 Me. SSB signal may be convertedto the desired operating frequency in a6BA 7 mixer stage. Eight volts of r-f from anexternal v-f-o injected on grid # 1 of the6BA 7 is sufficient for good conversion efficiencyand low distortion. The plate circuit ofthe 6BA 7 is tuned to the sum or differencemixing frequency and the resulting signal isamplified in a 6AG7 linear amplifier stage.Two "tweet" traps are incorporated in the6BA 7 stage to reduce unwanted responses ofthe mixer which are apparent when the unit isoperating in the 14 Me. band. Band-changingis accomplished by changing coils L, and L,and the frequency of the external mixing signal.Maximum power output is of the orderof 5 watts at any operating frequency.A Simple 80 MeterPhasing ExciterA SSB exciter employinsr-f and audio phasingcircuits is shownin figure 38. Since the r-f phasing circuits arebalanced only at one frequency of operation,the phasing exciter is necessarily a single frequencytransmitter unless provisions are madeto re-balance the phasing circuits every time afrequency shift is made. However for mobileoperation, or spot frequency operation a relativelysimple phasing exciter may be made toperform in a satisfactory manner.A 12AU7 is employed as a Pierce crystaloscillator, operating directly on the chosenSSB frequency in the 80 meter band. The secondsection of this tube is used as an isolationstage, with a tuned plate circuit, L,. The outputof the oscillator stage is link coupled to a90o r-f phase-shift network wherein the audiosignal from the audio phasing network is combinedwith the r-f signals. Carrier balanceis accomplished by adjustment of the two 1000ohm potentiometers in the r-f phase network.The output of the r-f phasing network is coupledthrough L, to a single 6CL6 linear amplifierwhich delivers a 3 watt peak SSB signalon 80 meters.A cascade 12AT7 and a single 6C4 comprisethe speech amplifier used to drive the audiophase shift network. A small inter-stage transformeris used to provide the necessary 180 oaudio phase shift required by the network. Theoutput of the audio phasing network is coupledto a 12AU7 dual cathode follower whichprovides the necessary low impedance circuitto match the r-f phasing network. A double-
AUDIO PHASE SHIFTNETWORKPS-112AT7wol:loCIOVlQ..(1)o-0::::sQ.."' n::c~~:!!-fiD-cn;;Ow.......-iz"'~"' >
<strong>HANDBOOK</strong>S.B. Exciters34912AU76CL6fc.3W.PEAK558L3L1, L2,L3= 24T. N-22 E. ONXR-SOFORM(0.5"01A.)l1-LINKCOILS ARE 4T. EACH.D1-D4" 1N7170KNOTE: UNLESS O<strong>THE</strong>RWISE SPECIFIED;ALL RESISTORS 0.5 WATTALL CAPACITORS /N.J./F.ASTERISK AFTER CAPACITOR ORRESISTOR VALUE INDICATESPRECISION UNIT. EXACT VALUECRITICAL ONLY IN THAT IT SHOULDMATCH <strong>THE</strong> MATINf; UNIT CLOSELY.+300 v.MIC,,.,21KIMFigure 38SIMPLE 3-WATT PHASING TYPE SSB EXCITERpole double-throw switch in the output circuitof the cathode follower permits sideband selection.A Filter-Type Exciterfor 80 and 40 MetersA simple SSB filtertypeexciter employingthe Collins mechanicalfilter illustrates many of the basicprinciples of sideband generation. Such an exciteris shown in figure 39. The exciter is designedfor operation in the 80 or 40 meterphone bands and delivers sufficient output todrive a class AB, tetrode such as the 2E26, 807,or 6146. A conversion crystal may be employed,or a separate conversion v-f-o can beused as indicated on the schematic illustration.The exciter employs five tubes, exclusiveof power supply. They are: 6U8 low frequencyoscillator and r-f phase inverter, 6BA6 i-f amplifier,6BA 7 high frequency mixer, 6AG 7linear amplifier, and 12AU7 speech amplifierand cathode follower. The heart of the exciteris the balanced modulator employing two1N81 germanium diodes and the 455 kc.,3500-cycle bandwidth mechanical filter. Theinput and output circuits of the filter are resonatedto 455 kc. by means of small paddingcapacitors.A series-tuned Clapp oscillator covers therange of 452 kc. - 457 kc. permitting thecarrier frequency to be adjusted to the "20decibel" points on the response curve of thefilter, as shown in figure 40. Proper r-f signalbalance to the diode modulator may beobtained by adjustment of the padding capacitorin the cathode circuit of the triode sectionof the 6U8 r-f tube. Carrier balance is set bymeans of a 50K potentiometer placed acrossthe balanced modulator.One half of a 12AU7 serves as a speech amplifierdelivering sufficient output from ahigh level crystal microphone to drive the secondhalf of the tube as a low impedance cathodefollower, which is coupled to the balancedmodulator. The two 1N81 diodes act as anelectronic switch, impressing a double sideband,suppressed-carrier signal upon the mechanicalfilter. By the proper choice of frequencyof the beating oscillator, the unwantedsideband may be made to fall outside the passbandof the mechanical filter. Thus a singlesideband suppressed-carrier signal appears atthe output of the filter. The 455 kc. SSB signalis amplified by a 6BA6 pentode stage, andis then converted to a frequency in the 80meter or 40 meter band by a 6BA 7 mixerstage. Either a crystal or an external v-f-o maybe used for the mixing signal.To reduce spurious signals, a double tuned
MAjAliAP~TPOINTa:w, 0II j .-PfSSBANOOFM£CHANICA~-FILTER~SB5581or---~ N.J-- -- --~ POl fl:..u.u.030 /1 llzQ401-~,zw.0If-uzw'" '" '"::J '" Cl';' I Iaw "' .. .. "'a:"-..~"' "'zQ a:II) .. "'a:1-w..>z " 0..0ua:~"'~I"'..a:..:;;0"..a:0..~Figure 40<strong>THE</strong> "TWENTY DB" CARRIERPOINTS ON <strong>THE</strong> FILTER CURVEThe beating oscillator should be adjusted sothat its frequency corresponds to the 20 dbattenuation points of the mechanical filterpassband. The carrier of the SSB signal isthus attenuated 20 db in addition to theInherent carrier attenuation of the balancedmixer. A total carrier attenuation of 50 dbis achieved. Unwanted sideband rejection isof the same order.6BE6RFC"'
S.B. Exciters 351transformer is placed between the mixer stageand the 6AG 7 output stage. A maximum signalof 3 watts may be obtained from the 6AG 7linear amplifier.Selection of the upper or lower sideband isaccomplished by tuning the 6U8 beating oscillatoracross the passband of the mechanicalfilter, as shown in figure 40. If the 80 meterconversion oscillator is placed on the low frequencyside of the SSB signal, placing the6U8 beating oscillator on the low frequencyside of the passband of the mechanical filterwill produce the upper sideband on 80 meters.When the beating oscillator is placed on thehigh frequency side of the passband of themechanical filter the lower sideband will begenerated on 80 meters. If the 80 meter conversionoscillator is placed on the high frequencyside of the SSB signal, the sidebandswill be reversed from the above. The variableoscillator should be set at approximately the20 db suppression point of the passband ofthe mechanical filter for best operation, asshown in figure 40. If the oscillator is closerin frequency to the filter passband than this,carrier rejection will suffer. If the oscillator ismoved farther away in frequency from thepassband, the lower voice frequencies will beattenuated, and the SSB signal will sound highpitchedand tinny. A little practice in settingthe frequency of the beating oscillator whilemonitoring the 80 meter SSB signal in thestation receiver will quickly acquaint the operatorwith the proper frequency setting of thebeating oscillator control for transmission ofeither sideband.If desired, an amplitude modulated signalwith full carrier and one sideband may betransmitted by placing the 6U8 low frequencyoscillator just inside either edge of the passbandof the filter (designated "AM point",figure 40).After the 6U8 oscillator is operating overthe proper frequency range it should be possibleto tune the beating oscillator tuning capacitoracross the passband of the mechanical filterand obtain a reading on the S-meter of areceiver tuned to the filter frequency and coupledto the input grid of the 6BA6 i-f amplifiertube. The two carrier balance controls ofthe 6U8 phase inverter section should be adjustedfor a null reading of the S-meter whenthe oscillator is placed in the center of thefilter passband. The 6BA6 stage is nowchecked for operation, and transformed T1aligned to the carrier frequency. It may benecessary to unbalance temporarily potentio-meter # 2 of the 6U8 phase inverter in orderto obtain a sufficiently strong signal for properalignment of T,.A conversion crystal is next plugged in the6BA 7 conversion oscillator circuit, and the operationof the oscillator is checked by monitoringthe crystal frequency with a nearby receiver.The SSB "carrier" produced by the unbaianceof potentiometer # 2 should be heardat the proper sideband frequency in either the80 meter or 40 meter b~nd. The coupled circuitbetween the 6BA 7 and the 6AG 7 is resonatedfor maximum carrier voltage at thegrid of the amplifier stage. Care should betaken that this circuit is tuned to the sidebandfrequency and not to the frequency of the conversionoscillator. Finally, the 6AG 7 stage istuned for maximum output. When these adjustmentshave been completed, the 455 kc.beating oscillator should be moved just outof the passband of the mechanical filter. The80 meter "carrier" will disappear. If it doesnot, there is either energy leaking around thefilter, or the amplifier stages are oscillating.Careful attention to shielding (and neutralization)should cure this difficulty.Audio excitation is now applied to the exciter,and the S-meter of the receiver shouldkick up with speech, but the audio output ofthe receiver should be unintelligible. As thefrequency of the beating oscillator is adjustedso as to bring the oscillator frequency withinthe passband of the mechanical filter the modulationshould become intelligible. A singlesideband a.m. signal is now being generated.The BFO of the receiver should now be turnedon, and the beating oscillator of the excitermoved out of the filter passband. When thereceiver is correctly tuned, clean, crisp speechshould be heard. The oscillator should be setat one of the "20 decibel" points of the filtercurve, as shown in figure 40 and all adjustmentstrimmed for maximum carrier suppression.17-8 Reception ofSingle Sideband SignalsSingle-sideband signals may be received,after a certain degree of practice in the technique,in a quite adequate and satisfactorymanner with a good communications receiver.However, the receiver must have ·quite goodfrequency stability both in the high-frequencyoscillator and in the beat oscillator. For thisreason, receivers which use a crystal-controlledfirst oscillator are likely to offer a
352 Sideband Transmission<strong>THE</strong> <strong>RADIO</strong>greater degree of satisfaction than the morecommon type which uses a self-controlled oscillator.Beat oscillator stability in most receiversis usually quite adequate, but many receiversdo not have a sufficient amplitude of beat oscillatorinjection to allow reception of strongSSB signals without distortion. In such receiversit is necessary either to increase theamount of beat-oscillator injection into thediode detector, or the manual gain control ofthe receiver must be turned down quite low.The tuning procedure for SSB signals is asfollows: The SSB signals may first be locatedby tuning over the band with receiver setfor the reception of c-w.; that is, with the manualgain at a moderate level and with the beatoscillator operating. By tuning in this mannerSSB signals may be located when they are farbelow the amplitude of conventional AM signalson the frequency band. Then after a signalhas been located, the beat oscillator shouldbe turned off and the receiver put on a.v.c.Following this the receiver should be tunedfor maximum swing of the S meter with modulationof the SSB signal. It will not be possibleto understand the SSB signal at this time, butthe receiver may be tuned for maximum deflection.Then the receiver is put back on manualgain control, the beat oscillator is turned onagain, the manual gain is turned down untilthe background noise level is quite low, andthe beat oscillator control is varied until thesignal sounds natural.The procedure in the preceding paragraphmay sound involved, but actually all the stepsexcept the last one can be done in a moment.However, the last step is the one which willrequire some practice. In the first place, it isnot known in advance whether the upper orlower sideband is being transmitted. So it willbe best to start tuning the beat oscillator fromone side of the pass band of the receiver tothe other, rather than starting with the beatoscillator near the center of the pass band asis normal for c-w reception.With the beat oscillator on the wrong sideof the sideband, the speech will sound inverted;that is to say that low-frequency modulationtones will have a high pitch and high-frequencymodulation tones will have a low pitchandthe speech will be quite unintelligible.With the beat oscillator on the correct side ofthe sideband but too far from the correct position,the speech will have some intelligibilitybut the voice will sound quite high pitched.Then as the correct setting for the beat oscilla-tor is approached the voice will begin to soundnatural but will have a background growl oneach syllable. At the correct frequency for thebeat oscillator the speech will clear completelyand the voice will have a clean, crisp quality.It should also be mentioned that there isa narrow region of tuning of the beat oscillatora small distance on the wrong side of the sidebandwhere the voice will sound quite bassyand difficult to understand.With a little experience it will be possibleto identify the sound associated with impropersettings of the beat-oscillator control so thatcorrections in the setting of the control can bemade. Note that the main tuning control of thereceiver is not changed after the sidebandonce is tuned into the pass band of the receiver.All the fine tuning should be done withthe beat oscillator control. Also, it is very importantthat the r-f gain control be turned toquite a low level during the tuning process.Then after the signal has been tuned properlythe r-f gain may be increased for good signallevel, or until the point is reached where bestoscillator injection becomes insufficient andthe signal begins to distort.Single-SidebandReceivers andAdaptersGreatly simplified tuning,coupled with strong attenuationof undesired signals,can be obtainedthrough the use of a single-sideband receiveror receiver adapter. The exalted carrier principleusually is employed in such receivers, witha phase-sensitive system sometimes includedfor locking the local oscillator to the frequencyof the carrier of the incoming signal. In orderfor the locking system to operate, some carriermust be transmitted along with the SSB signal.Such receivers and adapters include a meansfor selecting the upper or lower sideband bythe simple operation of a switch. For the receptionof a single-sideband signal the switchobviously must be placed in the correct position.But for the reception of a conventionalAM or phase-modulated signal, either sidebandmay be selected, allowing the sideband withthe least interference to be used.The Product Detector An unusually satisfactoryform of demodulatorfor SSB service is the product detector,shown in one form in figure 41. Thiscircuit is preferred since it reduces intermodulationproducts and does not require a largelocal carrier voltage, as contrasted to the morecommon diode envelope detector. This productdetector operates much in the same manner as
<strong>HANDBOOK</strong>S.S.B. Reception 3536L6MODUL.ATORSB+ 4000 v.Figure 43HIGH-LEVEL DSB BALANCED MODULATORa multi-grid mixer tube. The SSB signal isapplied to the control grid of the tube andthe locally generated carrier is impressed uponthe other control grid. The desired audio outputsignal is recovered across the plate resistanceof the demodulator tube. Since the cathodecurrent of the tube is controlled by thesimultaneous action of the two grids, the currentwill contain frequencies equal to the sumand difference between the sideband signal andthe carrier. Other frequencies are suppressed bythe low-pass r-f filter in the plate circuit of thestage, while the audio frequency is recoveredfrom the i-f sideband signal.17-9 Double SidebandTransmissionMany systems of intelligence transmissionlie in the region between amplitude modulationon the one hand and single sideband suppressed-carriertransmission on the other hand.One system of interest to the amateur is theSynchronous Communications System, popularlyknown as "double sideband" (DSB-) transmission,wherein a suppressed-carrier doublesideband signal is transmitted (figure 42) .Reception of such a signal is possible by utilizinga local oscillator phase-control systemwhich derives carrier phase information fromthe sidebands alone and does not require theuse of any pilot carrier.The DSB Transmitter A balanced modulatorof the type shown infigure 8 may be employed to create a DSBsignal. For higher operating levels, a pair ofclass-C type tetrode amplifier tubes may bescreen modulated by a push-pull audio systemand excited from a push-pull r-f source. Theplates of such a modulator are connected inparallel to the tank circuit, as shown in figure43. This DSB modulator is capable of !-kilowattpeak power output at a plate potential of4000 volts. The circuit is self-neutralizing andthe tune-up process is much the same as withany other class-C amplifier stage. As in thecase of SSB, the DSB signal may also be generatedat a low level and amplified in linearstages following the modulator.SynchronousDetectionA DSB signal may be receivedwith difficulty on aconventional receiver, andone of the two sidebands may easily be receivedon a single sideband receiver. For best reception,however, a phase-locked local oscillatorand a synchronous detector should be employed.This operation may be performed eitherat the frequency of reception or at a convenientintermediate frequency. A block dia-Figure 44BLOCK DIAGRAM OF DSBRECEIVING ADAPTER
354 Sideband Transmissiongram of a DSB synchronous receiver is shownin figure 44. The DSB signal is applied totwo detectors having their local oscillator conversionvoltages in phase quadrature to eachother so that the audio contributions of theupper and lower sidebands reinforce one another.The in-phase oscillator voltage is adjustedto have the same phase as the suppressedcarrier of the transmitted signal. The !-amplifieraudio output, therefore, will contain thedemodulated audio signal, while the Q-amplifier(supplied with quadrature oscillator voltage)will produce no output due to the quadraturenull. Any frequency change of the localoscillator will produce some audio output inthe Q-amplifier, while the !-amplifier is relativelyunaffected. The Q-amplifier audio willhave the same polarity as the !-channel audiofor one direction of oscillator drift, and oppositepolarity for oscillator drift in the oppositedirection. The Q-amplifier signal level is proportionalto the magnitude of the local oscillatorphase angle error (the oscillator drift) forsmall errors. By combining the !-signal and.the Q-signal in the audio phase discriminatora d-e control voltage is developed which automaticallycorrects for local oscillator phase errors.The reactance tube therefore locks thelocal oscillator to the correct phase. Phase controlinformation is derived entirely from thesideband component of the signal .and thecarrier (if present) is not employed. Phasecontrol ceases with no modulation of the signaland is reestablished with the reappearanceof modulation.Interference Rejection Interference fallingwithin the passbandof the receiver can be reduced by proper combinationof the I- and Q- audio signals. Underphase lock conditions, the !-signal is composedof the audio signal plus the undesiredinterference, whereas the Q-signal containsonly the interference component. Phase cancellationobtained by combining the two signalswill reduce the interference while stilladding the desired information contained inboth side-bands. The degree of interference rejectionis dependent upon the ratio of interferencefalling upon the two sidebands of thereceived signal and upon the basic design ofthe audio networks. A schematic and descriptionof a complete DSB receiving adapter isshown in the June, 1957 issue of CQ magazine.
E 1!. E TECHNI-SHEETAI RWOUND INDUCTORSCOIL DIA. TURNS PER B&W AIR DUXINDUCTANCE COIL DIA. TURNS PER B&W AIR DUXINDUCTANCEINCHES INCH .UH INCHES INCH .UH4 3001 404T 0.18 4 - 1004 2..7!!>8 - 408T 0.40 6 - 1006 6.301 8 3002. 408T o. 72. 1_!_ 8 - 1008 11. 2.2410 - 410T 1.1 2. 10 - 1010 17.516 3003 416T 2..90 16 - 1016 42..!!>32. 3004 432T 12..0 4 - 12.04 3.94 3005 504T 0.2.8 6 - 1208 8.88 - 506T 0.62. 1.L 8 - 12.08 15.62.5 8 3006 508T 1.1 10 - 12.10 2.4.!!>8 10 - !!>lOT 1.7 18 - 12.18 63.016 3007 516T 4.4 4 - 1404 !!>.2.32. 3008 !!>32.T 18.0 6 - 1406 11.84 3009 604T 0.39 1.A. 8 - 1408 2.1.046 - 606T 0.87 10 - 1410 33.03 8 3010 608T 1. !!>7 16 - 1416 85.04 10 - 6101 2..45 4 - 1604 6.6116 3011 616T 6.40 8 - 1606 1.5.032. 3012. 632.T 2.8.0 2 8 3900 1608 2.6.54 3013 804T 1.0 10 3907-1 1610 42..08 - 806T 2..3 18 - 1816 108.08 3014 808T 4.2. 4 - 2.004 10.110 - 810T 6.62.!.._6 3905-1 2.008 2.3.018 301!!> 816T 18.8 2. 8 3906-1 2.008 41.032. 3016 832.T 68.0 10 - 2010 108.0NOTE• 4 - 2.404 14.0COIL INDUCTANCE APPROXIMATELY 8 - 2.406 31.5PROPORTIONAL TO LENGTH. I.E., FOR 1/2. 3- 58.0INOUCrANCE VALUE, TRIM COIL TO 1/2 LEN~TH. 8 240810 - 2.410 89.0
CHAPTER EIGHTEENTransmitter DesignThe excellence of a transmitter is a functionof the design, and is dependent upon theexecution of the design and the proper choiceof components. This chapter deals with thestudy of transmitter circuitry and of the basiccomponents that go to make up this circuitry.Modern components are far from faultless. Resistorshave inductance and distributed capacity.Capacitors have inductance and resistance,and inductors have resistance anddistributed capacity. None of these residualattributes show up on circuit diagrams, yetthey are as much responsible for the successor failure of the transmitter as are the necessaryand vital bits of resistance, capacitanceand inductance. Because of these unwantedattributes, the job of translating a circuit onpaper into a working piece of equipment oftenbecomes an impossible task to those individualswho disregard such important trivia.Rarely do circuit diagrams show such pitfallsas ground loops and residual inductive couplingbetween stages. Parasitic resonant circuitsare rarely visible from a study of theschematic. Too many times radio equipment isrushed into service before it has been entirelychecked. The immediate and only too apparentresults of this enthusiasm are transmitter instability,difficulty of neutralization, r.f. wanderingall over the equipment, and a general"touchiness" of adjustment. Hand in glovewith these problems go the more serious onesof TVI, key-clicks, and parasitics. By payingattention to detail, with a good working knowledgeof the limitations of the components,and with a basic conception of the actions ofground currents, the average amateur will beable to build equipment that will work "justlike the book says."The twin problems of TVI and parasiticsare an outgrowth of the major problem of overallcircuit design. If close attention is paidto the cardinal points of circuitry design, thesecondary problems of TVI and parasiticswill in themselves be solved.18-1 ResistorsThe resistance of a conductor is a functionof the material, the form the material takes,the temperature of operation, and the frequencyof the current passing through the resistance.In general, the variation in resistancedue to temperature is direct! y proportional tothe temperature change. With most wire-woundresistors, the resistance increases with temperatureand returns to its original value whenthe temperature drops to normal. So-calledcomposition or carbon resistors have lessreliable temperature/ resistance characteristics.They usually have a positive temperaturecoefficient, but the retrace curve as theresistor is cooled is often erratic, and in356
Resistors 357+ 5"+4+ 3wu+ 2z;!+ I~ /U) 0wa:I:'!:w-2z
358 Transmitter Design<strong>THE</strong> R AD I 060,;-,------,-------r------.-----~1/)~ •of---+-----P.-7"-=~f+----\---H-~.-----10"-01/)0z0If- IOI--+-----b''----"tr---.ft-'""--""'--15 10FREQUENCY (MC.)Figure 4CURVES OF <strong>THE</strong> IMPEDANCE OF WIRE·WOUND RESISTORS AT <strong>RADIO</strong>FREQUENCIESthe impedance of the resistor may be thoughtof as a series reactance (X 8) as shown in fig·ure 2B. This reactance may be either indue·tive or capacitive depending upon whether theresidual inductance or the distributed capaci·tance of the resistor is the dominating factor.As a rule, skin effect tends to increase thereactance with frequency, while the capacitybetween turns of a wire·wound resistor, or ca·pacity between the granules of a compositionresistor tends to cause the reactance and re·sistance to drop with frequency. The behaviorof various types of composition resistors overa large frequency range is shown in figure 3.By proper component design, non-inductive re·sistors having a minimum of residual react·ance characteristics may be constructed. Eventhese have reactive effects that cannot be ig·nored at high frequencies.Wirewound resistors act as low·Q inductorsat radio frequencies. Figure 4 shows typicalcurves of the high frequency characteristicsof cylindrical wirewound resistors. In additionto resistance variations wirewound resistorsexhibit both capacitive and inductive react·ance, depending upon the type of resistor andthe operating frequency. In fact, such resis·tors perform in a fashion as low·Q r·f chokesbelow their parallel self·resonant frequency.18-2 CapacitorsThe inherent residual characteristics of ca·pacitors include series resistance, series in·ductance and shunt resistance, as shown infigure 5. The series resistance and inductanceRSHUNT~C L RSERIESFigure 5EQUIVALENT CIRCUIT OF A CAPACITORdepend to a large extent upon the physicalconfiguration of the capacitor and upon thematerial of which it is made. Of great interestto the amateur constructor is the series in·ductance of the capacitor. At a certain fre·quency the series inductive reactance of thecapacitor and the capacitive reactance areequal and opposite, and the capacitor is initself series resonant at this frequency. Asthe operating frequency of the circuit in whichthe capacitor is used is increased above theseries resonant frequency, the effectivenessof the capacitor as a by-passing element de·teriorates until the unit is about as effectiveas a block of wood.By-PassCapacitorsThe usual forms of by·pass ca·pacitors have dielectrics of paper,mica, or ceramic. For audio work,and low frequency r·f work up to perhaps 2 Me.or so, the paper capacitors are satisfactoryas their relatively high internal inductance haslittle effect upon the proper operation of thecircuit. The actual amount of internal induct·ance will vary widely with the manufacturingprocess, and some types of paper capacitorshave satisfactory characteristics up to a fre·quency of 5 Me. or so.When considering the design of transmittingequipment, it must be remembered that whilethe transmitter is operating at some relativelylow frequency of, say, 7 Me., there will beharmonic currents flowing through the variousby·pass capacitors of the order of 10 to 20times the operating frequency. A capacitorthat behaves properly at 7 Me. however, mayoffer considerable impedance to the flow ofthese harmonic currents. For minimum har·monic generation and radiation, it is obviouslyof greatest importance to employ by-pass ca·pacitors having the lowest possible internalinductance.Mica dielectric capacitors have much lessinternal inductance than do most paper con·densers. Figure 6 lists self-resonant frequen·cie s of various mica capacitors having variouslead lengths. It can be seen from inspectionof this table that most mica capacitors be·come self-resonant in the 12·Mc. to 50-Mc.region. The inductive reactance they wouldoffer to harmonic currents of 100 Me. or so
<strong>HANDBOOK</strong>Capacitors 35 9CONDENSER LEAD LENGTHS RESONANT FREQ..02 .Uf MICA NONE «.~ MC..002. .UF MICA NONE 23.~ MC..01 .UF MICA 10 MC.t". 0009 .UF MICA ~~f'MC ..002 .UF CERAMIC ~· 24 MC.•.001 .UF CERAMIC f' .. MC ..500 JJJJF BUTTON NONE 22.0 MC..001 JJF CERAMIC cJ.: 90 MC.4.01 JJF CERAMICf14.!:1 MC.Figure 6SELF-RESONANT FREQUENCIES OFVARIOUS CAPACITORS WITHRANDOM LEAD LENGTHwould be of considerable magnitude. In certaininstances it is possible to deliberately seriesresonatea mica capacitor to a certain frequencysomewhat below its normal self-resonantfrequency by trimming the leads to a criticallength. This is sometimes done for maximumby-passing effect in the region of 40 Me. to60 Me.The recently developed button-mica capacitorsshown in figure 7 are especially designedto have extremely low internal inductance.Certain types of button-mica capacitors ofsmall physical size have a self-resonant frequencyin the region of 600 Me.Ceramic dielectric capacitors in generalhave the lowest amount of series inductanceper unit of capacitance of these three universallyused types of by-pass capacitors. Typicalresonant frequencies of various ceramicunits are listed in figure 6. Ceramic capacitorsare available in various voltage and capacityratings and different physical configurations.Stand-off types such as shown in figure7 are useful for by-passing socket and transformerterminals. Two of these capacitors maybe mounted in close proximity on a chassisand connected together by an r-f choke to forma highly effective r-f filter. The inexpensive"clamshell" type of ceramic capacitor isrecommended for general by-passing in r-f circuitry,as it is effective as a by-pass unit towell over 100 Me.The large TV "doorknob" capacitors areuseful as by-pass units for high voltage lines.These capacitors have a value of 500 micromicrofarads,and are available in voltage ratingsup to 40,000 volts. The dielectric ofthese capacitors is usually titanium-dioxide.This material exhibits piezo-electric effects,and capacitors employing it for a dielectricwill tend to "talk-back" when a-c voltagesare applied across them. When these capaci-tors are used as plate bypass units in a modulatedtransmitter they will cause acousticalnoise. Otherwise they are excellent for generalr-f work.A recent addition to the varied line of capacitorsis the coaxial _or "Hyp~s~" type _ofcapacitor. These capacitors ex~Iblt supenorby-passing qualities at frequencies up to 200Me. and the bulkhead type are especially effectivewhen used to filter leads passingthrough partition walls between two stages.Variable Air Even though air is the perfectCapacitors dielectric, air capacitors exhibitlosses because of the inherentresistance of the metallic parts that make upthe capacitor. In addition, the leakage lossacross the insulating supports may become ofsome consequence at high frequencies. Ofgreater concern is the inductance of the capacitorat high frequencies. Since the capacitormust be of finite size, it will have tie-rodsand metallic braces and end plates, all of whichcontribute to the inductance of the unit. Theactual amount of the inductance will dependupon the physic,al size of the capacitor andthe methbd used to make contact to the statorand rotor plates. This inductance may be cutto a minimum value by using as small a capacitoras is practical, by using insulated tierodsto prevent the formation of closed inductiveloops in the frame of the unit, and bymaking connections to the centers of the plateassemblies rather than to the ends as is com-Figure 7TYPES OF CERAMIC AND MICA CAPACITORS SUITABLE FOR HIGH-FREQUENCYBYPASSINGThe Centralab 858S (7000 ppfd) Is recommendedfar screen and plate circuits of tetrodetubes.
360 Transmitter Design<strong>THE</strong> R AD I 0monly done. A large transmuting capacitormay have an inherent inductance as large as0.1 microhenry, making the capacitor susceptibleto parasitic resonances in the 50 Me. to150 Me. range of frequencies.The question of optimum C/L ratiq and caPacitor plate spacing is covered in ChapterThirteen. For all-band operation of a high powerstage, it is recommended that a capacitor justlarge enough for 40-meter phone operation bechosen. (This will have sufficient capacitancefor phone operation on all higher frequencybands.) Then use fixed padding capacitors foroperation on 80 meters. Such padding capacitorsare available in air, ceramic, and vacuumtypes.Specially designed variable capacitors arerecommended for u-h-f work; ordinary capacitorsoften have "loops" in the metal framewhich may resonate near the operating frequency.Variable VacuumCapacitarsVariable vacuum capacitorsbecause of their small physicalsize have less inherentinductance per unit of capacity than dovariable air capacitors. Their losses are extremelylow, and their dielectric strength ishigh. Because of increased production thecost of such units is now within the reach ofthe designer of amateur equipment, and theiruse is highly recommended in high power tankcircuits.18-3 Wire and InductorsAny length of wire, no matter how short,has a certain value of inductance. This propertyis of great help in making coils and inductors,but may be of great hindrance whenit is not taken into account in circuit designand construction. Connecting circuit elements(themselves having residual inductance) togetherwith a conductor possessing additionalinductance can often lead to puzzling difficulties.A piece of no. 10 copper wire ten incheslong (a not uncommon length for a plate leadin a transmitter) can have a self-inductance of0.15 microhenries. This inductance and thatof the plate tuning capacitor together with theplate-to-ground capacity of the vacuum tubecan form a resonant circuit which may lead toparasitic oscillations in the v-h-f regions. Tokeep the self-inductance at a minimum, all r-fcarrying leads should be as short as possibleand should be made out of as heavy materialas possible.At the higher frequencies, solid enamelledcopper wire is most efficient for r-f leads.Tinned or stranded wire will show greaterlosses at these frequencies. Tank coil andtank capacitor leads should be of heavier wirethan other r-f leads.The best type of flexible lead from the envelopeof a tube to a terminal is thin copperstrip, cut from thin sheet copper. Heavy, rigidleads to these terminals may crack the envelopeglass when a tube heats or cools.Wires carrying only a.f. or d.c. should bechosen with the voltage and current in mind.Some of the low-filament-voltage transmittingtubes draw heavy current, and heavy wire mustbe used to avoid voltage drop. The voltage islow, and hence not much insulation is required.Filament and heater leads are usually twistedtogether. An initial check should be made onthe filament voltage of all tubes of 25 wattsor more plate dissipation rating. This voltageshould be measured right at the tube sockets.If it is low, the filament transformer voltageshould be raised. If this is impossible, heavieror parallel wires should be used for filamentleads, cutting down their length if possible.Coaxial cable may be used for high voltageleads when it is desirable to shield them fromr-f fields. RG-8/U cable may be used at d-epotentials up to 8000 volts, and the lighterRG-17 /U may be used to potentials of 3000volts. Spark-plug type high-tension wire maybe used for unshielded leads, and will withstand10,000 volts.If this cable is used, the high-voltage leadsmay be cabled with filament and other lowvoltageleads. For high-voltage leads in lowpowerexciters, where the plate voltage is notover 450 volts, ordinary radio hookup wire ofgood quality will serve the purpose.No r-f leads should be cabled; in fact it isbetter to use enamelled or bare copper wirefor r-f leads and rely upon spacing for insulation.All r-f joints should be soldered, and thejoint should be a good mechanical junctionbefore solder is applied.The efficiency and Q of air coils commonlyused in amateur equipment is a factor of theshape of the coil, the proximity of the coil toother objects (including the coil form) and thematerial of which the coil is made. Dielectriclosses in so-called "air wound" coils arelow and the Q of such coils runs in the neighborhoodof 300 to 500 at medium frequencies.Unfortunately, most of the transmitting typeplug-in coils on the market designed for linkcoupling have far too small a pick up link forproper operation at 7 Me. and 3.5 Me. The coefficientof coupling of these coils is about0.5, and additional means must be employedto provide satisfactory coupling at these lowfrequencies. Additional inductance in serieswith the pick up link, the whole being reso-
<strong>HANDBOOK</strong> Inductors 361~C DISTRIBUTED@ ©Figure 8ELECTRICAL EQUIVALENT OF R-F CHOKE AT VARIOUS FREQUENCIESnated to the operating frequency will oftenpermit satisfactory coupling.Coil Placement For best Q a coil should bein the form of a solenoid withlength from one to two times the diameter. Forminimum interstage coupling, coils should bemade as small physically as is practicable.The coils should then be placed so that adjoiningcoils are oriented for minimum mutualcoupling. To determine if this condition exists,apply the following test: the axis of one of thetwo coils must lie in the plane formed by thecenter turn of the other coil. If this conditionis not met, there will be appreciable couplingunless the unshielded coils are very small indiameter or are spaced a considerable distancefrom each other.Insulation On frequencies above 7 Me., cera·mic, polystyrene, or Mycalex insulationis to be recommended. Cold flow mustbe considered when using polystyrene (Amphenol912, etc.). Bakelite has low losses onthe lower frequencies but should never beused in the field of high-frequency tank circuits.Lucite (or Plexiglas), which is availablein rods, sheets, or tubing, is satisfactory foruse at all radio frequencies where the r·f volt·ages are not especially high. It is very easyto work with ordinary tools and is not expensive.The loss factor depends to a consider·able extent upon the amount and kind of plasticizerused.The most important thing to keep in mindregarding insulation is that the best insulationis air. If it is necessary to reinforce air-woundcoils to keep turns from vibrating or touching,use strips of Lucite or polystyrene cementedin place with Amphenol 912 coil dope. Thiswill result in lower losses than the commonlyused celluloid ribs and Duco cement.Radio Frequency R-f chokes may be considChokesered to be special induct·ances designed to have ahigh value of impedance over a large range offrequencies. A practical r·f choke has induct·ance, distributed capacitance, and resistance.At low frequencies, the distributed capacityhas little effect and the electrical equivalentcircuit of the r·f choke is as shown in figureSA. As the operating frequency of the chokeis raised the effect of the distributed capacitybecomes more evident until at some particularfrequency the distributed capacity resonateswith the inductance of the choke and a parallelresonant circuit is formed. This point is shownin figure 88. As the frequency of operationis further increased the overall reactance ofthe choke becomes capacitive, and finally apoint of series resonance is reached (figureSC.). This cycle repeats itself as the operatingfrequency is raised above the series resonantpoint, the impedance of the choke rapidly becominglower on each successive cycle. Achart of this action is shown in figure 9. Itcan be seen that as the r·f choke approachesand leaves a condition of series resonance,the performance of the choke is seriously im·paired. The condition of series resonancemay easily be found by shorting the terminalsof the r-f choke in question with a piece ofwire and exploring the windings of the chokewith a grid-dip oscillator. Most commercialtransmitting type chokes have series reso·nances in the vicinity of 11 Me. or 24 Me.~2~rr----,-----~~TT----.-----.----.J:0~61.5wu~ 1.0wa.::; ·•H-----+----\-t-1--+------+\r-V-+--/'-=--lw.::0~ .I t!====!===t===J;==±~::!±==:::j10 I~ 20 30FREQUENCY (MC.)Figure 9FREQUENCY-IMPEDANCE CHARACTERISTICS FOR TYPICAL PIE-WOUNDR-F CHOKES
362 Transmitter Design<strong>THE</strong> R AD I 0®oxBOX@Figure 10GROUND LOOPS IN AMPLIFIER STAGESA. Using chassis returnB. Common grounc/ point18-4 GroundsAt frequencies of 30 Me. and below, a chassismay be considered as a fixed ground reference,since its dimensions are only a fractionof a wavelength. As the frequency is increasedabove 30 Me., the chassis must be consideredas a conducting sheet on which there arepoints of maximum current and potential. However,for the lower amateur frequencies, anobject may be assumed to be at ground potentialwhen it is affixed to the chassis.In transmitter stages, two important currentloops exist. One loop consists of the grid circuitand chassis return, and the other loopconsists of the plate circuit and chassis return.These two loops are shown in figure lOA.It can be seen that the chassis forms a returnfor both the grid and plate circuits, and thatground currents flow in the chassis towardsthe cathode circuit of the stage. For someyears the theory has been to separate theseground currents from the chassis by returningall ground leads to one point, usually thecathode of the tube for the stage in question.This is well and good if the ground leads areof minute length and do not introduce crosscouplings between the leads. Such a techniqueis illustrated in figure lOB. wherein all stagecomponents are grounded to the cathode pinof the stage socket. However, in transmitterconstruction the physical size of the compon·ents prevent such close grouping. It is necessaryto spread the components of such astage over a fairly large area. In this case itis best to ground items directly to the chassisat the nearest possible point, with short, directgrounding leads. The ground currents willflow from these points through the low induct·ance chassis to the cathode return of thestage. Components grounded on the top of thechassis have their ground currents flow throughholes to the cathode circuit which is usuallylocated on the bottom of the chassis, sincesuch currents travel on the surface of the chassis.The usual "top to bottom" ground pathis through the hole cut in the chassis for thetube socket. When the gain per stage is relativelylow, or there are only a small numberof stages on a chassis this universal groundingsystem is ideal. It is only in high gainstages (i-f strips) where the "gain per inch"is very high that circulating ground currentswill cause operational instability.lntercoupl ing of It is important to prevent inGround Currents tercoupling of various differentground currents when thechassis is used as a common ground return.To keep this intercoupling at a minimum, thestage should be completely shielded. Thiswill prevent external fields from generatingspurious ground currents, and prevent theground currents of the stage from upsettingthe action of nearby stages. Since the groundcurrents travel on the surface of the metal,the stage should be enclosedin an electricallytight box. When this is done, all ground currentsgenerated inside the box will remain inthe box. The only possible means of escapefor fundamental and harmonic currents are imperfectionsin this electrical! y tight box. Wheneverwe bring a wire lead into the box, makea ventilation hole, or bring a control shaftthrough the box we create an imperfection. Itis important that the effect of these imperfectionsbe reduced to a minimum.18-5 Holes, Leads and ShaftsLarge size holes for ventilation may be putin an electrically tight box provided they areproperly screened. Perforated metal stock havingmany small, closely spaced holes is thebest screening material. Copper wire screenmay be used provided the screen wires arebonded together every few inches. As the wirecorrodes, an insulating film prevents contactbetween the individual wires, and the attenuationof the screening suffers. The screeningmaterial should be carefully soldered to the
<strong>HANDBOOK</strong>Shielding 363HOLES FORMETER STUDStAETERLEADFigure llASIMPLE METER SHIELDHOLEbox, or bolted with a spacing of not less thantwo inches between bolts. Mating surfaces ofthe box and the screening should be clean.A screened ventilation opening should beroughly three times the size of an eq~ivalentunscreened opening, since the screenmg representsabout a 70 per cent coverage of thearea. Careful attention must be paid to equipmentheating when an electrically tight boxis used.Commercially available panels having halfinchventilating holes may be used as part ofthe box. These holes have much less attenuationthan does screening, but will perform in asatisfactory manner in all but the areas ofweakest TV reception. If it is desired to reduceleakage from these panels to a minim~m,the back of the grille must be covered wahscreening tightly bonded to the pa~el. .Doors may be placed in electncally ttghtboxes provided there is no r-f lea~age aroundthe seams of the door. Electromc weatherstrippingor metal "finger stock" may be ~sedto seal these doors. A long, narrow slot In aclosed box has the tendency to act as a slotantenna and harmonic energy may pass morereadily through such an opening than it wouldthrough a much larger circular hole ..Variable capacitor shafts or switch sha~tsmay act as antennas, picking up curren~s Insidethe box and re-radiating them outside ofthe box. It is necessary either to ground theshaft securely as it leaves the box, or else tomake the shaft of some insulating material.A two or three inch panel meter requires alarge leakage hole if it is mounte~ ~n _the wallof an electrically tight box. To mmimize leakage,the meter leads should be by-passe~ andshielded. The meter should be encased With ametal shield that makes contact to the boxentirely around the meter. The connectingstuds of the meter may project through theback of the metal shield. Such a shield maybe made out of the end of a tin can of correctINTERNAL GROUND / jJ IllCURRENTS L_ dV' II' INTERNAL .. EXTERNAL CURRENTS~ ON. EXTERIOR OF BOXFigure 11 BUse of cooxiol connectors on electricallytight box prevents escape of ground currentsfrom interior of box. At th·e same time externalfields are not conducted into the interiorof the box.diameter cut to fit the depth of the meter.This co:Oplete shield assembly is shown infigure llA.Careful attention should be paid to leadsentering and leaving the electric~ly_ tightbox. Harmonic currents generated wtthm thebox can easily flow out of the box on poweror control leads, or even on the outer shieldsof coaxially shielded wires. Figure llB illustratesthe correct method of bringing shieldedcables into a box where it is desired to preservethe continuity of the shielding.Unshielded leads entering the box must becarefully filtered to prevent fundamental andharmonic energy from escaping down the lead.Combinations of r-f chokes and low inductanceby-pass capacitors should be used in powerleads. If the current in the lead is high, thechokes must be wound of large gauge wire.Composition resistors may be substituted forthe r-f chokes in high impedance circuits.Bulkhead or feed-through type capacitors arepreferable when passing a lead through ashield partition. A summary of lead leakagewith various filter arrangements is shown infigure 12.Internal Leads Leads that connect two pointswithin an electrically tight box
364 Transmitter Design<strong>THE</strong> R AD I 0FIELD STRENGTHT~~T IN.UV ,--C~~D!Q~C~A~R----------,410II1212000 B+ r=SMALL HOLE IN SHIELD O~Oc.l10000830800ISO7014060011025TRACER 1- 10001l. CARBONRFC- OHM ITE Z-50CJ- 75JJ.UFCERAMICFEED-THROUGH..LCIIJ......reLDED HOOK-uP WIRE Ij_ iJ~R :1: "'f{Cz ~TC1 RFC :;:Cz I~I..L ~1: SHIELD I~I~I~I_LCI RFC ..,LS:I RFC Ilc4 '!!!' Ic4 ~ fez ' I~I...lC3 ,m, ..J..S:3 ;.tELDEDiWI~T T ~ czlL -----"-----_::__ _JFigure 12C2 -.005 DISC CERAMICC3 - .0 I SPRAGUE HI-PASSC4- .005 CERAMICFEED-THROUGHLEAD LEAKAGE WITH VARIOUSLEAD FILTERING SYSTEMS(COURTESY W1DBM)may pick up fundamental and harmonic currentsif they are located in a strong field offlux. Any lead forming a closed loop with itselfwill pick up such currents, as shown infigure 13. This effect is enhanced if the leadhappens to be self-resonant at the frequencyat which the exciting energy is supplied. Thesolution for all of this is to by-pass all internalpower leads and control leads at eachend, and to shield these leads thei! entirelength. All filament, bias, and meter leadsshould be so treated. This will make the jobof filtering the leads as they leave the boxmuch easier, since normally "cool" leadswithin the box will not have picked up spuriouscurrents from nearby "hot" leads.was operated near this frequency marked instabilitywas noted, and the filaments of the810 tubes increased in brilliance when platevoltage was applied to the amplifier, indicatingthe presence of r.f. in the filament circuit.Changing the filament by-pass capacitors to.01-(Lfd. lowered the filament resonance frequencyto 2.2 Me. and cured this effect. Aceramic capacitor of .01-fLfd. used as a filamentby-pass capacitor on each filament legseems to be satisfactory from both a resonantand a TVI point of view. Filament by-passcapacitors smaller in value than .01-fLfd.should be used with caution.Various parasitic resonances are also foundin plate and grid tank circuits. Push-pull tankcircuits are prone to double resonances, asshown in figure 14. The parasitic resonancecircuit is usually several megacycles higherthan the actual resonant frequency of the fulltank circuit. The cure for such a double resonanceis the inclusion of an r-f choke in thecenter tap lead to the split coil.Chassis Material From a point of view of electricalproperties, aluminumis a poor chassis material. It is difficult tomake a soldered joint to it, and all groundsmust rely upon a pressure joint. These pres-SHIELDEDCOMPARTt.AENFigure 13ILLUSTRATION OF HOW A SUPPOSEDLYGROUNDED POWER LEAD CAN COUPLEENERGY FROM ONE COMPARTMENTTO ANO<strong>THE</strong>R®18-6 Parasitic ResonancesFilament leads within vacuum tubes mayresonate with the filament by-pass capacitorsat some particular frequency and cause instabilityin an amplifier stage. Large tubes ofthe 810 and 250TH type are prone to this spuriouseffect. In particular, a push-pull 810 amplifierusing .001-(Lfd. filament by-pass capacitorshad a filament resonant loop that fell inthe 7-Mc. amateur band. When the amplifierY TIGHTTELECTRICALLY-TIGHTCOMPARTMENT® RADIATIONFl ELD \ BULKHEAD TYPE" ~Y-PASS CAPACITOR"'IZS\T\-:l •cLOOP r II 1----f II fILLUSTRATION OF LEAD ISOLATION BYPROPER USE OF BULKHEAD BYPASSCAPACITOR
<strong>HANDBOOK</strong>Parasitic Oscillations 365Figure 14DOUBLE RESONANCE EFFECTS IN PUSHPULL TANK CIRCUIT MAY BE ELIMINATED BY <strong>THE</strong> INSERTION OF ANYR-F CHOKE IN <strong>THE</strong> COIL CENTERTAP LEADsure joints are prone to give trouble at a laterdate because of high resistivity caused bythe formation of oxides from eletrolytic actionin the joint. However, the ease of workingand forming the aluminum material far outweighsthe electrical shortcomings, and aluminumchassis and shielding may be usedwith good results provided care is taken inmaking all grounding connections. Cadmiumand zinc plated chassis are preferable from acorrosion standpoint, but are much more difficultto handle in the home workshop.18-7 Parasitic Oscillationin R-F AmplifiersParasitics (as distinguished from self-oscillationon the normal tuned frequency of theamplifier) are undesirable oscillations eitherof very high or very low frequencies whichmay occur in radio-frequency amplifiers.They may cause spurious signals (whichare often rough in tone) other than normal harmonics,hash on each side of a modulated carrier,key clicks, voltage breakdown or flashover,instability or inefficiency, and shortenedlife or failure of the tubes. They may be dampedand stop by themselves after keying or modulationpeaks, or they may be undamped andbuild up during ordinary unmodulated transmission,continuing if the excitation is removed.They may result from series or parallelresonant circuits of all types. Due to neutralizinglead length and the nature of mostparasitic circuits, the amplifier usually is notneutralized for the parasitic frequency.Sometimes the fact that the plate supply iskeyed will obscure parasitic oscillations in a+final amplifier stage that might be very severeif the plate voltage were left on and the excitationwere keyed.In some cases, an all-wave receiver willprove helpful in locating v-h-f spurious oscillations,but it may be necessary to check fromseveral hundred megacycles downward in frequencyto the operating range. A normal harmonicis weaker than the fundamental but ofgood tone; a strong harmonic or a rough noteat any frequency generally indicates a parasitic.In general, the cure for parasitic oscillationis two-fold: The oscillatory circuit is dampeduntil sustained oscillation is impossible, orit is detuned until oscillation ceases. An examinationof the various types of parasitic oscillationsand of the parasitic oscillatory circuitswill prove handy in applying the correctcure.Low FrequencyParasitic OscillationsOne type of unwantedoscillation often occursin shunt-fed circuits inwhich the grid and plate chokes resonate, coupledthrough the tube's interelectrode capacitance.This also can happen with series feed.This oscillation is generally at a much lowerfrequency than the operating frequency andwill cause additional carriers to appear, spacedfrom perhaps twenty to a few hundred kilocycleson either side of the main wave. Such acircuit is illustrated in figure 15. In this case,RFC, and RFC 2 form the grid and plate inductancesof the parasitic oscillator. The neutralizingcapacitor, no longer providing out-ofphasefeedback to the grid circuit actually enhancesthe low frequency oscillation. Becauseof the low Q of the r-f chokes, they will usuallyrun warm when this type of parasitic oscillationis present and may actually char andburn up. A neon bulb held near the oscillatorycircuit will glow a bright yellow, the colorappearing near the glass of the neon bulb andnot between the electrodes.One cure for this type of oscillation is tochange the type of choke in either the plateor the grid circuit. This is a marginal cure,because the amplifier may again break into thesame type of oscillation when the plate voltageis raised slightly. The best cure is to removethe grid r-f choke entirely and replaceit with a wirewound resistor of sufficient wattageto carry the amplifier grid current. If theinclusion of such a resistor upsets the operatingbias of the stage, an r-f choke may beused, with a 100-ohm 2-watt carbon resistorin series with the choke to lower the operatingQ of the choke. If this expedient does noteliminate the condition, and the stage underinvestigation uses a beam-tetrode tube, negativeresistance can exist in the screen circuit
366 Transmitter Design<strong>THE</strong> R AD I 0NCRFC1GRID"TANK•RFC2.PLATE"TANK.,R.F. CIRCUIT+PARASITIC CIRCUIT FORLOW FREQ. OSCILLATION® ® ©Figure 15<strong>THE</strong> CAUSE AND CURE OF LOW FREQUENCY PARASITICSCUREof such tubes. Try larger and smaller screenby-pass capacitors to determine whether or notthey have any effect. If the condition is comingfrom the screen circuit an audio choke witha resistor across it in series with the screenfeed lead will often eliminate the trouble.Low-frequency parasitic oscillations canoften take place in the audio system of an AMtransmitter, and their presence will not beknown until the transmitter is checked on areceiver. It is easy to determine whether ornot the oscillations are coming from the modulatorsimply by switching off the modulatortubes. If the oscillations are coming from themodulator, the stage in which they are beinggenerated can be determined by removing tubessuccessively, starting with the first speechamplification stage, until the oscillation stops.When the stage has been found, remedial stepscan be taken on that stage.If the stage causing the oscillation is a lowlevelspeech stage it is possible that thetrouble is coming from r-f or power-supplyfeedback, or it may be coming about as a resultof inductive coupling between two transformers.If the oscillation is taking place ina high-level audio stage, it is possible thatinductive or capacitive coupling is takingplace back to one of the low-level speechstages. It is also possible, in certain cases,that parasitic push-pull oscillation can takeplace in a Class B or Class AB modulator asa result of the grid-to-plate capacitance withinthe tubes and in the stage wiring. This conditionis more likely to occur if capacitorshave been placed across the secondary of thedriver transformer and across the primary ofthe modulation transformer to act in the reductionof the amplitude of the higher audio frequencies.Relocation of wiring or actual neutralizationof the audio stage in the mannerused for r-f stages may be required.It may be said in general that the presenceof low-frequency parasitics indicates thatsomewhere in the oscillating circuit there isan impedance which is high at a frequency inthe upper audio or low r-f range. This impedancemay include one or more r-f chokes ofthe conventional variety, power supply chokes,modulation components, or the high impedancemay be presented simply by an RC circuitsuch as might be found in the screen-feed circuitof a beam-tetrode amplifier stage.18-8 Elimination of V-H-FParasitic OscillationsV-h-f parasitic oscillations are often difficultto locate and difficult to eliminate sincetheir frequency often is only moderately abovethe desired frequency of operation. But it maybe said that v-h-f parasitics always may beeliminated if the operating frequency is appreciablybelow the upper frequency limit for thetubes used in the stage. However, the eliminationof a persistent parasitic oscillation ona frequency only moderately higher than thedesired operating frequency will involve asacrifice in either the power output or thepower sensitivity of the stage, or in both.Beam-tetrode stages, particularly thoseusing 807 type tubes, will almost invariablyhave one or more v-h-f parasitic oscillationsunless adequate precautions have been takenin advance. Many of the units described inthe constructional section of this edition hadparasitic oscillations when first constructed.But these oscillations were eliminated in eachcase; hence, the expedients used in theseequipments should be studied. V-h-f parasiticsmay be readily identified, as they cause a
<strong>HANDBOOK</strong>Parasitic Oscillations 367neon lamp to have a purple glow close to theelectrodes when it is excited by the parasiticenergy.Parasitic Oscillationswith TriodesTriode stages are lesssubject to parasitic oscillationsprimarily becauseof the much lower power sensitivity ofsuch tubes as compared to beam tetrodes. Butsuch oscillations can and do take place. Usually,however, it is not necessary to incorporatelosser resistors as normally is the case withbeam tetrodes, unless the triodes are operatedquite near to their upper frequency limit, orthe tubes are characterized by a relativelyhigh transconductance. Triode v-h-f parasiticoscillations normally may be eliminated by adjustmentof the lengths and effective inductanceof the leads to the elements of the tubes.In the case of triodes, v-h-f parasitic oscillationsoften come about as a result of inductancein the neutralizing leads. This is particularlytrue in the case of push-pull amplifiers.The cure for this effect will usually be foundin reducing the length of the neutralizing leadsand increasing their diameter. Both the reductionin length and increase in diameter willreduce the inductance of the leads and tendto raise the parasitic oscillation frequencyuntil it is out of the range at which the tubeswill oscillate. The use of straightforward circuitdesign with short leads will assist inforestalling this trouble at the outset. Butterfly-typetank capacitors with the neutralizingcapacitors built into the unit (such as theB&W type) are effective in this regard.V-h-f parasitic oscillations may take placeas a result of inadequate by-passing or longby-pass leads in the filament, grid-return andplate-return circuits. Such oscillations alsocan take place when long leads exist betweenthe grids and the grid tuning capacitor or betweenthe plates and the plate tuning capacitor.The grid and plate leads should be keptshort, but the leads from the tuning capacitorsto the tank coils can be of any reasonablelength insofar as parasitic oscillations areconcerned. In an amplifier where oscillationshave been traced to the grid or plate leads,their elimination can often be effected by makingthe grid leads much longer than the plateleads or vice versa. Sometimes parasitic oscillationscan be eliminated by using iron ornichrome wire for the grid or plate leads, orfor the neutralizing leads. But in any eventit will always be found best to make the neutralizingleads as short and of as heavy conductoras is practicable.In cases where it has been found that increasedlength in the grid leads for an amplifieris required, this increased length can oftenbe wound into the form of a small coil and stillPC=er. # 18 E. ON 10011,2W.CARSON RESISTORFigure 16GRID PARASITIC SUPPRESSORS IN PUSHPULL TRIODE STAGEobtain the desired effect. Winding these smallcoils of iron or nichrome wire may sometimesbe of assistance.To increase losses at the parasitic frequency,the parasitic coils may be wound on 100-ohm2-watt resistors. These "lossy" suppressorsshould be placed in the grid leads of the tubesclose to the grid connection, as shown in figure16.Porosities withBeam TetrodesWhere beam-tetrode tubes areused in the stage which hasbeen found to be generatingthe parasitic oscillation, all the foregoingsuggestions apply in general. However, thereare certain additional considerations involvedin elimination of parasitics from beam-tetrodeamplifier stages. These considerations involvethe facts that a beam-tetrode amplifier stagehas greater power sensitivity than an equivalenttriode amplifier, such a stage has a certainamount of screen-lead inductance whichmay give rise to trouble, and such stages havea small amount of feedback capacitance.Beam-tetrode stages often will require theinclusion of a neutralizing circuit to eliminateoscillation on the operating frequency. However,oscillation on the operating frequencynormally is not called a parasitic oscillation,and different measures are required to eliminatethe condition.Basically, parasitic oscillations in beamtetrodeamplifier stages fall into two classes:cathode-grid-screen oscillations, and cathodescreen-plateoscillations. Both these types ofoscillation can be eliminated through the useof a parasitic suppressor in the lead betweenthe screen terminal of the tube and the screenby-pass suppressor, as shown in figure 17.Such a suppressor has negligible effect on theby-passing effect of the screen at the operatingfrequency. The method of connecting this+
368 Transmitter Design<strong>THE</strong> R AD I 0+ +PC=3 T. #18£. ON52.11,2 W. CARBON RESISTORRFC=OHMITE Z-50 OREQUIVALENTFigure 17SCREEN PARASITIC SUPPRESSION CIR·CUlT FOR TETRODE TUBESsuppressor to tubes having dual screen leadsis shown in figure 18. At the higher frequen·cies at which parasitics occur, the screen isno longer at ground potential. It is thereforenecessary to include an r-f choke by-pass condenserfilter in the screen lead after the parasiticsuppressor. The screen lead, in addition,should be shielded for best results.During parasitic oscillations, considerabler-f voltage appears on the screen of a tetrodetube, and the screen by-pass condenser caneasily be damaged. It is best, therefore, toemploy screen by-pass condensers whose d-eworking voltage is equal to twice the maximumapplied screen voltage.The grid-screen oscillations may occasionallybe eliminated through the use of a parasiticsuppressor in series with the grid leadof the tube. The screen plate oscillations mayalso be eliminated by inclusion of a parasiticsuppressor in series with the plate lead of thetube. A suitable grid suppressor may be madeof a 22-ohm 2-watt Ohmite or Allen-Bradleyresistor wound with 8 turns of no. 18 enameledwire. A plate circuit suppressor is more of aproblem, since it must dissipate a quantity ofpower that is dependent upon just how closethe parasitic frequency is to the operating frequencyof the tube. If the two frequencies areclose, the suppressor will absorb some of thefundamental plate circuit power. For kilowattstages operating no higher than 30 Me. a satisfactoryplate circuit suppressor may be madeof five 570-ohm 2-watt carbon resistors 1n parallel,shunted by 5 turns of no. 16 enameledwire, X inch diameter and %1 inch long (figure19A and B).The parasitic suppressor for the plate circuitof a small tube such as the 5763, 2E26,807, 6146 or similar type normally may consistof a 47-ohm carbon resistor of 2-watt sizewith 6 turns of no. 18 enameled wire woundaround the resistor. However, for operationabove 30 Me., special tailoring of the valueFigure 18PHOTO OF APPLICATION OF SCREENPARASITIC SUPPRESSION CIRCUITOF FIGURE 17of the resistor and the size of the coil woundaround it will be required in order to attainsatisfactory parasitic suppression without excessivepower loss in the parasitic suppressor.Tetrode Screening Isolation between the gridand plate circuits of a retrodetube is not perfect. For maximum stability,it is recommended that the tetrode stagebe neutralized. Neutralization is absolutelynecessary unless the grid and plate circuitsof the tetrode stage are each completely isolatedfrom each other in electrical! y tightboxes. Even when this is done, the stage willshow signs of regeneration when the plateand grid tank circuits are tuned to the samefrequency. Neutralization will eliminate thisregeneration. Any of the neutralization circuitsdescribed in the chapter Generation ofR-F Energy may be used.18-9 Checking for-ParasiticOscillationsIt is an unusual transmitter which harborsno parasitic oscillations when first constructed
<strong>HANDBOOK</strong>Parasitic Oscillations 369PC=ar.~t-JBE.ON22.D.,2W. COMPOSITION RESISTORFOR 60 7, ETC.PC=~r:#te£. ON41.n.~zw.COMPOSITION RESISTORFOR 4-250A, ETC.PC:..5-.570J2., 2 W. COMPOSITIONRESISroRS IN PARALLEL WITHST.IIIG£. /AI" DIA.®+ +®+ +Figure 19PLATE AND GRID PARASITIC SUPPRESSION IN TETRODE TUBESand tested. Therefore it is always wise to followa definite procedure in checking a newtransmitter for parasitic oscillations.Parasitic oscillations of all types are mosteasily found when the stage in question isrunning by itself, with full plate (and screen)voltage, sufficient protective bias to limit theplate current to a safe value, and no excitation.One stage should be tested at a time,and the complete transmitter should never beput on the air until all stages have been thoroughlychecked for parasitics.To protect tetrode tubes during tests forparasitics, the screen voltage should be appliedthrough a series resistor which will limitthe screen current to a safe value in case theplate voltage of the tetrode is suddenly removedwhen the screen supply is on. The correctprocedure for parasitic testing is as follows(figure 20):1. The stage in question should be coupledto a dummy load, and tuned up in correct operatingshape. Sufficient protective bias shouldbe applied to the tube at all times. For protectionof the stage under test, a lamp bulbshould be added in series with one leg of theprimary circuit of the high voltage power supply.As the plate supply load increases duringa period of parasitic oscillation, the voltagedrop across the lamp increases, and the effectiveplate voltage drops. Bulbs of varioussize may be tried to adjust the voltage undertesting conditions to the correct amount. If aVariac or Powerstat is at hand, it may be usedin place of the bulbs for smoother voltage control.Don't test for parasitics unless some typeof voltage control is used on the high voltagesupply! When a stage breaks into parasiticoscillations, the plate current increases violently,and some protection to the tube undertest must be used.2. The r-f excitation to the tube should nowbe removed. When this is done, the grid, screenand plate currents of the tube should drop tozero. Grid and plate tuning condensers shouldbe tuned to minimum capacity. No change inresting grid, screen or plate current shouldbe observed. If a parasitic is present, grid currentwill flow, and there will be an abrupt increasein plate current. The size of the lampbulb in series with the high voltage supply maybe varied until the stage can oscillate continuously,without exceeding the rated plate dissipationof the tube.3. The frequency of the parasitic may nowbe determined by means of an absorption wavemeter, or a neon bulb. Low frequency oscillationswill cause a neon bulb to glow yellow.High frequency oscillations will cause thebulb to have a soft, violet glow. Once the frequencyof oscillation is determined, the curessuggested in this chapter may be applied tothe stage.4. When the stage can pass the above testwith no signs of parasitics, the bias supply ofthe tube in question should be decreased untilthe tube is dissipating its full plate ratingwhen full plate voltage is applied, with no r-fFigure 20SUGGESTED TEST SETUP FOR PARASITICTESTS
370 Transmitter Designexcitation. Excitation may now be applied andthe stage loaded to full input into a dummyload. The signal should now be monitored ina nearby receiver which has the antenna terminalsgrounded or otherwise shorted out. Aseries of rapid dots should be sent, and thefrequency spectrum for several megacycleseach side of the carrier frequency carefullysearched. If any vestige of parasitic is left,it will show up as an occasional "pop" on akeyed dot. This "pop" may be enhanced by aslight detuning of either the grid or plate circuit.5. If such a parasitic shows up, it meansthat the stage is still not stable, and furthermeasures must be applied to the circuit. Parasiticsuppressors may be needed in both screenand grid leads of a tetrode, or perhaps in bothgrid and neutralizing leads of a triode stage.As a last resort, a 10,000-ohm 25-watt wirewoundresistor may be shunted across the gridcoil, or grid tuning condenser of a high poweredstage. This strategy removed a keyingpop that showed up in a commercial transmitter,operating at a plate voltage of 5000.Test for ParasiticTendency in TetrodeAmplifiersIt is common experienceto develop an engineeringmodel of a newequipment that is apparentlyfree of parasitics and then find troublesomeoscillations showing up in productionunits. The reason for this is that the equipmenthas a parasitic tendency that remains belowthe verge of oscillation until some change ina component, tube gain, or operating conditionraises the gain of the parasitic circuit enoughto start oscillation.In most high frequency transmitters thereare a great many resonances in the tank circuitsat frequencies other than the desiredFigure 21PARASITIC GAIN MEASUREMENTGrid-dip oscillator and vacuum tubevoltmeter may be used to measure parasiticstage gain over 100kc-200mcregion.operating frequency. Most of these parasltrcresonant circuits are not coupled to the tubeand have no significant tendency to oscillate.A few, however, are coupled to the tube insome form of oscillatory circuit. If the regenerationis great enough, oscillation at the parasiticfrequency results. Those spurious circuitsexisting just below oscillation must be foundand suppressed to a safe level.One test method is to feed a signal from agrid-dip oscillator into the grid of a stage andmeasure the resulting signal level in the platecircuit of the stage, as shown in figure 21. Thetest is made with all operating voltages appliedto the tubes. Class C stages should have biasreduced so a reasonable amount of static platecurrent flows. The grid-dip oscillator is tunedover the range of 100 kc to 200 me. and therelative level of the r·f voltmeter is watchedand the frequencies at which voltage peaksoccur are noted. Each significant peak in volt·age gain in the stage must be investigated. Circuitchanges or suppression must then be addedto reduce all peaks by 10 db or more in amplitude.
CHAPTER NINETEENTelevision and BroadcastInterferenceThe problem of interference to televisionreception is best approached by the philosophydiscussed in Chapter Eighteen. By correctdesign procedure, spurious harmonic generationin low frequency transmitters may be heldto a minimum. The remaining problem is twofold:to make sure that the residual harmonicsgenerated by the transmitter are not radiated,and to make sure that the fundamental signalof the transmitter does not overload the televisionreceiver by reason of the proximity ofone to the other.In an area of high TV-signal field intensitythe TVI problem is capable of complete solutionwith routine measures both at the amateurtransmitter and at the affected receivers. Butin fringe areas of low TV-signal field strengththe complete elimination of TVI is a difficultand challenging problem. The fundamentalsillustrated in Chapter Fifteen must be closelyfollowed, and additional antenna filtering ofthe transmitter is required.19-1 Types of TelevisionInterferenceThere are three main types of TVI whichmay be caused singly or in combination by theemissions from an amateur transmitter. Thesetypes of interference are:(1) Overloading of the TV set by the trans·mitter fundamental(2) Impairment of the picture by spuriousemissions(3) Impairment of the picture by the radiationof harmonics371TV SetOverloadingEven if the amateur transmitterwere perfect and had no harmonicr ad i at ion or spuriousemissions whatever, it still would be likely tocause overloading of TV sets whose antennaswere within a few hundred feet of the transmittingantenna. This type of overloading isessentially the same as the common type ofBCI encountered when operating a medium·power or high-power amateur transmitter withina few hundred feet of the normal type ofBCL receiver. The field intensity in the immediatevicinity of the transmitting antennais sufficiently high that the amateur signalwill get into the BC or TV set either throughoverloading of the front end, or through thei-f, video, or audio system. A characteristicof this type of interference is that it alwayswill be eliminated when the transmitter tem·porarily is operated into a dummy antenna.Another characteristic of this type of overloadingis that its effects will be substantiallycontinuous over the entire frequencycoverage of the BC or TV receiver. Channels2 through 13 will be affected in approximatelythe same manner.With the overloading type of interferencethe problem is simply to keep the fundamentalof the transmitter out of the affected receiver.Other types of interference may or may notshow up when the fundamental is taken out ofthe TV set (they probably will appear), but atleast the fundamental must be eliminated first.The elimination of the transmitter fundamentalfrom the TV set is normally the only operationperformed on or in the vicinity of the TVreceiver. After the fundamental has been elimi-
372 TV and Broadcast Interference<strong>THE</strong> R AD I 0@FOR 300-0HM LINE, SHIELDED OR UNSHIELDEDTO TV ANTENNACOAXFITTING®L2 }C2TO ANTENNATERMINALSOF TV SET@ FOR 50-75 OHM COAXIAL LINEFigure 1TUNED TRAPS FOR <strong>THE</strong> TRANSMITTERFUNDAMENTALThe arrangement at (A) has proven to be effectivein eliminating the condition of gen·era/ blocking as cousec/ by a 28-Mc. transmitterin the vicinity of a TV receiver. Thetunecl circuits L1-C1 are resonated separate·ly to the frequency of transmission. The adjustmentmay be clone at the station, or itmay be occomp/ishec/ at the TV receiver bytuning for minimum interference on the TVscreen.Shown at (B) is on alternative arrangementwith a series-tunecl circuit across the antennaterminals of the TV set. The tunecl circuitshould be resonated to the operatingfrequency of the transmitter. This arrangementgives less attenuation of the interferingsignal than that at (A); the circuit hasproven effective with interference from transmitterson the 50-Mc. bone/, one! with lowpower28-Mc. transmitters.Figure 2HIGH-PASS TRANSMISSION LINE FILTERSThe arrangement of (A) will stop the passingof all signals below about 45 Me. from theantenna transmission I ine into the TV set.Coils Lt ore each 1.2 microhenrys (17 turnsno. 24 enom. c/osewounc/ on '4-inch clio. polystyreneroc/) with the center top grounc/ecl.It will be founc/ best to scrape, twist, one/so/c/er the center top before winding the coil.The number of turns each sic/e of the top maythen be voriec/ until the top is in the exactcenter of the winding. Coil L2 is 0.6 microhenry(12 turns no. 24 enom. c/osewounc/ on'4-inch clio. polystyrene roc/). The capacitorsshoulc/ be about 16.5 !1{-Lfc/., but either 15 or20 f-411c/. ceramic capacitors will give satisfactoryresults. A simi/or filter for coaxialantenna transmission line is shown at (B).Both coils should be 0.12 microhenry (7 turnsno. 18 enom. spacec/ to !1 inch on '4-inch eli a.polystyrene roc!).7S !1{-Lfc!.Capacitors C2 shoulcl bemic/get ceramics, while c3 shoulclbe a 40-f-411cl. ceramic.nated as a source of interference to reception,work may then be begun on or in the vicinityof the transmitter toward eliminating the othertwo types of interference.Taking Outthe FundamentalMore or less standard SCItypepractice is most commonlyused in taking outfundamental interference. Wavetraps and filtersare installed, and the antenna system mayor may not be modified so as to offer less responseto the signal from the amateur transmitter.In regard to a comparison betweenwavetraps and filters, the same considerationsapply as have been effective in regard to BCIfor many years; wavetraps are quite effectivewhen properly installed and adjusted, but theymust be readjusted whenever the band of operationis changed, or even when moving fromone extreme end of a band to the other. Hence,wavetraps are not recommended except whenoperation will be confined to a relatively narrowportion of one amateur band. However, figure1 shows two of the most common signaltrapping arrangements.High-Pass Filters High-pass filters in theantenna lead of the TV sethave proven to be quite sat is factory as ameans of eliminating TVI of the overloadingtype. In many cases when the interfering transmitteris operated only on the bands below7.3 Me., the use of a high-pass filter in theantenna lead has completely eliminated all
<strong>HANDBOOK</strong>Harmonic Radiation 3 73TVI. In some cases the installation of a highpassfilter in the antenna transmission lineand an a-c line filter of a standard variety hasproven to be completely effective in eliminatingthe interference from a transmitter operatingin one of the lower f r e que n c y amateurbands.In general, it is suggested that commerciallymanufactured high-pass filters be purchased.Such units are available from a number of manufacturersat a relatively moderate cost. However,such units may be home constructed;suggested designs are given in figures 2 and3. Types for use both with coaxial and withbalanced transmission lines have been shown.In most cases the filters may be constructedin one of the small shield boxes which arenow on the market. Input and output terminalsmay be standard connectors, or the inexpensivetype of terminal strips usually used onBC and TV sets may be employed. Coaxialterminals should of course be employed whena coaxial feed line is used to the antenna. Inany event the leads from the filter box to theTV set should be very short, including boththe antenna lead and the ground lead to thebox itself. If the leads from the box to the sethave much length, they may pick up enoughsignal to nullify the effects of the high-passfilter.Blocking from50-Mc. SignalsOperation on the 50-Mc. amateurband in an area wherechannel 2 is in use for TVimposes a special problem in the matter ofblocking. The input circuits of most TV setsare sufficiently broad so that an amateur signalon the 50-Mc. band will ride through withlittle attenuation. Also, the normal TV antennawill have a quite large response to a signalin the 50-Mc. band since the lower limit ofchannel 2 is 54 Me.High-pass filters of the normal type simplyare not capable of giving sufficient attenuationto a signal whose frequency is so closeto the necessary pass band of the filter. Hence,a resonant circuit element, as illustrated infigure 1, must be used to trap out the amateurfield at the input of the TV set. The trap mustbe tuned or the section of transmission linecut, if a section of line is to be used for aparticular f r e que n c y in the 50-Me. band.This frequency will have to be near the lowerfrequency limit of the 50-Mc. band to obtainadequate rejection of the amateur signal whilestill not materially affecting the response ofthe receiver to channel 2.Elimination ofSpurious Emissionsfor the time being)All spurious emissionsfrom amateur transmitters(ignoring harmonic signalsmust be eliminated to com-Figure 3SERIES-DERIVED HIGH-PASS FILTERThis filter Is designee/ for use in the300-ohm transmission line from the TVantenna to the TV receiver. Nominal cutofffrequency is 36 Me. anc/ maximum rejectionis at about 29 Me.C 1 ,C 6 -15-,L4.Lfd. zero-coefficient ceramicc2,c3,c4,c5 -20-p.,u.fd. zero-coefficient cera·micL 1 ,L 3 -2.0 fLh. About 24 turns no. 28 d.c.c.wound to '%, 11 on %, 11 diameter polystyrenerod. Turns should be adjusted until thecoil resonates to 29 Me. with the associ·ated 15·!L!Lfd. capacitor.L 2 -0.66 fLh., 14 turns no. 28 d.c.c. woundto 5 /s" on% 11 dio. polystyrene rod. Adjustturns to resonate externally to 20 Me.with an auxiliary 100-J.LI.Lfd. capacitorwhose value is accurately known.ply with FCC regulations. But in the pastmany amateur transmitters have emitted spurioussignals as a result of key clicks, parasides,and overmodulation transients. In mostcases the operators of the transmitters werenot aware of these emissions since they wereradiated only for a short distance and hencewere not brought to his attention. But withone or more TV sets in the neighborhood itis probable that such spurious signals willbe brought quickly to his attention.19-2 Harmonic RadiationAfter any condition of blocking at the TVreceiver has been eliminated, and when thetransmitter is completely free of transientsand parasitic oscillations, it is probable thatTVI will be eliminated in certain cases. Certainlygeneral interference should be eliminated,particularly if the transmitter is a welldesigned affair operated on one of the lowerfrequency bands, and the station is in a highsignalTV area. But when the transmitter isto be operated on one of the higher frequencybands, and particularly in a marginal TV area,the job of TVI-proofing will just have begun.The elimination of harmonic radiation fromthe transmitter is a difficult and tedious jobwhich must be done in an orderly manner ifcompletely satisfactory results are to be obtained.
374 TV and Broadcast Interference <strong>THE</strong> R AD I 0~~~~~~~~E2ND 3RD 4TH 5TH 6TH 7TH 8TH 9TH 10TH7.0 21-21.9 42-44 56-~8.4 63-65.7 70-73I7.3TV I.FNEWTV LF.CH®NE @NEL C@NEL14.0 42-43 56-57.6 70-72 84-86.4 98-100.8INEW CH®NEL CHANNEL CHANNEL14.4TV I.F'. @) ® ~R~~-CAST21.0 63-64.35 84-85.8 105-1p7.25 189-193 210-214.5ICHANNEL21.45 H~NEL FMCH@ELBROAD-(TV I. F.) ®CAST26.96 3.92- 80.86- 107.84- 189 21827 1 54.46 81.69 108.9223 CH~NEL CH(SNEL FM CH®EL CH~NELBROAD-ABOVE 2.7CASTt.AC..ONLY28.0 56-59.4 84-89.1 168-178.2 196-207.9I29.7 CH~NEL CH®NEL CHANNEL0 ®>oo~50.0 100-108 200-216 450-4e6 500-540IFM54.0 BROAD- ~·@~CAST~E INTERFERENCETO U H-F CHANNELSFigure 4HARMONICS OF <strong>THE</strong> AMATEUR BANDSShown are the harmoni~ frequency ranges of the amateur bancls between 7 one/ 54 Me., with theTV ~hannels (one/ TV 1-f systems) which are most likely to receive interference from these harmomcs.Uncler certain conclitions amateur signals in the 1.8 one/ 3.5 Me. bancls can cause interferenceas a result of clirect pickup in the vic/eo systems of TV receivers which are not aclequatelyshielclecl.First it is well to become familiar with theTV channels presently assigned, with the TVintermediate frequencies commonly used, andwith the channels which will receive interferencefrom harmonics of the various amateurbands. Figures 4 and 5 give this information.Even a short inspection of figures 4 and 5will make obvious the seriousness of the interferencewhich can be caused by harmonicsof amateur signals in the higher frequencybands. With any sort of reasonable precautionsin the design and shielding of the transmitterit is not likely that harmonics higher than the6th will be encountered. Hence the main offendersin the way of harmonic interferencewill be those bands above 14-Mc.Nature ofHarmonic InterferenceInvestigations into thenature of the interferencec au s e d by amateursignals on the TV screen, assuming thatblocking has been eliminated as describedearlier in this chapter, have revealed the followingfacts:1. An unmodulated carrier, such as a c-wsignal with the key down or an AM signalwithout modulation, will give across-hatch or herringbone pattern onthe TV screen. This same general typeof picture also will occur in the case ofa narrow-band FM signal either with orwithout modulation.2. A relatively strong AM signal will givein addition to the herringbone a veryserious succession of light and darkbands across the TV picture.3. A moderate strength c-w signal withouttransients, in the absence of overloadingof the TV set, will result merely inthe turning on and off of the herringboneon the picture.To discuss condition (1) above, the herringboneis a result of the beat note between theTV video carrier and the amateur harmonic.Hence the higher the beat note the less obviouswill be the resulting cross-hatch. Further,it has been shown that a much strongersignal is required to produce a discernibleherringbone when the interfering harmonic isas far away as possible from the video carrier,without running into the sound carrier.Thus, as a last resort, or to eliminate the lastvestige of interference after all correctivemeasures have been taken, operate the transmitteron a frequency such that the interfer-
<strong>HANDBOOK</strong>Harmonic Interference 375VIDEOIt:~I TV I1 cHANNEL 1I@ II66r TV I!CHANNELlI@)IIILOW BANDSOLJ.ND~ ~~ N"' "' "'"' ·" "':.., "'"' "'"' "'_
376 TV and Broadcast Interference<strong>THE</strong> R AD I 0relatively low powered and adequately shielded,it will be found that the interference dropsmaterially when the antenna is removed and adummy load substituted. It will also be foundin such an average case that the interferencewill stop when the exciter only is operating.TransmitterPower LevelIt should be made clear at thispoint that the level of powerused at the transmitter is not ofgreat significance in the basic harmonic reductionproblem. The difference in power levelbetween a 20-watt transmitter and one ratedat a kilowatt is only a matter of about 17 db.Yet the degree of harmonic attenuation requiredto eliminate interference caused byharmonic radiation is from 80 to 120 db, dependingupon the TV signal strength in thevicinity. This is not to say that it is not asimpler job to eliminate harmonic interferencefrom a low-power transmitter than from a kilo·waa equipment. It is simpler to suppress harmonicradiation from a low-power transmittersimply because it is a much easier problem toshield a low-power unit, and the filters for theleads which enter the transmitter enclosuremay be constructed less expensively andsmaller for a low-power unit.19-3 Low-Pass FiltersAfter the transmitter has been shielded, andall power leads have been filtered in such amanner that the transmitter shielding has notbeen rendered ineffective, the only remainingavailable exit for harmonic energy lies in theantenna transmission line. Hence the mainburden of harmonic attenuation will fall on thelow-pass filter installed between the outputof the transmitter and the antenna system.Experience has shown that the low-passfilter can best be installed externally to themain transmitter enclosure, and that the transmissionline from the transmitter to the lowpassfilter should be of the coax i a 1 type.Hence the majority of low-pass filters are designedfor a characteristic impedance of 52ohms, so that RG-8/U cable (or RG-58/U fora small transmitter) may be used between theoutput of the transmitter and the antenna transmissionline or the antenna tuner.Transmitting-type low-pass filters for amateuruse usually are designed in su1:h a manneras to pass frequencies up to about 30 Me.without attenuation. The nominal cutoff fre·quency of the filters is usually between 38and 45 Me., and m·derived sections with maxi·mum attenuation in channel 2 usually are included.Well-designed filters capable of carryingany power level up to one k i 1 ow at t are®Figure 7LOW·PASS FILTER SCHEMATIC DIAGRAMSThe filter illustrated at (A) uses m·derived terminating half sections at eachencl, with three constant-k mid-sections.The filter at (B) is essentially the someexcept that the center section has beenchanged to act as an m·clerivecl sectionwhich can be designee/ to offer maximumattenuation to channels 2, 4, S, or 6 inaccorclance with the constants given below.Cutoff frequency is 45 Me. in allcases. All coils, except L4 in (B) above,ore wound ~ 11 i.d. with 8 turns per inch.The (A) FilterC 1,C, -41.5!-L!-Lfd. (40 1-L/Lfd. will be found suit·able.)c;, c,, c.-136 1-L!-Lfd. (130 to 140 1-L/Lfd. may beused.)L 1, L 6 -0.2 /Lh; 3 Y.z t. no. 14L 2, Ls -0.3 1-Lh; 5 t. no. 12L 3 , L 4 ,-0.37 !-Lh; 6 Y.z t. no. 12The (B) Filter with Mid-Section tuned to Channel2 (58 Me .. )c,, Cs -41.5 1-Lfd.c;. c 4 -136 1-L!-Lfd.C,-87 1-L!-Lfd. (50 1-LI-'fd. fixed and 75 1-LI-'fd. vari·able in parallel.)L 1 , L 7 -0.2 1-Lh; 3 Y.z t. no. 14L2, L,, Ls, L,-0.3 !-Lh; 5 t. no. 12L 4 -0.09 !-Lh; 2 t. no. 14 \l 11 dia. by ~ 11 longThe (B) Filter with Mid-Section tuned to Channel4 (71 Me.). All components same except that:C.-106 1-L!-Lfd.L 3 , L 5 -0.33 /Lh; 6 t. no. 12L.-0.05 /Lh; 1 Y.z t. no. 14, 318 11 dla. by 3/8 11long.The (B) Filter with Mid-Section tuned to Channel5 (81 Me.). Change the following:C,-113 1-L!-Lfd.L 3, L 5 -0.34 ,Uh; 6 t. no. 12L 4 -0.033 ,Uh; 1 t. no. 14 3/8 11 dia.The (B) Filter with Mid-Section tuned to Channel6 (86 Me.). All components are essentiallythe some except that the theoretical value ofL 4 Is changed to 0.03 ,uh., and the capac ita neeof C. Is changed to 117 ,U,Ufd.
<strong>HANDBOOK</strong>Low Pass Filters 377available commercially from several manufacturers.Alternatively, filters in kit form areavailable from sever a 1 manufacturers at asomewhat lower price. Effective filters maybe home constructed, if the test equipment isavailable and if sufficient care is taken in theconstruction of the assembly.Construction of Figures 7, 8 and 9 illustrateLow·Pass Filters high-performance low-passfilters which are suitablefor home construction. All are constructed inslip-cover aluminum boxes (ICA no. 29110)with dimensions of 17 by 3 by 2% inches. Fivealuminum baffle plates have been installed inthe chassis to make six shielded sectionswithin the enclosure. Feed-through bushingsbetween the shielded sections are Johnson no.135-55.Both the (A) and (B) filter types are designedfor a nominal cut-off frequency of 45Me., with a frequency of maximum rejectionat about 57 Me. as established by the terminatinghalf-sections at each end. Characteristicimpedance is 52 ohms in all cases. Thealternative filter designs diagrammed in figure7B have provision for an additional rejectiontrap in the center of the filter unit which maybe designed to offer maximum r e j e c t ion inchannel 2, 4, 5, or 6, depending upon whichchannel is likely to be received in the area inquestion. The only components which must bechanged when changing the frequency of themaximum rejection notch in the center of thefilter unit are inductors L 3 , L 4 , and L 5 , andcapacitor C 3 • A trimmer capacitor has been includedas a portion of C 3 so that the frequencyof maximum rejection can be tuned accuratelyto the desired value. Reference to figures 5and 6 will show the amateur bands which areFigure 8PHOTOGRAPH OF <strong>THE</strong> (B) FILTER WITH<strong>THE</strong> COVER IN PLACEmost likely to cause interference to specificTV channels.Either high-power or low-power componentsmay be used in the filters diagrammed in figure7. With the small Centralab TCZ zero-coefficientceramic capacitors used in the f i 1 t e runits of figure 7 A or figure 7B, power levelsup to 200 watts output may be used withoutdanger of damage to the capacitors, providedthe filter is feeding a 52-ohm resistive load.It may be practicable to use higher levels ofpower with this type of ceramic capacitor inthe filter, but at a power level of 200 watts onthe 28-Mc. band the capacitors run just perceptiblywarm to the touch. As a point of interest,it is the current rating which is of significanceto the capacitors used in fi 1 t e r ssuch as illustrated. Since current ratings forsmall capacitors such as these are not readilyavailable, it is not possible to establish anaccurate power rating for such a unit. Thehigh-power unit illustrated in figure 9, whichuses Centralab type 850S and 854S capacitors,Figure 9PHOTOGRAPH OF <strong>THE</strong> (B) FILTER WITH COVER REMOVEDThe mic/-section in this filter is ocljustecl for maximum rejection of channel 4. Note that the maincoils of the filter are mountec/ at an angle of about 45 clegrees so that there will be minimuminductive coupling from one settion to the next through the holes in the aluminum partitions.Mounting the coifs in this mann"' was founc/ to give a m"asurab/., improv.,ment in th" attenuationcharacteristics of the filter.
378 TV and Broadcast Interference<strong>THE</strong> R AD I 0has proven quite suitable for power levels upto one kilowatt.Capacitors C., C 2 , C 4 , and C 5 can be standardmanufactured units with normal 5 per centtolerance. The coils for the end sections canbe wound to the dimensions given (L, L 6 , andL 7). Then the resonant frequency of the seriesresonant end sections should be checked witha grid-dip meter, after the adjacent input oroutput terminal has been shorted with a veryshort lead. The coils should be squeezed orspread until resonance occurs at 57 Me.The intermediate m-derived section in thefilter of figure 7B may also be checked with agrid-dip meter for resonance at the correct rejectionfrequency, after the hot end of L 4 hasbeen temporarily grounded with a low-induct·ance lead. The variable capacitor portion ofC 3 can be tuned until resonance at the correctfrequency has been obtained. Note that thereis so little difference between the constantsof this intermediate section for channels 5 and6 that variation in the setting of C, will tuneto either channel without materially changingthe operation of the filter.The coils in the intermediate sections ofthe filter (L 2 , L,, L 4 , and L 5 in figure 7 A, andL 2 , L 3 , L 5 , and L 6 in figure 7B) may be checkedmost conveniently outside the filter unit withthe aid of a small ceramic capacitor of knownvalue and a grid-dip meter. The ceramic capacitoris paralleled across the small coilwith the shortest possible leads. Then the assemblyis placed atop a cardboard box and theresonant frequency checked with a grid-dipmeter. A Shure reactance slide rule may beused to ascertain the correct resonant frequencyfor the desired L-C combination and thecoil altered until the desired resonant frequencyis attained. The coil may then be installedin the filter unit, making sure that it is notsqueezed or compressed as it is being installed.However, if the coils are wound exactlyas given under figure 10, the filter maybe assembled with reasonable assurance that ·it will operate as designed.Using Low-PassFiltersThe low-pass f i 1 t e r connectedin the output transmissionline of the transmitteris capable of affording an enormous degreeof harmonic attenuation. However, thefilter must be operated in the correct manneror the results obtained will not be up to expectations.In the first place, all direct radiation fromthe transmitter and its control and power leadsmust be suppressed. This subject has beendiscussed in the previous section. Secondly,the filter must be operated into a load impedanceapproximately equal to its design characteristicimpedance. The filter itself willFigure 10SCHEMATIC OF <strong>THE</strong> SINGLE-SECTIONHALF·WAVE FILTERThe constants given below are for a char·acteristic impedance of 52 ohms, for use withRG-8/U ancl RG-58/U cable. Coil L 1 shoulclbe checlcecl for resonance at the operatingfrequency with C z, ancl the some with L2 anclC4. This check can be macle by soldering alow-inductance grounding strap to the leaclbetween Lz ancl L2 where it passes throughthe shielcl. When the coils have been trimmeclto resonance with a gricl-clip meter, thegrounding strap shoulcl of course be removecl.This filter type will give an attenuation ofabout 30 clb to the secane/ harmonic, about48 clb to the thircl, about 60 clb to the fourth,67 to the fifth, ancl so on increasing at arate of about 30 clb per octave.G,C:z,~,C4 -Silver mica or small ceramic for lowpower, transmitting type ceramic for high power.Capacitance for different bands is given below:160 meters-1700 f.J.f.J.fd.80 meters-850 f.J.f.J.fd.40 meters-440 f.J.f.J.fd.20 meters-220 f.J.f.J.fd.10 meters-110 f.J.f.J.fd.6 meters-60 f.J.f.J.fd.L 1 ,L 2 -May be made up of sections of B&W Mini·ductor for power levels below 250 watts, or ofno. 12 enam. for power up to one kilowatt. Approximatedimensions for the coils are givenbelow, but the coils should be trimmed to resonateat the proper frequency with a grid-dip meteras discussed above. All coils except theones for 160 meters are wound 8 turns per inch.160 meters-4.2 f.J.h; 22 turns no. 16 enam., 1 11dla. 2 11 long80 meters-2.1 f.J.h; 13 t. 1 11 dla. (No. 3014 Mini·ductor or no. 12)40 meters-1.1 tJ.h; 8 t. 1 11 dla. (No. 3014 or no.12 at 8 t.p.i.)20 meters-0.55 !Lh; 7 t.• % 11 dla. (No. 3010 orno. 12 at 8 t.p.l.)10 meters-0.3 tJ.h; 6 t. ~ 11 dla. (No. 3002 orno. 12 at 8 t.p.l.)6 meters-0.17 tJ.h; 4 t. ~ 11 dla. (No. 3002 orno. 12 at 8 t.p.i.)have very low losses (usually less than 0.5db) when operated into its nominal value of resistiveload. But if the filter is mis-terminatedits losses will become excessive, and it willnot present the correct value of load impedanceto the transmitter.If a f i 1 t e r, being fed from a high-powertransmitter, is operated into an incorrect terminationit may be damaged; the coils may becwerheated and the capacitors destroyed as aresult of excessive r-f currents. Hence it iswise, when first installing a low-pass filter,
<strong>HANDBOOK</strong>Broadcast Interference 379Figure 11HALF-WAVE FILTERFOR <strong>THE</strong> 28-MC. BANDShowing one possible typeof construction of a 52-ohmhalf-wave filter for relative•ly low power operation onthe 28-Mc. bane/.to check the standing-wave ratio of the loadbeing presented to the output of the filter witha standing-wave meter of any of the conventionaltypes. Then the antenna termination orthe antenna coupled should be adjusted, withlow power on the transmitter, until the s.w.r.of the load being presented to the filter isless than 2.0, and preferably below 1.5.Half-Wave Filters Half-wave f i 1 t e r s ("Hacmonikers")have been discussedin various publications including theNov.-Dec. 1949 GE Ham News. Such filtersare relatively simple and offer the advantagethat they present the same value of impedanceat their input terminals as appears as loadacross their output terminals. Such filters normallyare used as one-band affairs, and theyoffer high attenuation only to the third andhigher harmonics. Design data on the halfwavefilter is given in figure 10. Constructionof half-wave filters is illustrated in figure 11.19-4 BroadcastInterferenceInterference to the reception of signals inthe broadcast band (540 to 1600 kc.) or in theFM broadcast band (88 to 108 Me.) by amateurtransmissions is a serious matter to thoseamateurs living in densely populated areas.Although broadcast interference has recentlybeen overshadowed by the seriousness of televisioninterference, the condition of BCI isstill present.In general, signals from a transmitter operatingproperly are not picked up by receiverstuned to other frequencies unless the receiveris of inferior design, or is in poor condition.Therefore, if the receiver is of good designand is in good repair, the burden of rectifyingthe trouble rests with the owner of the interferingstation. Phone and c-w stations bothare capable of causing broadcast interference,key-click annoyance from the code transmittersbeing particularly objectionable.A knowledge of each of the several typesof broadcast interference, their cause, andmethods of eliminating them is necessary forthe successful disposition of this trouble. Aneffective method of combating one variety ofinterference is often of no value whatever inthe correction of another type. Broadcast interferenceseldom can be cured by "rule ofthumb" procedure.Broadcast interference, as covered in thissection refers primarily to standard (amplitudemodulated, 550-1600 kc.) broadcast. Interferencewith FM broadcast reception is muchless common, due to the wide separation infrequency between the FM broadcast band andthe more popular amateur bands, and due alsoto the limiting action which exists in all typesof FM receivers. Occasional interference withFM broadcast by a harmonic of an amateurtransmitter has been reported; if this conditionis encountered, it may be eliminated bythe procedures discussed in the first portionof this chapter under Television Interference.The use of frequency-modulation transmissionby an amateur station is likely to resultin much less interference to broadcast receptionthan either amplitude-modulated telephonyor straight keyed c.w. This is true because,insofar as the broadcast receiver is concerned,the amateur FM transmission will consist of aplain unmodulated carrier. There will be nokey clicks or voice reception picked up bythe b-c-1 set (unless it happens to be an FMreceiver which might pick up a harmonic ofthe signal), although there might be a slightclick when the transmitter is put on or taken
380 TV and Broadcast Interference<strong>THE</strong> R AD I 0BROADCASTRECEIVER®AGBROADCASTRECEIVERBROADCASTRECEIVERFigure 12WAVE-TRAP CIRCUITS®The circuit at (A) is the most common or·rongement, but the circuit at (B) may giveimproved results under certain conditions.Manufactured wove traps for the desired bondof operation may be purchased or the trapsmay be assembled from the data given infigure !4.off the air. This is one reason why narrowbandFM has become so popular with phoneenthusiasts who reside in densely populatedareas.InterferenceClassificationsDepending upon whether it istraceable directly to causeswithin the station or withinthe receiver, broadcast interference may bedivided into two main classes. For example,that type of interference due to transmitterover-modulation is at once listed as beingcaused by improper operation, while an inter·fering signal that tunes in and out with abroadcast station is probably an indication ofcross modulation or image response in the receiver,and the poorl y·designed input stageof the receiver is held liable. The varioustypes of interference and recommended cureswill be discussed in the following paragraphs.Blanketing This is not a tunable effect, buta total blocking of the receiver.A more or less complete "washout" coversthe entire receiver range when the carrier isswitched on. This produces either a completeblotting out of all broadcast stations, or elseknocks down their volume several decibelsdependingupon the severity of the interference.Voice modulation of the carrier causingthe blanketing will be highly distorted or evenFigure 13HIGH-ATTENUATION WAVE- TRAPCIRCUITThe two circuits may be tuned to the somefrequency for highest attenuation of a strongsigna/, or the two traps may be tuned separatelyfor different bonds of operation.unintelligible. Keying of the carrier whichproduces the blanketing will cause an annoy·ing fluctuation in the volume of the broadcastsignals.Blanketing generally occurs in the immediateneighborhood (inductive field) of a powerfultransmitter, the affected area being directlyproportional to the power of the transmitter.Also it is more prevalent with trans·mitters which operate in the 160-meter andSO-meter bands, as compared to those on thehigher frequencies.The remedies are to (1) shorten the receiv·ing antenna and thereby shift its resonant fre·quency, or ( 2) remove it to the interior of thebuilding, (3) change the direction of eitherthe receiving or transmitting antenna to minimizetheir mutual coupling, or ( 4) keep theinterfering signal from entering the receiverinput circuit by installing a wavetrap tunedto the signal frequency (see figure 12) or alow-pass filter as shown in figure 21.A suitable wave-trap is quite simple in con·struction, consisting only of a coil and midgetvariable capacitor. When the trap circuit istuned to the frequency of the interfering signal,little of the interfering voltage reachesthe grid of the first tube. Commercially manu·factured wave-traps are available from severalconcerns, including the J. W. Miller Co. inLos Angeles. However, the majority of amateursprefer to construct the traps from sparecomponents selected from the "junk box."The circuit shown in figure 13 is particu·larly effective because it consists of twotraps. The shunt trap blocks or rejects thefrequency to which it is tuned, w hi 1 e theseries trap across the antenna and ground terminalsof the receiver provides a very low im·pedance path to ground at the frequency to
<strong>HANDBOOK</strong>Wavetraps381BAND COIL, L CAPACITOR, C1.8 Me. inch no. 30 enam. 75-JJ.JJ.fd. var.closewound on I" form3.5 Me. 42 turns no. 30 enam. 50-jJ.jJ.fd. var.closewound on 1 11form7.0 Me. 23 turns no. 24 enam. 50-JJ.JJ.fd. var.closewound on 1" form14 Me. 10 turns no. 24 enam. 50-JJ.JJ.fd. var.closewound on 1 11form21 Me. 7 turns no. 24 enam. 50-""fd. var.closewound on 1" form28 Me. 4 turns no. 24 enam. 25-JJ.JJ.fd. var.closewound on 1" formso Me. 3 turns no. 24 enam. 25-jJ.jJ.fd. var.spaced v2u on 1" formFigure 14COIL AND CAPACITOR TABLE FORAMATEUR-BAND WAVETRAPSAGFigure 15BROADCASTRECEIVERMODIFICATION OF <strong>THE</strong> FIGURE 13CIRCUITIn this circuit arrangement the parallel-tunecltank is inductively coup/eel to the ante~na/eacl with a 3 to 6 turn link instead of berngplacecl directly in series with the antennaleacl.which it is tuned and by-passes the signal toground. In moderate interference cases, eitherthe shunt or series trap may be used alone,while similarly, one trap may be tuned to oneof the frequencies of the interfering transmitterand the other trap to a different interferingfrequency. In either case, each trap iseffective over but a small frequency rangeand must be readjusted for other frequencies.The wave-trap must be installed as closeto the receiver antenna terminal as practicable,hence it should be as small in size aspossible. The variable capacitor may be amidget air-tuned trimmer type, and the coilmay be wound on a l-inch dia. form. The tableof figure 14 gives winding data for wave-trapsbuilt around standard variable capacitors. Forbest results, both a shunt and a series trapshould be employed as shown.Figure 15 shows a two-circuit coupledwave-trap that is somewhat sharper in tuningand more efficacious. The specifications forthe secondary coil L, may be obtained fromthe table of figure 14. The primary coil of theshunt trap consists of 3 to 5 closewound turnsof the same size wire wound in the same directionon the same form as L 1and separatedfrom the latter by 'Ia of an inch.Overmodulation A carrier modulated in excessof 100 per cent acquiressharp cutoff periods which give rise to transients.These transients create a broad signaland generate spurious responses. Transientscaused by overmodulation of a radio-telephonesignal may at the same time bring about impactor shock excitation of nearby receivingantennas and power lines, generating interferingsignals in that manner.Broadcast interference due to overmodulationis frequently encountered. The remedy isto reduce the modulation percentage or to usea clipper-filter system or a high-level splattersuppressor in the speech circuit of the transmitter.CrossModulationCross modulation or cross talk ischaracterized by the amateur signalriding in on top of a strongbroadcast signal. There is usually no heterodynenote, the amateur signal being tuned inand out with the program carriers.This effect is due frequently to a faulty inputstage in the affected receiver. Modulationof the interfering carrier will swing the operatingpoint of the input tub e. This type oftrouble is seldom experienced when a variable-ptube is used in the input stage.Where the receiver is too ancient to incorporatesuch a tube, and is probably poorlyshielded at the same time, it will be better toattach a wave-trap of the type shown in figure12 rather than to attempt rebuilding of the receiver.The addition of a good ground and ashield can over the input tube often adds tothe effectiveness of the wave-trap.Transmission viaCapacitive CouplingA s m a 11 amount of capacitivecoupling is nowwidely used in receiverr.f. and antenna transformers as a gain boosterat the high-frequency end of the tuning range.The coupling capacitance is obtained bymeans of a small loop of wire cemented closeto the grid end of the secondary winding, withone end directly connected to the plate or antennaend of the primary winding. (See figure16.)
38 2 TV and Broadcast Interference<strong>THE</strong> R AD I 0AVCFigure 16CAPACITIVE BOOST COUPLINGCIRCUITSuch circuits, lncluclec/ within the broac/castreceiver to bring up the stage gain at thehigh-frequency encl of the tuning range, havea tenclency to Increase the susceptibility ofthe receiver to interference From amateurbane/transmissions.It is easily seen that a small capacitor atthis position will favor the coupling of thehigher frequencies. This type of capacitivecoupling in the receiver co i 1 s will tend topass amateur high-frequency signals into a receivertuned to broadcast frequencies.The amount of capacitive coupling may bereduced to eliminate interference by movingthe coupling turn further away from the secondarycoil. However, a simple wave-trap ofthe type shown in figure 12, inserted at theantenna input terminal, will generally accomplishthe same result and is more to be recommendedthan reducing the amount of capacitivecoupling (which lowers the receiver gainat the high-frequency end of the broadcastband). Should the wave-trap alone not suffice,it will be necessary to resort to a reductionin the coupling capacitance.In some simple broadcast receivers, capacitivecoupling is obtained by closely coupledprimary and secondary coils, or as a result ofrunning a long primary or antenna lead closeto the secondary coil of an unshielded antennacoupler.Phantoms With two strong local carriers appliedto a non-linear impedance,the beat note resulting from cross-modulationbetween them may fall on some frequencywithin the broadcast band and will be audibleat that point. If such a "phantom" signal fallson a local broadcast frequency, there will beheterodyne interference as well. This is acommon occurence with broadcast receivers inthe neighborhood of two amateur stations, oran amateur and a police station. It also sometimesoccurs when only one of the stations islocated in the immediate vicinity.As an example: an amateur signal on 3514kc. might beat with a local 2414-kc. policecarrier to produce a 1100-kc. phantom. If thetwo carriers are strong enough in the vicinityof a circuit which can cause rectification, the1100-kc. phantom will be heard in the broadcastband. A poor contact between two oxidizedwires can produce rectification.Two stations must be transmitting simultaneouslyto produce a phantom signal; wheneither station go e s off the air the phantomdisappears. Hence, this type of interferenceis apt to be reported as highly intermittent andmight be difficult to duplicate unless a testoscillator is used "on location" to simulatethe missing station. Such interference cannotbe remedied at the transmitter, and often therectification takes place some distance fromthe receivers. In such occurrences it is mostdifficult to locate the source of the trouble.It will also be apparent that a ph an tommight fall on the intermediate frequency of asimple superhet receiver and cause interferenceof the untunable variety if the manufacturerhas not provided an i-f wave-trap in theantenna circuit.This particular t y p e of phantom m a y, inaddition to causing i-f interference, generateharmonics which may be tuned in and out withheterodyne whistles from one end of the receiverdial to the other. It is in this mannerthat birdies often result from the operation ofnearby amateur stations.When one component of a phantom is asteady, unmodulated carrier, only the intelligencepresent on the other carrier is conveyedto the broadcast receiver.Phantom signals almost always may beidentified by the suddenness with which theyare interrupted, signalizing withdrawal of oneparty to the union. This is especially bafflingto the inexperienced interference-locater, whoobserves that the interference suddenly disappears,even though his own transmitter remainsin operation.If the mixing or rectification is taking placein the receiver itself, a phantom signal maybe eliminated by removing either one of thecontributing signals from the receiver inputcircuit. A wave-trap of the type shown in figure12, tuned to either signal, w i 11 do thetrick. If the rectification is taking place outsidethe receiver, the wave-trap should betuned to the frequency of the phantom, insteadof to one of its components. 1-f wave-trapsmay be built around a 2.5-millihenry r-f chokeas the inductor, and a compression-type micapadding capacitor. The capacitor should havea capacitance range of 250-525 Jlp.fd. for the175- and 206-kc. intermediate frequencies;65-175 JLJLfd. for 260-kc. and other intermedi-
<strong>HANDBOOK</strong>Auto Rectification383ates lying between 250- and 400-kc; and17-80 ppfd. for 456-, 465-, 495-, and 500-kc.Slightly more capacitance will be required forresonance with a 2.1 millihenry choke.SpuriousEmissionsThis sort of interference arisesfrom the transmitter itself. Theradiation of any signal (other thanthe intended carrier frequency) by an amateurstation is prohibited by FCC regulations. Spuriousradiation may be traced to imperfect neutralization,parasitic oscillations in the r-f ormodulator stages, or to "broadcast-band"variable-frequency oscillators or e.c.o.'s.Low-frequency parasitics may actually occuron broadcast frequencies or their near subharmonics,causing direct interference to programs.An all-wave monitor operated in thevicinity of the transmitter will detect thesespurious signals.The remedy will be obvious in individualcases. Elsewhere in this book are discussedmethods of complete neutralization and thesuppression of parasitic oscillations in r-fand audio stages.A-c/d-c Receivers Inexpensive tab 1 e- modela-c/ d-e receivers are particularlysusceptible to interference from amateurtransmissions. In fact, it may be saidwith a fair degree of assurance that the majorityof BCI encountered by amateurs operatingin the 1.8-Mc. to 29-Mc. range is a result ofthese inexpensive receivers. In most casesthe receivers are at fault; but this does notabsolve the amateur of his responsibility inattempting to eliminate the interference.Stray ReceiverRectificationIn most cases of interferenceto inexpensive receivers, particularlythose of the a-c/ d-etype, it will be found that stray receiver rectificationis causing the trouble. The offendingstage usually will be found to be a high-mutriode as the first audio stage following thesecond detector. Tubes of this type are quitenon-linear in their grid characteristic, andhence will readily rectify any r-f signal appearingbetween grid and cathode. The r-f signalmay get to the tube as a result of directsignal pickup due to the lack of shielding, butmore commonly will be fed to the tube fromthe power line as a result of the series heaterstring.The remedy for this condition is simply toinsure that the cathode and grid of the high-muaudio tube (usually a 12SQ7 or equivalent) areat the same r-f potential. This is accomplishedby placing an r-f by-pass capacitor with theshortest possible leads directly from grid tocathode, and then adding an impedance in thelead from the volume control to the grid of the~2 TO fOMEG.®Figure 17CIRCUITS FOR ELIMINATING AUDIOSTAGE RECTIFICATIONaudio tube. The impedance may be an amateurband r-f choke (such as a National R-100U)for best results, but for a majority of casesit will be found that a 47,000-ohm Yz-watt resistorin series with this lead will give satisfactoryoperation. Suitable circuits for suchan operation on the receiver are given in figure17.In many a.c.-d.c. receivers there is no r-fby-pass included across the plate supply rectifierfor the set. If there is an appreciablelevel of r-f signal on the power line feedingthe receiver, r-f rectification in the powerrectifier of the receiver can cause a particularlybad type of interference which may bereceived on other broadcast receivers in thevicinity in addition to the one causing therectification. The soldering of a 0.0 1-JLfd. discceramic capacitor directly from anode to cathodeof the power rectifier (whether it is of thevacuum-tube or selenium-rectifier type) usuallywill by-pass the r-f signal across the rectifierand thus eliminate the difficulty."Floating" VolumeControl ShaftsSeveral sets have beenencountered where therewas only a slightly interferingsignal; but, upon placing one's hand upto the volume control, the signal would greatlyincrease. Investigation revealed that thevolume control was installed with its shaftinsulated from ground. The control itself wasconnected to a critical part of a circuit, inmany instances to the grid of a high-gain audiostage. The cure is to install a volume controlwith all the terminals insulated from theshaft, and then to ground the shaft.
384 TV and Broadcast Interference<strong>THE</strong> R AD I 0BANDCOIL, LCAPACITOR, CMETAL 60X~------~i-~~-,~~~~~S~H~IE~~~O~B~R~A~IO~~~----+•3.5 Me.7.0 Me.17 turnsno. 14 enameled 100-tt.ufd.variablel-inch diameterl-1j4-inch length11 turnsno. 14 enameled2Vrinch 100-JL.ufd. variablediameter1 V2 -inch length4 turns14 and no. 10 enameled 1 00-JLJLfd. variable21 Me. 3-inch diameter1 Va-inch length3 turns27 and 1j4-inch o.d.28 Me. copper tubing 100-,u.ufd. variable'2.-inch diameterl-inch lengthTO A. C.LINEccITO TRANSMITTE.ROR RECEIVER~------~~~~~~~~~SH=I~E=~~O=B~R=A~IO~~~~--·L--}-JFigure 19RESONANT POWER-LINEWAVE-TRAP CIRCUITThe resonant type of power-line filter ismore effective than the more conventional"brute force .. type of I ine filter, but requirestuning to the operating frequency of thetransmitter.Power-LinePickupFigure 18COIL AND CAPACITOR TABLEFORA-CLINE TRAPSW h en radio-frequency energyfrom a radio transmitter enters abroadcast receiver through thea-c power lines, it has either been fed backinto the lighting system by the offending transmitter,or picked up from the air by over-headpower lines. Underground lines are seldom responsiblefor spreading this interference.To check the path whereby the interferingsignals reach the line, it is only necessary toreplace the transmitting antenna with a dummyantenna and adjust the transmitter for maximumoutput. If the interference then ceases,overhead lines have been picking up the energy.The trouble can be cleared up by installinga wave-trap or a commercial line filter inthe power lines at the receiver. If the receiveris reasonably close to the transmitter, it isvery doubtful that changing the direction ofthe transmitting antenna to right angles withthe overhead lines will eliminate the trouble.If, on the contrary, the interference continueswhen the transmitter is connected tothe dummy antenna, radio-frequency energy isbeing fed directly into the power line by thetransmitter, and the station must be inspectedto determine the cause.One of the following reasons for the troublewill usually be found: ( 1) the r-f stages arenot sufficiently bypassed and/ or choked, (2)the antenna coupling system is not performingefficiently, (3) the power transformers haveno electrostatic shields; or, if shields are present,they are ungrounded, (4) power lines arerunning too close to an antenna or r-f circuitscarrying high currents. If none of these causesapply, wave-traps must be installed in thepower lines at the transmitter to remove r-fenergy passing back into the lighting system.The wave-traps used in the power lines attransmitter or receiver must be cap a b 1 e ofpassing relatively high current. The coils areaccordingly wound with heavy wire. Figure 18lists the specifications for power line wavetrapcoils, while figure 19 illustrates the methodof connecting these wave-traps. Observethat these traps are enclosed in a shield boxof heavy iron or steel, well grounded.All-WoveReceiversEach complete-coverage home receiveris a potential source of annoyanceto the transmitting amateur.The novice short-wave broadcast-listenerwho tunes in an amateur station often considersit an interfering signal, and complainsaccordingly.Neither selectivity nor image rejection inmost of these sets is comparable to thoseproperties in a communication receiver. Theresult is that an amateur signal will occupytoo much dial space and appear at more thanone point, giving rise to interference on adjacentchannels and distant channels as well.If carrier-frequency harmonics are presentin the amateur transmission, serious interferencewill result at the all-wave receiver. Theharmonics may, if the carrier frequency hasbeen so unfortunately chosen, fall directlyupon a favorite short-wave broadcast stationand arouse warranted objection.The amateur is apt to be blamed, too, fortransmissions for which he is not responsible,so great is the public ignorance of short-waveallocations and signals. Owners of all-wavereceivers have been quick to ascribe to amat-eurstations all signals they hear from tapemachines and V-wheels, as well as stray tonesand heterodyne flutters.
<strong>HANDBOOK</strong>lmaqe Interference 38 5The amateur cannot be held responsiblewhen his carrier is deliberately tuned in onan all-wave receiver. Neither is he account·able for the width of his signal on the receiverdial, or for the strength of image repeat points,if it can be proven that the receiver designdoes not afford good selectivity and image re·jection.If he so desires, the amateur (or the ownerof the receiver) might sharpen up the receivedsignal somewhat by shortening the receivingantenna. Set retailers often supply quite asizeable antenna with all-wave receivers, butmost of the time these sets perform almost aswell with a few feet of inside antenna.The amateur is accountable for harmonicsof his carrier frequency. Such emissions areunlawful in the first place, and he must takeall steps necessary to their suppression. Prac·tical suggestions for the elimination of har·monies have been given earlier in this chap·ter under Television Interference.Image Interference In addition to those typesof interference aIr e ad ydiscussed, there are two more which are com·mon to superhet receivers. The prevalence ofthese types is of great concern to the ama·teur, although the responsibility for their ex·istence more properly rests with the broadcastreceiver.The mechanism whereby image productiontakes place may be explained in the followingmanner: when the first detector is set to thefrequency of an incoming signal, the high-fre·quency oscillator is operating on another fre·quency which differs from the signal by thenumber of kilocycles of the intermediate fre·quency. Now, with the setting of these twostages undisturbed, there is another signalwhich will beat with the high-frequency oscillatorto produce an i-f signal. This other signalis the so-called image, which is separatedfrom the desired signal by twice the inter·mediate frequency.Thus, in a receiver with 175-kc. i.f., tunedto 1000 kc.: the h·f oscillator is operating on1175 kc., and a signal on 1350 kc. (1000 kc.plus 2 x 175 kc.) will beat with this 1175 kc.oscillator frequency to produce the 175-kc. i-fsignal. Similarly, when the same receiver istuned to 1400 kc., an amateur signal on 1750kc. can come through.If the image appears only a few cycles orkilocycles from a broadcast carrier, heterodyneinterference will be present as well. Other·wise, it will be tuned in and out in the mannerof a station operating in the broadcast band.Sharpness of tuning will be comparable to thatof broadcast stations producing the same a-v·cvoltage at the receiver.The second variety of superhet interferenceis the result of harmonics of the receiver h-foscillator beating with amateur carriers to pro·duce the intermediate frequency of the receiv·er. The amateur transmitter will always befound to be on a frequency equal to some har·monic of the receiver h·f oscillator, plus orminus the intermediate frequency.As an example: when a broadcast superhetwith 465-kc. i.f. is tuned to 1000 kc., its highfrequencyoscillator operates on 1465 kc. Thethird harmonic of this oscillator frequency is4395 kc., which will beat with an amateur signalon 3930 kc. to send a signal through thei-f amplifier. The 3930 kc. signal would betuned in at the 1000-kc. point on the dial.Some oscillator harmonics are so related toamateur frequencies that more than one pointof interference will occur on the receiver dial.Thus, a 3500-kc. signal may be tuned in atsix points on the dial of a nearby broadcastsuperhet having 175 kc. i.f. and no r-f stage.Insofar as remedies for image and harmonicsuperhet interference are concerned, it is wellto remember that if the amateur signal did notin the first place reach the input stage of thereceiver, the annoyance would not have beencreated. It is therefore good policy to try toeliminate it by means of a wave-trap or low·pass filter. Broadcast superhets are not alwaysthe acme of good shielding, however,and the amateur signal is apt to enter the cir·cuit through channels other than the input circuit.If a wave-trap or filter will not cure thetrouble, the only alternative will be to attemptCONSTANT K TYPE:-H:UTo-J.___J_oM- DERIVED TYPEFigure 20TYPES OF LOW-PASS FILTERSFilters such as these may be used in thecircuit between the antenna and the inputof the receiver.
3 8 6 TV and Broadcast InterferenceANT. L1 L2~TO RECEIVER ANT. POSTI' !2 C3 !40 0 TO RECEIVER ~NO. POSTGND.Figure 21COMPOSITE LOW-PASS FILTERCIRCUITThis filter is highly effective in reducingbroadcast interference from all high frequencystations, one/ requires no tuning. Constantsfor 400 ohm terminal impedance and1600 kc. cutoff ore os follows: Lt, 65 turnsno. 22 d. c. c. c/osewound on 1!1 in. clio. form.L2, 41 turns ditto, not coupled to Lt. Ct,250 !l+'fd. fixed mica capacitor. c2. 400 !l+'fd.fixed mica capacitor. c3 and c4. 150 !l+'fd.fixed mica capacitors, former of 5% tolerance.With some receivers, better resultswill be obtained with o 200 ohm carbon re•sistor inserted between the filter and on·tenno post on the rece1"ver, With other receiversthe effectiveness will be improvedwith o 600 ohm carbon resistor placed fromthe antenna post to the ground post on thereceiver. The filter s h o u I d be placed asclose to the receiver terminals as possible.to select a transmitter frequency such thatneither image nor harmonic interference will beset up on favorite stations in the susceptiblereceivers. The equation given earlier may beused to determine the proper frequencies.Low Pass Filters The greatest drawback ofthe wave-trap is the factthat it is a single-frequency device; i.e.-itmay be set to reject at one time only one fre·quency (or, at best, an extremely narrow bandof frequencies). Each time the frequency ofthe interfering transmitter is changed, everywave-trap tuned to it must be retuned. A muchmore satisfactory device is the wave filterwhich requires no cuning. One type, the low·pass filter, passes all frequencies below onecritical frequency, and eliminates all higherfrequencies. It is this property that makes thedevice ideal for the task of removing amateurfrequencies from broadcast receivers.A good low-pass filter designed for maxi·mum attenuation around 1700 kc. will passall broadcast carriers, but will reject signalsoriginating in any amateur band. Naturallysuch a device should be installed only instandard broadcast receivers, never in allwavesets.Two types of low-pass filter sections areshown in figure 20. A composite arrangementcomprising a section of each type is moreeffective than either type operating alone. Acomposite filter composed of one K·sectionand one shunt-derived M·section is shown infigure 21, and is highly recommended. TheM·section is designed to have maximum atten·uation at 1700 kc., and for that reason C,should be of the "close tolerance" variety.Likewise, C, should not be stuffed down insideL 2 in the interest of compactness, asthis will alter the inductance of the coil appre·ciably, and likewise the resonant frequency.If a fixed 150 /-(f!fd. mica capacitor of 5 percent tolerance is not available for C., a com·pression trimmer covering the range of 125-175 /-(f!fd. may be substituted and adjusted togive maximum attenuation at about 1700 kc.19-5 HI-FI InterferenceThe rapid growth of high-fidelity sound systemsin the home has brought about many cases ofinterference from a nearby amateur transmitter.In most cases, the interference is caused bystray pickup of the r·f signal by the interconnect·ing leads of the hi·fi system and audio rectifi·cation in the 'low level stages of the amplifier.The solution to this difficulty, in general, is tobypass and filter all speaker and power leadsto the hi-fi amplifier and preamplifier. A com·bination of a VHF choke and 5 00 f!f!Id ceramicdisc capacitors in each power and speaker leadwill eliminate r·f pickup in the high level sectionof the amplifier. A filter such as shown in figure17 A placed in the input circuit of the first audiostage of the preamplifier will reduce the level ofthe r-f signal reaching the input circuit of theamplifier. To prevent loss of the higher audiofrequencies it may be necessary to decrease thevalue of the grid bypass capacitor to 50 f!f!Idor so.Shielded leads should be employed between theamplifier and the turntable or f-m tuner. Theshield should be. grounded at both ends of theline to the chassis of the equipment, and careshould be taken to see that the line does notapproach an electrical half-wave length of theradio signal causing the interference. In someinstances, shielding the power cable to the hi·fiequipment will aid in reducing interference. Theframework of the phonograph turntable should begrounded to the chassis of the amplifier to reducestray r-f pickup in the turntable equipment.
CHAPTER TWENTYTransmitter Keying and Control20-1 Power SystemsIt is probable that the average amateur stationthat has been in operation for a numberof years will have at least two transmittersavailable for operation on different frequencybands, at least two receivers or one receiverand a converter, at least one item of monitoringor frequency measuring equipment andprobably two, a v.f.o., a speech amplifier, adesk light, and a clock. In addition to theabove 8 or 10 items, there must be an outletavailable for a soldering iron and there shouldbe one or two additional outlets available forplugging in one or two pieces of equipmentwhich are being worked upon.It thus becomes obvious that 10 or 12 outletsconnected to the 115-volt a-c line shouldbe available at the operating desk. It may bepracticable to have this number of outlets installedas an outlet strip along the baseboardat the time a new home is being planned andconstructed. Or it might be well to installthe outlet strip on the operating desk so asto have the flexibility of moving the operatingdesk from one position to another. Alternatively,the outlet strip might be wallmountedjust below the desk top.Power DrainPer OutletWhen the power drain of all theitems of equipment, other thantransmitters, used at the operatingposition is totalled, you probably willfind that 350 to 600 watts will be required.387Since the usual home outlet is designed tohandle only about 600 watts maximum, thetransmitter, unless it is of relatively low power,should be powered from another source. Thisprocedure is desirable in any event so that thevoltage supplied to the receiver, frequency control,and frequency monitor will be substantiallyconstant with the transmitter on or offthe air.So we come to two general alternative planswith their variations. Plan (A) is the more desirableand also the most expensive since itinvolves the installation of two separate linesfrom the meter box to the operating positioneither when the house is constructed or as analteration. One line, with its switch, is for thetransmitters and the other line and switch isfor receivers and auxiliary equipment. Plan(B) is the more practicable for the average amateur,but its use requires that all cords be removedfrom the outlets whenever the stationis not in use in order to comply with the electricalcodes.Figure 1 shows a suggested arrangement forcarrying out Plan (A). In most cases an installationsuch as this will require approval of theplans by the city or county electrical inspector.Then the installation itself will also requireinspection after it has been completed. It willbe necessary to use approved outlet boxes atthe rear of the transmitter where the cable isconnected, and also at the operating benchwhere the other BX cable connects to the outletstrip. Also, the connectors at the rear ofthe transmitter will have to be of an approved
388 Transmitter Keying and Control<strong>THE</strong> R AD I 0SHORT CORDS FROMRECEIVE~ V.f.O .. CLOCK6~~~E:~~T~~~-ETC. TO IFROM LINEPLAN®PLAN @Figure 1<strong>THE</strong> PLAN {A) POWER SYSTEMA-c line power from the main fuse box in thehouse is run separately to the receivingequipment ancl to the transmitting equipment.Separate switches ancl fuse blacks then areavailable for the transmitters ancl for theauxiliary equipment. Since the fuses in theboxes at the operating room will be in serieswith those at the main fuse box, those in theoperating room shaulcl have a lower ratingthan those at the main fuse box. Then it willalways be possible to replace blown fuseswithout leaving the operating room. The fuseboxes can conveniently be lacatecl alongsideone another on the wall of the operating room.Figure 2<strong>THE</strong> PLAN {B) POWER SYSTEMThis system is less convenient than the (A}system, hut does not require extensive re ..wiring of the electrical system within thehouse to accommoclate the a"angement. Thusit is better lor a temporary or semi .. permanentinstallation. In most cases it will be necessaryto run an extra conduit from the mainfuse box to the outlet from which the trans·mitter is powered, since the stone/arc/ arrangementin most houses is to run all the outletsin one room (ancl sometimes all in the house}from a single pair of fuses anclleacls.type. It is possible also that the BX cable willhave to be permanently affixed to the trans·mitter with the connector at the fuse-box end.These details may be worked out in advancewith the electrical inspector for your area.The general aspects of Plan (B) are shownin figure 2. The basic difference between thetwo plans is that (A) represents a permanentinstallation even though a degree of mobilityis allowed through the use of BX for powerleads, while plan (B) is definitely a temporarytype of installation as far as the electrical inspectoris concerned. While it will be permis·sible in most areas to leave the transmittercord plugged into the outlet even though it isturned off, the Fire Insurance Underwriterscodes will make it necessary that the cordwhich runs to the group of outlets at the backof the operating desk be removed whenever theequipment is not actually in use.Whether the general aspects of plans (A) or(B) are used it will be necessary to run a numberof control wires, keying and audio leads,and an excitation cable from the operating deskto the transmitter. Control and keying wirescan best be grouped into a multiple-wire rubber·covered cable between the desk and the trans·mitter. Such an arrangement gives a good ap·pearance, and is particularly practical if cableconnectors are used at each end. High-levelaudio at a moderate impedance level (600 ohmsor below) may be run in the same control cableas the other leads. However, low-level audiocan best be run in a small coaxial cable. Smallcoaxial cable such as RG-58/U or RG-59/Ualso is quite satisfactory and quite convenientfor the signal from the v .f.o. to the r-f stagesin the transmitter. Coaxial-cable connectors ofthe UG series are quite satisfactory for theterminations both for the v-f-o lead and for anylow-level audio cables.Checking anOutlet with aHeavy LoadTo make sure that an outlet willstand the full load of the entiretransmitter, plug in an electricheater rated at about 50 per centgreater wattage than the power you expect todraw from the line. If the line voltage does
<strong>HANDBOOK</strong>Transmitter Control 389not drop more than 5 volts (assuming a 117-volt line) under load and the wiring does notoverheat, the wiring is adequate to supply thetransmitter. About 600 watts total drain is themaximum that should be drawn from a 117-voltlighting outlet or circuit. For greater power,a separate pair of heavy conductors should berun right from the meter box. For a 1-kw. phonetransmitter the total drain is so great that a230-volt "split" system ordinarily will be required.Most of the newer homes are wired withthis system, as are homes utilizing electricityfor cooking and heating.With a three-wire system, be sure there isno fuse in the neutral wire at the fuse box. Aneutral fuse is not required if both "hot" legsare fused, and, should a neutral fuse blow,there is a chance that damage to the radiotransmitter will result.If you have a high power transmitter and doa lot of operating, it is a good idea to checkon your local power rates if you are on astraight lighting rate. In some cities a lowerrate can be obtained (but with a higher "minimum")if electrical equipment such as anelectric heater drawing a specified amountof current is permanently wired in. It is notrequired that you use this equipment, merelythat it be permanently wired into the electricalsystem. Naturally, however, there would be nosaving unless you expect to occupy the samedwelling for a considerable length of time.Outlet Strips The outlet strips which havebeen suggested for installationin the baseboard or for use on the rear of a deskare obtainable from the large electrical supplyhouses. If such a house is not in the vicinityit is probable that a local electrical contractorcan order a suitable type of strip from one ofthe supply house catalogs. These strips arequite convenient in that they are available invarying lengths with provision for insertinga-c line plugs throughout their length. Thea-c plugs from the various items of equipmenton the operating desk then may be insertedin the outlet strip throughout its length. Inmany cases it will be desirable to reduce theequipment cord lengths so that they will plugneatly into the outlet strip without an excessto dangle behind the desk.Contactors andRelaysThe use of power-control contactorsand relays often willadd considerably to the operatingconvenience of the station installation.The most practicable arrangement usually isto have a main a-c line switch on the front ofthe transmitter to apply power to the filamenttransformers and to the power control circuits.It also will be found quite convenient to havea single a-c line switch on the operating deskto energize or cut the power from the outletstrip on the rear of the operating desk. Throughthe use of such a switch it is not necessary toremember to switch off a large number of separateswitches on each of the items of equipmenton the operating desk. The alternativearrangement, and that which is approved by theUnderwriters, is to remove the plugs from thewall both for the transmitter and for the operating-deskoutlet strip when a period of operationhas been completed.While the insertion of plugs or operation ofswitches usually will be found best for applyingthe a-c line power to the equipment, thechanging over between transmit and receivecan best be accomplished through the use ofrelays. Such a system usually involves threerelays, or three groups of relays. The relaysand their functions are: (1) power control relayfor the transmitter-applies 115-volt line to theprimary of the high-voltage transformer andturns on the exciter; ( 2) control relay for thereceiver-makes the receiver inoperative byany one of a number of methods when closed,also may apply power to the v.f.o. and to akeying or a phone monitor; and (3) the antennachangeover relay-connects the antenna to thetransmitter when the transmitter is energizedand to the receiver when the transmitter is notoperating. Several circuits illustrating the applicationof relays to such control arrangementsare discussed in the paragraphs to follow inthis chapter.Controlling TransmitterPower OutputIt is necessary, inorder to comply withFCC regulations, thattransmitter power output be limited to the minimumamount necessary to sustain communication.This requirement may be met in severalways. Many amateurs have two transmitters;one is capable of relatively high power outputfor use when calling, or when interference issevere, and the other is capable of considerablyless power output. In many cases thelower powered transmitter acts as the exciterfor the higher powered stage when full poweroutput is required. But the majority of the amateursusing a high powered equipment havesome provision for reducing the plate voltageon the high-level stages when reduced poweroutput is desired.One of the most common arrangements forobtaining two levels of power output involvesthe use of a plate transformer having a doubleprimary for the high-voltage power supply. Themajority of the high-power plate transformersof standard manufacture have just such a dualprimaryarrangement. The two primaries aredesigned for use with either a 115-volt or 230-volt line. When such a transformer is to beoperated from a 115-volt line, operation of both
390 Transmitter Keying and Control<strong>THE</strong> R AD I 0s~~
<strong>HANDBOOK</strong>Transmitter Control 3 91115V.A.C.LINETO EXCITER POWER SUPPLIESTOH.V.POWER SUPPLY,--------,, III~~cci'~ 1INTERLOCKSIN TRANSMITTER]}}TRANSMITTERFILAMENTTRANSFORMERSTO FILAMENTTRANSFORMERSFigure 4EXTERNAL VARIACOR POWERSTATCIRCUIT WITH VARIABLE-RATIOAUTO-TRANSFORMERWhen the dummy plug is inserted into the re·ceptocle on the equipment, closing of thepower control relay will apply full voltageto the primaries. With the cab I e from theVariac or Powerstot plugged into the socketthe voltage output of the high-voltage powersupply may be varied from zero to about l 5per cent above normal.Figure 5PROTECTIVE CONTROL CIRCUITWith this circuit arrangement either switchmay be closed first to light the heaters ofall tubes and the filament pilot light. Thenwhen the second switch is closed the highvoltage will be oppl ied to the transmitter andthe red pilot will light. With a 30-second deloybetween the closing of the first switchand the closing of the second, the rectifiertubes will be adequately protected. Similarly,the opening of either swItch will removeplate voltage from the rectifiers while theheaters remain lighted.One convenient arrangement for using aVariac or Powerstat in conjunction with thehigh-voltage transformer of a transmitter isillustrated in figure 4. In this circuit a heavythree-wire cable is run from a plug on the trans·mitter to the Variac or Powerstat. The Variacor Powerstat then is installed so that it is ac·cessible from the operating desk so that theinput power to the transmitter may be con·trolled during operation. If desired, the cableto the Variac or Powerstat may be unpluggedfrom the transmitter and a dummy plug insertedin its place. With the dummy plug in place thetransmitter will operate at normal plate voltage.This arrangement allows the transmitter to bewired in such a manner that an external Variacor Powerstat may be used if desired, eventhough the unit is not available at the timethat the transmitter is constructed.Notes on the Useof the Variacor PowerstatPlate voltage to the modula·tors may be controlled at thesame time as the plate voltageto the final amplifier isvaried if the modulator stage uses beam tetrodetubes; variation it;t the plate voltage on suchtubes used as modulators causes only a moderatechange in the standing plate current.Since the final amplifier plate voltage is beingcontrolled simultaneously with the modulatorplate voltage, the conditions of impedancematch will not be serious! y upset. In severalhigh power transmitters using this system, andusing beam-tetrode modulator tubes, it is pos·sible to vary the plate input from about 50watts to one kilowatt without a change otherthan a slight increase in audio distortion atthe adjustment which gives the lowest poweroutput from the transmitter.With triode tubes as modulators it usuallywill be found necessary to vary the grid biasat the same time that the plate voltage ischanged. This will allow the tubes to be op·erated at approximately the same relative pointon their operating characteristic when the platevoltage is varied. When the modulator tubes areoperated with zero bias at full plate voltage, itwill usually be possible to reduce the modu·lator voltage along with the voltage on the.modulated stage, with no apparent change inthe voice quality. However, it will be necessaryto reduce the audio gain ar the same time thatthe plate voltage is reduced.20-2 TransmitterControl MethodsAlmost everyone, when getting a new trans·
392 Transmitter Keying and Control<strong>THE</strong> R AD I 0ft~ VOLT SUPPLY FORENTIRE TRANSMITTER51 HUSKY TOC.GLE SWITCHON TRANSMITTERr------QRECEIVER POWER,------oTRANSFORMER C.T.PROTECTIVEINTERLOCKSTRANSMITRECEIVE SWITCHr1r1INDICATOR LIGHTSALL FILAMENT TRANSFORMERS e.3v. EXCITER H.V.TRANSFORMERTUNE-UPSWITCHFigure 6TRANSMITTER CONTROL CIRCUITClosing S1 lights all filaments in the transmitter one/ starts the time-c/elay relay in its cycle.When the time-c/elay relay has operatecl, closing the transmit-receive switch at the operating po·sition will apply plate power to the transmitter one/ c/isable the receiver. A tune-up switch hasbeen p r o vi c/ e c/ so that the exciter stages may be tune c/ without p~ate voltage on the finalamplifier.mitter on the air, has had the experience ofhaving to throw several switches and pull orinsert a few plugs when changing from receiveto transmit. This is one extreme in the directionof how not to control a transmitter. At theother extreme we find systems where it is onlynecessary to speak into the microphone ortouch the key to change both transmitter andreceiver over to the transmit condition. Mostamateur stations are intermediate between thetwo extremes in the control provisions and usesome relatively simple system for transmittercontrol.In figure 5 is shown an arrangement whichprotects mercury-vapor rectifiers against prematureapplication of plate voltage withoutresorting to a time-delay relay. No matter whichswitch is thrown first, the filaments will beturned on first and off last. However, doublepoleswitches are required in place of the usualsingle-pole switches.When assured time delay of the proper intervaland greater operating convenience are desired,a group of inexpensive a·c relays maybe incorporated into the circuit to give a con·trol circuit such as is shown in figure 6. Thisarrangement uses a llS·volt thermal (or motor·operated) time-delay relay and a d-p-d·t ll5·volt control relay. Note that the protectiveinterlocks are connected in series with thecoil of the relay which applies high voltage tothe transmitter. A tune-up switch has been in·eluded so that the transmitter may be tuned upas far as the grid circuit of the final stage isconcerned before application of high voltageto the final amplifier. Provisions for operat·ing an antenna-changeover relay and for cut·ring the plate voltage to the receiver when thetransmitter is operating have been included.A circuit similar to that of figure 6 but incorporatingpush-button control of the trans·mitter is shown in figure 7. The circuit featuresa set of START-STOP and TRANSMIT-RECEIVE buttons at the transmitter and a separateset at the operating position. The controlpush buttons operate independently so thateither set may be used to control the trans·mitter. It is only necessary to push the START
<strong>HANDBOOK</strong>Safety Precautions3 9311~ VOLT SUPPLY FORENTIRE TRANSMITTER,---- -ATOPERAriNGPOSlTION-IISTOPTRANSMITI.LLI---,RECEIVEIIIPROTECTIVEINTERLOCKSOVERLOAD)'RELAYCONTACTS~~t-----1--------------t-----cHd~:N~N;J~:AQ--1~---i----~RELAYALL FILAMENT TRANSFORMERSEXCITER H.V.TRANSFORMERHIGH VOLTAGETRANSFORM£ RFigure 7PUSH-BUTTON TRANSMITTER-CONTROL CIRCUITPushing the START button either at the transmitter or at the operating position will light allfilaments one/ start the time-cleloy reI a y in its cycle. When the c y c I e has been completed, atouch of the TRANSMIT button will put the transmitter on the air one/ eli sable the receiver. Push·ing the RECEIVE button will clisoble the transmitter one/ restore the receiver. Pushing the STOPbutton will instantly clrop the entire transmitter from the o·c line. If clesirecl, a switch may beplocecl in series with the leocl from the RECEIVE button to the protective interlocks; openingthe switch will make it impossible for any person occiclentolly to put the transmitter on the air.Various other safety provisions, such as the protective-interlock arrangement clescribecl in thetext have been incorporated.With the circuit arrangement shown lor the overloacl-relay contacts, it is only necessary to usea simple normolly-closecl cl-c relay with a variable shunt across the coil of the relay. When thecurrent through the coil becomes great enough to open the normo/ly-closecl contacts the holclcircuiton the plate-voltage relay will be broken one/ the plate voltage will be removecl. If theoverloacl is only momentary, such as a moclulot/on peak or u tonk flashover, merely pushing theTRANSMIT button will again put the transmitter on the air. This simple circuit provision elimi·nates the requirement for expensive overloocl relays of the mechanically-latching type, but stillgives excellent overloocl protection.button momentarily to light the transmitter filamentsand start the time-delay relay in its cy·de. When the standby light comes on it is onlynecessary to touch the TRANSMIT button toput the transmitter on the air and disable thereceiver. Touching the RECEIVE button willturn off the transmitter and restore the receiver.After a period of operation it is only necessaryto touch the STOP button at either the trat\s·mitter or the operating position to shut downthe transmitter. This type of control arrange·ment is called an electrically-locking push-to·transmit control system. Such systems are fre·quently used in industrial electronic control.20-3 Safety PrecautionsThe best way for an operator to avoid ser·ious accidents from the high voltage suppliesof a transmitter is for him to use his head, actonly with deliberation, and not take unneces-
394 Transmitter Keying and Control<strong>THE</strong> R AD I 0sary chances. However, no one is infallible,and chances of an accident are greatly lessenedif certain factors are taken into considerationin the design of a transmitter, in orderto protect the operator in the event of a lapseof caution. If there are too many things onemust "watch out for" or keep in mind there isa good chance that sooner or later there willbe a mishap; and it only takes one. When designingor constructing a transmitter, the followingsafety considerations should be givenattention.Grounds For the utmost in protection, everythingof metal on the front panel ofa transmitter capable of being touched by theoperator should be at ground potential. Thisincludes dial set screws, meter zero adjusterscrews, meter cases if of metal, meter jacks,everything of metal protruding through the frontpanel or capable of being touched or nearlytouched by the operator. This applies whetheror not the panel itself is of metal. Do not relyupon the insulation of meter cases or tuningknobs for protection.The B negative or chassis of all plate powersupplies should be connected together, and toan external ground such as a waterpipe.Exposed Wiresand ComponentsIt is not necessary to resortto rack and panel constructionin order to provide completeenclosure of all components and wiringof the transmitter. Even with metal-chassisconstruction it is possible to arrange things soas to incorporate a protective shielding housingwhich will not interfere with ventilationyet will prevent contact with all wires andcomponents carrying high voltage d.c. or a.c.,in addition to offering shielding action.If everything on the front panel is at groundpotential (with respect to external ground) andall units are effectively housed with protectivecovers, then there is no danger except whenthe operator must reach into the interior partof the transmitter, as when changing coils,neutralizing, adjusting coupling, or shootingtrouble. The latter procedure can be made safeby making it possible for the operator to beabsolutely certain that all voltages have beenturned off and that they cannot be turned oneither by short circuit or accident. This can bedone by incorporation of the following systemof main primary switch and safety signal lights.Combined SafetySignal and SwitchThe common method ofusing red pilot lights toshow when a circuit is onis useless except from an ornamental standpoint.When the red pilot is not lit it usuallymeans that the circuit is turned off, but it canLJ riL. TRANS.MAIN 11~ v. suPPLY1..--{D.P.O.T. SWITCH------------------~Figure 8COMBINED MAIN SWITCH ANDSAFETY SIGNALWhen shutting clown the transmitter, throwthe main switch to neutral. If work is to beclone on the transmiHer, throw the switch allthe way to "pilot," thus turning on the greenpilot lights on the panel ana on each chassis,one/ insuring that no voltage can existon the primary of any transformer, even byvirtue of a short or acciclental grouncJ.mean that the circuit is on but the lamp isburned out or not making contact.To enable you to touch the tank coils inyour transmitter with absolute assurance thatit is impossible for you to obtain a shock exceptfrom possible undischarged filter condensers(see following topic for elimination ofthis hazard), it is only necessary to incorpo·rate a device similar to that of figure 8. It isplaced near the point where the main 110-voltleads enter the room (preferably near the door)and in such a position as to be inaccessibleto small children. Notice that this switch breaksboth leads; switches that open just one leaddo not afford complete protection, as it issometimes possible to complete a primary circuitthrough a short or accidental ground. Break·ing just one side of the line may be all rightfor turning the transmitter on and off, but whenyou are going to place an arm inside the trans·mitter, both 110-volt leads should be broken.When you are all through working your transmitterfor the time being, simply throw themain switch to neutral.When you find it necessary to work on thetransmitter or change coils, throw the switchso that the green pilots light up. These can beordinary 6. 3-volt pilot lamps behind greenbezels or dipped in green lacquer. One shouldbe placed on the front panel of the transmitter;others should be placed so as to be easilyvisible when changing coils or making adjust·ments requiring the operator to reach insidethe transmitter.
<strong>HANDBOOK</strong>Transmitter Keying 395For 100 per cent protection, just obey thefollowing rule: never work on the transmitteror reach inside any protective cover exceptwhen the green pilots are glowing. To avoidconfusion, no other green pilots should be usedon the transmitter; if you want an indicatorjewel to show when the filaments are lighted,use amber instead of green.Safety Bleeders Filter capacitors of good qualityhold their charge for sometime, and when the voltage is more than 1000volts it is just about as dangerous to get acrossan undischarged 4-p.fd. filter capacitor as it isto get across a high-voltage supply that isturned on. Most power supplies incorporatebleeders to improve regulation, but as theseare generally wire-wound resistors, and aswire-wound resistors occasionally open upwithout apparent cause, it is desirable to incorporatean auxiliary safety bleeder acrosseach heavy-duty bleeder. Carbon resistors willnot stand much dissipation and sometimeschange in value slightly with age. However,the chance of their opening up when run wellwithin their dissipation rating is very small.To make sure that all capacitors are bled, itis best to short each one with an insulatedscrewdriver. However, this is sometimes awkwardand al-ways inconvenient. One can be virtuallysure by connecting auxiliary carbonbleeders across all wire-wound bleeders usedon supplies of 1000 volts or more. For every500 volts, connect in series a 500,000-ohm1-watt carbon resistor. The drain will be negligible(l rna.) and each resistor will have todissipate only 0. 5 watt. Under these conditionsthe resistors will last indefinitely withlittle chance of opening up. For a 1500-voltsupply, connect three 500,000-ohm resistors inseries. If the voltage exceeds an integral numberof 500 volt divisions, assume it is the nexthigher integral value; for instance, assume1800 volts as 2000 volts and use four resistors.Do not attempt to use fewer resistors byusing a higher value for the resistors; not over500 volts should appear across any single1-watt resistor.In the event that the regular bleeder opensup, it will take several seconds for the auxiliarybleeder to drain the capacitor~ down to asafe voltage, because of the very high resistance.Therefore, it ill best to allow 10 or 15seconds after turning off the plate supply beforeattempting to work on the t'ransmitter.If a 0-1 d-e milliammeter is at hand, it maybe connected in series with the auxiliarybleeder to act as a high voltage voltmeter."Hot" AdjustmentsSome amateurs contendthat it is almost impossibleto make certain adjustments, such as couplingand neutralizing, unless the transmitter is running.The best. thing to do is to make all neutralizingand coupling devices adjustable fromthe front panel by means of flexible controlshafts which are broken with insulated couplingsto permit grounding of the panel bearing.If your particular transmitter layout is suchthat this is impracticable and you refuse tothrow the main switch to make an adjustment-throw the main switch-take a reading-throwthe main switch-make an adjustment-and soon, then protect yourself by making use of longadjusting rods made from ~-inch dowel stickswhich have been wiped with oil when perfectlyfree from moisture.If you are addicted to the use of pickup loopand flashlight bulb as a resonance and neutralizingindicator, then fasten it to the end of along dowel stick and use it in that manner.Protective lnoterlocks With the increasing tendencytoward constructionof transmitters in enclosed steel cabinetsa transmitter becomes a particularly lethal deviceunless adequate safety provisions havebeen incorporated. Even with a combined safetysignal and switch as shown in figure 8 it isstill conceivable that some person unfamiliarwith the transmitter could come in contact withhigh voltage. It is therefore recommended thatthe transmitter, wherever possible, be builtinto a complete metal housing or cabinet andthat all doors or access covers be providedwith protective interlocks (all interlocks mustbe connected in series) to remove the highvoltage whenever these doors or covers areopened. The term "high voltage" should meanany voltage above approximately 150 volts,although it is still possible to obtain a seriousburn from a 150-volt circuit under certain circumstances.The 150-volt limit usually willmean that grid-bias packs as well as highvoltagepacks should have their primary circuitsopened when any interlock is opened.20-4 Transmitter KeyingThe carrier from a c-w telegraph transmittermust be broken into dots and dashes for thetransmission of code characters. The carriersignal is of constant amplitude while the keyis closed, and is entirely removed when thekey is open. When code characters are beingtransmitted, the carrier may be considered asbeing modulated by the keying. If the changefrom the no-output condition to full-output, orvice versa, occurs too rapidly, the rectangularpulses which form the keying characters containhigh-frequency components which take up
396 Transmitter Keying and Control<strong>THE</strong> R AD I 0a wide frequency band as sidebands and areheard as clicks.The cure for transient key clicks is relativelysimple, although one would not believeit, judging from the hordes of dicky, "snappy"signals heard on the air.To be capable of transmitting code charactersand at the same time not splitting theeardrums of neighboring amateurs, the c-wtransmitter MUST meet two important specifications.1- It must have no parasitic oscillationseither in the stage being keyed or in anysucceeding stage.2- It must have some device in the keyingcircuit capable of shaping the leadingand trailing edge of the waveform.Both these specifications must be met beforethe transmitter is capable of c-w operation.Mere! y turning a transmitter on and offby the haphazard insertion of a telegraph keyin some power lead is an invitation to trouble.The two general methods of keying a transmitterare those which control the excitationto the keyed amplifier, and those which controlthe plate or screen voltage applied to thekeyed amplifier.Key-ClickEliminationKey-click elimination is ac-complishedby preventing a too-rapidmake-and-break of power to theantenna circuit, rounding off the keying charactersso as to limit the sidebands to a valuewhich does not cause interference to adjacentchannels. Too much lag will prevent fast keying,but fortunate! y key clicks can be practicallyeliminated without limiting the speedof manual (hand) keying. Some circuits whicheliminate key clicks introduce too much timelagand thereby add tails to the dots. Thesetails may cause the signals to be difficult tocopy at high speeds.Location ofKeyed StogeConsiderable thought should begiven as to which stage in atransmitter is the proper one tokey. If the transmitter is keyed in a stage closeto the oscillator, the change in r-f loading ofthe oscillator will cause the oscillator to shiftfrequency with keying. This will cause thesignal to have a distinct chirp. The chirp willbe multiplied. as many times as the frequencyof the oscillator is multiplied. A chirpy oscillatorthat would be passable on 80 meterswould be unusable on 28 Me. c.w.Keying the oscillator itself is an excellentway to run into keying difficulties. If no keyclick filter is used in the keying circuit, thetransmitter will have bad key clicks. If a keyclick filter is used, the slow rise and decayof oscillator voltage induced by the filter actionwill cause a keying chirp. This action isFigure 9CENTER-TAP KEYING WITH CLICKFILTERThe constants shown obove are suggested asstarting values; considerable variation inthese values can be expected for optimumkeying of amplifiers of different operatingconditions. It is suggested that a keying relaybe substituted for the key in the circuitabove wherever practicable.true of all oscillators, whether electron coupledor crystal controlled.The more amplifier or doubler stages thatfollow the keyed stage:;, the more difficult it isto hold control of the shape of the keyed waveform.A heavily excited doubler stage or classC stage acts as a peak clipper, tending tosquare up a rounded keying impulse, and thecumulative effect of several such stages cascadedis sufficient to square up the keyedwaveform to the point where bad clicks arereimposed on a clean signal.A good rule of thumb is to never key backfarther than one stage removed from the finalamplifier stage, and never key closer than onestage removed from the frequency controllingoscillator of the transmitter. Thus there willalways be one isolating stage between thekeyed stage and the oscillator, and one isolatingstage between the keyed stage and theantenna. At this point the waveform of thekeyed signal may be most easily controlled.Keyer CircuitRequirementsIn the first place it may be establishedthat the majority ofnew design transmitters, andmany of those of older design as well, use amedium power beam tetrode tube either as theoutput stage or as the exciter for the outputstage of a high power transmitter. Thus thetransmitter usually will end up with a tubesuch as type 2E26, 807, 6146, 813, 4-65A,4E27/257B, 4-125A or similar, or one of thesetubes will be used as the stage just ahead ofthe output stage.Second, it may be established that it is undesirableto key further down in the transmitterchain than the stage just ahead of the final
<strong>HANDBOOK</strong> Cathode Keying 3 97TOCATHODE OFKEYED STAGEESTANCOR PA8421®2.K,2WL_ ___ +H.V. SUPPLYFigure 10VACUUM TUBE KEYERS FOR CENTER-TAP KEYING CIRCUITSThe type A keyer is suitable far keying stages running up to 1250 volts on the plate. Two 2A3or 6A3 tubes can safely key 160 milliamperes of cathode current. The simple 6Y6 keyer in fig·ure B is for keying stages running up to 650 volts on the plate. A single 6Y6 can key 80 milliamperes.Two in parallel may be used for plate currents under 160 ma. If softer keying is clesirecl,the 500-!"-!"-fcl. mica condenser should be increased to .001 !"'fcl.amplifier. If a low-level stage, which is fol·lowed by a series of class C .amplifiers, iskeyed, serious transients will be generatedin the output of the transmitter even thoughthe keyed stage is being turned on and off verysmoothly. This condition arises as a result ofpulse sharpening, which has been discussedpreviously.Third, the output from the stage should becompletely cut off when the key is up, and thetime constant of the rise and decay of the key·ing wave should be easily controllable.Fourth, it should be possible to make therise period and the decay period of the keyingwave approximately equal. This type of keyingenvelope is the only one tolerable for commer·cial work, and is equally desirable for obtain·ing clean cut and easily readable signals inamateur work.Fifth, it is desirable that the keying circuitbe usable without a keying relay, even whena high-power stage is being keyed.Last, for the sake of simplicity and safety,it should be possible to ground the frame ofthe key, and yet the circuit should be suchthat placing the fingers across the key willnot result in an electrical shock. In other words,the keying circuit should be inherently safe.All these requirements have been met in thekeying circuits to be described.20-5 Cathode KeyingThe lead from the cathode or center-tap con·nection of the filament of an r·f amplifier canbe opened and closed for a keying circuit. Sucha keying system opens the plate voltage cir·cuit and at the same time opens the grid biasreturn lead. For this reason, the grid circuitis blocked at the same time the plate circuitis opened. This helps to reduce the backwavethat might otherwise leak through the keyedstage.The simplest cathode keying circuit is il·lustrated in figure 9, where a key-click filteris employed, and a hand key is used to breakthe circuit. This simple keying circuit is not
3 98 Transmitter Keying and Control<strong>THE</strong> R AD I 0PC-BLOCKINGBIAS+L.V.+H.V.Figure 11SIMPLE BLOCKED-GRID KEYINGSYSTEMThe blocking bias must be sufficient to cut·off plate current to the amplifier stage in thepresence of the excitation voltage. Rz Is nor·mal bios resistor for the tube. R2 one/ Czshould be acljustecl for correct keying wave·form.Figure 12SELF-BLOCKING KEYING SYSTEM FORHIGH-MU TRIODERz and Cz adjustecl for correct keying waveform.Rz is bias resistor of tube.recommended for general use, as considerablevoltage will be developed across the key whenit is open.An electronic switch can take the place ofthe hand key. This will remove the danger ofshock. At the same time, the opening and clos·ing characteristics of the electronic switchmay easily be altered to suit the particularneed at hand. Such an electronic switch iscalled a vacuum tube keyer. Low internal re·sistance triode tubes such as the 45, 6A3, or6AS7 are used in the keyer. These tubes actas a very high resistance when sufficientblocking bias is applied to them, and as avery low resistance when the bias is removed.The desired amount of lag or cushioning effectcan be obtained by employing suitable resist·ance and capacitance values in the grid of thekeyer tube(s). Because very little spark isproduced at the key, due to the small amountof power in the key circuit, sparking clicksare easily suppressed.One type 45 tube should be used for every50 rna. of plate current. Type 6B4G or 2A3tubes may also be used; allow one 6B4G tubefor every 80 rna. of plate current.Because of the series resistance of the keyertubes, the plate voltage at the keyed tube willbe from 30 to 60 volts less than the powersupply voltage. This voltage appears as cath·ode bias on the keyed tube, assuming the biasreturn is made to ground, and should be takeninto consideration when providing bias.Some typical cathode circuit vacuum tubekeying units are shown in figure 10.20-6 Grid Circuit KeyingFigure 13TWO-STAGE BLOCKED-GRID KEYERA separate filament transformer must be usedfor the 6J5, as its filament is at a potentialof -400 volts.Grid circuit, or blocked grid keying is anothereffective method of keying a c-w transmitter.A basic blocked grid keying circuit isshown in figure 11. The time constant of thekeying is determined by the RC circuit, whichalso forms part of the bias circuit of the tube.\\-hen the key is closed, operating bias is developedby the flow of grid current through R1.When the key is open, sufficient fixed bias isapplied to the tube to block it, preventing thestage from functioning. If an un-neutralized
<strong>HANDBOOK</strong>Screen Grid Keying 399preventing any backwave emission. The firstkeyed stage may be the oscillator, or a lowpowered buffer. The last keyed stage may bethe driver stage to the power amplifier, or theamplifier itself. Since the circuit is so proportionedthat the lower powered stage comeson first and goes off last, any keying chirp inthe oscillator is not emitted on the air. Keyinglag is applied to the high powered keyed stageonly.20-7 Screen Grid KeyingFigure 14SINGLE-STAGE SCREEN GRID KEYERFOR TETRODE TUBEStetrode is keyed by this method, there is thepossibility of a considerable backwave causedby r-f leakage through the grid-plate capacityof the tube.Certain hi-ll triode tubes, such as the 811-Aand the 805, automatically block themselveswhen the grid return circuit is opened. It ismerely necessary to insert a key and associatedkey click filter in the grid return lead ofthese tubes. No blocking bias supply is needed.This circuit is shown in figure 12.A more elaborate blocked-grid keying systemhas been developed by W1DX, and wasshown in the February, 1954 issue of QSTmagazine. This highly recommended circuitis shown in figure 13. Two stages are keyed,The screen circuit of a tetrode tube may bekeyed for c-w operation. Unfortunately, whenthe screen grid of a tetrode tube is brought tozero potential, the tube still delivers considerableoutput. Thus it is necessary to placea negative blocking voltage on the screen gridto reduce the backwave through the tube. Asuitable keyer circuit that will achieve thiswas developed by W6DTY, and was describedin the February, 1953 issue of CQ magazine.This circuit is shown in figure 14. A 6L6 isused as a combined damper tube and keyingtube. When the key is closed, the 6L6 tubehas blocking bias applied to its control grid.This bias is obtained from the rectified gridbias of the keyed tube. Screen voltage is appliedto the keyed stage through a screen droppingresistor and a VR-105 regulator tube.'W'hen the key is open, the 6L6 is no longercut-off, and conducts heavily. The voltagedrop across the dropping resistor caused bythe heavy plate current of the 6L6 lowers thevoltage on the VR-105 tube until it is extin-Figure 15TOP VIEW OF SCREENGRID KEYER SHOWN INFIGURE 16•
400 Transmitter Keying and Control <strong>THE</strong> <strong>RADIO</strong>LOW POWER BUFFER6AG7svomf~} ""'-._-110TIME-+c -110IOKlOWD 373 375E -27~ -275Figure 16TWO-STAGE SCREEN GRID KEYER UNITguished, removing the screen voltage from thetetrode r-f tube. At the same time, rectifiedgrid bias is applied to the screen of the tetrodethrough the 1 megohm resistor between screenand key. This voltage effectively cuts off thescreen of the tetrode until the key is closedagain. The RC circuit in the grid of the 6L6tube determines the keying characteristic ofthe tetrode tube.A more elaborate screen grid keyer is shownin figures 15 and 16. This keyer is designedto block-grid key the oscillator or a low poweredbuffer stage, and to screen key a mediumpowered tetrode tube such as an 807, 2E26 or6146. The unit described includes a simpledual voltage power supply for the positivescreen voltage of the tetrode, and a negativesupply for the keyer stages. A 6K6 is used asthe screen keyer, and a 12AU7 is used as acathode follower and grid block keyer. As inthe W1DX keyer, this keyer turns on the excitera moment before the tetrode stage isturned on. The tetrode stage goes off an instantbefore the exciter does. Thus any keyingchirp of the oscillator is effectively removedfrom the keyed signal.By listening in the receiver one can hearthe exciter stop operating a fraction of a secondafter the tetrode stage goes off. In fact,during rapid keying, the exciter may be heardas a steady signal in the receiver, as it hasappreciable time lag in the keying circuit. Theclipping effect of following stages has a definitehardening effect on this, however.20-8 Differential Keying CircuitsExcellent waveshaping may be obtained bya differential keying system whereby themaster oscillator of the transmitter is turned
<strong>HANDBOOK</strong>DifferentialKeying401VACUUMTUBEKEYER(F!G.lO)Figure 18I--- KEY IS DEPRESSED ---------lDURING THIS TIMEIIf.-TRANSMITTER IS "ON <strong>THE</strong>---!AIR" DURING THIS TIME.BLOCKING DIODES EMPLOYEDTO VARY TIME CONSTANT OF"MAKE"AND"BREAK"CHARAGTERISTICS OF VACUUM TUBEKEYERFigure 17TIME SEQUENCE OF ADIFFERENTIAL KEYERon a moment before the rest of the stages areenergized, and remains on a moment longerthan the other stages. The "chirp" or frequencyshift associated with abrupt switching ofthe oscillator is thus removed from the emittedsignal. In addition, the differential keyer canapply waveshaping to the amplifier sectionof the transmitter, eliminating the "click"caused by rapid keying of the latter stages.The ideal keying system would perform asillustrated in figure 17. When the key isclosed, the oscillator reaches maximum outputalmost instantaneously. The followingstages reach maximum output in a fashiondetermined by the waveshaping circuits ofthe keyer. When the key is released, the outputof the amplifier stages starts to decayin a predetermined manner, followed shortlythereafter by cessation of the oscillator. Theoverall result of these actions is to providerelatively soft "make" and "break" to thekeyed signal, meanwhile preventing oscillatorfrequency shift during the keying sequence.The rates of charge and decay in a typicalR-C keying circuit may be varied independent•ly of each other by the blocking diode systemof figure 18. Each diode permits the chargingcurrent of the timing capacitor to flow throughonly one of the two variable potentiometers,thus permitting independent adjustment ofthe "make" and "break" characteristics ofthe keying system.A practical differential keying system de-veloped by WliCP (Feb., 1956 QST) is shownin figure 19. A 6AL5 switch tube turns theoscillator on before the keying action starts,and holds it on until after the keying sequenceis completed. Time constant of thekeying cycle is determined by values of Cand R. When the key is open, a cut-off biasof about -110 volts is applied to the screengrid circuits of the keyed stages. When thekey is closed, the screen grid voltage risesto the normal value at a rate determined bythe time constant R-C. Upon opening the keyagain, the screen voltage returns to cut-offvalue at the predetermined rate.The potentiometer R 1 serves as an outputcontrol, varying the minimum internal resistanceof the 12BH7 keyer tube, and is auseful device to limit power input during tuneupperiods. Excitation to the final amplifierstage may be controlled by the screen potentiometerR3 in the second buffer stage.An external bias source of approximately -120volts at 10 milliamperes is required for operationof the keyer, in addition to the 300-voltscreen supply.Blocking voltage may be removed from theoscillator for "zeroing" purposes by closingswitch S1, rendering the diode switch inoperative.A second popular keying system is shownin figure 20, and is widely used in manyJohnson transmitters. Grid block keying isused on tubes V2 and V3. A waveshapingfilter consisting of R2, R3, and C1 is usedin the keying control circuit of V2 and V3.To avoid chirp when the oscillator (Vl) iskeyed, the keyer tube V4 allows the oscillatorto start quickly--before V2 and V3 start
402OSCILLATOR BUFFER# I BUFFER# 2.1 Kt-------~- +300 v.50 K2W-120V.ROUTPUTCONTROLR1'-----------'!; 100 K2WFigure 19DIFFERENTIAL KEYING SYSTEM WITHOSCILLATOR SWITCHING DIODEV2BUFFERV3DRIVER100 KVFO "HOLD"-sov.Figure 20DIFFERENTIAL KEYER EMPLOYED IN"JOHNSON" TRANSMITTERSconducting--and then continue operatinguntil atter V2 and V3 have stopped conducting.Potentiometer R 1 adjusts the "hold"time for VFO operation after the key is opened.This may be adjusted to cut off the VFObetween marks of keyed characters, thusallowing rapid break-in operation.
CHAPTER TWENTY-ONERadiation, Propagationand Transmission LinesRadio waves are electromagnetic wavessimilar in nature but much lower in frequencythan light waves or heat waves. Such wavesrepresent electric energy traveling throughspace. Radio waves travel in free space withthe velocity of light and can be reflected andrefracted much the same as light waves.21-1 Radiation from anAntennaAlternating current passing through a conductorcreates an alternating electromagneticfield around that conductor. Energy is alternate!y stored in the field, and then returnedto the conductor. As the frequency is raised,more and more of the energy does not returnto the conductor, but instead is radiated offinto space in the form of electromagneticwaves, called radio waves. Radiation from awire, or wires, is materially increased wheneverthere is a sudden change in the electricalconstants of the line. These sudden changesproduce reflection, which places standingwaves on the line.When a wire in space is fed radio frequencyenergy having a wavelength of approximately2.1 times the length of the wire in meters, thewire resonates as a half-wave dipole antennaat that wavelength or frequency. The greatest403possible change in the electrical constantsof a line is that which occurs at the open endof a wire. Therefore, a dipole has a great mismatchat each end, producing a high degree ofreflection. We say that the ends of a dipoleare terminated in an infinite impedance.A returning wave which has been reflectedmeets the next incident wave, and the voltageand current at any point along the antenna arethe vector sum of the two waves. At the endsof the dipole, the voltages add, while the cur·rents of the two waves cancel, thus producinghigh voltage and low current at the ends of thedipole or half wave section of wire. In thesame manner, it is found that the currents addwhile the voltages cancel at the center of thedipole. Thus, at the center there is high cur·rent but low voltage.Inspection of figure 1 will show that thecurrent in a dipole decreases sinusoidallytowards either end, while the voltage similarlyincreases. The voltages at the two ends of theantenna are 180° out of phase, which meansthat the polarities are opposite, one being pluswhile the other is minus at any instant. Acurve representing either the voltage or currenton a dipole represents a standing waveon the wire.Radiation fromSources otherthan AntennasRadiation can and does takeplace from sources other thanantennas. Undesired radiationcan take place from open-wire
4 04 Radiation, Propagation and Lines-- -,~oLTA!iE',CENTER', ----t---CURRENJ ... ""'(, I ........... ,c~RENT,.""" ' ' .......,.,"" ', ! ',,--ex»" ' ~ 'ICDI--t--HALF-wAVE ANTENNA i '', , •lSHOWIN~ HOW STANDIN!; WAVESEXIST ON A HORIZONTAL ANTENNA.CURRENT IS MAXIMUM AT CENTER~VOLTAGE IS MAXIMUM AT ENDS.Figure 1' ' ' ', 'VOLTAGE'...,--STANDING WAVES ON A RESONANTANTENNAtransmission lines, both from single-wire linesand from lines comprised of more than onewire. In addition, radiation can be made totake place in a very efficient manner from electromagnetichorns, from plastic lenses or fromelectromagnetic lenses made up of spaced conductingplanes, from slots cut in a piece ofmetal, from dielectric wires, or from the openend of a wave guide.Directivity of The radiation from any physRodiation ically practicable radiatingsystem is directive to a certaindegree. The degree of directivity can be enhancedor altered when desirable through thecombination of radiating elements in a prescribedmanner, through the use of reflectingplanes or curved surfaces, or through the useof such systems as mentioned in the precedingparagraph. The construction of directive antennaarrays is covered in detail in the chapterswhich follow.Polarization Like light waves, radio wavescan have a definite polarization.In fact, while light waves ordinarily have tobe reflected or passed through a polarizingmedium before they have a definite polarization,a radio wave leaving a simple radiatorwill have a definite polarization, the polarizationbeing indicated by the orientation ofthe electric-field component of the wave. This,in turn, is determined by the orientation of theradiator itself, as the magnetic-field componentis always at right angles to a linear radiator,and the electric-field component is always inthe same plane as the radiator. Thus we seethat an antenna that is vertical with respectto the earth will transmit a vertically polarizedwave, as the electrostatic lines of forcewill be vertical. Likewise, a simple horizontalantenna will radiate horizontally polarizedwaves.<strong>THE</strong> R AD I 0Because the orientation of a simple linearradiator is the same as the polarization of thewaves emitted by it, the radiator itself is referredto as being either vertically or horizontallypolarized. Thus, we say that a horizontalantenna is horizontally polarized.Figure 2A illustrates the fact that the polarizationof the electric field of the radiationfrom a vertical dipole is vertical. Figure 28,on the other hand, shows that the polarizationof electric-field radiation from a vertical slotradiator is horizontal. This fact has been utilizedin certain commercial F.M antennas whereit is desired to have horizontally polarizedradiation but where it is more convenient touse an array of vertically stacked slot arrays.If the metallic sheet is bent into a cylinderwith the slot on one side, substantially omnidirectionalhorizontal coverage is obtainedwith horizontally-polarized radiation when thecylinder with the slot in one side is orientedvertically. An arrangement of this type is shownin figure 2C. Several such cylinders may bestacked vertically to reduce high-angle radiationand to concentrate the radiated energyat the useful low radiation angles.In any event the polarization of radiationfrom a radiating system is parallel to the electricfield as it is set up inside or in the vicinityof the radiating system.21-2 General Characteristicsof AntennasAll antennas have certain general characteristicsto be enumerated. It is the result ofdifferences in these general characteristicswhich makes one type of antenna system mostsuitable for one type of application and anothertype best for a different application. Sixof the more important characteristics are: (1)polarization, (2) radiation resistance, (3) horizontaldirectivity, (4) vertical directivity,(5) bandwidth, and (6) effective power gain.The polarization of an antenna or radiatingsystem is the direction of the electric fieldand has been defined in Section 21-1.The radiation resistance of an antenna systemis normally referred to the feed point inan antenna fed at a current loop, or it is referredto a current loop in an antenna systemfed at another point. The radiation resistanceis that value of resistance which, if insertedin series with the antenna at a current loop,would dissipate the same energy as is actuallyradiated by the antenna if the antenna currentat the feed point were to remain the same.The horizontal and vertical directivity canbest be expressed as a directive pattern which
<strong>HANDBOOK</strong> Antenna Characteristics 405Figure 2ANTENNA POLARIZATIONThe polarization (electric fielc{} ofthe radiation from a resonant dipolesuch as shown at (A) above is parallelto the length of the radiator. Inthe case of a resonant slot cut in asheet of metal ancl usee! as a raclia·tor, the polarization (of the elec·tric fielcl) is perpendicular to thelength of the slot. In both cases,however, the polarization of theracliatecl fielcl is parallel to the po·tential gradient of the radiator; inthe case of the a/pole the electriclines of force are from end to encl,while in the case of the slot thefielcl is across the sicles of theslot. The metal/ ic sheet containingthe slot may be formed into a cylinderto make up the radiator shownat (C). With this type of radiatorthe racliatec! fielcl will be horizon•tally polarized even though theradiator is mounted vertically."'IELECTRICfiELD(POLARIZATION)lVERTICALMETALLIC SH!ET~I 10ELECTRIC FIELD(POLARIZATION)HORIZONTAL®8 "a:"'0z:::;1;IJis a graph showing the relative radiated fieldintensity against azimuth angle for horizontaldirectivity and field intensity against elevationangle for vertical directivity.The bandwidth of an antenna is a measureof its ability to operate within specified limitsover a range of frequencies. Bandwidth canbe expressed either "operating frequency plus·or-minus a specified per cent of operating fre·quency" or "operating frequency plus-or-minusa specified number of megacycles" for a cer·tain standing-wave-ratio limit on the trans·mission line feeding the antenna system.The effective power gain or directive gainof an antenna is the ratio between the powerrequired in the specified antenna and the powerrequired in a reference antenna (usually a half·wave dipole) to attain the same field strengthin the favored direction of the antenna undermeasurement. Directive gain may be expressedeither as an actual power ratio, or as is morecommon, the power ratio may be expressedin decibels.Physical Lengthof a Half-WaveAntennaIf the cross section of theconductor which makes upthe antenna is kept verysmall with respect to theantenna length, an electrical half wave is afixed percentage shorter than a physical half·wavelength. This percentage is approximately5 per cent. Therefore, most linear half-wave an·tennas are close to 95 per cent of a half wavelengthlong physically. Thus, a half-wave antennaresonant at exactly 80 meters would beone-half of 0.95 times 80 meters in length. An·other way of saying the same thing is that awire resonates at a wavelength of about 2.1times its length in meters. If the diameter ofthe conductor begins to be an appreciable fractionof a wavelength, as when tubing is usedas a v-h-f radiator, the factor becomes slightlyless than 0. 95. For the use of wire and nottubing on frequencies below 30 ~lc., however,the figure of 0.95 may be taken as accurate.This assumes a radiator removed from surroundingobjects, and with no bends.Simple conversion into feet can be obtainedby using the factor 1. 56. To find the physicallength of a half-wave 80-meter antenna, wemultiply 80 times 1. 56, and get 124.8 feet forthe length of the radiator.It is more common to use frequency thanwavelength when indicating a specific spot inthe radio spectrum. For this reason, the relationshipbetween wavelength and frequencymust be kept in mind. As the velocity of radiowaves through space is constant at the speedof light, it will be seen that the more wavesthat pass a point per second (higher frequency),the closer together the peaks of those wavesmust be (shorter wavelength). Therefore, thehigher the frequency, the lower will be thewavelength.A radio wave in space can be compared toa wave in water. The wave, in either case, haspeaks and troughs. One peak and one troughconstitute a full wave, or one wavelength.Frequency describes the number of wavecycles or peaks passing a point per second.Wavelength describes the distance the wavetravels through space during one cycle oroscillation of the antenna current; it is the
4 06 Radiation, Propagation and Lines<strong>THE</strong> R AD I 0distance in meters between adjacent peaksor adjacent troughs of a wave train.As a radio wave travels 300,000,000 metersa second (speed of light), a frequency of 1cycle per second corresponds to a wavelengthof 300,000,000 meters. So, if the frequency ismultiplied by a million, the wavelength mustbe divided by a million, in order to maintaintheir correct ratio.A frequency of 1,000,000 cycles per second(1,000 kc.) equals a wavelength of 300 meters.Multiplying frequency by 10 and dividing wavelengthby 10, we find: a frequency of 10,000 kc.equals a wavelength of 30 meters. Multiplyingand dividing by 10 again, we get: a frequencyof 100,000 kc. equals 3 meters wavelength.Therefore, to change wavelength to frequency(in kilocycles), simply divide 300,000 by thewavelength in meters (.:\).300,000Fkc =---A300,000A=--FkcNow that we have a simple conversion formulafor converting wavelength to frequencyand vice versa, we can combine it with ourwavelength versus antenna length formula, andwe have the following:Length of a half-wave radiator made fromwire (no. 14 to no. 10):3. 5-Mc. to 30-Mc. bands468Length in feet =Freq. in Me.50-Mc. band460Length in feet =Freq. in Me.Length in inches144-Mc. band5600Freq. in Me.5500Length in inches =Freq. in Me.Length·to·DiameterRatioWhen a half-wave radiatoris constructed from tubingor rod whose diameter isan appreciable fraction of the length of theradiator, the resonant length of a half-waveantenna will be shortened. The amount ofeJzzUJfa:0I3"' >- . ~~ua:~iI"'~...........1!--...II0I---40 !>0 60 200 300 400 600 800 1000 2000 3000RATIO OF LENGTH TO DIAMETERFigure 3CHART SHOWING SHORTENING OF ARESONANT ELEMENT IN TERMS OFRATIO OF LENGTH TO DIAMETERThe use of this chart Is basec/ on the basicformula where rac//ator length In feet isequal to 468/frequency in Me. This formulaapplies to frequencies below perhaps 30 Me.when the racliator Is macle from wire. Onhigher frequencies, or on 14 ancl 28 Me. whenthe racliator Is mac/e of iarge-c/iameter tubing,the racliator /s shortenec/ from the value obtalneclwith the above formula by an amountc/eterm/nec/ by the ratio of length to c/lameterof the racliator. The amount of this shorteningis obtainable from the chart shown above.shortening can be determined with the aid ofthe chart of figure 3. In this chart the amountof additional shortening over the values givenin the previous paragraph is plotted againstthe ratio of the length to the diameter of thehalf-wave radiator.The length of a wave in free space is somewhatlonger than the length of an antenna forthe same frequency. The actual free-spacehalf-wavelength is given by the followingexpressions:Half-wavelength"'Half-wavelength =HarmonicResonanceI'492Freq. in Me.5905Freq. in Me.Iin feetin inchesA wire in space can resonate atmore than one frequency. The lowestfrequency at which it resonatesis called its fundamental frequency, and atthat frequency it is approximately a half wavelengthlong. A wire can have two, three, four,five, or more standing waves on it, and thusit resonates at approximately the integral harmonicsof its fundamental frequency. However,the higher harmonics are not exactly integralmultiples of the lowest resonant frequency asa result of end el/ects.
<strong>HANDBOOK</strong>Radiation Resistance 407A harmonic operated antenna is somewhatlonger than the corresponding integral numberof dipoles, and for this reason, the dipolelength formula cannot be used simply by multiplyingby the corresponding harmonic. Theintermediate half wave sections do not haveend effects. Also, the current distribution isdisturbed by the fact that power can reachsome of the half wave sections only by flowingthrough other sections, the latter then actingnot only as radiators, but also as transmissionlines. For the latter reason, the resonant lengthwill be dependent to an extent upon the methodof feed, as there will be less attenuation ofthe current along the antenna if it is fed at ornear the center than if fed towards or at oneend. Thus, the antenna would have to be somewhatlonger if fed near one end than if fed nearthe center. The difference would be small,however, unless the antenna were many wavelengthslong.The length of a center fed harmonically operateddoublet may be found from the formula:L =(K-.05) x 492Freq. in Me.where K = number of ~ waves onantennaL = length in feetUnder conditions of severe current attenuation,it is possible for some of the nodes, orloops, actually to be slightly greater than aphysical half wavelength apart. Practice hasshown that the most practical method of resonatinga harmonically operated antenna accuratelyis by cut and try, or by using a feedsystem in which both the feed line and antennaare resonated at the station end as an integralsystem.A dipole or half-wave antenna is said tooperate on its fundamental or first harmonic.A full wave antenna, 1 wavelength long, operateson its second harmonic. An antenna withfive half-wavelengths on it would be operatingon its fifth harmonic. Observe that the fifthharmonic antenna is 2!6 wavelengths long, not5 wavelengths.Antenna Most types of antennas operateResonance most efficient! y when tuned orresonated to the frequency ofoperation. This consideration of course doesnot apply to the rhombic antenna and to theparasitic elements of arrays employing parasiticallyexcited elements. However, in practical!y every other case it will be found that increasedefficiency results when the entire antennasystem is resonant, whether it be a simpledipole or an elaborate array. The radiationefficiency of a resonant wire is many timesthat of a wire which is not resonant.~---------------0,--------------~~-----------------0,------------------~Figure 4EFFECT OF SERIES INDUCTANCE ANDCAPACITANCE ON <strong>THE</strong> LENGTH OF AHALF-WAVE RADIATORThe top antenna has been electrically lengthenedby placing o coil in series with the cen•ter. In other worcls, on antenna with o lumpeclinductance in its center can be macle shorterfor o given frequency than o plain wire rocliotor,The bottom antenna has been capacitive·ly shortened electrically. In other worcls, onantenna with a capacitor in series with Itmust be mocle Ianger for o given frequencysince its effective electrical length os comparee/to plain wire is shorter.If an antenna is slightly too long, it can beresonated by series insertion of a variablecapacitor at a high current point. If it is slightlytoo short, it can be resonated by means ofa variable inductance. These two methods,illustrated schematically in figure 4, are generallyemployed when part of the antenna isbrought into the operating room.With an antenna array, or an antenna fed bymeans of a transmission line, it is more commonto cut the elements to exact resonantlength by "cut and try" procedure. Exact antennaresonance is more important when theantenna system has low radiation resistance;an antenna with low radiation resistance hashigher Q (tunes sharper) than an antenna withhigh radiation resistance. The higher Q doesnot indicate greater efficiency; it simply indicatesa sharper resonance curve.21-3 Radiation Resistanceand Feed-Point ImpedanceIn many ways, a half-wave antenna is like atuned tank circuit. The main difference lies inthe fact that the elements of inductance, capacitance,and resistance are lumped in the tankcircuit, and are distributed throughout thelength of an antenna. The center of a half-waveradiator is effectively at ground potential asfar as r-f voltage is concerned, although thecurrent is highest at that point.
4 08Radiation, Propagation and Lines<strong>THE</strong> R AD I 010000900 0/DIAMETER= •loop of a ~Jarconi antenna is at the base ratherthan in the center. In either case it is a quarterwavelength from the end.A half-wave dipole far from ground and otherreflecting objects has a radiation resistancear the center of about 73 ohms. A Marconi ant60000U)::;I0~8000700 0~ 6000z~(f) sao 0U)w0:: 40001-z0a_ 30000UJw"- 20001000DIAMETER=•DIAMETER=-,&,\\IJ 1'\.\ If .\.v '-J./ ~00.15..\ 0.5>.. 1.0>. 1.5>.. 2.0A 2.5),OVERALL LENGTH OF RADIATORFigureSFEED POINT RESISTANCE OF A CENTERDRIVEN RADIATOR AS A FUNCTION OFPHYSICAL LENGTH IN TERMS OF FREESPACE WAVELENGTHWhen the antenna is resonant, ann it alwaysshould be for best results, the impedance atthe center is substantially resistive, and istermed the radiation resistance. Radiation resistanceis a fictitious term; it is that valueof resistance (referred to the current loop)which would dissipate the same amount ofpower as being radiated by the antenna, whenfed with the current flowing at the current loop.The radiation resistance depends on theantenna length and its proximity to nearbyobjects which either absorb or re-radiate power,such as the ground, other wires, etc.The MarconiAntennaBefore going too far with thediscussion of radiation resistance,an explanation of the Marconi(grounded quarter wave) antenna is inorder. The Marconi antenna is a special typeof Hertz antenna in which the earth acts as the"other half" of the dipole. In other words, thecurrent flows into the earth instead of into asimilar quarter-wave section. Thus, the current+400 0+300 0~ +200 0I0~ +100 0w0(1IIIII/(-DIAMETER= IO~OOOrIIIyoiAMETER=MII//}/_z o L -;;('\ PI~ v/ ..--'\ /?Z ~-4u~ -1000::1-r/DIAMETER=~ ~IIOIIIII IIz -2000 I II I~ -3000 I IwI I0a."--400 0 ll-500 0-fiCO0.15), 0.5).. 1.0.>.. 1.5).. 2.0>.. 2.5>..OVERALL LENGTH OF RADIATORFigure 6REACTIVE COMPONENT OF <strong>THE</strong> FEEDPOINT IMPEDANCE OF A CENTERDRIVEN RADIATOR AS A FUNCTION OFPHYSICAL LENGTH IN TERMS OF FREESPACE WAVELENGTHtenna is simply one-half of a dipole. For thatreason, the radiation resistance is roughlyhalf the 73-ohm impedance of the dipole or36.5 ohms. The radiation resistance of a Marconiantenna such as a mobile whip will belowered by the proximity of the automobilebody.AntennaImpedanceBecause the power throughout theantenna is the same, the impedanceof a resonant antenna at anypoint along its length merely expresses theratio between voltage and current at that point.Thus, the lowest impedance occurs where thecurrent is highest, namely, at the center of adipole, or a quarter wave from the end of aMarconi. The impedance rises uniformly towardeach end, where it is about 2000 ohms for adipole remote from ground, and about twice a,shigh for a vertical Marconi.If a vertical half-wave antenna is set up sothat its lower end is at the ground level, theeffect of the ground reflection is to increase
<strong>HANDBOOK</strong>Antenna Impedance 409I~ 805~ 80~ 70~If) 00iiia:"'a"'~ 40i5
410 Radiation, Propagation and Lines<strong>THE</strong> R AD I 0ic resistance of the wire, ground resistance(in the case of a Marconi), corona discharge,and insulator losses.The approximate effective radiation efficiency(expressed as a decimal) is equal to:N, = Ra!
<strong>HANDBOOK</strong>.... ov • r-- I-.....!""'--..·~ rE>L"'- ~.3'A7..•R\..~ _,I--•....11.1 1.3~ / \~~1.2~ i\Jg •.,1/ ~\ I\..~ 1.0g .•~ .7 ......_~da:I ~i/ ~\!'il f:(~ 1--- ... \7./I~• r- I 'I\... "I\ 1/ 1\\.\ J \.. :; \\\ J.......0~ a a ~ u ~ ~ • ~ 12 ~ a e 4 2 oWAVE ANGLE IN DEGREESFigure 8VERTICAL-PLANE DIRECTIONAL CHARACTERISTICS OF HORIZONTAL AND VERTICAL DOUBLETS ELEVATED 0.6 WAVELENGTH AND ABOVE TWO TYPES OFGROUNDH 1 represents a horizontal doublet over typi·cal farmland. H2 over salt water. V1 Is avertical pattern of radiation from a verticaldoublet over typical farmland, v2 over saltwater. A salt water ground is the closestapproach to an extensive ideally perfectground that will be met in actual practice.great-circle path, or within 2 or 3 degrees ofthat path under all normal propagation conditions.However, under turbulent ionosphereconditions, or when unusual propagation conditionsexist, the deviation from the great-circlepath for greatest signal intensity may be asgreat as 90°. Making the array rotatable overcomesthese difficulties, but arrays having extremelyhigh horizontal directivity become toocumbersome to be rotated, except perhapswhen designed for operation on frequenciesabove 50 Me.VerticalDirectivityVertical directivity is of the great·est importance in obtaining satisfactorycommunication above 14Me. whether or not horizontal directivity isused. This is true simply because only theenergy radiated between certain definite elevationangles is useful for communication. Ener·"'Antenna Directivity 4 11gy radiated at other elevation angles is lostand performs no useful function .Optimum Angle The optimum angle of radiationof Radiation for propagation of signals be·tween two points is dependentupon a number of variables. Among these sig·nificant variables are: (1) height of the iono·sphere layer which is providing the reflection,(2) distance between the two stations, (3) numberof hops for propagation between the twostations. For communication on the 14-Mc.band it is often possible for different modesof propagation to provide signals between twopoints. This means, of course, that more thanone angle of radiation can be used. If no eleva·tion directivity is being used under this conditionof propagation, selective fading willtake place because of interference between thewaves arriving over the different paths .On the 28-Mc. band it is by far the most com·mon condition that only one mode of propagationwill be possible between two points at anyone time. This explains, of course, the reasonwhy rapid fading in general and selective fad·ing in particular are almost absent from signalsheard on the 28-Mc. band (except for fad·ing caused by local effects).Measurements have shown that the anglesuseful for communication on the 14-Mc. bandare from 3° to about 30°; angles above about15 ° being useful only for local work. On the28-Mc. band measurements have shown thatthe useful angles range from about 3° to 18°;angles above about 12° being useful only forlocal (less than 3000 miles) work. These fig·ures assume normal propagation by virtue ofthe ~ layer.Angle of Radiationof Typical Antennasand ArraysIt now becomes of inter·est to determine the a·mount of radiation avail·able at these useful low·er angles of radiation from commonly used an·tennas and antenna arrays. Figure 8 showsrelative output voltage plotted against eleva·tion angle (wave angle) in degrees above thehorizontal, for horizontal and vertical doubletselevated 0.6 wavelength above two types ofground. It is obvious by inspection of thecurves that a horizontal dipole mounted at thisheight above ground (20 feet on the 28-Mc.band) is radiating only a small amount of ener·gy at angles useful for communication on the28-Mc. band. Most of the energy is being radi·ated uselessly upward. The vertical antennaabove a good reflecting surface appears muchbetter in this respect-and this fact has beenproven many times by actual installations.It might immediately be thought that the a·mount of radiation from a horizontal or vertical
412 Radiation, Propagation and Lines<strong>THE</strong> R AD I 0POWER OUTPUTFigure 9VERTICAL RADIATIONPATTERNSShowing the vertical racliationpatterns for half-wave antennas(or colinear half-wave or ex·tenclecl half-wave antennas) atdifferent heights above averagegrouncl one/ perfect grouncl. Notethat such antennas one-quarterwave above ground concentratemost racliation at the very highangles which are useful for com•municotion only on the lower fre•quency bancls. Antennas one-halfwave above g r o u n c/ are n o tshown, but the elevation patternshows one lobe on each sicle otan angIe of 30° above horizontal.dipole could be increased by raising the an·tenna higher above the ground. This is true toan extent in the case of the horizontal dipole;the low-angle radiation does increase slowlyafter a height of 0.6 wavelength is reachedbut at the expense of greatly increased highangleradiation and the formation of a numberof nulls in the elevation pattern. No signalcan be transmitted or received at the elevationangles where these nulls have been formed.Tests have shown that a center height of 0.6wavelength for a vertical dipole (0.35 wavelengthto the bottom end) is about optimum forthis type of array.Figure 9 shows the effect of placing a horizontaldipole at various heights above ground.It is easily seen by reference to figure 9 (andfigure 10 which shows the radiation from a dipoleat % wave height) that a large percentageof the total radiation from the dipole is beingradiated at relatively high angles which areuseless for communication on the 14-Mc. and28-Mc. bands. Thus we see that in order to obtaina worthwhile increase in the ratio of lowangleradiation to high-angle radiation it isnecessary to place the antenna high aboveground, and in addition it is necessary to useadditional means for suppressing high-angleradiation.Suppression ofHigh·angleRadiationHigh-angle radiation can besuppressed, and this radiationcan be added to that going outat low angles, only through theuse of some sort of directive antenna system.There are three general types of antenna arrayscomposed of dipole elements commonlyused which concentrate radiation at the lowermore effective angles for high-frequency communication.These types are: (1) The closespacedout-of-phase system as exemplified bythe "flat-top" beam or WSJK array. Such configurationsare classified as end fire arrays.(2) The wide-spaced in-phase arrays, as exemplifiedby the "Lazy H" antenna. These configurationsare classified as broadside arrays.(3) The close-spaced parasitic systems, asexemplified by the three element rotary beam.A comparison between the radiation from adipole, a "flat-top beam" and a pair of dipolesstacked one above the other (half of a "lazyH"), in each case with the top of the antennaat a height of % wavelength is shown in figure11. The improvement in the amplitude of low-Figure 10VERTICAL RADIATIONPATTERNSShowing vertical-plane racliationpatterns of a horizontal single·section flot·top beam with one·eighth wave spacing (soliclcurves) one/ a horizontal half.wave antenna ( clashecl curves)when both are 0.5 wavelength(A) one/ 0.75 wavelength (B) a·bove grouncl.GAIN IN FIELD STRENGTH
<strong>HANDBOOK</strong>Antenna Bandwidth41 3Figure 11COMPARATIVE VERTICALRADIATION PATTERNSShowing the vertical radiationpatterns of a horizontal singlesectionflat-top beam (A), anarray of two stacked horizontalin-phase half-wave elementshalfof a "Lazy H"-(B), anda horizontal dipole (C). In eachcase the top of the antenna systemis 0.75 wavelength aboveground, as shown to the left ofthe curves.angle radiation at the expense of the uselesshigh-angle radiation with these simple arraysas contrasted to the dipole is quite marked.Figure 12 compares the patterns of a 3 elementbeam and a dipole radiator at a height of0. 75 wavelength. It will be noticed that althoughthere is more energy in the lobe of thebeam as compared to the dipole, the axis ofthe beam is at the same angle above the horizontal.Thus, although more radiated energyis provided by the beam at low angles, theaverage angle of radiation of the beam is nolower than the average angle of radiation ofthe dipole.21-6Propagation ofRadio WavesThe preceding sections have discussed themanner in which an electromagnetic-wave orradio-wave field may be set up by a radiatingsystem. However, for this field to be usefulfor communication it must be propagated tosome distant point where it may be received,or where it may be reflected so that it may bereceived at some other point. Radio wavesmay be propagated to a remote point by eitheror both of two general methods. Propagation21-5 BandwidthThe bandwidth of an antenna or an antennaarray is a function primarily of the radiationresistance and of the shape of the conductorswhich make up the antenna system. For arraysof e~sentially similar construction the bandwidth(or the deviation in frequency which thesystem can handle without mismatch) is increasedwith increasing radiation resistance,and the bandwidth is increased with the useof conductors of larger diameter (smaller ratioof length to diameter). This is to say that ifan array of any type is constructed of largediameter tubing or spaced wires, its bandwidthwill be greater than that of a similar arrayconstructed of single wires.The radiation resistance of antenna arraysof the types mentioned in the previous para·graphs may be increased through the use ofwider spacing between elements. With increasedradiation resistance in such arrays the radiationefficiency increases since the ohmiclosses within the conductors become a smallerpercentage of the radiation resistance, and thebandwidth is increased proportionately.Figure 12VERTICAL RADIATION PATTE~NSShowing vertical radiation patterns of a horizontaldipole (A) and a horizontal 3-elementparasitic array (B) at a height above groundof 0.75 wavelength. Note that the axis of themain radiation lobes are at the same angleabove the horizontal. Note also the suppressionof high angle radiation by the parasiticarray.
414 Radiation, Propagation and Lines<strong>THE</strong> R AD I 0"'"~,, ~-. -..._.._@DIRECT WAVE':\.' -' '' ' -' ', '-. @GROUND-REFLECTE~ .._ -, ~ WAVE ----:::~ --~=----=~--=::::-=---=-=-@SURFACE WAVEfigure 13GROUND-WAVE SIGNAL PROPAGATIONThe illustration above shows the three componentsof the grounc/ wave: (A), the surfacewave; (B), the clirect wave; ancl (C), thegrouncl-reflectecl wave. The clirect wave anclthe grouncl-reflectecl wave combine at thereceiving antenna to make up the spacewave.may take place as a result of the ground wave,or as a result of the sky wave or ionosphericwave.The Ground Wave The term ground wave actu·ally includes several differenttypes ofwaves which usually are called:(1) the surface wave, (2) the direct wave, and(3) the ground-reflected wave. The latter twowaves combine at the receiving antenna toform the resultant wave or the space wave.The distinguishing characteristic of the com·ponents of the ground wave is that all travelalong or over the surface of the earth, so thatthey are affected by the conductivity and terrainof the earth's surface.The Ionospheric Waveor Sky WaveIntense bombardment ofthe upper regions ofthe atmosphere by radiationsfrom the sun results in the formationof ionized layers. These ionized layers, whichform the ionosphere, have the capability ofreflecting or refracting radio waves which impingeupon them. A radio wave which has beenpropagated as a result of one or more reflec·tions from the ionosphere is known as anionospheric wave or a sky wave. Such wavesmake possible long distance radio communication.Propagation of radio signals by iono·spheric waves is discussed in detail in Sec·tion 21-8.21-7 Ground-WoveCommunicationAs stated in the preceding paragraph, theterm ground wave applies both to the surfacewave and to the space wave (the resultantwave from the combination of the direct waveand the ground-reflected wave) or to a com·bination of the two. The three waves whichmay combine to make up the ground wave areillustrated in figure 13.The Surface Wave The surface wave is thatwave which we normallyreceive from a standard broadcast station. Ittravels directly along the ground and terminateson the earth's surface. Since the earth isa relatively poor conductor, the surface waveis attenuated quite rapidly. The surface waveis attenuated less rapidly as it passes oversea water, and the attenuation decreases fora specific distance as the frequency is decreased.The rate of attenuation with distancebecomes solarge as the frequency is increasedabove about 3 Me. that the surface wave be·comes of little value for communication.The Space Wave The resultant wave or spacewave is illustrated in figur~13 by the combination of (B) and (C). It is thiswave path, which consists of the combinationof the direct wave and the ground-reflectedwave at the receiving antenna, which is thenormal path of signal propagation for line-ofsightor near line-of-sight communication orFM and TV reception on frequencies aboveabout 40 Me.Below line-of-sight over plane earth orwater, when the signal source is effectivelyat the horizon, the ground-reflected wave doesnot exist, so that the direct wave is the onlycomponent which goes to make up the spacewave. But when both the signal source and thereceiving antenna are elevated with respect tothe intervening terrain, the ground-reflectedwave is present and adds vectorially to thedirect wave at the receiving antenna. The vee·torial addition of the two waves, which travelover different path lengths (sine e one of thewaves has been reflected from the ground) resultsin an interference pattern. The interferencebetween the two waves brings about acyclic variation in signal strength as the receivingantenna is raised above the ground.This effect is illustrated in figure 14. Fromthis figure it can be seen that best spacewavereception of a v-h-f signal often will beobtained with the receiving antenna quite closeto the ground. This subject, along with otheraspects of v-h-f signal propagation and reception,are discussed in considerable detail ina book on fringe-area TV reception.*The distance from an elevated point to thegeometrical horizon is ~en by .the approximateequation: d = 1.22y H where the d1stance* 11 Better TV Re~eption," by W. W. Smith and R. L. Dawley,published by Editors and Engineers, Ltd., Summerland,Calif.
<strong>HANDBOOK</strong> Ground Wave Communication 415GROUND- REFLECTED 0 2 03WAVESFigure 14RECEIVINGANTENNAAT 01 FFEREN<strong>THE</strong>IGHTSWAVE INTERFERENCE WITH HEIGHTWhen the source of a horizontally•palarizeclspace-wave signal is above the horizen, thereceivecl signal at a clistant location will gothrough a cyclic variation as the antennaheight is progressively raisecl. This is clueto the difference in total path length betweenthe direct wave and the ground-reflectedwave, and to the fact that this path lengthdifference changes with antenna height.When the path length cliHerence is such thatthe two waves arrive at the receiving anten·no with a phase difference of 360o or somemultiple of 360°, the two waves will appearto be in phase as lor as the antenna is con·cerned and maximum signal will be obtained.On the other hand, when the antenna heightis such that the path length difference forthe two waves causes the waves to a"lvewith a phose difference of an oclcl multipleof 180° the two waves will substantially can•eel, and a null will be obtained at that an•tenna height. The difference between D 1and D 2 plus D3 is the path-length difference.Note also that there is an additional 180°phose shift in the ground-reflected wave atthe point where it is reflected from theground. It is this latter phase shift whichcauses the space-wave fielcl intensity of ahorizontally polarized wave to be zero withthe receiving antenna at ground level.d is in miles and the antenna height H is infeet. This equation must be applied separatelyto the transmitting and receiving antennas andthe results added. However, refraction anddiffraction of the signal around the sphericalearth cause a smaller reduction in field strengththan would occur in the absence of such bend·ing, so that the average radio horizon is somewhatbeyond the li!:ometrical horizon. Theequation d = 1.4 y H is sometimes used fordetermining the radio horizon.Tropospheric Propagation by signal bendingPropagation in the lower atmosphere, calledtropospheric propagation, canresult in the reception of signals over a muchgreater distance than would be the case if thelower atmosphere were homogeneous. In ahomogeneous or well-mixed lower atmosphere,called a normal or standard atmosphere, thereis a gradual and uniform decrease in index ofrefraction with height. This effect is due tothe combined effects of a decrease in temper·ature, pressure, and water-vapor content withheight.This gradual decrease in refractive indexwith height causes waves radiated at very lowangles with respect to the horizontal to bebent downward slightly in a curved path. Theresult of this effect is that such waves will bepropagated beyond the true or geometricalhorizon. In a so-called standard atmospherethe effect of the curved path is the same asthough the radius of the earth were increasedby approximately one third. This condition extendsthe horizon by approximately 30 per centfor normal propagation, and the extended.horizonis known as the radio path horizon, men·tioned before.Conditions Leading toTroposphericStratificationWhen the temperature,pre .. sure, or water-vaporcontent of the atmos·phere does not changesmoothly with rising altitude, the discontinuityor stratification will result in the reflectionor refraction of incident v-h-f signals. Ordinarilythis condition is more prevalent at nightand in the summer. In certain areas, such asalong the west coast of North America, it isfrequent enough to be considered normal. Sig·nal strength decreases slowly with distanceand, if the favorable condition in the loweratmosphere covers sufficient area, the rangeis limited only by the transmitter power, an·tenna gain, receiver sensitivity, and signal-to·noise ratio. There is no skip distance. Usually,transmission due to this condition is accom·panied by slow fading, although fading can beviolent at a point where direct waves of aboutthe same strength are also received.Bending in the troposphere, which refers tothe region from the earth's surface up to about10 kilometers, is more likely to occur on dayswhen there are stratus clouds than on clear,cool days with a deep blue sky. The temperatureor humidity discontinuities may be brokenup by vertical convection currents over landin the daytime but are more likely to continueduring the day over water. This condition isin some degree predictable from weather informationseveral days in advance. It does notdepend on the sunspot cycle. Like direct communication,best results require similar an·tenna polarization or orientation at both thetransmitting and receiving ends, whereas intransmission via reflection in the ionosphere(that part of the atmosphere between about 50and 500 kilometers high) it makes little differencewhether antennas are similarly polar·ized.Duct FormationWhen bending conditions areparticularly favorable they
416 Radiation, Propagation and Lines<strong>THE</strong> R AD I 0IIwIREFRACTIVE INDEXFigure 15ILLUSTRATING DUCT TYPESShowing two types of variation in refractiveindex with height which will give rise to theformation of a duct. An elevated duct isshown at (A}, and a ground-based duct isshown at (B). Such ducts con propagateground-wave signals far beyond their normalrange.may give rise to the formation of a duct whichcan propagate waves with very little attenuationover great distances in a manner similarto the propagation of waves through a waveguide. Guided propagation through a du£t inthe atmosphere can give quite remarkabletransmission conditions (figure 15). However,such ducts usually are formed only on an overwaterpath. The depth of the duct over thewater's surface may be only 20 to 50 feet, orit may be 1000 feet deep or more. Ducts exhibita low-frequency cutoff characteristicsimilar to a wave guide. The cutoff frequencyis determined by depth of the duct and by thestrength of the discontinuity in refractive indexat the upper surface of the duct. The lowest·frequency that can be propagated by sucha duct seldom goes below 50 Me., and usuallywill be greater than 100 Me. even along thePacific Coast.StratosphericReflectionCommunication by virtue ofstratospheric reflection can bebrought about during magneticstorms, aurora borealis displays, and duringmeteor showers. Dx communication during extensivemeteor showers is characterized byfrequent bursts of great signal strength followedby a rapid decline in strength of thereceived signal. The motion of the meteorforms an ionized trail of considerable extentwhich can bring about effective reflection ofsignals. However, the ionized region persistsonly for a matter of seconds so that a showerof meteors is necessary before communicationbecomes possible.The type of communication which is possibleduring visible displays of the aurora borealisand during magnetic storms has been calledaurora-type dx. These conditions reach a maximumsomewhat after the sunspot cycle peak,possibly because the spots on the sun arenearer to its equator (and more directly in linewith the earth) in the latter part of the cycle.Ionospheric storms generally accompany magneticstorms. The normal layers of the ionospheremay be churned or broken up, makingradio transmission over long distances difficultor impossible on high frequencies. Unusualconditions in the ionosphere sometimesmodulate v-h-f waves so that a definite tone ornoise modulation is noticed even on transmitterslocated only a few miles away.A pecularity of this type of auroral propagationof v-h-f signals in the northern hemisphereis that directional antennas usually must bepointed in a northerly direction for best resultsfor transmission or reception, regardless ofthe direction of the other station being contacted.Distances out to 700 or 800 miles havebeen covered during magnetic storms, using30 and 50 Me. transmitters, with little evidenceof any silent zone between the stationscommunicating with each other. Generally,voice-modulated transmissions are difficult orimpossible due to the tone or noise modulationon the signal. Most of the communication ofthis type has taken place by c. w. or by tonemodulated waves with a keyed carrier.21-8 IonosphericPropagationPropagation of radio waves for communicationon frequencies between perhaps 3 and30 Me. is normally carried out by virtue ofionospheric reflection or refraction. Under conditionsof abnormally high ionization in theionosphere, communication has been knownto have taken place by ionospheric reflectionon frequencies higher than 50 Me.The ionosphere consists of layers of ionizedgas located above the stratosphere, and extendingup to possibly 300 miles above theearth. Thus we see that high-frequency radiowaves may travel over short distances in adirect line from the transmitter to the receiver,or they can be radiated upward into the ionosphereto be bent downward in an indirect ray,returning to earth at considerable distancefrom the transmitter. The wave reaching a receivervia the ionosphere route is termed asky wave. The wave reaching a receiver bytraveling in a direct line from the transmittingantenna to the receiving antenna is commonlycalled a ground wave.The amount of bending at the ionosphere
<strong>HANDBOOK</strong>Ionospheric Propagation 417200uo100w 50"'-':::;:0~ 2001-I'-' 150wI10050/0VD/F2/F1MIDDAY:JE/ MIDNIGHT) E)F2IONIZATION DENSITY-Figure 16IONIZATION DENSITY IN <strong>THE</strong> IONO-SPHEREShowing typical ionization density of theionosphere in mid-summer. Note that the F 1and D layers disappear at night, and that thedensity of the E Ioyer falls to such o lowvalue that it is ineffective.which the sky wave can undergo dc:'pends uponits frequency, and the amount of ionizationin the ionosphere, which is in turn dependentupon radiation from the sun. The sun increasesthe density of the ionosphere layers (figure 16)and lowers their effective height. For thisreason, the ionosphere acts very differentlyat different times of day, and at different timesof the year.The higher the frequency of a radio wave,the farther it penetrates the ionosphere, andthe less it tends to be bent back toward theearth. The "lower the frequency, the more easilythe waves are bent, and the less they penetratethe ionosphere. 160-meter and SO-metersignals will usually be bent back to eartheven when sent straight up, and may be consideredas being reflected rather than refract·ed. As the frequency is raised beyond about5,000 kc. (dependent upon the critical frequencyof the ionosphere at the moment), it isfound that waves transmitted at angles higherthan a certain critical angle never return toearth. Thus, on the higher frequencies, it isnecessary to confine radiation to low angles,since the high angle waves simply penetratethe ionosphere and are lost.The F 2LayerThe higher of the two majorreflection regions of the ionosphereis called the F 2 layer. This layer hasa virtual height of approximately 175 miles atnight, and in the daytime it splits up into twolayers, the upper one being called the F 2layerand the lower being called the F, layer. Theheight of the F 2 layer during daylight hours isnormally about 250 miles on the average andthe F, layer often has a height of as low as140 miles. It is the F 2 layer which supportsall nighttime dx communication and nearly alldaytime dx propagation.The E Loyer Below the F 2 layer is anotherlayer, called the E layer, whichis of importance in daytime communicationover moderate distances in the frequency rangebetween 3 and 8 Me. This layer has an almostconstant height at about 70 miles. Since there-combination time of the ions at this heightis rather short, the E layer disappears almostcompletely a short time after local sunset.The D Loyer Below the E layer at a height ofabout 35 miles is an absorbinglayer, called the D layer, which exists in themiddle of the day in the summertime. The layeralso exists during midday in the winter timeduring periods of high solar activity, but thelayer disappears completely at night. It is thislayer which causes high absorption of signalsin the medium and high-frequency range duringthe middle of the day.Critical Frequency The critical frequency ofan ionospheric layer is thehighest frequency which will be reflected whenthe wave strikes the layer at vertical incidence.The critical frequency of the most highlyionized layer of the ionosphere may be aslow as 2 Me-. at night and as high as 12 to 13Me. in the middle of the day. The critical frequencyis directly of interest in that a skip·distance zone will exist on all frequenciesgreater than the highest critical frequency atthat time. The critical frequency is a measureof the density of ionization of the reflectinglayers. The higher the critical frequency thegreater the density of ionization.Maximum UsableFrequencyThe maximum usable fre·quency or m. u. f. is of greatimportance in long-distancecommunication since this frequency is the highestthat can be used for communication betweenany two specified areas. The m.u.f. isthe highest frequency at which a wave projectedinto space in a certain direction willbe returned to earth in a specified region byionospheric reflection. The m.u.f. is highestat noon or in the early afternoon and is highestin periods of greatest sunspot activity,often going to frequencies higher than 50 Me.(figure 17).
418 Radiation, Propagation and Lines<strong>THE</strong> R A 0 I 03643202 8•24.22~2 0~·I 8u. •4& ~·120a: "''... .642--4~ ...... I'\. IIII 1\II \ WINTERy SUNSPOT- -MAXIMUM1\\SUMMER-SUNSPOT-"--+ MINIMUM/ \1\. ....... ,_....'\........... f.---10 12 14 16 16 20 22 24LOCAL TIMEFigure 17TYPICAL CURVES SHOWING CHANGE INM.U.F. AT MAXIMUM AND MINIMUMPOINTS IN SUNSPOT CYCLEThe m.u.f. often drops to frequencies below10 Me. in the early morning hours. The highm.u.f. in the middle of the day is brought aboutby reflection from the F 2 layer. M.u.f. data ispublished periodically in the magazines de·voted· to amateur work, and the m.u.f. can becalculated with the aid of Basic Radio PropagationPredictions, CRPL-D, published monthlyby the Government Printing Office, Washington,D.C.Absorption andOptimum WorkingFrequencyThe optimum working frequencyfor any particulardirection and distance isusually about 15 per centless than the m. u. f. for contact with that particularlocation. The absorption by the ionospherebecomes greater and greater as theoperating frequency is progressively loweredbelow the m.u.f. It is this condition whichcauses signals to increase tremendously instrength on the 14-Mc. and 28-Mc. bands justbefore the signals drop completely out. At thetime when the signals are greatest in amplitudethe operating frequency is equal to them.u.f. Then as the signals drop out the m.u.f.has become lower than the operating frequency.Skip Distance The shortest distance from atransmitting location at whichsignals reflected from the ionosphere can bereturned to the earth is called the skip distance.As was mentioned above under CriticalFrequency there is no skip distance for a frequencybelow the critical frequency of themost highly ionized layer of the ionosphereat the time of transmission. However, the skipdistance is always present on the 14-Mc. bandand is almost always present on the 3.5-Mc.and 7-Mc. bands at night. The actual measureof the skip distance is the distance betweenthe point where the ground wave falls to zeroand the point where the sky wave begins toreturn to earth. This distance may vary from40 to 50 miles on the 3.5-Mc. band to thousandsof miles on the 28-Mc. band.The Sporadic-ELayerOccasional patches of ex·tremel y high ionization densityappear at intervalsthroughout the year at a height approximatelyequal to that of the E layer. These patches,called the sporadic-E layer may be very smallor may be up to several hundred miles in extent.The critical frequency of the sporadic-Elayer may be greater than twice that of thenormal ionosphere layers which exist at thesame time.It is this sporadic-E condition which provides"short-skip" contacts from 400 to perhaps1200 "!iles on the 28-Mc. band in theevening. It is also the sporadic-E conditionwhich provides the more common type of "bandopening" experienced on the 50-Mc. band whenvery loud signals are received from stationsfrom 400 to 1200 miles distant.Cycles inIonosphere ActivityThe ionization density ofthe ionosphere is determinedby the amount ofradiation (probably ultra violet) which is beingreceived from the sun. Consequently, ionosphereactivity is a function of the amount ofradiation of the proper character being emittedby the sun and is also a function of the relativeaspect of the regions in the vicinity ofthe location under discussion to the sun. Thereare four main cycles in ionosphere activity.These cycles are: the daily cycle which isbrought about by the rotation of the earth, the27-day cycle which is caused by the rotationof the sun, the seasonal cycle which is causedby the movement of the earth in its orbit, andthe 11-year cycle which is a cycle in sunspotactivity. The effects of these cycles are superimposedinsofar as ionosphere activity is con·cerned. Also, the cycles are subject to shortterm variations as a result of magnetic stormsand similar terrestrial disturbances.The most recent minimum of the 11-yearsunspot cycle occured during the winter of1954-1955, and we are currently moving upthe slope of a new cycle, the maximum ofwhich will probably occur during the year1958. The current cycle is pictured in figure 18.FadingThe lower the angle of radiation ofthe wave, with respect to the hori-
<strong>HANDBOOK</strong>11-Year Sunspot Cycle 419170160150a: 140~ 130::
420 Radiation, Propagation and Lines<strong>THE</strong> R AD I 0dent, particularly a "flutter fade" and a char·acteristic "hollow" or echo effect.Deviations from a great circle path are especiallynoticeable in the case of great circlepaths which cross or pass near the auroralzones, because in such cases there often iscomplete or nearly complete absorption of thedirect sky wave, leaving off-path scatteredreflections the only mechanism of propagation.Under such conditions the predominant wavewill appear to arrive from a direction closerto the equator, and the signal will be noticeablyif not considerably weaker than a directsky wave which is received under favorableconditions.Irregular reflection of radio waves from"scattering patches" is divided into two categories:"short scatter" and "long scatter".Short scatter is the scattering that occurswhen a radio wave first reaches the scatteringpatches or media. Ordinarily it is of no parti·cular benefit, as in most cases it only servesto fill in the inner portion of the skip zonewith a weak, distorted signal.Long scatter occurs when a wave has beenrefracted from the F 2 layer and strikes scatter·ing patches or media on the way down. Whenthe skip distance exceeds several hundredmiles, long scatter is primarily responsiblefor reception within the skip zone, particularlythe outer portion of the skip zone. Dis·tortion is much less severe than in the caseof short scatter, and while the signal is like·wise weak, it sometimes can be utilized forsatisfactory communication.During a severe ionosphere disturbance inthe north auroral zone, it sometimes is possibleto maintain communication between the EasternUnited States and Northern Europe by the followingmechanism: That portion of the energywhich is radiated in the direction of the greatcircle path is completely absorbed upon reachingthe auroral zone. However, the portion ofthe wave leaving the United States in a southeasterlydirection is refracted downward fromthe F 2 layer and encounters scattering patchesor media on its downward trip at a distanceof approximately 2000 miles from the trans·mitter. There it is reflected by "long scatter"in all directions, this scattering region actinglike an isotropic radiator fed with a very smallfraction of the original transmitter power. Thegreat circle path from this southerly point tonorthern Europe does not encounter unfavorableionosphere conditions, and the wave ispropagated the rest of the trip as though it hadbeen radiated from the scattering region.Another type of scatter is produced whena sky wave strikes certain areas of the earth.Upon striking a comparatively smooth surfacesuch as the sea, there is little scattering, thewave being shot up again by what could beconsidered specular or mirror reflection. Butupon striking a mountain range, for instance,the reradiation or reflected energy is scattered,some of it being directed back towards thetransmitter, thus providing another mechanismfor producing a signal within the skip zone.Meteors and"Bursts"When a meteor strikes the earth'satmosphere, a cylindrical regionof free electrons is formed atapproximately the height of the E layer. Thisslender ionized column is quite long, and whenfirst formed is sufficiently dense to reflectradio waves back to earth most readily, includingv·h·f waves which are not ordinarilyreturned by the F 2 layer.The effect of a single meteor, of normal size,shows up as a sudden "burst" of signal ofshort duration at points not ordinarily reachedby the transmitter. After a period of from 10to 40 seconds, recombination and diffusionhave progressed to the point where the effectof a single fairly large meteor is not perceptible.However, there are many small meteorsimpinging upon earth's atmosphere every minute,and the aggregate effect of their transientionized trails, including the small amount ofresidual ionization that exists for severalminutes after the original flash butis too weakand dispersed to prolong a "burst", is believedto contribute to tbe existence of tbe"nighttime E" layer, and perhaps also tosporadic E patches.While there are many of these very smallmeteors striking the earth's atmosphere everyminute, meteors of normal size (sufficientlylarge to produce individual "bursts") do notstrike nearly so frequently except during someof the comparatively rare meteor "showers".During one of these displays a "quivering"ionized layer is produced which is intenseenough to return signals in the lower v-h-frange with good strength, but with a type of"flutter" distortion which is characteristicof this type of propagation.21-9 Transmission LinesFor many reasons it is desirable to placean antenna or radiating system as high and inthe clear as is physically possible, utilizingsome form of nonradiating transmission lineto carry energy with as little loss as possiblefrom the transmitter to the radiating antenna,and conversely from the antenna to the re·ceiver.There are many different types of transmissionlines and, generally speaking, practicallyany type of transmission line or feeder systemmay be used with any type of antenna. How-
<strong>HANDBOOK</strong>Transmission Lines 421ever, mechanical or electrical considerationsoften make one type of transmission line betteradapted for use to feed a particular type ofantenna than any other type.Transmission lines for carrying r-f energyare of two general types: non-resonant andresonant. A non-resonant transmission lineis one on which a successful effort has beenmade to eliminate reflections from the termination(the antenna in the transmitting caseand the receiver for a receiving antenna) andhence one on which standing waves do notexist or are relatively small in magnitude. Aresonant line, on the other hand, is a transmissionline on which standing waves of appreciablemagnitude do appear, either throughinability to match the characteristic impedanceof the line to the termination or through intentionaldesign.The principal types of transmission line inuse or available at this time include the openwireline (two-wire and four-wire types), two·wire solid-dielectric line ("Twin-Lead" andsimilar ribbon or tubular types), two-wire polyethylene-filledshielded line, coaxial line ofthe solid-dielectric, beaded, stub-supported,or pressurized type, rectangular and cylindricalwave guide, and the single-wire feeder operatedagainst ground. The significant characteristicsof the more popular types of transmissionline available at this time are givenin the chart of figure 21.21-10 Non-ResonantTransmission LinesA non-resonant or untuned transmission lineis a line with negligible standing waves.Hence, a non-resonant line is a line carryingr-f power only in one direction-from the sourceof energy to the load.Physically, the line itself should be identicalthroughout its length. There will be asmooth distribution of voltage and currentthroughout its length, both tapering off veryslightly towards the load end of the line as aresult of line losses. The attenuation (loss)in certain types of untuned lines can be keptvery low for line lengths up to several thousandfeet. In other types, particularly wherethe dielectric is not air (such as in the twistedpair"line), the losses may become excessiveat the higher frequencies, unless the line isrelatively short.Transmission-LineImpedanceance. Neglectingminor importanceAll transmission lines havedistributed inductance,capacitance and resist·the resistance, as it is ofin short lines, it is found-~ 6. .1 .8 .9 1.0 2.INCHES, CENTER TO CENTERFigure 20CHARACTERISTIC IMPEDANCE OF TYPI·CAL TWO-WIRE OPEN LINESthat the inductance and capacitance per unitlength determine the characteristic or surgeimpedance of the line. Thus, the surge impedancedepends upon the nature and spacingof the conductors, and the dielectric separatingthem.Speaking in electrical terms, the characteristicimpedance of a transmission line issimply the ratio of the voltage across the lineto the current which is flowing, the same asis the case with a simple resistor: Z "' 0E/1.Also, in a substantially loss-less line (onewhose attenuation per wavelength is small)the energy stored in the line will be equallydivided between the capacitive field and theinductive field which serve to propagate theenergy along the line. Hence the characteristicimpedance of a line may be expressed as:Two-WireOpen LineA two-wire transmission systemis easy to construct. Its surge impedancecan be calculated quiteeasily, and when properly adjusted and balancedto ground, with a conductor spacingwhich is negligible in terms of the wavelengthof the signal carried, undesirable feederradiation is minimized; the current flow inthe adjacent wires is in opposite directions,and the magnetic fields of the two wires arein opposition to each other. When a two-wireline is terminated with the equivalent of apure resistance equal to the characteristicimpedance of the line, the line becomes a nonresonantline.Expressed in physical terms, the character·istic impedance of a two-wire open line isequal to:
422 Radiation, Propagation and Lines <strong>THE</strong> R AD I 0CHARACTERISTICS OF COMMON TRANSMISSION LINESATTENUATIONdb/100FEETVSWR = t.OVELO-F~~~';?30 MC 100 MC 300 MC v FT.REMARKSOPEN WIRE LINE, NO 120.1!1 0.3 0.8 o.oe-o ... BASED UPON 4"' SPACING BELOW ~0 MC.; 2'" SPACING ABOVE~ MC. RADIATIONCOPPER.- LOSSES INCLUDED. CLEAN, LOW LOSS CERAMIC INSULATION ASSUMED. RADIATIONHIGH ABOVE 1~ MC.~~~i~NE, REC.TYPE,FOR CLEAN, DRY LINE. WET WEA<strong>THE</strong>R PERFORMANCE RA<strong>THE</strong>R POOR. BEST LINE IS0.88 2.2 ~.3 0.82.., 8.., SL.IGHTLY CONVEX. AVOID LINE THAT HAS CONCAVE DIELECTRIC. SUITABLE FOR(7/28 CONDUCTORS) L.OW POWER TRANSMITTING APPLICATIONS. LOSSES INCREASE AS LINE WEA<strong>THE</strong>RS.HANDLES400WATTS AT30MC.IF VSWR IS LOW.TUBULAR "TWIN-LEAD"CHARACTERISTICS SIMILAR TO RECEIVING TYPE RIBBON LINE EXCEPT FOR MUCH:~~-~~·(~~p~~rtJc. - - - - - BETTER WET WEA<strong>THE</strong>R PERFORMANCE.TYPE 14-271)RIBBON LINE, TRANS.CHARACTERISTICS VARY SOMEWHAT WITH MANUFACTURER, BUT APPROXIMATETYPE, 300 OHMS. - - - - - THOSE OF RECEIVING TYPE RIBBON EXCEPT FOR GREATER POWER HANDLINGCAPABILITY AND SLIGHTLY BETTER WET WEA<strong>THE</strong>R PERFORMANCE.TUBULAR "TWIN-LEAD..FOR USE WHERE RECEIVING TYPE TUBULAR •TWIN-LEAD.. DOES NOT HAVE SUfFI-(~~~~ETJ6C· ,3{~Z7~f·CIENT POWER HANDLING CAPABILITY. WILL HANDLE 1 KW AT 30 MC. IF VSWRIS LOW.o.e~ 2.3 ~.4 0.79 8.1RIBBON LINE, RECEIVETYPE, 150 OHMS.1.1 2.7 8.0 0.77 ... 10..,USEFUL FOR QUARTER WAVE MATCHING SECTIONS. NO LONGER WIDELY USEDAS A LINE.RIBBON UNE 1 RECEIVE.2.0 ~-0 11.0 0.88"4' 19.., USEFUL MAINLY IN <strong>THE</strong> H-F RANGE BECAUSE OF EXCESSIVE LOSSES AT V-H-FTYPE, 7~ OHMS.AND U-H·F. LESS AFFECTED BY WEA<strong>THE</strong>R THAN 300 OHM-RIBBON.RIBBON LINE, TRANS. 1.5 3.9• 8.0 0.71-t 10.., VERY SATISFACTORY FOR TRANSMITTING APPLICATIONS BELOW 30 MC. ATTYPE, 75 OHMS.POWERS UP TO 1 KW. NOT SIGNIFICANTLY AFfECTCO BY WET WEA<strong>THE</strong>R.RG-8/U COAX (52 0HMS) 1.0 2.1 4.2 o.ee 29.5 WILL HANDLE 2 KW AT aclMC. IF VSWR IS LOW. 0.4 ... 0.0. 7/21 CONDUCTOR.RG-1 1/UC:CWC.(75 OHMS) 0.94 1.9 3.0 o.&8 20.5 WILL HANDLE 1.4 KW AT 30 MC. IF VSWR IS LOW. 0.4"0.D. 7/26 CONDUCTOR .RG-17/U COAX (52 0HMS) 0.38 o.e5 ... o.ee 29 ..5 WILL HANDLE 7.0 KW. AT 30 MC. IF VSWR IS LOW. 0.87" 0.0. 0.19" DIA. CONDUCTORRG·~/U COAX (>3 OH~ 1.95 4.1 8.0 o.ee 28.5 WILL HANDLE 430 WATTS AT 30 MC. IF VSWR IS LOW. 0.2" 0.0. N"" 20 CONDUCTOR.RG-59/U COAX (73 OHMS) 1.9 3.8 7.0 o.ee 21 WILL HANDLE 080 WATTS AT 30 MC. IF VSWR IS LOW. Q.24"0.D. N° 22 CONDUCTOR.rv..:.sg COAX (720HMS) 2.0 4.0 7.0 o.ee 22 •COMMERCIAL" VERSION OF RG-.59/U FOR LESS EXACTING APPLICATIONS. LESSEXPENSIVE.RG-22/U SHIELDEDPAIR (9.5 OHMS)1.7 3.0 5.5 o.ee 18 -FOR SHIELDED, BALANCED-TO-GROUND APPLICATIONS. VERY LOW NOISEPICK UP. 0.4 II O.D.K-1 11 SHI ELOED PAl RDESIGNED FOR TV LEAD-IN IN NOISY LOCATIONS. LOSSES HIGHER THAN2.0 3.~ (300 OHMS)0.1 - 4 REGULAR 300 OHM RIBBON, BUT DO NOT INCREASE AS MUCH FROM WEA<strong>THE</strong>RING.t APPROXIMATE. EXACT fiGURE VARIES SLIGHTLY WITH MANUFACTURER.FIGURE 21Where:S is the exact distance between wire centersin some convenient unit of measurement, andd is the diameter of the wire measured in thesame up.its as the wire spacing, s_2SZ 0 = 276 log 10-d2SSince - expresses a ratio only, the unitsdof measurement may be centimeters, millimeters,or inches. This makes no differencein the answer, so long as the substitutedvalues for S and dare in the same units.The equation is accurate so long as thewire spacing is relatively large as comparedto the wire diameter.Surge impedance values of less than 200ohms are seldom used in the open-type twowireline, and, even at this rather high valueof Z 0 the wire spacing S is uncomfortablyclose, being only 5.3 times the wire diameter d_Figure 20 gives in graphical form the surgeimpedance of practicable two-wire lines. Thechart is self-explanatory, and is sufficientlyaccurate for practical purposes.Ribbon andTubular Trans·mission LineInstead of using spacer insulatorsplaced periodicallyalong the transmission lineit is possible to mold theline conductors into a ribbon or tube of flexiblelow·loss dielectric material. Such line,with polyethylene dielectric, is used in enormousquantities as the lead·in transmissionline for FM and TV receivers_ The line isavailable from several manufacturers in the
<strong>THE</strong> R AD I 0Transmission Lines 423ribbon and tubular configuration, with characteristicimpedance values from 75 to 300ohms. Receiving types, and transmitting typesfor power levels up to one kilowatt in the h-frange, are listed with their pertinent characteristics,in the table of figure 21.Coaxial Line Several types of coaxial cablehave come into wide use forfeeding power to an antenna system. A crosssectionalview of a coaxial cable (sometimescalled concentric cable or line) is shown infigure 22.As in the parallel-wire line, the power lostin a properly terminated coaxial line is thesum of the effective resistance losses alongthe length of the cable and the dielectriclosses between the two conductors.Of the two losses, the effective resistanceloss is the greater; since it is largely due tothe skin effect, the line loss (all other conditionsthe same) will increase directly as thesquare root of the frequency.Figure 22 shows that, instead of having twoconductors running side by side, one of theconductors is placed inside of the other. Sincethe outside conductor completely shields theinner one, no radiation takes place. The conductorsmay both be tubes, one within theother; the line may consist of a solid wirewithin a tube, or it may consist of a strandedor solid inner conductor with the outer conductormade up of one or two wraps of coppershielding braid.In the type of cable most popular for militaryand non-commercial use the inner conductorconsists of a heavy stranded wire, theouter conductor consists of a braid of copperwire, and the inner conductor is supportedwithin the outer by means of a semi-soliddielectric of exceedingly low loss characteristicscalled polyethylene. The Army-Navydesignation on one size of this cable suitablefor power levels up to one kilowatt at frequenciesas high as 30 Me. is AN/RG-8/U.The outside diameter of this type of cable isapproximately one-half inch. The characteristicimpedance of this cable type is 52 ohms,but other similar types of greater and smallerpower-handling capacity are available in impedancesof 52, 75, and 95 ohms.When using solid dielectric coaxial cableit is necessary that precautions be taken toinsure that moisture cannot enter the line. Ifthe better grade of connectors manufacturedfor the line are employed as terminations, thiscondition is automatically satisfied. If connectorsare not used, it is necessary thatsome type of moisture-proof sealing compoundbe applied to the end of the cable where itwill be exposed to the weather.Nearby metallic objects cause no loss, andcoaxial cable may be run up air ducts or ele-0Nvi::;J:0!:wuz~~:::;;u~a:wfu
424 Radiation, Propagation and Lines<strong>THE</strong> R AD I 0line fed by a transmitter. It is the reflectionfrom the antenna end which starts waves movingback toward the transmitter end. Whenwaves moving in both directions along a conductormeet, standing waves are set up.Semi-ResonantParallel-Wire LinesA well-constructed openwireline has acceptablylow losses when its lengthis less than about two wavelengths even whenthe voltage standing-wave ratio is as high as10 to 1. A transmission line constructed ofribbon or tubular line, however, should havethe standing-wave ratio kept down to not morethan about 3 to 1 both to reduce power lossand because the energy dissipation on the linewill be localized, causing overheating of theline at the points of maximum current.Because moderate standing waves can betoleratedon open-wire lines without much loss,a standing-wave ratio of 2/1 or 3/1 is consideredacceptable with this type of line, evenwhen used in an untuned system. Strictlyspeaking, a line is untuned, or non-resonant,only when it is perfectly flat, with a standingwaveratio of 1 (no standing waves). However,some mismatch can be tolerated with open-wireuntuned lines, so long as the reactance is notobjectionable, or is eliminated by cutting theline to approximately resonant length.21-11 Tuned orResonant LinesIf a transmission line is terminated in itscharacteristic surge impedance, there will beno reflection at the end of the line, and thecurrent and voltage distribution will be uniformalong the line. If the end of the line iseither open-circuited or short-circuited, thereflection at the end of the line will be 100per cent, and standing waves of very great amplitudewill appear on the line. There will stillbe practically no radiation from the line if it isclosely spaced, but voltage nodes will befound every half wavelength, the voltage loopscorresponding to current nodes (figure 23).If the line is terminated in some value ofresistance other than the characteristic surgeimpedance, there will be some reflection, theamount being determined by the amount of mismatch.With reflection, there will be standingwaves (excursions of current and voltage)along the line, though not to the same extentas with an open-circuited or short-circuitedline. The current and voltage loops will occurat the same points along the line as with theopen or short-circuited line, and as the terminatingimpedance is made to approach thecharacteristic impedance of the line, the cur-
<strong>HANDBOOK</strong>Tuned Lines 425amplitude, in turn, depends upon the mismatchat the line termination. A line of no. 12 wire,spaced 6 inches with good ceramic or plasticspreaders, has a surge impedance of approximately600 ohms, and makes an excellenttuned feeder for feeding anything between 60and 6000 ohms (at frequencies below 30 Me.).If used to feed a load of higher or lower impedancethan this, the standing waves becomegreat enough in amplitude that some loss willoccur unless the feeder is kept short. At frequenciesabove 30 Me., the spacing becomesan appreciable fraction of a wavelength, andradiation from the line no longer is negligible.Hence, coaxial line or close-spaced parallelwireline is recommended for v-h-f work.If a transmission line is not perfectly matched,it should be made resonant, even thoughthe amplitude of the standing waves (voltagevariation) is not particularly great. This preventsreactance from being coupled into thefinal amplifier. A feed system having moderatestanding waves may be made to present a nonreactiveload to the amplifier either by tuningor by pruning the feeders to approximate resonance.Usually it is preferable with tuned feedersto have a current loop (voltage minimum) at thetransmitter end of the line. This means thatwhen voltage- feeding an antenna, the tunedfeeders should be made an odd number of quarterwavelengths long, and when current-feedingan antenna, the feeders should be made aneven number of quarter wavelengths long. Actually,the feeders are made about 10 per cent ofa quarter wave longer than the 4:alculatedvalue (the value given in the tables) whenthey are to be series tuned to resonance bymeans of a capacitor, instead of being trimmedand pruned to resonance.When tuned feeders are used to feed an antennaon more than one band, it is necessaryto compromise and make provision for bothseries and parallel tuning, inasmuch as it isimpossible to cut a feeder to a length thatwill be optimum for several bands. If a voltageloop appears at the transmitter end of the lineon certain bands, parallel tuning of the feederswill be required in order to get a transferof energy. It is impossible to transfer energyby inductive coupling unless current is flowing.This is effected at a voltage loop by thepresence of the resonant tank circuit formedby parallel tuning of the antenna" coil.21-12 Line DiscontinuitiesIn the previous discussion we have assumeda transmission line which was uniform throughoutits length. In actual practice, this isusually not the case.Whenever there is any sudden change in thecharacteristic impedance of the line, partialreflection will occur at the point of discontinuity.Some of the energy will be transmittedand some reflected, which is essentially thesame as having some of the energy absorbedand some reflected in so far as the effect uponthe line frolll the generator to that point isconcerned. The discontinuity can by ascribeda reflection coefficient just as in the case ofan unmatched load.In a simple case, such as a finite length ofuniform line having a characteristic impedanceof 5 00 ohms feeding into an infinite length ofuniform line having a characteristic impedanceof 100 ohms, the behavior is easily predicted.The infinite 100 ohm lin~ will have no standingwaves and will accept the same power from the500 ohm line as would a 100 ohm resistor,and the rest of the energy will be reflected atthe discontinuity to produce standing wavesfrom there back to the generator. However, inthe case of a complex discontinuity placed atan odd distance down a line terminated in acomplex impedance, the picture becomes complicated,especially when the discontinuity isneither sudden nor gradual, but intermediatebetween the two. This is the usual case withamateur lines that must be erected aroundbuildings and trees.In any case, when a discontinuity existssomewhere on a line and is not a smooth,gradual change embracing several wavelengths,it is not possible to avoid standing wavesthroughout the entire length of the line. If thediscontinuity is sharp enough and is greatenough to be significant, standing waves mustexist on one side of the discontinuity, andmay exist on both sides in many cases.
CHAPTER TWENTY-TWOAntennas and Antenna MatchingAntennas for the lower frequency portion ofthe h-f spectrum (perhaps from 1.8 to 7.0 Me.),and temporary or limited use antennas for theupper portion of the h-f range, usually are ofa relatively simple type in which directivityis not a prime consideration. Also, it often isdesirable, in amateur work, that a single antennasystem be capable of operation at leaston the 3.5-Mc. and 7.0-Mc. range, and preferablyon other frequency ranges. Consequently,the first portion of this chapter will be devotedto a discussion of such antenna systems.The latter portion of the chapter is devotedto the general problem of matching theantenna transmission line to antenna systemsof the fixed type. Matching the antenna transmissionline to the rotatable directive arrayis discussed in Chapter Twenty-five.22-1 End-Fed Half-WaveHorizontal AntennasThe half-wave horizontal dipole is the mostcommon and the most practical antenna for the3.5-Mc. and 7-Mc. amateur bands. The form ofthe dipole, and the manner in which it is fedare capable of a large number of variations.Figure 2 shows a number of practicable formsof the simple dipole antenna along with methodsof feed.426Usually a high-frequency doublet is mountedas high and as much in the clear as possible,for obvious reasons. However, it is sometimesjustifiable to bring part of the radiating systemdirectly to the transmitter, feeding the antennawithout benefit of a transmission line.This is permissible when ( 1) there is insufficientroom to erect a 75- or SO-meter horizontaldipole and feed line, (2) when a long wireis also to be operated on one of the higherfrequency bands on a harmonic. In either case,it is usually possible to get the main portionof the antenna in the clear because of itslength. This means that the power lost bybringing the antenna directly to the transmitteris relatively small.Even so, it is not best practice to bring thehigh-voltage end of an antenna into the operatingroom because of the increased difficultyin eliminating BCI and TVI. For this reasonone should dispense with a feed line in conjunctionwith a Hertz antenna only as a lastresort.End-FedAntennasThe end-fed antenna has no formof transmission line to couple itto the transmitter, but brings theradiating portion of the antenna right down tothe transmitter, where some form of couplingsystem is used to transfer energy to the antenna.Figure 1 shows two common methods offeeding the Fuchs antenna or end-fed Hertz.
\FROM TRANSMITTERI_____ .,IANY NUMBER OF HALF-WAVES--!ANY EVEN NUMBl:R OF QUARTER-WAY SCenter-Fed Antennas 427retuning the feeders. The overall efficiency ofthe zepp antenna system is not quite as highfor long feeder lengths as for some of the an·tenna systems which employ non-resonanttransmission lines, but where space is limitedand where operation on more than one bandis de sired, the zepp has some decided advantages.As the radiating portion of the zepp antennasystem must always be some multiple of ahalf wave long, there is always high voltagepresent at the point where the live zepp feed·er attaches to the end of the radiating portionof the antenna. Thus, this type of zepp antennasystem is voltage fed.Figure 1<strong>THE</strong> END-FED HERTZ ANTENNAShowing the manner in which on encl-fec/ Hertzantenna may be feel through a low-impeclonceline one/ low-pass filter by using a resonanttonk circuit as at (A), or through the use ofa reverse-connectecl pi network os at (B).Some harmonic-attenuating provision (in additionto the usual low-pass TVI filter) must beincluded in the coupling system, as an endfedantenna itself offers no discriminationagainst harmonics, either odd or even.The end-fed Hertz antenna has rather highlosses unless at least three-quarters of theradiator can be placed outside the operatingroom and in the clear. As there is r·f voltageat the point where the antenna enters theoperating room, the insulation at that pointshould be several times as effective as theinsulation commonly used with low-voltagefeeder systems. This antenna can be operatedon all of its higher harmonics with good efficiency,and can be operated at half frequencyagainst ground as a quarter-wave Marconi.As the frequency of an antenna is raisedslightly when it is bent anywhere except at avoltage or current loop, an end-fed Hertz antennausually is a few per cent longer than astraight half-wave doublet for the same frequency,because, ordinarily, it is impracticalto bring a wire in to the transmitter withoutmaking several bends.The Zepp AntennaSystemThe zeppelin or zepp antennasystem, illustratedin figure 2A is very con·venient when it is desired to operate a singleradiating wire on a number of harmonically relatedfrequencies.The zepp antenna system is easy to tune,and can be used on several bands by merelyStub-Fed ZeppType RadiatorFigure 2C shows a modificationof the zepp-type antennasystem to allow the use ofa non-resonant transmission line between theradiating portion of the antenna and the transmitter.The zepp portion of the antenna isresonated as a quarter-wave stub and the nonresonantfeeders are connected to the stub ata point where standing waves on the feederare minimized. The procedure for making theseadjustments is described in detail in Section22-8 This type of antenna system is quitesatisfactory when it is necessary physicallyto end feed the antenna, but where it is neces·sary also to use non-resonant feeder betweenthe transmitter and the radiating system.22-2 Center-Fed HalfWave Horizontal AntennasThe center feeding of a half-wave antennasystem is usually to be desired over an endfedsystem since the center-fed system is inherentlybalanced to ground and is thereforeless likely to be troubled by feeder radiation.A number of center-fed systems are illustratedin figure 2.The Tuned The current-fed do u b 1 e t withDoublet spaced feeders, sometimescalled a center-fed zepp, is aninherently balanced system if the two legs ofthe radiator are electrically equal. This factholds true regardless of the frequency, or ofthe harmonic, on which the system is oper·ated. The system can successfully be operatedover a wide range of frequencies if thesystem as a whole (both tuned feeders and thecenter-fed flat top) can be resonated to theoperating frequency. It is usually possible totune such an antenna system to resonancewith the aid of a tapped coil and a tuning ca·pacitor that can optionally be placed either
428 Antennas and Antenna Matching<strong>THE</strong> R AD I 0t~-------1STUB-FED©END-FED TYPES@= TUNED DOUBLET~0.9> X/2-------l !--o.9> '-'2----t-300-600.C.. LINE~300-6000HMLINEOPENQUARTER-WAVESTUB-FEDt+--~0.9> X/2----ir------ 0.9> X/2 ______,f--- 0.9> X/2 ----115011 TWINLEAD0.193 OF FREESPACE WAVELENGTH OR0. 77 OF A/4Figure 2ALTERNATIVEMETHODS OFFEEDING AHALF-WAVE DIPOLE-,30011 TWIN LEAD~~~~~~~~PENED ®2"'0R6"FEEDERSPREADERSTWINLEAD 2-WIRE DOUBLET DELTA·MATCHEO"FOLDED DIPOLE" OR ~FOLDED DIPOLE" DOUBLETf---o.9> X/2 ----1- ......FOR DELTADIMENSIONSSEE CHAP.19300 OHM TWIN LEAD 3000HM TWIN LEAD 600 OHM LINEANY LENGTH ANY LENGTH ANY LENGTH~DOUBLET®CO-AX FEDDOUBLET@OFF-CENTFEDo=14':bOFTOTAL LENGTHi 75 fi TWIN LEAD. ANY LENGTHN°14WIRECENTER-FED TYPES
<strong>THE</strong> R AD I 0Multi-wire Doublets 429in series with the antenna coil or in parallelwith it. A series tuning capacitor can beplaced in series with one feeder leg withoutunbalancing the system.The tuned-doublet antenna is shown in figure2D. The antenna is a current-fed systemwhen the radiating wire is a half wave longelectrically, or when the system is operatedon its odd harmonics, but becomes a voltagefedradiator when operated on its even harmonics.The antenna has a different radiation patternwhen operated on its harmonics, as wouldbe expected. The arrangement u sed on thesecond harmonic is better known as the Franklincolinear array and is described in ChapterTwenty-three. The pattern is similar to a Yrwavedipole except that it is sharper in the broadsidedirection. On higher harmonics of operationthere will be multiple lobes of radiationfrom the system.Figures 2E and 2F show alternative arrangementsfor using an untuned transmission linebetween the transmitter and the tuned-doubletradiator. In figure 2E a half-wave shorted lineis used to resonate the radiating system,while in figure 2F a quarter-wave open line isutilized. The adjustment of quarter-wave andhalf-wave stubs is discussed in Section 19-8.Doublets withQuarter-WaveTransformersThe average value of feed impedancefor a center-fed halfwavedoublet .ls 75 ohms. Theactual value varies with heightand is shown in Chapter Twenty-one. Othermethods of matching this rather low value ofimpedance to a medium-impedance transmissionline are shown in (G), (H), and (I) of figure2. Each of these three systems uses aquarter-wave transformer to accomplish theimpedance transformation. The only differencebetween the three systems lies in the type oftransmission line used in the quarter-wavetransformer. (G) shows the Johnson Q systemwhereby a line made up of Yz-inch dural tubingis used for the low-impedance linear transformer.A line made up in this manner is frequentlycalled a set of Q bars. Illustration(H) shows the use of a four-wire line as thelinear transformer, and (I) shows the use of apiece of 150-ohm Twin-Lead electrically ~wave in length as the transformer between thecenter of the dipole and a piece of 300-ohmTwin-Lead. In any case the impedance of thequarter-wave transformer will be of the orderof 150 to 200 ohms. The use of sections oftransmission line as linear transformers isdiscussed in detail in Section 22-8.Multi-WireDoubletsAn alternative method for increasingthe feed-point impedance of adipole so that a medium-imped-ance transmission line may be used is shownin figures 2J and 2K. This system utilizesmore than one wire in parallel for the radiatingelement, but only one of the wires is brokenfor attachment of the feeder. The most commonarrangement uses two wires in the flattop of the antenna so that an impedance multiplicationof four is obtained.The antenna shown in figure 2J is the socalledTwin-Lead folded dipole which is acommonly used antenna system on the mediumfrequencyamateur bands. In this arrangementboth the antenna and the transmission line tothe transmitter are constructed of 300-ohmTwin-Lead. The flat top of the antenna ismade slightly less than the conventionallength (462/FMc. instead of 468/FMc. for asingle-wire flat top) and the two ends of theTwin-Lead are joined together at each end.The center of one of the conductors of theTwin-Lead flat top is broken and the two endsof the Twin-Lead feeder are spliced into theflat top leads. As a protection against moisturepieces of flat polyethylene taken fromanother piece of 300-ohm Twin-Lead may bemolded over the joint between conductors withthe aid of an electric iron or soldering iron.Better bandwidth characteristics can be obtainedwith a folded dipole made of ribbon lineif the two conductors of the ribbon line areshorted a distance of 0.82 (the velocity factorof ribbon line) of a free-space quarter wavelengthfrom the center or feed point. This procedureis illustrated in figure 3A. An alternativearrangement for a Twin-Lead foldeddipole is illustrated in figure 3B. This type ofhalf-wave antenna system is convenient foruse on the 3.5-Mc. band when the 116 to 132foot distance required for a full half-wave isnot quite available in a straight line, since thesingle-wire end pieces may be bent away ordownward from the direction of the main sec·tion of the antenna.Figure 2K shows the basic type of 2-wiredoublet or folded dipole wherein the radiatingsection of the system is made up of standardantenna wire spaced by means of feederspreaders. The feeder again is made of 300-ohm Twin-Lead since the feed-point impedanceis approximately 300 ohms, the same asthat of the Twin-Lead folded dipole.The folded-dipole type of antenna has thebroadest response characteristic (greatestbandwidth) of any of the conventional halfwaveantenna systems constructed of smallwires or conductors. Hence such an antennamay be operated over the greatest frequencyrange without serious standing waves of anycommon half-wave antenna type.The increased bandwidth of the multi-wiredoublet type of radiator, and the fact that thefeed-point resistance is increased sever a 1
l430 Antennas and Antenna Matching<strong>THE</strong>R AD I 0i·@r-300-0HM RIBBONr~4iIlL.l VERTICALii}~ED®STUB-FEDVERTICALTII094>.12©L-C- FEDVERTICAL1-------- 404 ---------1fMc.@Figure 3----300-0HM RIBBONFOLDED DIPOLE WITH SHORTINGSTRAPS\30FMC.lThe impeclance match one/ banclwiclth characteristicsof a folclecl clipole may be improveclby shorting the two wires of the ribbon a c/is·tance out from the center equal to the veloci·ty factor of the ribbon times the half-lengthof the clipole as shown at (A}. An alternativearrangement with bent clown encls for spaceconservation is illustratecl at (B).times over the radiation resistance of the ele·ment, have both contributed to the frequentuse of the multi-wire radiator as the drivenelement in a parasitic antenna array.Delta-MatchedDoublet andStandard DoubletThese two types of radiat·ing elements are shown infigure 2L and figure 2M. Thedelta-matched do ubI e t isdescribed in detail in Section 22·8 of thischapter. The standard doublet, shown in figure2M, is fed in the center by means of 75·ohm Twin-Lead, either the transmitting or thereceiving type, or it may be fed by means oftwisted-pair feeder or by means of parallel·wire lamp-cord. Any of these types of feedline will give an approximate match to thecenter impedance of the dipole, but the 75-ohm Twin-Lead is far to be preferred over theother types of low-impedance feeder due tothe much lower losses of the polyethylene·dielectric transmission line.The coaxial-cable-fed doublet shown in figure2N is a variation on the system shown infigure 2M. Either 52-ohm coaxial cable or 75-ohm coaxial cable may be used to feed thecenter of the dipole, although the 75-ohm typeFigure 4HALF-WAVE VERTICAL ANTENNA SHOWING ALTERNATIVE METHODS OF FEEDwill give a somewhat blj,tter impedance matchat normal antenna heights. Due to the asym·metty of the coaxial feed system difficultymay be encountered with waves traveling onthe outside of the coaxial cable. For this reasonthe use of Twin-Lead is normally to bepreferred over the use of coaxial c a b I e forfeeding the center of a half-wave dipole.Off-CenterFed DoubletThe system shown in figure2 (0) is sometimes used tofeed a half-wave dipole, especiallywhen it is desired to use the same antennaon a number of harmonically-related fre·quencies. The feeder wire (no. 14 enamelledwire should be used) is tapped a distance of14 per cent of the total length of the antennaeither side of center. The feeder wire, operatingagainst ground for the return current, hasan impedance of approximately 600 ohms. Thesystem works well over high I y conductingground, but will introduce rather high losseswhen the antenna is located above rocky orpoorly conducting soil. The off-center fed antennahas a further disadvantage that it ishighly responsive to harmonics fed to it fromthe transmitter.The effectiveness of the antenna system inradiating harmonics is of course an advantagewhen operation of the antenna on a number offrequency bands is desired. But it is neces·sary to use a harmonic filter to insure thatonly the desired frequency is fed from thetransmitter to the antenna.22-3 The Half-WaveVertical AntennaThe half-wave vertical antenna with its bot·tom end from 0.1 to 0.2 wavelength above
<strong>HANDBOOK</strong>Vertical Antennas 4 31~2 OHM COAXIAL. LINE.CENTER CONDUCTOR CONNECTSTO VERTICAL WHIP52 OHM COAXIAL LINE_.!1 FEET LON«;Figure 5<strong>THE</strong> LOW-FREQUENCY GROUND PLANEANTENNAThe radials of the ground plane antennashould lie in a horizontal plane, althoughslight departures from this caused by nearbyobjects is allowable. The whip may bemounted on a short post, or on the roof of abuilding, The wire radials may slope clown•wards towards their tips, acting as guywires for the installation,ground is an effective transmitting antenna forlow-angle radiation, where ground conditionsin the vicinity of the antenna are good. Suchan antenna is not good for short-range skywavecommunication, such as is the normalusage of the 3. 5-Mc. amateur band, but is excellentfor short-range ground-wave communicationsuch as on the standard broadcast bandand on the amateur 1.8-Mc. band. The verticalantenna normally will cause greater BCI thanan equivalent horizontal antenna, due to themuch greater ground-wave field intensity. Also,the vertical antenna is poor for receivingunder conditions where man-made interferenceis severe, since such interference is predominantlyvertically polarized.Three ways of feeding a half-wave verticalantenna from an untuned transmission line areillustrated in figure 4. The J-fed system shownin figure 4A is obviously not practicable excepton the higher frequencies where the extralength for the stub may easily be obtained.However, in the normal case the ground-planevertical antenna is to be recommended overthe J -fed system for high frequency work.22-4 The Ground Plane AntennaAn effective low angle radiator for any ama-Figure 680 METER LOADED GROUND PLANEANTENNANumber of turns in loacllng coil to be acljustecluntil antenna system resonates at cleslreclfrequency in 80 meter bane/,teur band is the ground-plane antenna, shownin figure 5. So called because of the radialground wires, the ground-plane antenna is notaffected by soil conditions in its vicinity dueto the creation of an artificial ground systemby the radial wires. The base impedance ofthe ground plane is of the order of 30 to 35ohms, and it may be fed with 52-ohm coaxialline with o.Vy a slight impedance mis-match.For a more exact match, the ground-plane antennamay be fed with a 72-ohm coaxial lineand a quarter-wave matching section made of52-ohm coaxial line.The angle of radiation of the ground-planeantenna is quite low, and the antenna will befound less effective for contacts under 1000miles or so on the 80 and 40 meter bands thana high angle radiator, such as a dipole. However,for DX contacts of 1000 miles or more,the ground-plane antenna will prove to behighly effective.The SO-MeterLoadedGround-PlaneA vertical antenna of 66 feetin height presents quite a problemon a small lot, as the supportingguy wires will tend totake up quite a large portion of the lot. Undersuch conditions, it is possible to shorten thelength of the vertical radiator of the groundplaneby the inclusion of a loading coil in thevertical whip section. The ground-plane antennamay be artificially loaded in this mannerso that a 25-foot vertic a 1 whip may be
432 Antennas and Antenna Matching<strong>THE</strong> R AD I 0used for the radiator. Such an antenna isshown in figure 6. The loaded ground-planetends to have a rather high operating Q andoperates only over a narrow band of frequencies.An operating range of about 100 kilocycleswith a low SWR is possible on 80 me·ters. Operation over a larger frequency rangeis possible if a .higher standing wave ratio istolerated on the transmission line. The radiationresistance of a loaded SO-meter groundplaneis about 15 ohms. A quarter wavelength( 45 feet) of 52-ohm coaxial line will act as anefficient feed line, presenting a load of approximately180 ohms to the transmitter.22-5 The MarconiAntennaA grounded quarter-wave Marconi antennawidely used on frequencies below 3 Me., i~sometimes used on the 3.5-Mc. band, and isalso used in v-h-f mobile services where acompact antenna is required. The Marconi typeantenna allows the use of half the length ofwire that would be required for a half-wave!fertz radiator. The ground acts as a mirror,In effect, and takes the place of the additionalquarter-wave of wire that would be requiredto reach resonance if the end of the wire werenot returned to ground.The fundamental practical form of the Marconiantenna system is shown in figure 7.Other Marconi antennas differ from this typeprimarily in regard to the method of feedingthe energy to the radiator. The feed methodshown in figure 7B can often be used to advantage,particularly in mobile work.Variations on the basic Marconi antennaare shown in the illustrations of figure 8. FiguresSB and SC show the "L "-type and "T"type Marconi antennas. These arrangementshave been more or less superseded by the toploadedforms of the Marconi antenna shown infigures SD, BE, and SF. In each of these latterthree figures an antenna somewhat lessthan one quarter wave in 1 eng t h has beenloaded to increase its effective length by theinsertion of a loading coil at or near the topof the radiator. The arrangement shown at figureSD gives the least loading but is the mostpractical mechanically. The system shown atfigure BE gives an intermediate amount ofloading, while that shown at figure SF, utilizinga "hat" just above the loading coil, givesthe greatest amount of loading. The object ofall the top-loading methods shown is to producean increase in the effective length ofthe radiator, and thus to raise the point ofmaximum current in the radiator as far as pos-Figure 7TICOAX. f'ROM TRANS.®FEEDING A QUARTER-WAVE MARCONIANTENNAWhen an open-wire line is to be usee/ it maybe I ink coupled to a series-resonan; circuitbetween the boHom encl of the Marconi anclground, as at (A). Alternatively, a reason·ably goocl impedance match may be obtainedbetween 52-ohm coaxial line ancl the bottomof a resonant quarter-wave antenna, as ill liS•tratecl at (B) above.sible above ground. Raising the maximum-currentpoint in the radiator above ground hastwo desirable results: The percentage of lowangleradiation is increased and the amount ofground current at the base of the radiator isreduced, thus reducing the ground losses.To estimate whether a 1 o ad in g coil willproba?ly be required, it is necessary only tonote If the length of the antenna wire andground lead is over a quarter wavelength; ifso, no loading coil is needed, provided theseries tuning capacitor has a high maximumcapacitance.Amateurs primarily interested in the higherfrequency bands, but who like to work 80 metersoccasionally, can usually manage to resonateone of their antennas as a Marconi byworking the whole system, feeders and allagainst a water pipe ground, and resorting t~a loading coil if necessary. A high-frequencyrotary,zepp, doublet, or single-wire-fed antennawill make quite a good SO-meter Marconiif high and in the clear, with a rather longfeed line to act as a radiator on 80 meters.Where two-wire feeders are used, the feedersshould be tied together for Marconi operation.Importance ofGround ConnectionWith a quarter-wave antennaand a ground, the antennacurrent generally ismeasured with a meter placed in the antennacircuit close to the ground connection. If this
<strong>HANDBOOK</strong>Marconi Antenna 4 33Figure 8LOADING <strong>THE</strong>MARCONI ANTENNAThe various loading systemsare cliscussec/ in the accomponyingtext.I.!>..4l-::'®r I T.l>. .l>.4 4LESSl lTHAN-tjLOADING.COILS.. HAT"'-= -::'-= -= -::'@ © @ © ®current flows through a r e s i s tor, or if theground itself presents some resistance, therewill be a power loss in the form of heat. Improvingthe ground connection, therefore, providesa definite means of reducing this powerloss, and thus increasing the radiated power.The best possible ground consists of asmany wires as possible, each at least a quarterwave long, buried just below the surfaceof the earth, and extending out from a commonpoint in the form of radials. Copper wire ofany size larger than no. 16 is satisfactory,though the larger sizes will take longer to disintegrate.In fact, the radials need not evenbe buried; they may be supported just abovethe earth, and insulated from it. This arrangementis called a counterpoise, and operatesby virtue of its high capacitance to ground.If the antenna is physically shorter than aquarter wavelength, the antenna current ishigher, due to lower radiation resistance. Consequently,the power lost in resistive soil isgreater. The importance of a good ground withshort, inductive-loaded Marconi radiators is,therefore, quite obvious. With a good groundsystem, even very short (one-eighth wavelength)antennas can be expected to give ahigh percentage of the efficiency of a quarterwaveantenna used with the same ground system.This is especially true when the shortradiator is top loaded with a high Q (low loss)coil.Water-PipeGroundsWater pipe, because of its comparativelylarge surface and crosssection, has a relatively low r-fresistance. If it is possible to attach to ajunction of several water pipes (where theybranch in several directions and run for somedistance under ground), a satisfactory groundconnection will be obtained. If one of thepipes attaches to a lawn or garden sprinklersystem in the immediate vicinity of the antenna,the effectiveness of the system will approachthat of buried copper radials.The main objection to water-pipe groundsis the possibility of high resistance joints inthe pipe, due to the "dope" put on the couplingthreads. By attaching the ground wireto a junction with three or more legs, the possibilityof requiring the main portion of ther-f current to flow through a high resistanceconnection is greatly reduced.The presence of water in the pip e addsnothing to the conductivity; therefore it doesnot relieve the problem of high resistancejoints. Bonding the joints is the best insurance,but this is, of course, impracticablewhere the pipe is buried. Bonding togetherwith copper wire the various water faucetsabove the surface of the ground will improvethe effectiveness of a water-pipe ground systemhampered by high-resistance pipe couplings.MarconiDimensionsA Marconi antenna is an oddnumber of electrical quarterwaves long (usually only onequarter wave in length), and is always resonatedto the operating frequency. The correctloading of the final amplifier is accomplishedby varying the coupling, rather than by detuningthe antenna from resonance.Physically, a quarter-wave Marconi may bemade anywhere from one-eighth to three-eighthswavelength overall, meaning the total length ofthe antenna wire and ground lead from the endof the antenna to the point where the groundlead attaches to the junction of the radials orcounterpoise wires, or where the water pipeenters the ground. The longer the antennais made physically, the lower will be the currentflowing in the ground connection, and thegreater will be the overall radiation efficiency.However, when the antenna length exceedsthree-eighths wavelength, the antenna becomesdifficult to resonate by me an s of aseries capacitor, and it begins to take shapeas an end-fed Hertz, requiring a method offeed such as a pi network.A radiator physically much shorter than a
434 Antennas and Antenna Matching<strong>THE</strong>R AD I 0®6"FEEOER'SPREAOERS-.2"Ft:EOERSPREADERSIWIRES SHORTED TOGE<strong>THE</strong>R AT ENDRESONANT ORNON-RESONANT LINERESONANT L!NI!©~FT.T_£!_FT.Ft.~c._LFigure 9300 .0. TWINLEAOTHREE EFFECTIVE SPACE CONSERVINGANTENNASThe arrangements shown at (A) and (B) aresatisfactory where resonant leerl line can beused. However, non-resonant 75-ohm feedline may be used in the arrangement at (A)when the dimensions in wavelengths are asshown. In the arrangement shown at (B) lowstanding waves will be obtained on the feedline when the overall length of the antennais a half wave. The arrangement shown at(C) may be tuned for any reasonable lengthof flat top to give a minimum of standingwaves on the transmission I in e.quarter wavelength can be lengthened elec·trically by means of a series loading coil, andused as a quarter-wave Marconi. However, ifthe wire is made shorter than approximatelyone-eighth wavelength, the radiation resist·ance will be quite low. This is a special prob·lem in mobile work below about 20-Mc.22-6 Space-ConservingAntennasIn many cases it is desired to undertake aconsiderable amount of operation on the SOmeteror 40-meter band, but sufficient spaceis simply not available for the installation ofa half-wave radiator for the desired frequencyof operation. This is a common experience ofapartment dwellers. The shortened Marconiantenna operated against a good ground canbe used under certain conditions, but the shortenedMarconi is notorious for the productionof broadcast interference, and a good groundconnection is usually completely unobtainablein an apartment house.Figure 10TWIN-LEAD MARCONI ANTENNA FOR <strong>THE</strong>80 AND 160 METER BANDSEssentially, the problem in producing anantenna for lower frequency operation in re·stricted space is to erect a short radiatorwhich is balanced with respect to ground andwhich is therefore independent of ground forits operation. Several antenna types meetingthis set of conditions are shown in figure 9.Figure 9A shows a conventional center-feddoublet with bent-down ends. This type of an·tenna can be fed with 75-ohm Twin-Lead in thecenter, or it may be fed with a resonant linefor operation on several bands. The overalllength of the radiating wire will be a few percent greater than the normal length for suchan antenna since the wire is bent at a posi·tion intermediate between a current loop anda voltage loop. The actual length will have tobe determined by the cut-and-try process becauseof the increased effect of interfering objectson the effective electrical length of anantenna of this type.Figure 9B shows a method for using a two·wire doublet on one half of its normal operat·ing frequency. It is recommended that spacedopen conductor be used both for the radiatingportion of the folded dipole and for the feedline. The reason for this recommendation liesin the fact that the two wires of the flat topare not at the same potential throughout theirlength when the antenna is operated on one·half frequency. Twin-Lead may be used forthe feed line if operation on the frequencywhere the flat top is one-half wave in lengthis most common, and operation on one-half fre·quency is infrequent. However, if the antennais to be used primarily on one-half frequencyas shown, it should be fed by me an s of anopen-wire line. If it is desired to feed the an·tenna with a non-resonant line, a quarter-wavestub may be connected to the antenna at thepoints X, X in figure 9B. The stub should betuned and the transmission line connected toit in the normal manner.The antenna system shown in figure 9C maybe used when not quite enough length is avail·able for a full half-wave radiator. The dimen-
<strong>HANDBOOK</strong> Space Conserving Antennas 4351------------ 64.$' ------------~FIGURE ACUTOFF SHIELD AND OUTERJACKET A.S SHOWN. ALLOWDIELECTRIC TO EXTEND PARTWAY TO O<strong>THE</strong>R CABLE. COVERALL EXPOSED SHIEL.D ANDDIELECTRIC ON BOTH CABLESWITH A CONTINUOUS WRAPPING OF SC.OTCH ELECTRICALTAPE TO EXCLUDE MOISTURE.FOR DETAIL SEE FIGURE 852 OHhA R~-8/U, ANY LENGTHREMOVE OUTER JACKETFROM A SHORT LENGTH OFCABLE AS SHOWN HERE.UNBRAIO <strong>THE</strong> SHIELD OFCOAX C. CUTOFF <strong>THE</strong> DIELECTRIC AND INNER CONDUCTOR fLUSH WITH <strong>THE</strong>OUTER JACKET. 00 NOT CUT<strong>THE</strong> SHIELD. WRAP SHIELDOF COAX C AROUND SHIELDOF COAX D. SOLDER <strong>THE</strong>CONNECTION, BEING VERYCAREfUL NOT TO DAMAGE<strong>THE</strong> DIELECTRIC MATERIALHOLD CABL.E 0 STRAIGHTWHILE SOLDERING. COVER<strong>THE</strong> AREA WITH A CONTINUOUS WRAPPING OF SCOTCHELECTRICAL TAPE. NO CONNECTION TO INNER CONDUCTORS.FOR DETAIL SEE FIGURE BDIMENSIONS SHOWN HERE ARE FOR <strong>THE</strong> 40 METER BAND. THIS ANTENNA MAY BE BUILT FOR O<strong>THE</strong>R BANDS BY USING DIMENSIONS THATARE MULTIPLES OR SUBMULTIPLES OF <strong>THE</strong> DIMENSIONS SHOWN~BALUN SPACING IS 1.5• ON ALL BAND$.Figure 11HALF-WAVE ANTENNA WITH QUARTER·WAVE UNBALANCED TO BALANCEDTRANSFORMER (BALUN) FEED SYSTEMFOR 40-METER OPERATIONDIMENSIONS SHOWN HERE ARE FOR <strong>THE</strong> MJ METER BAND. THIS ANTENNA MofY BE BUILT FOR CT<strong>THE</strong>R BANOS BY USING DIMENSIONS THATARE MULTIPLES OR SUBMULTIPLES OF <strong>THE</strong> DIMENSIONS SHOWN.BALUNSPACIN~ IS 1.5•0N ALL BANDS.Figure 12BROADBAND ANTENNA WITH QUARTER·WAVE UNBALANCED TO BALANCEDTRANSFORMER (BALUN) FEED SYSTEMFOR 80-METER OPERATIONsions in terms of frequency are given on thedrawing. An antenna of this type is 93 feetlong for operation on 3600 kc. and 86 feet longfor operation on 3900 kc. This type of antennahas the additional advantage that it may beoperated on the 7-Mc. and 14-Mc. bands, whenthe flat top has been cut for the 3.5-Mc. band,simply by changing the position of the shortingbar and the feeder line on the stub.A sacrifice which must be made when usinga shortened radiating system, as for examplethe types shown in figure 9, is in the bandwidthof the radiating system. The frequencyrange which may be covered by a shortenedantenna system is approximately in proportionto the amount of shortening which has beenemployed. For example, the antenna systemshown in figure 9C may be operated over therange from 3800 kc. to 4000 kc. without seriousstanding waves on the feed line. If theantenna had been made full length it wouldbe possible to cover about half again as muchfrequency range for the same amount of mismatchon the extremes of the frequency range.The Twin-LeadMarconi AntennaMuch of the power loss inthe Marconi antenna is a resultof low radiation resistanceand high ground resistance. In somecases, the ground resistance may even bebe higher than the radiation resistance, causinga loss of 50 per cent or more of the transmitterpower output. If the radiation resistanceof the Marconi antenna is raised, the amountof power lost in the ground resistance is proportionatelyless. If a Marconi antenna is madeout of 300 ohm TV-type ribbon line, as shownin figure 10, the radiation resistance Qf theantenna is raised from a low value of 10 or 15ohms to a more reasonable value of 40 to 60
436 Antennas and Antenna Matching<strong>THE</strong> R A 0 I 0~:1 f'bl bfd3.5 3.0 3. 7 3.8 3.8 4.0FREQUENCY ("c)r----ANT£NNA~PHENOL.IC Bl..OCKS,-t-tiJ"'·/,1SEE nc;. 12.80 METERSL = 13'&"C = 400.lJJJF40 t.4£TERSL=7'3"C ::: 2.00 .U.U FFigure 13SWR CURVE OF 80-METER BROAD-BANDDIPOLESEE FIG.I2 FOR CONNECTIONohms. The ground losses are now reduced bya factor of 4. In addition, the antenna may bedirectly fed from a 50-ohm coaxial line, or di·rectly from the unbalanced output of a pi· net·work transmitter.Since a certain amount of power may stillbe lost in the ground connection, it is still ofgreatest importance that a good, low resist·ance ground be used with this antenna.The Collins Shown in figures 11 and 12Broad-band are broad-band dipoles forDipole System the 40 and 80 meter amateurbands, designed by CollinsRadio Co. for use with the Collins 32V·3 andKW-1 transmitters. These fan-type dipoleshave excellent broad-band response, and aredesigned to be fed with a 52-ohm unbalancedcoaxial line, making them suitable for use withmany of the other modern transmitters, suchas the Barker and Williamson 5100, JohnsonRanger, and Viking. The antenna system con·sists of a fan-type dipole, a balun matchingsection, and a suitable coaxial feedline. TheQ of the half-wave 80 meter doublet is low·ered by decreasing the effective length-todiameterratio. The frequency range of operationof the doublet is increased considerablyby this change. A typical SWR curve for the80 meter doublet is shown in figure 13.The balanced doublet is matched to the un·balanced coaxial line by the one-quarter wavebalun. If desired, a shortened balun may beused (figure 14). The short balun is capacityloaded at the junction between the balun andthe broad-band dipole.22-7 Multi-Band AntennasThe availability of a multi-band antenna isa great operating convenience to an amateurstation. In most cases it will be found best toinstall an antenna which is optimum for theband which is used for the m a j or it y of theFigure 14SHORT BALUN FOR 40 AND 80 METERSavailable operating time, and then to have anadditional multi-band antenna which may bepressed into service for operation on anotherband when propagation conditions on the mostfrequently used band are not suitable. Mostamateurs use, or plan to install, at least onedirective array for one of the higher-frequencybands, but find that an addi tiona! antennawhich may be used on the 3. 5-Mc. and 7 .0-Mc.band, or even up through the 28-Mc. band isalmost indispensable.The choice of a multi-band antenna dependsupon a number of factors such as the amountof space available, the band which is to beused for the majority of operating with the antenna,the radiation efficiency which is desired,and the type of antenna tuning networkto be used at the transmitter. A number ofrecommended types are shown in the nextpages.The l4-WaveFolded DoubletFigure 15 shows an antennatype which will be found tobe very effective when amoderate amount of space is available, whenmost of the operating will be done on one bandwith occasional operation on the second harmonic.The system is quite satisfactory foruse with high-power transmitters since a 600-ohm non-resonant line is used from the antennato the transmitter and since the antennasystem is balanced with respect to ground.With operation on the fundamental frequencyof the antenna where the flat top is % wavelong the switch SW is left open. The systemaffords a very close match between the 600-ohm line and the feed point of the antenna.Kraus has reported a standing-wave ratio ofapproximately 1.2 to 1 over the 14-Mc. bandwhen the antenna was located approximatelyone-half wave above ground.For operation on the second harmonic theswitch SW is closed. The antenna is still an
<strong>HANDBOOK</strong>Multi-band Antennas 4 37L =HI!>' FOR 3~!>0 KC. ANO 71!>0 KC.L::: 98' FOR 7100 KC. AND 142!>0 KC.L"49.6' FOR 14200 KC. AND 26MC6000HM LINE:TO TRANSMITTERFigure 15<strong>THE</strong> THREE-QUARTER WAVE FOLDEDDOUBLETThis antenna arrangement will give verysatisfactory operation with a 600-ohm feeclline for operation with the switch open onthe fundamental frequency ancl with theswitch closed on twice frequency.effective radiator on the second harmonic butthe pattern of radiation will be different fromthat on the fundamental, and the standing-waveratio on the feed line will be greater. The flattop of the antenna must be made of open wirerather than ribbon or tubular line.For greater operating convenience, the shortingswitch may be replaced with a section oftransmission line. If this transmission line ismade one-quarter wavelength long for the fundamentalfrequency, and the free end of theline is shorted, it will act as an open circuitacross the center insulator. At the second harmonic,the transmission line is one-half wavelengthlong, and reflects the low impedanceof the shorted end across the center insulator.Thus the switching action is automatic as thefrequency of operation is changed. Such aninstallation is shown in figure 16.The End-FedHertzThe end- f e d Hertz antennashown in figure 17 is not aseffective a radiating system asSHORTED ENDl =67FT. WHEN ANTENNA IS 195 FT.L = 33 FT. " 96FT.L= 16.5 FT. " "A9.6 FT.Figure 16600 OHM LINETO TRANSMIT.,-ERAUTOMATIC BANDSWITCHING STUB FOR<strong>THE</strong> THREE-QUARTER WAVE FOLDEDDOUBLETThe antenna of Figure 15 may be usecl witha shortecl stub line in place of the switchnormally usee/ for seconcl harmonic operation.many other antenna types, but it is particularlyconvenient when it is desired to install anantenna in a hurry for a test, or for field-daywork. The flat top of the radiator should beas high and in the clear as possible. In anyevent at least three quarters of the total wirelength should be in the clear. Dimensions foroptimum operation on various amateur bandsare given in addition in figure 17.The End-FedZeppThe end- fed Zepp has longbeen a favorite for multi-bandoperation. It is shown in figure18 along with recommended dimensionsfor operation on various amateur band groups.~Ll" ~~C FORt·SEETABLEBELOW-·------oi2FROMXMTR.L-SEE BELOW3.5,7,1-t. A.NO Z& MC.3.5, 7 AND 14 MC.3.5 AND 7 MC.3.9 MC. AND 28 MC.L=1ae•L= 137'L= 138'L•1zo•BANDS14 ' "' Me2e Me14 ' "' MeL1L, TYPE OF 1TUNINGSERIES IPARALLEL ISERIES ~IPARALLELPARALLELFigure 17RECOMMENDED LENGTHS FOR <strong>THE</strong> END·-.FED HERTZEND-FED ZEPPFIGURE 18
4 38 Antennas and Antenna MatchingJ'H......L= 90' FOR 10-40 METER OPERATIONJJJJF<strong>THE</strong> R AD I 0~-----------------L~------------------~BANDS L1 L,3.9 MCPHONE' MC14 MCTYPE OFTUNINGPARALLELPARALLELPARALLEL"PARALLELIF 300tl. HIANSMISSION LINE ISUSED FOR L2 <strong>THE</strong> IMPEDANCE ATI~~ET~A~~~~~:~~:TNEDL~:F r-17--1200 OHMS1200 OHMSL~,~~"~'~'~'"~"~'----~~SERIESI~Figure 19A TWO-BAND MARCONI ANTENNA FOR160-80 METER OPERATION' MC14 MC1, MC14 MCPARALLELPARALLELPARALLEL~1200 OHMS1200 OHMSPARALLELPARALL.£1..1a00 OH~S1200 OHMS1200 OHMSSince this antenna type is an unbalanced radiatingsystem, its use is not recommended withhigh-power transmitters where interference tobroadcast listeners is likely to be encountered.The r-f voltages encountered at the end ofzepp feeders and at points an electrical halfwave from the end are likely to be quite high.Hence the feeders should be supported an adequatedistance from surrounding objects andsufficiently in the clear so that a chance encounterbetween a passerby and the feeder isunlikely.The coupling coil at the transmitter end ofthe feeder system should be link coupled tothe output of the low-pass TVI filter in orderto reduce harmonic radiation.The Two-BandMarconi AntennaA three-eighths wavelengthMarconi antenna may beo p era ted on its harmonicfrequency, providing good two band performancefrom a simple wire. Such an arrangementfor operation on 160-80 meters, and 80-40 metersis shown in figure 19. On the fundamental(lowest) frequency, the antenna acts as athree-eighths wavelength series-tuned Marconi.On the second harmonic, the antenna is a current-fedthree-quarter wavelength antenna operatingagainst ground. For proper operation,the antenna should be resonated on its secondharmonic by means of a grid-dip oscillator tothe operating frequency most used on this particularband. The Q of the antenna is relativelylow, and the antenna will perform well over afrequency range of several hundred kilocycles.The overall length of the antenna may bevaried slightly to place its self-resonant frequencyin the desired region. Bends or turnsin the antenna tend to make it resonate higherin frequency, and it may be necessary tolengthen it a bit to resonate it at the chosenfrequency. For fundamental operation, theseries condenser is inserted in the circuit, andthe antenna may be resonated to any point inthe lower frequency band. As with any MarconiDIMENSIONSCENTER-FED ANTENNAFigure 20FOR CENTER-FED MULTIBAND ANTENNAtype antenna, the use of a good ground is essential.This antenna works well with transmittersemploying coaxial antenna feed, sinceits transmitting impedance on both bands is inthe neighborhood of 40 to 60 ohms. It may beattached directly to the output terminal of suchtransmitters as the Collins 32V and the VikingII. The use of a low-pass TVI f i 1 t e r is ofcourse recommended.The Center-FedMulti-Band AntennaFor multi-band operation,the center fed antenna iswithout doubt the bestcompromise. It is a balanced system on allbands, it requires no ground return, and whenproperly tuned has good rejection propertiesfor the higher harmonics generated in the transmitter.It is well suited for use with the variousmulti-band 150-watt transmitters that are currentlyso popular. For proper operation withthese transmitters, an antenna tuning unitmust be used with the center-fed antenna. Infact, some sort of tuning unit is necessary forany type of efficient, multi-band antenna. Theuse of such questionable antennas as the "offcenterfed" doublet is an invitation to TVItroubles and improper operation of the transmitter.A properly balanced antenna is thebest solution to multi-band operation. Whenused in conjunction with an antenna tuningunit, it will perform with top efficiency on allof the major amateur bands.Several types of center-fed antenna systemsare shown in figure 20. If the feed line is madeup in the conventional manner of no. 12 or no.
<strong>HANDBOOK</strong>Multi-band Antennas 4 39~-----------------134' ----------------~If-- I I SPREADER-I' SPREADER,~-;ANTENNA TUNER I L RlOR - !TRANSMITTER"'MATCHBOX• I Cf~:~ALFigure 21'MUL Tl· BAND ANTENNA USING FAN·DIPOLE TO LIMIT IMPEDANCE EXCUR·SIONS ON HARMONIC FREQUENCIESFigure 22FOLDED· TOP DUAL-BAND ANTENNA14 wire spaced 4 to 6 inches the antenna sys·tern is sometimes called a center-fed zepp.With this type of feeder the impedance at thetransmitter end of the feeder varies from about70 ohms to approximately 5000 ohms, the sameas is encountered in an end-fed zepp antenna.This great impedance ratio requires provisionfor either series or parallel tuning of the feedersat the transmitter, and involves quite highr-f voltages at various points along the feedline.If the feed line between the transmitter andthe antenna is made to have a characteristicimpedance of approximately 300 ohms the excursionsin end-of-feeder impedance are greatlyreduced. In fact the impedance then variesfrom approximately 75 ohms to 1200 ohms.With this much lowered impedance variationit is usually possible to use series tuning onall bands, or merely to couple the antenna directlyto the output tank circuit or the harmonicreduction circuit without any separatefeeder tuning provision.There are several practicable types of transmissionline which can give an impedance ofapproximately 300 ohms. The first is, obviously,300-ohm Twin-Lead. Twin-Lead of the receivingtype may be used as a resonant feedline in this case, but its use is not recommendedwith power levels greater than perhaps150 watts, and it should not be used whenlowest loss in the transmission line is desired.For power levels up to 250 watts or so, thetransmitting type tubular 300-ohm line may beused, or the open-wire 300-ohm TV line maybe employed. For power levels higher thanthis, a 4- wire transmission line, or a linebuilt of one-quarter inch tubing should beused.Even when a 300-ohm transmission line isused, the end-of-feeder impedance may reacha high value, particularly on the second harmonicof the antenna. To limit the impedanceexcursions,. a two-wire flat-top may be em·ployed for the radiator, as shown in figure 21.The use of such a radiator will limit the impedanceexcursions on the harmonic frequenciesof the antenna and make the operation ofthe antenna matching unit much less critical.The use of a two-wire radiator is highly recommendedfor any center-fed multi-band antenna.Folded Flat-TopDual- Band AntennaAs has been mentionedearlier, there is an increasingtendency among amateuroperators to utilize rotary or fixed arraysfor the 14-Mc. band and those higher in frequency.In order to afford complete coverageof the amateur bands it is then desirable tohave an additional system which will operatewith equal effectiveness on the 3.5-Mc. and7-Mc. bands, but this low-frequency antennasystem will not be required to operate on anybands higher in frequency than the 7-Mc. band.The antenna system shown in figure 22 hasbeen developed to fill this need.This system consists essentially of anopen-line folded dipole for the 7-Mc. band witha special feed system which allows the antennato be fed with minimum standing waveson the feed line on both the 7-Mc. and 3. 5-Mc.bands. The feed-point impedance of a foldeddipole on its fundamental frequency is approximately300 ohms. Hence the 300-ohm TwinLead shown in figure 22 can be connected directlyinto the center of the system for operationonly on the 7-Mc. band and standing waveson the feeder will be very small. However, itis possible to insert an electrical half-wave
440 Antennas and Antenna Matching<strong>THE</strong> R AD I 0of transmission line of any characteristic im·pedance into a feeder system such as this andthe impedance at the far end of the line willbe exactly the same value of impedance whichthe half-wave line sees at its termination.Hence this has been done in the antenna systemshown in figure 22; an electrical halfwave of line has been inserted between thefeed point of the antenna and the 300-ohmtransmission line to the transmitter.The characteristic impedance of this additionalhalf-wave section of transmission linehas been made about 715 ohms (no. 20 wirespaced 6 inches), but since it is an electricalhalf wave long at 7 Me. and operates into aload of 300 ohms at the antenna the 300-ohmTwin-Lead at the bottom of the half-wave sectionstill sees an impedance of 300 ohms. Theadditional half-wave section of transmissionline introduces a negligible amount of losssince the current flowing in the section of lineis the same which would flow in a 300-ohmline at each end of the half-wave section, andat all other points it is less than the currentwhich would flow in a 300-ohm line since theeffective impedance is greater than 300 ohmsin the center of the half-wave section. Thismeans that the loss is less than it would be inan equivalent length of 300-ohm Twin•Leaasince this type of manufactured transmissionline is made up of conductors which are equiv·alent to no. 20 wire.So we see that the added section of 715-ohmline has substantially no effect on the operationof the antenna system on the 7-Mc. band.However, when the flat top of the antenna isoperated on the 3.5-Mc. band the feed-pointimpedance of the flat top is approximately3500 ohms. Since the section of 715-ohm transmissionline is an electrical quarter-wave inlength on the 3.5-Mc. band, this section ofline will have the effect of transforming theapproximately 3500 ohms feed-point impedanceof the antenna down to an impedance ofabout 150 ohms which will res u 1 t in a 2:1standing-wave ratio on the 300-ohm Twin-Leadtransmission line from the transmitter to theantenna system.The antenna system of figure 22 operateswith very low standing waves over the entire7-Mc. band, and it will operate with moderatestanding waves from 3500 to 3800 kc. in the3.5-Mc. band and with sufficiently low standing-waveratio so that it is quite usable overthe entire 3.5-Mc. band.This antenna system, as well as all othertypes of multi-band antenna systems, must beused in conjunction with some type of harmonic-reducingantenna tuning network eventhough the system does present a convenientimpedance value on both bands.~-------------L--------------~US0-80 METERSL=-70'V=>2'Figure 23<strong>THE</strong> MULTEE TWO-BAND ANTENNAThis compact antenna can be used with excellentresults on 160/80 and 80/40 meters.The feedline should be helcl as vertical aspossible, since it radiates when the antennais operated on its fundamental frequency.The "Multee"AntennaAn antenna that works wellon 160 and 80 meters, or 80and 40 meters and is sufficient!y compact to permit erection on the averagecity lot is the W6BCX Multee antenna,illustrated in figure 23. The antenna evolvesfrom a vertical two wire radiator, fed on oneleg only. On the low frequency band the topportion does little radiating, so it is foldeddown to form a radiator for the higher frequencyband. On the lower frequency band, the antennaacts as a top loaded vertical radiator,while on the higher freqbency band, the flattopdoes the radiating rather than the verticalportion. The vertical portion acts as a quarterwavelinear transformer, matching the 6000ohm antenna impedance to the 50 ohm impedanceof the coaxial transmission line.The earth below a vertical radiator must beof good conductivity not only to provide a lowresistance ground connection, but also to providea good reflecting surface for the wavesradiated downward towards the ground. Forbest results, a radial system should be installedbeneath the antenna. For 160-80 meteroperation, six radials 50 feet in length,made of no. 16 copper wire should be buriedjust below the surface of the ground. While anordinary water pipe ground system with noradials may be used, a system of radials willprovide a worthwhile increase in signalstrength. For 80-40 meter operation, the length
<strong>HANDBOOK</strong>Low Frequency Discone 441s~-HQ4.0i 3.5"' >~ 3.0..z 2. 51izc( 2.0E..1.5ICUT -OFF FREQUENCY'< ~t::1: ['- ~6 ~ ~ ~ U U ~ M ~ ~ M ~ M ~FREQUENCY (Me)1Figure 25SWR CURVE FOR A 13.2 MC. DISCONEANTENNA. SWR IS BELOW 1.5 TO 1 FROM13.0 MC. TO 58 MC.20, 15,11,10,6 METERSD= 12' L= 18'S= JON R= 18'H= 15'7"DIMENSIONS15, 11,10,15 METERSD:: 8' L= 12•S = 6" R= 12'H= IO' 5"Figure 2411.10, 6, 2 METERSD= 6' L=9'6"S=4" R;9'6 11H=e•3"DIMENSIONS OF LOW-FREQUENCY DISCONE ANTENNA FOR LOW FREQUENCYCUTOFF AT 13.2 MC., 20.1 MC., AND26 MC.The Discone is a vertically polarized radiator,proclucing an omniclirectional patternsimilar to a ground plane. Operation on sev ..era/ amateur bands with low SWR on the coaxialfeed line is possible. Additional in·formation on L·F Discone by W2RYI in July,1950 CQ magazine.of the radials may be reduced to 25 feet. Aswith all multi-band antennas that employ nolumped tuned circuits, this antenna offers noattenuation to harmonics of the transmitter.When operating on the lower frequency band,it would be wise to check the transmitter forsecond harmonic emission, since this antennawill effectively radiate this harmonic.The Low-FrequencyDisconeThe discone antenna iswidely used on the v-h-fbands, but until recentlyit has not been put to any great use on thelower frequency bands. Since the discone is abroad-band device, it may be used on severalharmonically related amateur bands. Size isthe limiting factor in the use of a discone, andthe 20 meter band is about the lowest practi·cal frequency for a discone of reasonable di·mensions. A discone designed for 20 meteroperation may be used on 20, 15, 11, 10 and6 meters with excellent results. It affords agood match to a 50 ohm coaxial feed systemon all of these bands. A practical discone an·tenna is shown in figure 24, with a SWR curvefor its operation over the frequency range of13·55 Me. shown in figure 25. The disconeantenna radiates a vertically polarized waveand has a very low angle of radiation. Forv-h-f work the discone is constructed of sheetmetal, but for low frequency work it may bemade of copper wire and aluminum anglestock. A suitable mechanical layout for a lowfrequency disco n e is shown in figure 26.Smaller versions of this antenna may be con·structed for 15, 11, 10 and 6 meters, or for 11,10, 6 and 2 meters as shown in the chart offigure 24.For minimum wind resistance, the top "hat"of the discone is constructed from three·quar·ter inch aluminum angle stock, the rods beingbolted to an aluminum plate at the center ofthe structure. The tips of the rods are all connectedtogether by lengths of no. 12 enamelledcopper wire. The cone elements are made ofno. 12 copper wire and act as guy wires forthe discone structure. A very rigid arrange·ment may be made from this design; one thatwill give no trouble in high winds. A 4 11 x 4 11post can be used to support the discone struc·ture.The discone antenna may be fed by a lengthof 50-ohm coaxial cable directly from the trans·mitter, with a very low SWR on all bands.The Single-Wire-Fed AntennaThe old favorite single-wire·fed antenna system is quitesatisfactory for an impromp·tu all band antenna system. It is widely usedfor portable installations and "Field Day"contests where a simple, multi-band antennais required. A single wire feeder has a characteristicimpedance of some 500 ohms, de-
442 Antennas and Antenna Matching<strong>THE</strong> R AD I 0VOLTAGE CURVESCENTER----------r~--+~~AN~TE~NN~AW~IR~E--~--FEEDERFigure 27SINGLE-WIRE-FED ANTENNA FOR ALL·BAND OPERATIONAn antenna of this type for 40-, 20- ancl 10-meter operotjon woulcl have a racliator 67feet long, with the feecler tappecl 11 feet offcenter. The feecler can be 33, 66 or 99 feetlong. The same type of antenna for 80-, 40-,20- ancl 10-meter operation woulcl have aracliator 134 feet long, with the feecler tappecl22 feet off center. The feecler can be either66 or 132 feet long. This system shoulcl beusee/ only with those coupling methocls whichprovicle g oocl harmonic suppression.Figure 26MECHANICAL CONSTRUCTION OF 20.METER DISCONEpending upon the wire size and the point ofattachment to the antenna. The earth lossesare comparatively low over ground of good conductivity.Since the single wire feeder radiates,it is necessary to bring it away from the antennaat right an g 1 e s to the ante n n a wirefor at 1 e as t one-half the 1 eng t h of the antenna.The correct point for best impedance matchon the fundamental frequency is not suitablefor harmonic operation of the antenna. In addition,the correct 1 en g t h of the antenna forfundamental operation is not correct for harmonicoperation. Consequendy, a compromisemust be made in antenna length and point offeeder connection to enable the single-wirefedantenna to operate on more than one band.Such a compromise introduces additional reactanceinto the single wire feeder, and mightcause loading difficulties with pi-networktransmitters. To minimize this trouble, thesingle wire feeder should be made a multipleof 33 feet long.Two typical single-wire-fed antenna systemsare shown in figure 27 with dimensionsfor multi-band operation.22-8 Matching Non-ResonantLines to the AntennaPresent practice in regard to the use oftransmission lines for feeding antenna systemson the amateur bands is about equally divided
<strong>HANDBOOK</strong>Matching Systems 44 31+---------L ---------o{NON-RESONANTLINEFigure 28MATCHING SECTION<strong>THE</strong> DELTA-MATCHED DIPOLEANTENNA7 he dimensions for the portions of the an·tenna are given in the text.transmitter end of the feed line which willchange the magnirude of the standing waveson the antenna transmission line.Delta-MatchedAntenna SystemThe delta type matched-im·pedance antenna system isshown in figure 28. The im·pedance of the transmission line is trans·formed gradually into a higher value by thefanned-out Y portion of the feeders, and theY portion is tapped on the antenna at pointswhere the antenna impedance is a compromisebetween the impedance at the ends of the Yand the impedance of the unfanned portion ofthe line.The constants of the system are rather critical,and the antenna must r e son ate at theoperating frequency in order to minimize standingwaves on the line. Some slight readjust·ment of the taps on the antenna is desirable,if appreciable standing waves persist in appearingon the line.The constants for a doublet are determinedby the following formulas:between three types of transmission line: {1)Ribbon or tubular m o 1 de d 300-ohm line iswidely used up to moderate power levels (the"transmitting" type is useable up to the kilowattlevel). ( 2) Open-wire 400 to 600 ohm lineis most commonly used when the antenna issome distance from the transmitter, because ofthe low attenuation of this type of line. ( 3) Coaxialline ( u sua 11 y RG-8/U with a 52-ohmcharacteristic impedance) is widely used inv-h-f work and also on the lower frequencieswhere the feed line must run underground orthrough the walls of a building. Coaxial linealso is of assistance in TVI reduction sincethe r·f fie 1 d is entirely enclosed within theline. Molded 75-ohm line is sometimes usedto feed a doublet antenna, but the doublet hasbeen largely superseded by the folded-dipoleantenna fed by 300-ohm ribbon or tubular linewhen an antenna for a single band is required.Standing Waves As was discussed earlier,standing waves on the antennatransmission line, in the transmitting case,are a result of reflection from the point wherethe feed line joins the antenna system. Themagnitude of the standing waves is determinedby the degree of mismatch between thecharacteristic impedance of the transmissionline and the input impedance of the antennasystem. When the feed-point impedance of theantenna is resistive and of the same value asthe characteristic impedance of the feed line,standing waves will not exist on the feeder.It may be well to repeat at this time that thereis no adjustment which can be made at theLreetDreetEreet467.4F megacycles175F megacycles147.6F megacyclesWhere L is antenna length; D is the distance infrom each end at which the Y taps on; E is theheight of the Y section.Since these constants are correct only for a600-ohm transmission line, the spacing S ofthe line must be approximately 75 times thediameter of the wire used in the transmissionline. For no. 14 B & S wire, the spacing willbe slightly less than 5 inches. This systemshould never be used on either its even or oddharmonics, as entirely different constants arerequired when more than a single half wave·length appears on the radiating portion of thesystem.Multi-Wire Doublets When a doublet antennaor the driven element inan array consists of more than one wire ortubing conductor the radiation resistance ofthe antenna or array is increased slightly as aresult of the increase in the effective diameterof the element. Further, if we split just one
444 Antennas and Antenna Matching<strong>THE</strong> R AD I 01--------- -''·--·--- -·-- --10 +DRIVEN ELEMENT®r--"TWIN-LEAD" FLAT-TOP ORCONDUCTORS SPACED 1" TO 12"MOVEABLE CLAMP\_GAMMA RODRESONATING CONDENSER.50-70 OHM COAXIAL FEED LINE©3000HMSANY LENGTH) ) D"'" ELEM"'300-600OHM FEEDERSFigure 30TME GAMMA MATCH FOR CONNECTINGAN UNBALANCED COAXIAL LINE TO ABALANCED DRIVEN ELEMENT1RESONATING--" ,jCDNDE'5'RSII't.j·300-600OHM FEEDERSFigure 29"T" MATCHINi; SECTIONFOLDED-ELEMI:NT MATCHING SYSTEMSDrawing (A) above shows o half-wove madeup to two parallel wires. If one of the wiresis broken as in (B) oncl the feeder connected,the feed-point impedance is multipl iecl byfour; such an antenna is commonly callecl a11folclecl doublet. 11 The feed-point impedancefor o simple hoff-wove doublet feel in thismanner is approximately 300 ohms, dependingupon antenna height. Drawing (C) showshow the feed-point impedance can be multipiiecl by o factor greater than four by makingthe hoff of the element that is broken smallerin diameter than the unbroken hoff. An extensionof the principles of (B) and (C) isthe arrangement shown at (D) where the sectioninto which the feeders ore connected isconsiderably shorter than the driven element.This system is most convenient when thedriven element is too long (such as for o28-Mc. or 14-Mc. array) for o convenientmechanical arrangement of the system shownat (C).wire of such a radiator, as shown in figure 29,the effective feed-point resistance of the antennaor array will be increased by a factorof N 2 where N is equal to the number of conductors,all in parallel, of the same diameterin the array. Thus if there are two conductorsof the same diameter in the driven element orthe antenna the feed-point resistance will bemultiplied by 2 2 or 4. If the antenna has a radiationresistance of 75 ohms its feed-pointresistance will be 300 ohms, this is the caseof the conventional folded-dipole as shown infigure 29B.If three wires are used in the driven radiatorthe feed-point resistance is increased bya factor of 9; if four wires are used the impedanceis increased by a factor of I6, andso on. In certain cases when feeding a parasiticarray it is desirable to have an impedancestep up different from the value of 4: Iobtained with two elements of the same diameterand 9: I with three elements of the samediameter. Intermediate values of impedancestep up may be obtained by using two elementsof different diameter for the complete drivenelement as shown in figure 29C. If the conductorthat is broken for the feeder is of smallerdiameter than the other conductor of theradiator, the impedance step up will be greaterthan 4: I. On the other hand if the larger of thetwo elements is broken for the feeder the impedancestep up will be less than 4: I.The "T" Match A method of matching a balancedlow-impedance transmissionline to the driven element of a para·sitic array is the T match illustrated in figure29D. This method is an adaptation of themulti-wire doublet principle which is morepracticable for lower-frequency parasitic arrayssuch as those for use on the I4-Mc. and28-Mc. bands. In the system a section of tubingof approximately one-half the diameter ofthe driven element is spaced about four inchesbelow the driven element by means ofclamps which hold the T-section mechanicallyand which make electrical connection to thedriven element. The length of the T-sectionis normally between I5 and 30 inches eachside of the center of the dipole for transmissionlines of 300 to 600 ohms impedance, assuming28-Mc. operation. In series with eachleg of the T-section and the transmission lineis a series resonating capacitor. These twocapacitors tune out the reactance of the T-
<strong>HANDBOOK</strong>Matching Systems 445section. If they are not used, the T-sectionwill detune the dipole when the T-section isattached to it. The two capacitors may beganged together, and once adjusted for minimumdetuning action, they may be locked. Asuitable housing should be devised to protectthese capacitors from the weather. Additionalinformation on the adjustment of the T-matchis given in the chapter covering rotary beamantennas.The "Gamma" Match An unbalanced version ofthe T -match may be usedto feed a dipole from an unbalanced coaxialline. Such a device is called a Gamma Match,and is illustrated in figure 30.The length of the Gamma rod and the spacingof it from the dipole determine the impedancelevel at the transmission line end of therod. The series capacitor is used to tune outthe reactance introduced into the system bythe Gamma rod. The adjustment of the GammaMatch is discussed in the chapter coveringrotary beam antennas.Matching Stubs By connecting a resonantsection of transmission line(called a matching stub) to either a voltage orcurrent loop and attaching parallel-wire nonresonantfeeders to the resonant stub at asuitable voltage (impedance) point, standingwaves on the line may be virtually eliminated.The stub is made to serve as an auto-transformer.Stubs are particularly adapted tomatching an open line to certain directionalarrays, as will be described later.Voltage Feed Wben the stub attaches to theantenna at a voltage loop, thestub should be a quarter wavelength longelectrically, and be shorted at the bottom end.The stub can be resonated by sliding theshorting bar up and down before the non-resonantfeeders are attached to the stub, the antennabeing shock-excited from a separateradiator during the process. Slight errors inthe length of the radiator can be compensatedfor by adjustment of the stub if both sides ofthe stub are connected to the radiator in asymmetrical manner. Where only one side ofthe stub connects to the radiating system, asin the Zepp and in certain antenna arrays, theradiator length must be exactly right in orderto prevent excessive unbalance in the untunedline.A dial lamp may be placed in the center ofthe shorting stub to act as an r-f indicator.Current Feed When a stub is used to currentfeeda radiator, the stub shouldeither be left open at the bottom end insteadof shorted, or else made a half wave long.SHORTING BARr-o------t ANTENNA----------J1---------j-ANTENNA--------iSHORTINC'; BARt STUB1-------- tANTENNA --------1r---t----tFigure 31~EEDER TAPS NEAREND Of STUBtsTUBSHORTING BARMATCHING-STU"B APPLICATIONSAn end-fed half-wove antenna with a quarter·wave shorted stub is shown at (A). (B) showsthe use of a half-wave shorted stub to feeda relatively low impedance point such as thecenter of the driven element of a parasiticarray, or the center of a half-wave dipole.The use of an open-ended quarter-wave stubto feed a low impedance is illustrated at(C). (D) shows the conventional use of ashorted quarter-wave stub to voltage feedtwo half-wave antennas with a lBOo phasedifference.®©®
446 Antennas and Antenna Matching<strong>THE</strong> R AD I 0The open stub should be resonated in thesame manner as the shorted stub before at·taching the transmission 1 in e; however, inthis case, it is necessary to prune the stubto resonance, as there is no shorting bar.Sometimes it is handy to have a stub hangfrom the radiator to a point that can be reachedfrom the ground, in order to facilitate adjust·ment of the position of the transmission-lineattachment. For this reason, a quarter-wavestub is sometimes made three-quarters wavelengthlong at the higher frequencies, in orderto bring the bottom nearer the ground. Opera·tion with any odd number of quarter waves isthe same as for a quarter-wave stub.Any number of half waves can be added toeither a quarter-wave stub or a half-wave stubwithout disturbing the operation, though lossesand frequency sensitivity will be lowest ifthe shortest usable stub is employed. See figure31.Stub Length Current-Fed Voltage-Fed(Electrical) Radiator Radiator!4-*-1 !4-etc. Open Shortedwavelengths Stub Stubl-2-1-1 l-2-2-etc. Shorted Openwavelengths Stub StubLinear R.FTransformersA resonant quarter-wave linehas the unusual property ofacting much as a transformer.Let us take, for example, a section consistingof no. 12 wire spaced 6 inches, which happensto have a surge impedance of 600 ohms. Letthe far end be terminated with a pure resist·ance, and let the near end be fed with radiofrequencyenergy at the frequency for whichthe line is a quarter wavelength long. If animpedance measuring set is used to measurethe impedance at the near end while the impedanceat the far end is varied, an interestingrelationship between the 600-ohm charac·teristic surge impedance of this particularquarter-wave matching line, and the impedanceat the ends will be discovered.When the impedance at the far end of theline is the same as the characteristic surgeimpedance of the line itself (600 ohms), theimpedance measured at the near end of thequarter-wave line will also be found to be600 ohms.Under these conditions, the line would nothave any standing waves on it, since it isterminated in its characteristic impedance.Now, let the resistance at the far end of theline be doubled, or changed to 1200 ohms.The impedance measured at the near end ofthe line will be found to have been cut inhalf, to 300 ohms. If the resistance at the farend is made half the original v a 1 u e of 600ohms or 300 ohms, the impedance at the nearend doubles the original value of 600 ohms,and becomes 1200 ohms. As one resistancegoes up, the other goes down proportionately.It will always be found that the characteristicsurge impedance of the quarter-wavematching line is the geometric mean betweenthe impedance at both ends. This relationshipis shown by the following formula:ZMs = V ZA ZLwhereZMs =Impedance of matching section.ZA = Antenna resistance.ZL = Line impedance.Quarter-WaveMatchingTransformersThe impedance inverting characteristicof a quarter-wavesection of transmission line iswidely used by making such asection of line act as a quarter-wave trans·former. The Johnson Q feed system is a wide·ly known application of the quarter-wave transformerto the feeding of a dipole antenna andarray consisting of two dipoles. However, thequarter-wave transformer may be u s e d in awide number of applications wherever a transformeris required to match two impedanceswhose geometric mean is somewhere betweenperhaps 25 and 750 ohms when transmissionline sections can be used. Paralleled coaxiallines may be used to obtain the lowest impedancementioned, and open-wire lines com·posed of small conductors spaced a moderatedistance may be used to obtain the higher impedance.A short list of impedances, whichmay be matched by quarter-wave sections oftransmission line having specified impedances,is given below.Load or Ant. ~Impedance20_10so75100Johnson-QFeed System~Feed-Line300 480 600 Impedance77 98 110 Quarter•95 120 134 Wave110 139 155 Transformer150 190 212 Impedance173 220 245The standard form of Johnson•Q feed to a doublet is shownin figure 32. An impedancematch is obtained by utilizing a matching section,the surge impedance of which is the geometricmean between the transmission linesurge impedance and the radiation resistanceof the radiator. A sufficiently good match usu-
<strong>HANDBOOK</strong>Matching Systems 447----- L FEET=~ ----------1Za=~Center to Impedance ImpedanceCenter in Ohms in OhmsSpacing for Y:!" for Y4 11in Inches Diameters Diameters1 170 2501.25 188 2771.5 207 2981.75 225 3182 248 335ZzPARALLEL TUBING SURGE IMPEDANCE FORMATCHING SECTIONSSLUNTUNED LINEANY LENGTHFigure 32HALF-WAVE RADIATOR FEDBY "Q BARS"The Q matching section is simply a quarterwavetransformer whose impedance is equalto the geometric mean between the impec/.once at the center of the antenna one/ theimpedance of the transmission line to beusee/ to feecl the bottom of the transformer.The transformer may be made up of paralleltubing, a four-wire line, or any other typeof transmission line which has the correctvalue of impedance.ally can be obtained by either designing oradjusting the matching section for a dipole tohave a surge impedance that is the geometricmean between the line impedance and 72 ohms,the latter being the theoretical radiation re·sistance of a half-wave doublet either infi·nitely high or a half wave above a perfectground.Though the radiation resistance may departsomewhat from 72 ohms under actual condi·tions, satisfactory results will be obt~nedwith this assumed value, so long as the dipoleradiator is more than a quarter wave aboveeffective earth, and reasonably in the clear.The small degree of standing waves intro·duced by a slight mismatch will not increasethe line losses appreciably, and any smallamount of reactance present can be tuned outat the transmitter termination with no bad ef·fects. If the reactance is objectionable, it maybe minimized by making the untuned line anintegral number of quarter waves long.A Q-matched system can be adjusted pre·cisely, if desired, by constructing a matchingsection to the calculated dimensions with pro·vision for varying the spacing of the Q sec·tion conductors slightly, after the untunedline has been checked for standing waves.The CollinsTransmission LineMatching SystemThe advantage of unbalancedoutput networksfor transmitters are numerous;however this out·put system becomes awkward when it is de·sired to feed an antenna system utilizing abalanced input. For some time the CollinsRadio Co. has been experimenting with a balunand tapered line system for matching a co·axial output transmitter to an open-wire balancedtransmission line. Considerable successhas been obtained and matching systems goodover a frequency range as great as four to onehave been developed. Illustrated in figure 33is one type of matching system which is prov·ing satisfactory over this range. Z, is thetransmitter end of the system and may be anylength of 52-ohm coaxial cable. Z 2is one·quarter wavelength long at the mid-frequencyof the range to be covered and is made of 75ohm coaxial cable. ZA is a quarter-wavelengthshorted section of cable at the mid-frequency.Z 0 (ZA and Z 2 ) forms a 200-ohm quarter-wavesection. The ZA section is formed of a con·ductor of the same diameter as Z 2• The differ·ence in length between ZA and Z 2 is accountedfor by the fact that Z 2 is a coaxial conductorwith a solid dielectric, whereas the dielectricfor Z 0 is air. Z 3 is one-quarter wavelengthlong at the mid-frequency and has an imped·Z1INNER & OUTER CONDUCTORSSHORTED AT EACH ENDFigure 33COLLINS TRANSMISSION LINE MATCHINGSYSTEMA wide-band system for matching a 52-ohmcoaxial I ine to a ba/ancec/ 300-ohm I ine overa 4: I wicle frequency range.
448 Antennas and Antenna Matching<strong>THE</strong> R AD I 0ance of 123 ohms. Z 4 is one-quarter wavelengthlong at the mid-frequency and has an impedanceof 224 ohms. Z 5 is the balanced line tobe matched (in this case 300 ohms) and may beany length.Other system parameters for different outputand input impedances may be calculated fromthe following:Transformation ratio (r) for each section is:r=~Z;,Where N is the number of sections. In theabove case,r=~zlImpedance between sections, as Z 2• 3, is rtimes the preceding section. Z 2• 3 = r X Z, andZ 3 • 4 = r X Z 2 • 3 •Mid-frequency (m):F, + F 2m=27 + 30For 40-20-10 meters=218.5 Me.and one-quarter wavelength = 12 feet.For 20-10-6 meters=14 +54234 Me.and one-quarter wavelength= 5.5 feet.The impedances of the sections are:Z 2 = V Z, ~< Z 2 • 3Z 3 = V Z 2 • 3 X Z 3 • 4Z 4 = V Z 3 • 4 X Z 5Generally, tbe larger number of taper sectionsthe greater will be the bandwidth of thesystem.22-9 AntennaConstructionThe foregoing portion of this chapter hasbeen concerned primarily with the electricalcharacteristics and considerations of antennas.Some of the physical aspects and mechanicalproblems incident to the actual erectionof antennas and arrays will be discussedin the following section.Up to 60 feet, there is little point in usingmast-type antenna supports unless guy wireseither must be eliminated or kept to a minimum.While a little more difficult to erect, becauseof their floppy nature, fabricated woodpoles of tbe type to be described will be justas satisfactory as more rigid types, providedmany guy wires are used.Rather expensive when purchased throughthe regular channels, 40- and 50-foot telephonepoles sometimes can be obtained quitereasonably. In the latter case, they are hardto beat, inasmuch as they require no guyingif set in the ground six feet (standard depth),and the resultant pull in any lateral directionis not in excess of a hundred pounds or so.For heights of 80 to 100 feet, either threesidedor four-sided lattice type masts are mostpracticable. They can be made self-sup~orting,but a few guys will enable one to use asmaller eros s section without danger fromhigh winds. The torque exerted on the baseof a high self-supporting mast is terrific duringa strong wind.The "A-Frame" Figures 34A and 34B showMastthe standard method of constructionof the A-frametype of mast. This type of mast is quite frequentlyused since there is only a moderateamount of work involved in the constructionof the assembly and since the material cost isrelatively small. The three pieces of selected2 by 2 are first set up on three sawhorses orboxes and the holes drilled for the three ~inch bolts through the center of the assembly.Then the base legs are spread out to about 6feet and the bottom braces installed. Then theupper braces and the cross pieces are installedand the assembly given several coatsof good-quality paint as a protection againstweathering.Figure 34C shows another common type ofmast which is made up of sections of 2 by 4placed end-to-end with stiffening sections of1 by 6 bolted to tbe edge of the 2 by 4 sections.Both types of masts will require a setof top guys and another set of guys about onethirdof the way down from the top. Two guysspaced about 90 to 100 degrees and pullingagainst the load of the antenna will normallybe adequate for the top guys. Three guys areusually used at the lower level, with one directlybehind the load of the antenna and twomore spaced 120 degrees from the rear guy.The raising of the mast is made much easier
<strong>HANDBOOK</strong>Figure 34TWO SIMPLE WOOD MASTSShown at (A) is the methocl of as·sembly, anrl at (B) is the completed11structure, of the conventional Aframe" antenna mast. At (C) isshown a structure which is heavierbut more stable than the A-framefor heights above about 40 feet.Antenna Construction 449T r" t1 ,..T 1 2.X4ltO'1XO..." l1~~TS®® ©if a gin pole about 20 feet high is installedabout 30 or 40 feet to the rear of the directionin whi-ch the antenna is to be raised. A linefrom a pulley on the top of the gin pole is thenrun to the top of the pole to be raised. Thegin pole comes into play when the center ofthe mast has been raised 10 to 20 feet abovethe ground and an additional elevated pull isrequired to keep the top of the mast comingup as the center is raised further aboveground.Using TV Masts Steel tubing masts of thetelescoping variety are wide·ly available at a moderate price for use in sup·porting television antenna arrays. These mastsusually consist of several 10-foot lengths ofelectrical metal tubing (EMT) of sizes suchthat the sections will telescope. The 30-footand 40-foot lengths are well suited as mastsfor supporting antennas and arrays of thetypes used on the amateur bands. The mastsare constructed in such a manner that the bot·tom 10-foot length may be guyed permanentlybefore the other sections are raised. Then theupper sections may be extended, beginningwith the top-mast section, until the mast is atfull length (provided a strong wind is not blow·ing) following which all the guys may be an·chored. It is important that there be no loadon the top of the mast when the "vertical"raising method is to be employed.Guy Wires Guy wires should never be pulledtaut; a small amount of slack isdesirable. Galvanized wire, somewhat heavierthan seems sufficient for the job, should beused. The heavier wire is a little harder tohandle, but costs only a little more and takeslonger to rust through. Care should be takento make sure that no kinks exist when the poleor tower is ready for erection, as the wire willbe greatly weakened at such points if a kinkis pulled tight, even if it is later straightened.If "dead men" are used for the guy wireterminations, the wire or rod reaching from thedead men to the surface should be of non-rust·ing material, such as brass, or given a heavycoating of asphalt or other protective substanceto prevent destructive action by thedamp soil. Galvanized iron wire will last onlya short time when buried in moist soil.Only strain-type (compression) insulatorsshould be used for guy wires. Regular onesmight be sufficiently strong for the job, but itis not worth taking chances, and egg-typestrain halyard in s u 1 at or s are no more expensive.Only a brass or bronze p u 11 e y should beused for the halyard, as a high pole with arusted pulley is truly a sad affair. The bear·ing of the pulley should be given a few dropsof heavy machine oil before the pole or toweris raised. The halyard itself should be of goodmaterial, preferably water-proofed. Hemp ropeof good quality is better than window sashcord from several standpoints, and is less ex·pensive. Soaking it thoroughly in engine oil ofmedium viscosity, and then wiping it off witha rag, will not only extend its life but minimizeshrinkage in wet weather. Because of thedifficulty of replacing a broken halyard it isa good idea to replace it periodically, without
450 Antennas and Antenna Matching<strong>THE</strong> R AD I 0waiting for it to show excessive wear or deterioration.It is an excellent idea to tie both ends ofthe halyard line together in the manner of aflag-pole line. Then the antenna is tied ontothe place where the two ends of the halyardare joined. This procedure of making the halyardinto a loop prevents losing the top endof the halyard should the antenna break nearthe end, and it also prevents losing the halyardcompletely should the end of the halyardcarelessly be allowed to go free and be pulledthrough the pulley at the top of the mast bythe antenna load. A somewhat longer pieceof line is required but the insurance is wellworth the cost of the additional length of rope.Trees asSupportsOften a tall tree can be called uponto support one end of an antenna,but one should not attempt toattach anything to the top, as the swaying ofthe top of the tree during a heavy wind willcomplicate matters.If a tree is utilized for support, provisionshould be made for keeping the antenna tautwithout submitting it to the possibility of be·ing severed during a heavy wind. This can bedone by the simple expedient of using a pulleyand halyard, with weights attached to thelower end of the halyard to keep the antennataut. Only enough weight to avoid excessivesag in the antenna should be tied to the halyard,as the continual swaying of the tree submitsthe pulley and halyard to considerablewear.Galvanized iron pipe, or steel-tube conduit,is often used as a vertical radiator, and isquite satisfactory for the purpose. However,when used for supporting antennas, it shouldbe remembered that the grounded supportingpoles will distort the field pattern of a verticallypolarized antenna unless spaced somedistance from the radiating portion.Painting The life of a wood mast or pole canbe increased several hundred percent by protecting it from the elements with acoat or two of paint. And, of course, the appearanceis greatly enhanced. The wood shouldfirst be given a primer coat of flat white outsidehouse paint, which can be thinned downa bit to advantage with second-grade linseedoil. For the second coat, which should not beapplied until the first is thoroughly dry, aluminumpaint is not only the best from a preservativestandpoint, but looks very well. This typeof paint, when purchased in quantities, is considerablycheaper than might be gathered fromthe price asked for quarter-pint cans.Portions of posts or poles below the surfaceof the soil can be protected from termites andmoisture by painting with creosote. While notso strong initially, redwood will deterioratemuch more slowly when buried than will thewhite woods, such as pine.Antenna Wire The antenna or array itselfpresents no especial problem.A few considerations should be borne in mind,however. For instance, soft-drawn coppershould not be used, as even a short span willstretch several per cent after whipping aroundin the wind a few weeks, thus affecting theresonant frequency. Enameled-copper wire,as ordinarily available at radio stores, is usallysoft drawn, but by tying one end to someobject such as a telephone pole and the otherto the frame of an auto, a few husky tugs canbe given and the wire, after stretching a bit,is equivalent to hard drawn.Where a long span of wire is required, orwhere heavy insulators in the center of thespan result in considerable tension, coppercladsteel wire is somewhat better than harddrawncopper. It is a bit more expensive,though the cost is far from prohibitive. Theuse of such wire, in conjunction with straininsulators, is advisable, where the antennawould endanger persons or property should itbreak.For transmission I in e s and tuning stubssteel-core or hard-drawn wire will prove awkwardto handle, and soft-drawn copper should,therefore, be used. If the I in e is Ion g, thestrain can be eased by supporting it at severalpoints.More important from an electrical standpointthan the actual size of wire used is the solderingof joints, especially at current loopsin an antenna of low radiation resistance. Infact, it is good practice to solder all joints,thus insuring quiet operation when the antennais used for receiving.lnsulatlor• A question that often arises isthat of insulation. It depends, ofcourse, upon the r-f voltage at the point atwhich the insulator is placed. The r-f voltage,in turn, depends upon the distance from a currentnode, and the radiation resistance of theantenna. Radiators having low radiation resistancehave very high voltage at the voltageloops; consequently, better than usual insulationis advisable at those points.Open-wire lines operated as non-resonantlines have little voltage across them; hencethe most inexpensive ceramic types are sufficientlygood electrically. With tuned lines, thevoltage depends upon the amplitude of thestanding waves. If they are very g r e at, thevoltage will reach high values at the voltageloops, and the best spacers available are nonetoo good. At the current loops the voltage isquite low, and almost anything will suffice.
<strong>HANDBOOK</strong>Antenna Coupling Systems 451When in51ulators are subject to very high r·fvoltages, they should be cleaned occasionallyif in the vicinity of sea water or smoke. Saltscum and soot are not readily dislodged byrain, and when the coating becomes heavyenough, the efficiency of the insulators isgreatly impaired.If a very pretentious installation is to bemade, it is wise to check up on both underwriter'srules and local ordinances whichmight be applicable. If you live anywhere nearan airport, and are contemplating a tall pole,it is best to investigate possible regulationsand ordinances pertaining to towers in the district,before starting construction.22-10 Coup I ing to theAntenna SystemWhen coupling an antenna feed system to atransmitter the most important considerationsare as follows: ( 1) means should be providedfor varying the load on the amplifier; (2) thetwo tubes in a push-pull amplifier should beequally loaded; (3) the load presented to thefinal amplifier should be resistive (non-reactive)in character; and (4) means should beprovided to reduce harmonic coupling betweenthe final amplifier plate tank circuit and theantenna or antenna transmission line to an extremelylow value.The Transmitter-Loading ProblemThe problem of coupling thepower output of a high-frequencyor v-h-f transmitterto the radiating portion of the antenna systemhas been materially complicated by the virtualnecessity for eliminating interference to TV reception.However, the TVI-elimination portionof the problem may always be accomplishedby adequate shielding of the transmitter, byfiltering of the control and power leads whichenter the transmitter enclosure, and by the inclusionof a harmonic-attenuating filter be-tween the output of the transmitter and the antennasystem.Although TVI may be eliminated through inclusionof a filter between the output of ashielded transmitter and the antenna system,the fact that such a filter must be included inthe link between transmitter and antenna makesit necessary that the transmitter-loading problembe re-evaluated in terms of the necessityfor inclusion of such a filter.Harmonic-attenuating filters must be operatedat an impedance level which is close totheir design value; therefore they must operateinto a resistive termination substantially equalto the characteristic impedance of the filter.If such filters are operated into an impedancewhich is not resistive and approximately equalto their characteristic impedance: (1) the capacitorsused in the filter sections will besubjected to high peak voltages and may bedamaged, (2) the harmonic-attenuating propertiesof the filter will be decreased, and (3) theimpedance at the input end of the filter willbe different from that seen by the filter at theload end (except in the case of the half-wavetype of filter). It is therefore important thatthe filter be included in the transmitter-to-antennacircuit at a point where the impedanceis close to the nominal value of the filter, andat a point where this impedance is likely toremain fairly constant with variations in frequency.Block Diagrams ofTransmitter-to-AntennaCoupling SystemsThere are two basicarrangements w h i c hinclude all the provisionsrequired in thetransmitter-to-antenna coupling system, andwhich permit the harmonic-attenuating filter tobe placed at a position in the coupling systemwhere it can be operated at an impedancelevel close to its nominal value. These arrangementsare illustrated in block-diagramform in figures 35 and 36.The arrangement of figure 35 is recommendedfor use with a single-band antenna system,such as a dipole or a rotatable array, whereinan impedance matching system is includedFigure 35ANTENNA COUPLING SYSTEMThe harmonic suppressing antenna coupling system illustratecl above is lor use when the antennatransmission I ine has a low stanJing-wave ratio, anJ when the characteristic impeclance ofthe antenna transmission line is the same as the nominal impeclonce ol the low-pass harmonicattenuatingfilter.
452 Antennas and Antenna Matching <strong>THE</strong> R AD I 0Figure 36ANTENNA COUPLING SYSTEMThe antenna coupling system illustrated above is for use when the antenna transmis~ion linedoes not have the same characteristic impedance as the TV/ filter, and when the stanclmg·waveratio on the antenna transmission line may or may not he low.within or adjacent to the antenna. The feedline coming down from the antenna systemshould have a characteristic impedance equalto the nominal impedance of the harmonic filter,and the impedance matching at the antennashould be such that the standing-wave ratioon the antenna feed line is less than 2 to 1over the range of frequency to be fed to theantenna. Such an arrangement may be usedwith open-wire line, ribbon or tubular line, orwith coaxial cable. The use of coaxial cableis to be recommended, but in any event theimpedance of the antenna transmission lineshould be the same as the nominal impedanceof the harmonic filter. The arrangement of figure35 is more or less standard for commerciallymanufactured equipment for amateur andcommercial use in the h-f and v-h-f range.The arrangement of figure 36 merely addsan antenna coupler between the output of theharmonic attenuating filter and the antennatransmission line. The antenna coupler willhave some harmonic-attenuating action, but itsmain function is to transform the impedanceat the station end of the antenna transmissionline to the nominal value of the harmonic filter.Hence the arrangement of figure 36 is moregeneral than the figure 35 system, since theinclusion of the antenna coupler allows thesystem to feed an antenna transmission lineof any reasonable impedance value, and alsowithout regard to the standing-wave ratiowhich might exist on the antenna transmissionline. Antenna couplers are discussed in a followingsection.Output Coup I ingAdjustmentIt will be noticed by referenceto both figure 35 andfigure 36 that a box labeledCoupling Adjustment is included in the blockdiagram. Such an element is necessary in thecomplete system to afford an adjustment in thevalue of load impedance presented to the tubesin the final amplifier stage of the transmitter.The impedance at the input terminal of theharmonic filter is established by the antenna,through its matching system and the antennacoupler, if used. In any event the impedanceat the input term in a 1 of the harmonic filtershould be very close to the nominal impedanceof the filter. Then the Coupling Adjustmentprovides means for transforming this impedancevalue to the correct operating value ofload impedance which should be presented tothe final amplifier stage.There are two common ways for accomplishingthe antenna coupling adjustment, as illustratedin figures 37 and 38. Figure 37 showsthe variable-link arrangement most commonlyused in home-constructed equipment, while thepi-netowrk coup 1 in g arrangement commonlyused in commercial equipment is illustrated infigure 38. Either method may be used, and eachhas its advantages.Variable-LinkCoupl ingThe variable-link method illustratedin figure 37 has theadvantage that standard manufacturedcomponents may be used with nochanges. However, for greatest bandwidth ofoperation of the coupling circuit, the reactanceof the link coil, L, and the reactance of thelink tuning capacitor, C, should both be between3 and 4 times the nominal load impedanceof the harmonic filter. This is to say thatthe inductive reactance Qf the coupling link Lshould be tuned out or resonated by capacitorC, and the operating Q of the L-C link circuitshould be between 3 and 4. If the link coil isnot variable with respect to the tank coil ofthe final amplifier, capacitor C may be usedas a loading control; however, this system isnot recommended since its use will requireadjustment of C whenever a frequency changeis made at the transmitter. If L and C are maderesonant at the center of a band, with a linkcircuit Q of 3 to 4, and coupling adjustment ismade by physical adjustment of L with respectto the final amplifier tank coil, it usually willbe possible to operate over an entire amateurband without change in the coupling system.Capacitor C normally may have a low voltagerating, even with a high power transmitter, dueto the low Q and low impedance of the couplingcircuit.
<strong>HANDBOOK</strong> Pi-Network Coupling Systems 453COAX. TORECEIVERTO ANTENNAFEEDUNE ORTO ANTENNACOUPLERFigure 37TUNED-LINK OUTPUT CIRCUITCapacitor C should be acljustecl so as to tune out the lncluctlve reactance of the coupling link,L. Loading of the ompllfler then Is var/ecl by physically varying the coupling between the platetank of the final amp/ lfler ancl the antenna coup/ ing I Ink,PI-NetworkCouplingThe pi-network coupling systemoffers two advantages: (1) a me·chanica/ coupling variation isnot required to vary the loading of the final amplifier,and (2) the pi network (if used with anoperating Q of about 15) offers within itself aharmonic attenuation of 40 db or more, in additionto the harmonic attenuation provided bythe additional harmonic attenuating filter. Somecommercial equipments (such as the Collinsamateur transmitters) incorporate an L networkin addition to the pi network, for accomplish·ing the impedance transformation in two stepsand to provide additional harmonic attenuation.Tuning thePi-Section CouplerTuning of a pi-networkcoupling circuit such asillustrated in figure 38 isaccomplished in the following manner: Firstremove the connection between the output ofthe amplifier and the harmonic filter (load).Tune C 2 to a capacitance which is large forthe band in use, adding suitable additional capacitanceby switch S if operation is to be onone of the lower frequency bands. Apply re·duced plate voltage to the stage and dip toresonance with C,. It may be necessary to varythe inductance in coil L, but in any event resonanceshould be reached with a setting of C,which is approximately correct for the desiredvalue of operating Q of the pi network.Next, couple the load to the amplifier(through the harmonic filter), apply reducedplate voltage again and dip to resonance withC,.lf the plate current dip with load is too low(taking into consideration the reduced platevoltage), decrease the capacitance of C 2 andagain dip to resonance, repeating the proce-·dure until the correct value of plate current isobtained with full plate voltage on the stage.There should be a relatively small change requiredin the setting of C, (from the originalsetting of C, without load) if the operating Qof the network is correct and if a large valueof impedance transformation is being employed-aswould be the case when transformingfrom the plate imp e dan c e of a single-COAX TORECEIVERTO FEED LINEOR ANTENNACOUPLERFigure 38P~NETWORK ANTENNA COUPLERThe design of,pl·network output circuits is cliscussecl in Chapter Thirteen, The aclclitiona/ output·end shunting capacitors selected by switch S are for use on the fower frequency ronges. /ncluc·tor L may be selectee/ by a tap switch, it may be continuously variable, or plug-in Inductorsmay be usee/.
454 Antennas and Antenna Matching<strong>THE</strong>R A 0 I 0lPARALLEL-WIRELINE TO ANTENNAZEPPFEEDERSSINGLE-WIREHERTZ ANTENNAOR FEEDER®©51 N!;L.E-WIRENTENNA ORFEEDL.INECOAX. LINETO ANTENNA®®Figure 39ALTERNATIVE ANTENNA-COUPLER CIRCUITSPlug-in coils, one or two variable capacitors of the split-stator variety, one/ a system ofswitches or plugs one/ jocks may be usee/ in the antenna coupler to accomrlish the feecling ofclifferent types of antennas one/ antenna transmission lines from the coax/a input I ine from thetransmitter or from the antenna changeover relay. Link L shou/c/ be resonotec/ with capacitorC at the operating frequency of the transmitter so that the harmonic filter will operate Into aresistive loacl impeclance of the correct nom/no/ value.ended output stage down to the 50-ohm impedanceof the usual harmonic filter and its subsequentload.In a pi network of this type the harmonicattenuation of the section will be adequatewhen the correct value of C, and L are beingused and when the r e sonant dip in C 1issharp. If the dip in C 1 is broad, or if the platecurrent persists in being too high with C 2atmaximum setting, it means that a greater valueof capacitance is required at C 2, assuming thatthe values of C 1 and L are correct.22-11 Antenna CouplersAs stated in the previous section, an antennacoupler is not required when the impedanceof the antenna transmission line is the sameas the nominal impedance of the harmonic fil-
<strong>HANDBOOK</strong>Antenna Couplers455ter, assuming that the antenna feed line is beingoperated with a low standing-wave ratio.However, there are many cases where it is desirableto feed a multi-band antenna from theoutput of the harmonic filter, where a tunedline is being used to feed the antenna, orwhere a long wire without a separate feed lineis to be fed from the output of the harmonicfilter. In such cases an antenna coupler is required.Some harmonic attenuation will be providedby the antenna coupler, particularly if it iswell shielded. In certain cases when a pi networkis being used at the output of the transmitter,the addition of a shielded antenna couplerwill provide sufficient harmonic attenuation.But in all normal cases it will be necessaryto include a harmonic filter between theoutput of the transmitter and the antenna coupler.When an adequate harmonic filter is beingused, it will not be necessary in normalcases to shield the antenna coupler, exceptfrom the standpoint of safety or convenience.Function of anAntenna CouplerThe function of the antennacoupler is, basically, totransform the impedance ofthe antenna system being used to the correctvalue of resistive impedance for the harmonicfilter, and hence for the transmitter. Thus theantenna coupler may be used to resonate thefeeders or the radiating portion of the antennasystem, in addition to its function of impedancetransformation.It is important to remember that there isnothing that can be done at the antenna couplerwhich will eliminate standing waves onthe antenna transmission line. Standing wavesare the result of reflection from the antenna,and the coupler can do nothing about this condition.However, the antenna coupler can resonatethe feed line (by introducing a conjugateimpedance) in addition to providing an impedancetransformation. Thus, a resistive impedanceof the correct value can be presented tothe harmonic filter, as in figure 36, regardlessof any reasonable value of standing-wave ratioon the antenna transmission line.Types ofAntenna CouplersAll usual types of antennacouplers fall into two classifications:(1) inductivelycoupled resonant systems as exemplified bythose shown in figure 39, and (2) conductivelycoupled pi-network systems such as shown infigure 40. The inductively-coupled system ismuch more commonly used, since it is convenientfor feeding a balanced line from the coaxialoutput of the usual harmonic filter. Thepi-network system is most useful for feedinga length of wire from the output of a transmitter.Figure 40PI-NETWORKANTENNA COUPLERAn arrangement such as illustratedabove is convenient for feecling anencl-fecl Hertz antenna, or a ranc/omlength of wire for portable or emer·gency operation, from the nominalvalue of impedance of the harmonicfilter.Several general methods for using the inductively-coupledresonant type of antenna couplerare illustrated in figure 39. The couplingbetween the link coil L and the main tuned circuitneed not be variable; in fact it is preferablethat the correct link size and placementbe determined for the tank coil which will beused for each band, and then that the link bemade a portion of the plug-in coil. CapacitorC then can be adjusted to a pre-determinedvalue for each band such that it will resonatewith the link coil for that band. The reactanceof the link coil (and hence the reactance of thecapacitor setting which will resonate the coil)should be about 3 or 4 times the impedance ofthe transmission line between the aritenna couplerand the harmonic filter, so that the linkcoupling circuit will have an operating Q of3 or 4. The use of capacitor C to resonate withthe inductance of the link coil L will make iteasier to provide a low standing-wave ratioto the output of the harmonic filter, simply byadjustment of the antenna-coupler tank circuitto resonance. If this capacitor is not included,the system still will operate satisfactorily, butthe tank circuit will have to be detuned slightlyfrom resonance so as to cancel the inductivereactance of the coupling link and thusprovide a resistive load to the output of theharmonic filter. Variations in the loading ofthe final amplifier should be made by the couplingadjustment at the final amplifier, not atthe antenna coupler.The pi-network type of antenna coupler, asshown in figure 40 is useful for certain applications,but is primarily useful in feeding asingle-wire antenna from a low-impedancetransmission line. In such an application theoperating Q of the pi network may be somewhatlower than that of a pi network in the plate circuitof the final amplifier of a transmitter, asshown in figure 38. An operating Q of 3 or 4
456 Antennas and Antenna Matching<strong>THE</strong><strong>RADIO</strong>PARALLEL-WIRE TO40·00 M, ANTENNAr------------1ITORECEIVER~2.1lCOAXIAL~~~~ ::jF===:=JXMTR~INGLE WIREANTENNA""t:.....+III115 V. TO RELAY ICOIL______ jTOTRANSMITTER THROUGHHARMONIC FILTERFigure 41ALTERNATIVE COAXIAL ANTENNACOUPLERThis circuit is recommenclecl not only as beIng most desirable when coaxial lines withlow s. w. r. are being userl to feerl antennasystems such as rotatable beams, but whenIt also Is cles/recl to feecl through open-wireline to some sort of multi-bancl antenna forthe lower frequency ranges. The tunecl clr·cult of the antenna coupler is operative onlywhen using the open-wire feerl, ancl then ItIs In operation both for transmit anrl receive.in such an application will be found to be ade·quate, since hatmonic attenuation has beenaccomplished ahead of the antenna coupler.However, the circuit will be easier to tune;although it will not have as great a bandwidth,if the operating Q is made higher.An alternative arrangement shown in figure41 utilizes the antenna coupling tank circuitonly when feeding the coaxial output of thetransmitter to the open-wire feed line (or similatmulti-band antenna) of the 40- 80 meter antenna.The coaxial lines to the 10-meter beamand to the 20-meter beam would be fed directlyfrom the output of the coaxial antenna change·over relay through switch S.22-12 A Single-Wire Antenna TunerOne of the simplest and least expensiveantennas for transmission and reception isthe single wire, end-fed Hertz antenna. Whenused over a wide range of frequencies, thistype of antenna exhibits a very great range ofinput impedance. At the low frequency end ofthe spectrum such an antenna may present aresistive load of less than one ohm to thetransmitter, combined with a large positive ornegative value of reactance. As the frequencyof operation is raised, the resistive load mayFigure 42ANTENNA TUNER AND SWRINDICATOR FOR RANDOMLENGTH HERTZ ANTENNArise to several thousand ohms (near half-waveresonance) and the reactive component of theload can rapidly change from positive to nega·tive values, or vice-versa.It is possible to match a 52·ohm trans·mission line to such an antenna at almost anyfrequency between 1.8 me and 30 me with theuse of a simple tuner of the type shown infigure 42. A vatiable series inductor L, and avariable shunt capacitor CI permit circuitresonance and impedance transformation tobe established for most antenna lengths.Switch SI permits the selection of seriescapacitor C for those instances when thesingle wire antenna exhibits large values ofpositive reactance.To provide indication for the tuning of thenetwork, a radio frequency bridge (SWR meter)is included to indicate the degree of mis·match (standing wave ratio) existing at theinput to the tuner. All adjustments to thetuner are made with the purpose of reachingunity standing wave ratio on the coaxial feedsystem between the tuner and the transmitter.A PracticalAntenna TunerA simple antenna tuner foruse with transmitters of250 watts power or lessis shown in figures 43 through 46. A SWRbridge circuit is used to indicate tuner re·sonance. The resistive arm of the bridge con·sists of ten 10·ohm, l·watt carbon resistorsconnected in parallel to form a 1·ohm resistor(R1). The other pair of bridge arms are ca·paCIUve rather than resistive. The bridgedetector is a simple r·f voltmeter employing a1N56 crystal diode and a 0·1 d.c. milliammeter.A sensitivity control is incorporated to preventoverloading the meter when power is firstapplied to the tuner. Final adjustments aremade with the sensitivity control at its maxi·
<strong>HANDBOOK</strong>Single-wire Antenna Tuner 45752 n INPUTFROM XMTRL1Figure 43ANTENNA TUNER IS HOUSEDIN METAL CABINET 7" x 8" INSIZE.L1- ;~5~!-'~~J
Figure 45REAR VIEW OFTUNER SHOWINGPLACEMENT OFMAJOR COMPON-ENTS.Rotary inductor is drivenby Johnson 116-208-4 counter dial. Coaxialinput receptacle J lIs mounted directly belowrotary inductor.ohm termination. The transmitter is turned on(preferably at reduced input) and resonance isestablished in the amplifier tank circuit. Thesensitivity control of the tuner is adjusted toprovide near full scale deflection on the bridgemeter. Various settings of Sl, L2, and Clshould be tried to obtain a reduction of bridgereading. As tuner resonance is approached,the meter reading will decrease and the sensitivitycontrol should be advanced. When thesystem is in resonance, the meter will readzero. All loading adjustments may then bemade with the transl,lli.tter controls. The tunershould be readjusted whenever the frequencyof the transmitter is varied by an appreciableamount.Figure 46CLOSE-UP OF SWR BRIDGESimple SWR briclge is mounteclbelow the chassis of the tuner.Carbon resistors are mounted totwo copper rings to form lowinductance one-ohm resistor.Bridge capacitors lorm triangularconfiguration for lowest I eocllncluctance. Balancing capacitorC2 is at lower right.
CHAPTER TWENTY-THREEHigh frequencyAntenna ArraysIt is becoming of increasing importance inmost types of radio communication to be capableof concentrating the radiated signal fromthe transmitter in a certain desired directionand to be able to discriminate at the receiveragainst reception from directions other thanthe desired one. Such capabilities involve theuse of directive antenna arrays.Few simple antennas, except the single verticalelement, radiate energy equally well inall azimuth (horizontal or compass) directions.All horizontal antennas, except those specificallydesigned to give an omnidirectional azimuthradiation pattern such as the turnstile,have some directive properties. These propertiesdepend upon the length of the antenna inwavelengths, the height above ground, and theslope of the radiator.The various forms of the half-wave horizontalantenna produce maximum radiation at rightangles to the wire, but the directional effectis not great. Nearby objects also minimize thedirectivity of a dipole radiator, so that it hardlyseems worth while to go to the trouble torotate a simple half-wave dipole in an attemptto improve transmission and reception in anydirection.The half-wave doublet, folded dipole, zepp,single-wire-fed, matched impedance, and JohnsonQ antennas all have practically the sameradiation pattern when properly built and ad-459justed. They all are dipoles, and the feedersystem, if it does not radiate in itself, willhave no effect on the radiation pattern.23-1 Directive AntennasWhen a multiplicity of radiating elements islocated and phased so as to reinforce the radiationin certain desired directions and to neutralizeradiation in other directions, a directiveantenna array is formed.The function of a directive antenna whenused for transmitting is to give an increase insignal strength in some direction at the expenseof radiation in other directions. For reception,one might find useful an antenna givinglittle or no gain in the direction from whichit is desired to receive signals if the antennais able to discriminate against interfering signalsand static arriving from other directions.A good directive transmitting antenna, however,can also be used to good advantage for reception.If radiation can be c o n f i n e d to a narrowbeam, the signal intensity can be increased agreat many times in the desired direction oftransmission. This is equivalent to increasingthe power output of the transmitter. On thehigher frequencies, it is more economical to
460 High Frequency Directive Antennas<strong>THE</strong> R AD I 0;--...~--.~. ~"DOUBLE HOPl~r-- ~~--'ll f---~~"" T
<strong>HANDBOOK</strong> Long Wire Radiators 461Figure 3DIRECTIVE GAIN OFLONG-WIRE ANTENNAS1/):X:••...t; 12zIU.J~ 10~~:X:t;zIU.JIUa:i•LONG STRAIGHT WIRE ANTENNAS•.. ~~2........0...... ~""Lv1/v/lL~lL_~/'12.3"' 5 e 1 a •10DB POWER RATIO Of MAIN LOBE TO A DIPOLEvTypes ofDirective ArraysThere is an enormous varietyof directive antenna arraysthat can give a substantialpower gain in the de sired direction of transmissionor reception. However, some are moreeffective than others requiring the same space.In general it may be stated that long-wire antennasof various types, such as the singlelong wire, the V beam, and the rhombic, areless effective for a given space than arrayscomposed of resonant elements, but the longwirearrays h ave the significant advantagethat they may be used over a relatively largefrequency range while resonant arrays are usableonly over a quite narrow frequency band.23-2 Long Wire RadiatorsHarmonically operated long wires radiatebetter in certain directions than others, butcannot be considered as having appreciabledirectivity unless several wavelengths long.The current in adjoining half-wave elementsflows in opposite directions at any instant,and thus, the radiation from the various elementsadds in certain directions and cancelsin others.A half-wave do u b 1 e t in free space has a"doughnut" of radiation surrounding it. A fullwave has 2 lobes, 3 half waves 3, etc. Whenthe radiator is made more than 4 half wavelengthslong, the end lobes (cones of radiation)begin to show noticeable power gain overa half-wave doublet, while the broadside lobesget smaller and smaller in amplitude, eventhough numerous (figure 2).The horizontal radiation pattern of such antennasdepends upon the vertical angle of radiationbeing considered. If the wire is morethan 4 wavelengths long, the maximum radiationat vertical angles of 15° to 20° (usefulfor dx) is in line with the wire, being slightlygreater a few degrees either side of the wirethan directly off the ends. The directivity ofthe main lobes of radiation is not particularlysharp, and the minor lobes fill in between themain lobes to permit working stations in nearlyall directions, though the power radiatedbroadside to the radiator will not be great ifthe radiator is more than a few wavelengthslong. The directive gain of long-wire antennas,in terms of the wire length in wavelengths isgiven in figure 3.To maintain the out-of-phase condition inadjoining half-wave elements throughout thelength of the radiator, it is necessary that aharmonic antenna be fed either at one end orat a current loop. If fed at a voltage loop, theadjacent sections will be fed in phase, and adifferent radiation pattern will result.The directivity of a long wire does not increasevery much as the length is increasedbeyond about 15 wavelengths. This is due tothe fact that all long-wire antennas are adverselyaffected by the r-f resistance of thewire, and because the current amplitude beginsto become unequal at different currentloops, as a result of attenuation along the wirecaused by radiation and losses. As the lengthis increased, the tuning of the antenna becomesquite broad. In fact, a long wire about15 waves long is practically aperiodic, andworks almost equally well over a wide rangeof frequencies.
462 High Frequency Directive Antennas <strong>THE</strong> R AD I 0LONG- ANTENNA DESIGN TABLEAPPROXIMATE LENGTH IN FEET-END-FED ANTENNASFREQUENCY 1.>- 1t>- 2>. 2t>- 3>. Jt>- 4>. 4-k->-IN MC.2.9 33 50 87 84 101 118 135 152.2.8 34 52. 89 87 104 12.2 140 1572.1.4 45 88 91 112 1 14 1/2 138 f/2. teo 1/2. 185 1/2. 209 1/221.2 45 1/4 88 1/4 91 3/4 114 3/4 138 3,A. 11503/4 185 3/4 209 3/42.1.0 45 112 88 112 92. 115137 181 186 2.1014.2 87 112. 102. 137 17114.0 158 1/2 103 1/2. 139 1747.3 138 2.08 278 3487.15 138 1/2 207 277 3477.0 137 2.07 1/2 2.77 1/2. 3484.0 240 362 485 8183.6 2.52 381 511 8403.6 268 403 540 6783.5 2.74 414 555 8902.0 480 725 972. 12.30... 504 783 1020 12801.8 532 605 10802.08 2.40 275 310209 2.44 270 314418 488 555 825417 487 557 82.7418 468 558 828730 853 971 1100770 900 1030 1180812 950 1090 1220835 877 11201475One of the most practical methods of feedinga long-wire antenna is to bring one end ofit into the radio room for direct connection toa tuned antenna circuit which is link-coupledthrough a harmonic-attenuating filter to thetransmitter. The antenna can be tuned effec·tively to resonance for operation on any hat·monic by means of the tuned circuit which isconnected to the end of the antenna. A groundis sometimes connected to the center of thetuned coil.If desired, the antenna can be opened andcurrent-fed at a point of maximum current btmeans of low-impedance ribbon line, or by aquarter-wave matching section and open line.23-3 The V AntennaIf two long-wire antennas are built in theform of a V, it is possible to make two of themaximum lobes of one leg shoot in the samedirection as two of the maximum lobes of theother leg of the V. The resulting antenna isbidirectional (two opposite directions) for themain lobes of radiation. Each side of the Vcan be made any odd or even number of quarterwavelengths, depending on the method of feedingthe apex of the V. The complete systemmust be a multiple of half waves. If each legis an even number of quarter waves long, theantenna must be voltage-fed at the apex; if anodd number of quarter waves long, current feedmust be used.By choosing the proper apex angle, figure4 and figure 5, the lobes of radiation from thetwo long-wire antennas aid each other to forma bidirectional beam. Each wire by itselfwould have a radiation pattern similar to that18 014 0.,..,12. 0~100a:\~. 0zo/) • 0402.000f\',~~r----2 .. •LENGTH IN "l" WAVELENGTHSto-12Figure 4INCLUDED ANGLE FOR A"V" BEAMShowing the incluclecJ angle betweenthe legs of a V beam forvarious leg lengths. For opt/•mum alignment of the racliotionlobe at the c:orrec:t vertic:alangle with leg lengths less thanthree wavelengths, the optimuminc:luclecJ angle is shown by theclashecJ c:urve.
<strong>HANDBOOK</strong>Figure 5TYPICAL "V" BEAM ANTENNAThe V Antenna 463132II&J> 10- L=2>- L=4>. L =8>-IN KILOCYCLES 6 = oo• 6 = 70° 6 = ~2.· 6= 39.2.8000 34'8 11 89'8" 140' 2.80'2.9000 33'6"' 67'3" 13~' 271 12.1100 4!1'9" 91'9' 1 183' 306'2.1300 45'4" 91'4 .. 182.' e• 36!1 114050 69' 139' 279 1 558'141 so 68'6" 138' 277' 555'142.50 68'2 1 137' 2.75' 552'7020 I 38 1 2' 278 1 558 1 1120 17100 138'8"' 2.75' 552.' 1106'72.00 134'10" 271 I 545' f 090 I
464 High Frequency Directive Antennas23-4 The Rhombic:AntennaThe terminated rhombic or diamond is probablythe most effective directional antennathat is practical for amateur communication.This antenna is non-resonant, with the resultthat it can be used on three amateur bands,such as 10, 20, and 40 meters. When the antennais non-resonant, i.e., properly terminated,the system is undirectional, and thewire dimensions are not critical.2.0... '·',_,'·' '-'1.4'-'::I:1.2'-'1.00.9o.•0.,o .•0.,64• 12J::TILTANGU:1!16• SIN 0:: COS ill.6&- 0= 90'" .0.<strong>THE</strong>H:: HEIGHT IN WAVELENGTHS = 4S~'N aR A 0 I 0RhombicTerminationWhen the free end is terminatedwith a resistance of a valuebetween 700 and 800 ohms thebackwave is eliminated, the forward gain isincreased, and the antenna can be used onseveral bands without changes. The terminatingresistance should be capable of dissipatingone-third the power output of the transmitter,and should have very little reactance.For medium or low power transmitters, thenon-inductive plaque resistors will serve asa satisfactory termination. Several manufacturersoffer special resistors suitable for terminatinga rhombic antenna. The terminating deviceshould, for technical reasons, present asmall amount of inductive reactance at thepoint of termination.A compromise terminating device commonlyused consists of a terminated 250-foot orlonger length of line, made of resistance wirewhich does not have too much resistance perunit length. If the latter qualification is notmet, the reactance of the line will be excessive.A 250-foot line consisting of no. 25nichrome wire, spaced 6 inches and terminatedwith 800 ohms, will serve satisfactorily. Becauseof the at ten u at ion of the line, thelumped resistance_ at the end of the line needdissipate but a few watts even when high poweris used. A half-dozen 5000-ohm 2-watt carbonresistors in parallel will serve for all exceptvery high power. The attenuating linemay be folded back on itself to take up lessroom.The determination of the best value of terminatingresistor may be made while receiving,if the input impedance of the receiver is approximately800 ohms. The value of resistorwhich gives the best directivity on receptionwill not give the most gain when transmitting,but there will be little difference between thetwo conditions.The input resistance of the rhombic whichis reflected into the transmission line thatfeeds it is always somewhat less than theterminating resistance, and is around 700 to750 ohms when the terminating resistor is 800ohms.24 "2322"~:! ~~!\!111 ~~~~~~i·~:~4_ ~~4-~T~1 e• tz• te• ze• 2.&"'WAVE ANGLE L::.Figure 7RHOMBIC ANTENNA DESIGN TABLEDesign data is given in terms of the waveangle (vertical angle of transmission and re•ception) of the antenna. The lengths I arelor the" maximum output" clesign; the shorterlengths I 1 are for the "alignment" methodwhich gives approximately 1.5 db less gainwith a considerable reduction In the spacerequired for the antenna. The values of sidelength, tilt angle, and height for a givenwave angle are obtainecJ by drawing a verticalline upward from the desired waveangle.The antenna should be fed with a non-resonantline having a characteristic impedanceof 650 to 700 ohms. The four corners of therhombic should be at I east one-half wavelengthabove ground for the lowest frequencyof operation. For three-band operation theproper tilt angle ¢ for the center band shouldbe observed.The rhombic antenna transmits a horizontally-polarizedwave at a relatively low angleabove the horizon. The an g I e of radiation(wave angle) decreases as the height aboveground is increased in the same manner aswith a dipole antenna. The rhombic shouldnot be tilted in any plane. In other words, thepoles should all be of the same height andthe plane of the antenna should be parallelwith the ground.
<strong>HANDBOOK</strong>The Rhombic Antenna 465Figure 8TYPICAL RHOMBICANTENNA DESIGNThe antenna system illustrateclabove may be usee/over the frequency range from7 to 29 Me. without change.The directivity of the systemmay be reversed by the sys•tem c/iscussec/ in the text.SPACIN~ BETWEEN SlOES S,= 214 FEETTOTAL LEN~TH =-rtt2 FEETTERMINATING LINEOF 250' OF N• 2.&NICHROME SPACED 6"AND &00-0HM 16 -WATTCARBON RESISTOR ATEND. 8 2-WATT 100-0HMRESISTORS IN SERIESA considerable amount of directivity is lostwhen the terminating resistor is left off theend and the system is operated as a resonantantenna. If it is desired to reverse the direc·tion of the antenna it is much better practiceto run transmission lines to both ends of theantenna, and then run the terminating line tothe operating position. Then with the aid oftwo d·p·d·t switches it will be possible to con·nect either feeder to the antenna changeoverswitch and the other feeder to the terminatingline, thus rever sing the direction of the arrayand maintaining the same termination foreither direction of operation.Figure 7 gives curves for optimum-designrhombic antennas by both the maximum-out·put method and the alignment method. Thealignment method is about 1.5 db down fromthe maximum output method but requires onlyabout 0.74 as much leg length. The height andtilt angle is the same in either case. Figure8 gives construction data for a recommendedrhombic antenna for the 7.0 through 29.7 Me.bands. This antenna will give about 11 dbgain in the 14.0-Mc. band. The approximategain of a rhombic antenna over a dipole, bothabove normal soil, is given in figure 9.23-5 Stacked-DipoleArraysThe characteristics of a half-wave dipolealready have been described. When anotherdipole is placed in the vicinity and excitedeither directly or parasitically, the resultantradiation pattern will depend upon the spacingand phase differential, as well as the relativemagnitude of the currents. With spacings lessthan 0.65 wavelength, the radiation is mainlybroadside to the two wires (bidirectional) whenthere is no phase difference, and through thewires (end fire) when the wires are 180° outof phase. With phase differences between 0°Figure 9RHOMBIC ANTENNA GAINShowing the theoretical gain of a rhombicantenna, in terms of the side length, over ahalf-wave antenna mounted at the sameheight above the same type of soil.f7.J"''"~"Ci ,..~ 13~12IL1 f~ 10J:a:IU 0>0 7Ill0 •:;; .z 4IILI'/vL~ 3o 1 2 3 ,. s e 1 a v to tt 12 t3 14 1s te 11 11 tt zoLI-'I"""r- ~1--::: !-"•.Q." =LENGTH OF EACH LEG OF RHOMBIC IN WAVELENGTH-"
466High Frequency Directive Antennas<strong>THE</strong> R A 0 I 0I-s ---1... ...PLANE OF WIRESEND VIEWL1 Lz Lz L1S=i-1ao• OUT OF PHASE(FLAT-TOP BEAM, ETC.)~-.....----s=+\ I IN PHASE,_ ... /Figure 10{LAZY H. STERBA CURTAIN)RADIATION PATTERNS OF A PAIR OFDIPOLES OPERATING WITH IN-PHASEEXCITATION, AND WITH EXCITATION180° OUT OF PHASEIf the clipoles are orientecl horizontally mostof the directivity will be In the verticalplane; if they are orientecl vertically mostof the directivity will be in the horizontalplane.and 180° (45°, 90°, and 135° for instance),the pattern is unsymmetrical, the radiation be·ing greater in one direction than in the oppositedirection.With spacings of more than 0.8 wavelength,more than two main lobes appear for all phasingcombinations; hence, such spacings areseldom used.In-PhaseSpacingWith the dipoles driven so as tobe in phase, the most effectivespacing is between 0. 5 and 0. 7wavelength. The latter provides greater gain,but minor lobes are present which do not appearat 0.5-wavelength spacing. The radiationis broadside to the plane of the wires, and thegain is slightly greater than can be obtainedfrom two dipoles out of phase. The gain fallsoff rapidly for spacings less than 0.375 wavelength,and there is little point in using spacingof 0. 25 wavelength or less with in-phasedipoles, except where it is desirable to increasethe radiation resistance. (See MultiWire Doublet.)Out of Phase When the dipoles are fed 180°Spacing out of phase, the directivity isthrough the plane of the wires,and is greatest with close spacing, thoughthere is but l i ttl e difference in the patternafter the spacing is made less than 0.125wavelength. The radiation resistance becomesso low for spacings of less than 0.1 wavelengththat such spacings are not practicable.Figure 11<strong>THE</strong> FRANKLIN OR COLINEARANTENNA ARRAYAn antenna of this type, regardless of thenumber of elements, attains all of its clirectlvltythrough sharpening of the horizontalor azimuth radiation pattern; no vertical directivityIs praviclecl, Hence a long antennaof this type has on extremely sl-.arp azimuthpaHern, but no vertical directivity.In the three foregoing examples, most of thedirectivity provided is in a plane at a rightangle to the wires, though when out of phase,the directivity is in a line through the wires,and when in phase, the directivity is broadsideto them. Thus, if the wires are oriented vertically,mostly horizontal directivity will beprovided. If the wires are oriented horizontally,most of the directivity obtained will be verti·cal directivity.. To increase the sharpness of the directivitym all planes that include one of the wires,additional identical elements are added in theline of the wires, and fed so as to be in phase.The familiar H array is one array utilizingboth types of directivity in the manner prescribed.The two-section Kraus flat-top beamis another.These two antennas in their various formsare directional in a horizontal plane in additionto being low-angle radiators, and are perhapsthe most practicable of the bidirectionalstacked-dipole arrays for amateur use. Morephased elements can be used to provide greaterdirectivity in planes including one of theradiating elements. The H then becomes aSterba·curtain array.For unidirectional work the most practicablestacked-dipole arrays for amateur-band useare parasitically-excited systems using relativelyclose spacing between the reflectorsand the directors. Antennas of this type aredescribed in detail in a later chapter. Thenext most practicable unidirectional array isan H or a Sterba curtain with a similar systemplaced approximately one-quarter wave behind.The use of a reflector system in conjunctionwith any type of stacked-dipole broadside arraywill increase the gain by 3 db.
<strong>HANDBOOK</strong>Colinear Arrays 467FREQUENCYIN MC.2.&.S2.1.2.COLI NEAR ANTENNA DESIGN CHARTL1 L2. L316'8N 17' 8' eu2.2.'8" 23'3" 11'8"lo------nk---~800 _ __j1------ F(MC) -------,nfff.fuA BA- B= 150ft FEED POINT GAl N APPRO X. 3 DB14. 2.7, I~4.03.833'8"' 34 1 7 11 17' 3•67' 88'8'' 34'4"12.0' 12a' 81 I 8N133' 138'5" 88'2"Figure 12DOUBLE EXTENDED ZEPP ANTENNAFor best results, antenna sloou/c/ be tunec/ faop&ratlng fr&qu&ncy by m&ans of gric/-c/lposcillator.Co linearArraysThe simple colinear antenna arrayis a very effective radiating systemfor the 3. 5-Mc. and 7 .0-Mc. bands,but its use is not recommended on higher fre·quencies since such arrays do not possessany vertical directivity. The elevation radia·tion pattern for such an array is essentiallythe same as for a half-wave dipole. This considerationapplies whether the elements are ofnormal length or are extended.The colin ear antenna consists of two ormore radiating sections from 0.5 to 0.65 wavelengthslong, with the current in phase in eachsection. The necessary phase reversal betweensections is obtained through the use of resonanttuning stubs as illustrated in figure 11.The gain of a colinear array using half-waveelements (in decibels) is approximately equalto the number of elements in the array. Theexact figures are as follows:Number of Elements 2 3 4 5 6Gain in Decibels 1.8 3.3 4.5 5.3 6.2As additional in-phase colinear elementsare added to a doublet, the radiation resistancegoes up much faster than when additional halfwaves are added out of phase (harmonic operatedantenna).For a colinear array of from 2 to 6 elements,1--Ne)---ool~- + ----- ~------~TWO72.Jl. BALANCED LINESOF EQUAL UNGTHthe terminal radiation resistance in ohms atany current looP. is approximately 100 timesthe number of elements.It should be borne in mind that the gain froma colin ear antenna depends upon the sharpnessof the horizontal directivity since no verticaldirectivity is provided. An array with seve~alcolinear elements will give considerable gam,but will have a s h a r p horizontal radiationpattern.Double ExtendedZeppThe gain of a conventionaltwo-element Franklin colinearantenna can be increasedto a value approaching that obtained from athree-element Franklin, simply by making thetwo radiating elements 230° long instead of180° long. The phasing stub is shortened co_rrespondinglyto maintain the whole array 10resonance. Thus, instead of having 0.5-wavelengthelements and 0.25-wavelength stub, theelements are made 0.64 wavelength long andthe stub is approximately 0.11 wavelengthlong.Dimensions for the double extended Zeppare given in figure 12.The vertical directivity of a colinear antennahaving 230° elements is the same as forone having 180° elements. There is little advantagein using extended sections when thetotal length of the array is to be greater thanabout 1. 5 wavelength overall since the gainFigure 13TWO COLINEAR HALF-WAVE ANTENNASIN PHASE PRODUCE A 3 DB GAIN WHENSEPARATED ONE-HALF WAVELENGTHGAIN APPRO X. 3 DBFigure 14PRE-CUT LINEAR ARRAY FOR 40-METEROPERATION
468 High Frequency Directive Antennas<strong>THE</strong> R AD I 0of a colinear antenna is proportional to theoverall length, whether the individual radiatingelements are ~ wave, ~ wave or % wave inlength.L1L1Spaced HalfWave AntennasThe gain of two colinear halfwaves may be increased byincreasing the physical spacingbetween the elements, up to a maximum ofabout one half wavelength. If the half waveelements are fed with equal lengths of transmissionline, poled correctly, a gain of about3.3 db is produced. Such an antenna is shownin figure 13. By means of a phase reversingswitch, the two elements may be operated outof phase, producing a cloverleaf pattern withslightly less maximum gain.A three element "precut" array for 40 meteroperation is shown in figure 14. It is fed directlywith 300 ohm "ribbon line," and maybe matched to a 52 ohm coaxial output transmitterby means of a Balun, such as the Barker& Williamson 3975. The antenna has a gain ofabout 3.2 db, and a beam width at half-powerpoints of 40 degrees.L1L3L2L1QUARTER-WAVE STUBGAIN APPROX. S.~ DBL223-6 Broadside ArraysColinear elements may be stacked above orbelow another string of colinear elements toproduce what is commonly called a broadsidearray. Such an array, when horizontal elementsare used, possesses vertical directivity inproportion to the number of broadsided (verticallystacked) sections which have beenused. Since broadside arrays do have good verticaldirectivity their use is recommended onthe 14-Mc. band and on those higher in frequency.One of the most popular of simplebroadside arrays is the "Lazy H" array of figure15. Horizontal colinear elements stackedtwo above two make up this antenna systemwhich is highly recommended for work on frequenciesabove perhaps 14-Mc. when moderategain without too much directivity is desired.It has high radiation resistance and a gain ofapproximately 5.5 db. The high radiation resistanceresults in low voltages and a broadresonance curve, which permits use of inexpensiveinsulators and enables the array to beused over a fairly wide range in frequency.For dime'nsions, see the stacked dipole designtable.StackedDipolesVertical stacking may be appliedto strings of colinear elementslonger than two half waves. Insuch arrays, the end quarter wave of eachstring of radiators usually is bent in to meetL1RESONANT FEEO LINE®Figure 15<strong>THE</strong> "LAZY H" ANTENNA SYSTEMStacking the co/ /near pairs gives both horizontalone/ vertic a/ directivity. As shown thearray wJ/1 give about 5.5 clb gain. Note 'thatthe array may be feel either at the center ofthe phasing section or at the bottom; if feelat the bottom the phasing section must betwisted through 180°.a similar bent quarter wave from the oppositeend radiator. This provides better balance andbetter coupling between the upper and lowerelements when the array is current-fed. Arraysof this type are shown in figure 16, and arecommonly known as curtain arrays.Correct length for the elements and stubscan be determined for any stacked dipole arrayfrom the Stacked-Dipole Design Table.In the sketches of figure 16 the arrowheadsrepresent the direction of current flow at anygiven instant. The dots on the radiators repre-L1
<strong>HANDBOOK</strong> Broadside Arrays 469L> L3tsTUB®GAl N APPRO X. 6 DBNON-RESONANT ,EEDER®GAIN APPROX. 8 DBFigure 16<strong>THE</strong> STERBA CURTAIN ARRAYApproximate directive gains along with alternativefeed methods are shown.GAIN APPROX. 6 DBsent points of maximum current. All arrowsshould point in the same direction in each portionof the radiating sections of an antenna inorder to provide a field in phase for broadsideradiation. This condition is satisfied for thearrays illustrated in figure 16. Figures 16Aand 16C show simple methods of feedinga short Sterba curtain, while an alternativemethod of feed is shown in the higher gain antennaof figure 16B.In the case of each of the arrays of figure16, and also the "Lazy H" of figure 15, thearray may be made unidirectional and the gainincreased by 3 db if an exactly similar arrayis constructed and p 1 aced approximately ~wave behind the driven array. A screen or meshof wires slightly greater in area than the antennaarray may be used instead of an additionalarray as a reflector to obtain a unidirectionalsystem. The spacing between the reflectingwires may vary from 0.05 to 0.1 wavelengthwith the spacing between the reflectingwires the smallest directly behind the drivenelements. The wires in the untuned reflectingsystem should be parallel to the radiating elementsof the array, and the spacing of the completereflector system should be approximately0.2 to 0.25 wavelength behind the driven elements.On frequencies below perhaps 100 Me. itnormally will be impracticable to use a wirescreenreflector behind an antenna array suchas a Sterba curtain or a "Lazy H." Parasiticelements may be used as reflectors or directors,but parasitic elements have the disadvantagethat their operation is selective with respectto relatively small changes in frequency.Nevertheless, parasitic reflectors for such arraysare quite widely used.The X-Array In section 23-5 it was shGwnhow two dipoles may be arrangedin phase to provide a power gain of (some) 3db. If two such pairs of dipoles are stackedLAZY-H AND STERBA(STACKED DIPOLE) DESIGN TABLEFREQUENCYIN MC. L, Lo L.,7.0 68'2" 70' 35'7.3 65'10" 67'6" 33'9"14.0 34'1" 35' 17'6"14.2 33'8" 34'7" 17'3"14.4 33'4" 34'2" 17'21.0 22'9" 23'3" 11 '8"21.5 22'3" 22'9" 11 '5"27.3 17'7" 17'10" 8'11"28.0 17' 17'7" 8'9"29.0 16'6" 17' 8'6"50.0 9'7" 9'10" 4'11"52.0 9'3" 9'5" 4'8"54.0 8'10" 9'1" 4'6"144.0 39.8" 40.5" 20.3"146.0 39" 40" 20"148.0 38.4" 39.5" 19.8"
470 High Frequency Directive Antennas<strong>THE</strong> R AD I 01--r --L -----.ofe-~ 0 -+~i:::::=::::;::;:,;:=:::::JsDIMENSIONS10M. ISM. 20t.4. GAIN APPROX. I DBL 10'3* 2.2' 32'10"s 20' 30' 40' Jp 14'2· 21'3., 28'4 ..0 3·~ S•J• 7•0• 7SA TRANSMISSION LINEFigure 17<strong>THE</strong> X-ARRAY FOR 28 MC., 21 MC.,OR 14 MC.The entire array (with the exception of the75-ohm feecl I ine) is constructed of 300-ohmribbon line. Be sure phasing lines (P) arepolecl correctly, as shown.in a vertical p 1 an e and properly phased, asimplified form of in-phase curtain is formed,providing an overall gain of about 6 db. Suchan array is shown in figure 17. In this X-array,the four dipoles are all in phase, and are fedby four sections of 300-ohm line, each onehalfwavelength long, the free ends of all fourlines being connected in parallel. The feedimpedance at the junction of these four linesis about 75 ohms, and a length of 75-ohmTwin-Lead may be used for the feedline tothe array.An array of this type is quite small for the28-Mc. band, and is not out of the questionfor the 21-Mc. band. For best results, the bottomsection of the array should be one-halfwavelength above ground.The Double-BruceArrayThe Bruce Beam consistsof a long wire folded sothat vertical elementscarry in-phase currents while the horizontalelements carry out of phase currents. Radiationfrom the horizontal sections is low sinceonly a small current flows in this part of thewire, and it is largely phased-out. Since theheight of the Bruce Beam is only one-quarterwavelength, the gain per linear foot of arrayis quite low. Two Bruce Beams may be combinedas shown in figure 18 to produce theDouble Bruce array. A four section DoubleBruce will give a vertically polarized emission,with a power gain of 5 db over a simpleDIMENSION L10M. ISM. 2.0/IOM.e•t• 13' t7•euFigure 18<strong>THE</strong> DOUBLE-BRUCE ARRAY FOR 10,15, AND 20 METERSIf a 600-ohm feecl line Is usecl, the 20-meterarray will a/so perform on 10 meters as aSterba curtain, with an approximate gain of9 clb.dipole, and is a very simple beam to construct.This antenna, like other so-called "broadside"arrays, radiates maximum power at right anglesto the plane of the array.The feed impedance of the Double Bruce isabout 750 ohms. The array may be fed with aone-quarter wave stub made of 300-ohm ribbonline and a feedline made of 150-ohm ribbonline. Alternatively, the array may be fed di·rectly with a wide-spaced 600-ohm transmis·sion line (figure 18). The feed 1 in e shouldbe brought away from the Double Bruce for ashort distance before it drops downward, toprevent interaction between the feedline andthe lower part of the center phasing section ofthe array. For best results, the bottom sectionsof the array should be one-half wavelengthabove ground.Arrays such as the X-array and the DoubleBruce are essentially high impedance devices,and exhibit relatively broad-band characteristics.They are less critical of adjustment thana parasitic array, and they work well over awide frequency range such as is encounteredon the 28-29.7 Me. band.The "Bi-Square"Broadside ArrayIllustrated in figure 19 is asimple method of feeding asmall broadside array firstdescribed by W6BCX several years ago as apractical method of suspending an effectivearray from a single pole. As two arrays of thistype can be supported at right angles from asingle pole without interaction, it offers asolution to the problem of suspending two arraysin a restricted space with a minimum oferection work. The free space directivity gainis slightly less than that of a Lazy H, but is
<strong>HANDBOOK</strong>INTS OF MAXIMUMCURRENTBroadside Arrays 471...r~8'7• ----EACH //SL EFLECTOR'-- 8' 7" _.J/ ~- EACH$10£~ !' 'C=50JJJJFTUNE FOR MINIMUM150 OHM BALANCED LINEPICKUP OFF REAR STUB=10* TO TUNING UNIT OROF BEAM. ..14 E. TRANSMITTER.SPACED 3"NOrE: SIDE LENGTH= 17'4" FOR 21 MC.11' 7• FOR 14 MC.ELEMENT SPACING 16" FOR EACH BAND,STUB LENGTH APPRO X, 15" FOR 2 7 MC.zo• FOR 14 MC.GAl N APPRO X. 4 DBFigure 1915>o.A LINE TO TRANSMITTER<strong>THE</strong> "BI-SQUARE" BROADSIDE ARRAYThis bic/irectionol orroy is relatec/ to the"Lazy H," ancl in spite of the oblique ele·ments, is horizontally polarizec/. It has slight·ly less goln one/ c/irectlvlty than the LazyH, the free space clirectivity gain being ap·proximately 4 clb. Its chief aclvantage is thefact that only a single pole Is requlrec/ forsupport, ancl two such arrays may be sup·portecl from a single pole without interactionif the planes of the elements ore at rightangles. A 600-ohm line may be substitutec/for the Twin-Leacl, one/ either operatec/ as aresonant line, or made non-resonant by theincorporation of o matching stub.still worthwhile, being approximately 4 db overa half-wave horizontal dipole at the same aver·age elevation.When two Bi·Square arrays are suspendedat right angles to each other (for general cov·erage) from a sing 1 e pole, the Q sectionsshould be well separated or else symmetricallyarranged in the form of a square (the diagonalconductors forming one Q section) in order tominimize coupling between them. The sameapplies to the line if open construction is usedinstead of Twin-Lead, but if Twin-Lead isused the coupling can be made negligible sim·ply by separating the two Twin-Lead lines byat least two inches and twisting one Twin·Lead so as to effect a transposition every footor so.Figure 20<strong>THE</strong> CUBICAL-QUAD ANTENNA FOR<strong>THE</strong> l~METER BANDWhen tuned feeders are employed, the Bi·Square array can be used on half frequency asan end-fire vertically polarized array, givinga slight practical dx signal gain over a verticalhalf-wave dipole at the same height.A second Bi-Square serving as a reflectormay be placed 0.15 wavelength behind this an·tenna to provide an overall gain of 8.5 db. Thereflector may be tuned by means of a quarterwavestub which has a moveable shorting barat the bottom end. The stub is used as a sub·stitute for the Q-section, since the reflectoremploys no feed line.The "Cubical·Quad" AntennaA smaller version of the Bi-Square antenna is the Cubi·cal-Quad antenna. Two halfwavesof wire are folded into a square that isone-quarter wavelength on a side, as shown infigure 20. The array radiates a horizontallypolarized signal. A reflector placed 0.15 wave·length behind the antenna provides an overallgain of some 6 db. A shorted stub with a paralleledtuning capacitor is used to resonate thereflector.The Cubical-Quad is fed with a 150-ohmline, and should employ some sort of antennatuner at the transmitter eQd of the line if a pinetworktype transmitter is used. There is asmall standing wave on the line, and an open
472 High Frequency Directive Antennas<strong>THE</strong> R AD I 0D2 021 lDIMENSIONS10M. l~t.A. 20M,GAIN APPROX. 7.5 DB52ll.COAXIAL LINEFigure 21<strong>THE</strong> "SIX-SHOOTER" BROADSIDE ARRAYwire line should be employed if the antenna isused with a high power transmitter.To tune the reflector, the back of the an·tenna is aimed at a nearby field-strength meterand the reflector stub capacitor is adjustedfor minimum received signal at the operatingfrequency.This antenna provides high gain for its smallsize, and is recommended for 28-Mc. work.The elements may be made of number 14 en·amel wire, and the array may be built on a lightbamboo or wood framework.The "Six-Shooter" As a good compromise be·Broadside Array tween gain, directivity,compactness, mechanicalsimplicity, ease of adjustment, and band widththe array of figure 21 is recommended for the10 to 30 Me. range when the additional arraywidth and greater directivity are not obtain·able. The free space directivity gain is ap·proximately 7.5 db over one element, and thepractical dx signal gain over one element atthe same average elevation is of about thesame magnitude when the array is sufficientlyelevated. To show up to best advantage thearray should be elevated sufficiently to putthe lower elements well in the clear, and pre·ferably at least 0.5 wavelength above ground.The "Bobtail"BidirectionalBroadside CurtainAnother application of ver·tical orientation of the ra·diating elements of an ar·ray in order to obtain low·angle radiation at the I ower end of the h·frange with low pole heights is illustrated infigure 22. When precut to the specified dimen·sions this single pattern array will performwell over the 7-Mc. amateur band or the 4-Mc.amateur phone band. For the 4·Mc. band therequired two poles need be only 70 feet high,and the array will provide a practical signalFigure 22"BOBTAIL" BIDIRECTIONAL BROAD·SIDE CURTAIN FOR <strong>THE</strong> 7-MC. OR <strong>THE</strong>4.0-MC. AMATEUR BANDSThis simple vertically .polarized array pro·vides low angle radiation and response withcomparatively low pole heights, and is veryeffective far dx work on the 7-Mc. band orthe 4.0-Mc. phone bond. Because of thephose relationships, radiation from the hori·zontol portion of the antenna is effectivelysuppressed. Very little current flows in theground lead to the coupling tonk; so an elob·orate ground system is not required, and thelength of the ground lead. is not critical solong as it uses heavy wife ancl is reason·ably short.gain averaging from 7 to 10 db over a horizon·tal half·wave dipole utilizing the same poleheight when the path I eng t h exceeds 2500miles.The horizontal directivity is only moderate,the beam width at the half power points beingslight! y greater than that obtained from threecophased vertical radiators fed with equal cur·rents. This is explained by the fact that thecurrent in each of the two outer radiators ofthis array carries only about half as much cur·rent as the center, driven element. While this"binomial" current distribution suppressesthe end·fire lobe that occurs when an odd num·her of parallel radiators with half-wave spac·ing are fed equal currents, the array still ex·hibits some high·angle radiation and responseoff the ends as a result of imperfect cancella·tion in the flat top portion. This is not suffi·cient to affect the power gain appreciably, butdoes degrade the discrimination somewhat.A moderate amount of sag can be toleratedat the center of the flat top, where it connectsto the driven vertical element. The poles andantenna tank should be so located with respectto each other that the driven vertical elementdrops approximately straight down from theflat top.
<strong>HANDBOOK</strong>Endfire Arrays 473Normally the antenna tank will be locatedin the same room as the transmitter, to facilitateadjustment when changing frequency. Inthis case it is recommended that the link COU"pled tank be located across the room from thetransmitter if much power is used, in order tominimize r·f feedback difficulties which mightoccur as a result of the asymmetrical high im·pedance feed. If tuning of the antenna tankfrom tbe transmitter position is desired, flexibleshafting can be run from the antenna tankcondenser to a control knob at the transmitter.The lower end of the driven element is quite"hot" if much power is used, and the lead-ininsulator should be chosen with this in mind.The ground connection need not have very lowresistance, as the current flowing in theground connection is comparatively small. Astake or pipe driven a few feet in the groundwill suffice. However, the ground lead shouldbe of heavy wire and preferably the lengthshould 'not exceed about 10 feet at 7 Me. orabout 20 feet at 4 Me. in order to minimizereactive effects due to its inductance. If it isimpossible to obtain this short a ground lead,a piece of screen or metal sheet about fourfeet square may be placed parallel to the earthin a convenient location and used as an arti·ficial ground. A fairly high C/L ratio ordinari·ly will be required in the antenna tank in orderto obtain adequate coupling and loading.23-7 End-Fire DirectivityBy spacing two half-wave dipoles, or colineararrays, at a distance of from 0.1 to 0.25wavelength and driving the two 180° out ofphase, directivity is obtained through the twowires at right angles to them. Hence, this typeof bidirectional array is called end fire. A betteridea of end-fire directivity can be obtainedby referring to figure 10.Remember that end· fire refers to the radiationwith respect to the two wires in the arrayrather than with respect to the array as awhole.The vertical directivity of an end-fire bi·directional array which is oriented horizontal·ly can be increased by placing a similar end·fire array a half wave below it, and excited inthe same phase. Such an array is a combina·tion broadside and end-fire affair.Kraus Flat· TopBeamA very effective bidirectionalend-fire array is the Kraus or8JK Flat-Top Beam. Essentially,this antenna consists of two close·spaced dipoles or colinear arrays. Because ofthe close spacing, it is possible to obtain theproper phase relationships in multi-sectionflat tops by crossing the wires at the voltageloops, rather than by resorting to phasingstubs. This greatly simplifies the array. (Seefigure 23.) Any number of sections may beused, though the one· and two-section arrange·ments are the most popular. Little extra gainis obtained by using more than four sections,and trouble from phase shift may appear.A center-fed single-section flat-top beamcut according to the table, can be used quitesuccessfully on its second harmonic, the pat·tern being similar except that it is a littlesharper. The single-section array can also beused on its fourth harmonic with some success,though there then will be four cloverleaf lobes,much the same as with a full-wave antenna.If a flat-top beam is to be used on more thanone band, tuned feeders are necessary.The radiation resistance of a flat-top beamis rather low, especially when only one sectionis used. This means that the voltage willbe high at the voltage loops. For this reason,especially good insulators should be used forbest results in wet weather.The exact lengths for the radiating elementsare not especially critical, because slight deviationsfrom the correct lengths can be compensatedin tbe stub or tuned feeders. Properstub adjustment is covered in Chapter Twentyfive.Suitable radiator lengths and approximatestub dimensions are given in the accompanyingdesign table.Figure 23 shows top views of eight typesof flat-top beam antennas. The dimensions forusing these antennas on different bands aregiven in the design table. The 7- and 28-Mc.bands are divided into two parts, but the dimensionsfor either the low- or high-frequencyends of these bands will be satisfactory foruse over the entire band.In any case, the antennas are tuned to thefrequency used, by adjusting the shorting wireon the stub, or tuning the feeders, if no stubis used. The data in the table may be extendedto other bands or frequencies by applying theproper factor. Thus, for 50 to 52 Me. operation,the values for 28 to 29 Me. are divided by 1.8.All of the antennas have a bidirectional horizontalpattern on their fundamental frequency.The maximum signal is broadside to the flattop. The single-section type has this patternon both its fundamental frequency and secondharmonic. The other types have four main lobesof radiation on the second and higher harmonics.The nominal gains of the different typesover a half-wave comparison antenna are asfollows: single-section, 4 db; two-section, 6db; four-section, 8 db.The maximum spacings given make thebeams less critical in their adjustments. Up
474 High Frequency Directive Antennas <strong>THE</strong> R AD I 0CENTER FEDTO CENTEROr I'I.AT TOP1-SECTION liNG !>.TuJ lT··=:dtJr,1-SECTION2-SECTIONSTU& OF FEEDERSCONNECT AT F F3-SECTION4-SECTIONFIGURE 23FLAT-TOP BEAM !BJK ARRAYJ DESIGN DATA.Spac-FREQUENCY~ __ S ___ L_, __ L_• ___ La ____ X.. ____ M__ DA ('hl A ('h) A ( 3 4) Xapprox. ~ approx. approx~7.0-7.2Mc. "A/8 17'4' ~~ 52'8' ~ 8'10" ~26'60'""96'_4_'_7.2~ "A/8 17'0' 33'6' ~ 51'8' 43'1' ~~26'59'94'_4_'_14.0-14.4 "A/8 ~~~ 26'4' ~--:rs-·-~13'~41i'-2-,-14.0-14.4 .15">. 10'5' ~ ~ 25'3' ~~y----rF--w-47'-2-,-14.0-14.4 .20"A 13'11'~~ 22'1o·----=r2'~10'27'45'_3_'_14.0-14.4 "A/4 17'4' ~3"ii" 20'8' --- 8'10" ~--8-'-25'43'--:v-28.0-29.0 .15">. 5¥~ ~ 12'7" ~ ~ 1'6' __ 7_'_----r5'24'--1,-28.0-29.0 "A/4 ~~~ 10'4' ---~ 1'6' __ 5_'_13'22'--2-,-29.0-30.0 .15">. 5'i)'"' ~ 14'6" 12'2' 9'8'- ----zr7' 1'6' __ 7_'_15' 23' --1-,-29.0-30.0 V4 ----sr4" ~ 14'6" li)i(jT ---~ 1'6' --5-, -13' """21' --2,-Dimension chart for flat-top beam antennas. The meanings of the symbols are as follows:L,, L, L, and L., the lengths o/ the sides of the flat-top sections as shown. L 1 is leniCthof the sides of single-section center-fed, L, single-section end-fed and 2-section center-fed, L, 4-sectioncenter-fed and end-sections of 4-section end-/ed,' and L, middle sections of 4-section end-fed.S, the spacing between the flat-top wires.M, the wire length from the outside to the center o/ each cross-o,er.D, the spacing lengthwise between sections.A (Y.), the approximate length for a quarter-wa"e stub.A ('/ 2 ), the approximate length /or a hal/-wa"e stub.A (%),the approximate length for a three-quarter wa"e stub.X, the approximate distance abo"e the shorting wire o/ the stub /or the connection of a 600-ohmline. This distance, as gi,en in the table, is approximately correct only /or 2-section flat-tops.For single-section types it will be smaller and /or J. and 4-section types it will be larger.The lengths gi,en /or a hal/-wa"e stub are applicable only to single-section center-fed flat-tops. Tobe certain o/ sufficient stub length, it is ad,isable to make the stub a fool or so longer than shown inthe table, especially with the end-fed types. The lengths, A, are measured from the point where thestub connects to the flat-top.Both the center and end-fed types may be used hori~ontally. Howe,er, where a "ertical antenna isdesired, the flat-tops can be turned on end. In this case, the end-fed types may be more con,enient,feeding from the lower end.
<strong>HANDBOOK</strong>Triplex Beam475Figure 24<strong>THE</strong> TRIPLEX FLAT-TOP BEAMANTENNA FOR 10, IS AND 20METERS,'/,lo-4...-''"---- sMAX. RADIATION-4.5 DBMAXIMUMRADIATION-4.& DB///IIIDIMENSIONSI OM. ISM. 20M MATERIALL ... ,~ 2.1'5" 32.'2." ~.n:~u.~ub 3·s 5'0" 7•&• II'D 1•a• 10'7• 14'4" 3000HM FUBBONto one-quarter wave spacing may be used onthe fundamental for the one-section types andalso the two-section center-fed, but it is notdesirable to use more than 0.15 wavelengthspacing for the other types.Although the center-fed type of flat-top gen·erally is to be preferred because of its sym·metry, the end-fed type often is convenient ordesirable. For example, when a flat-top beamis used vertically, feeding from the lower endis in most cases more convenient.If a multisection flat-top array is end-fedinstead of center-fed, and tuned feeders areused, stations off the ends of the array can beworked by tying the feeders together and work·ing the whole affair, feeders and all, as a long·wire harmonic antenna. A single-pole double·throw switch can be used for changing thefeeders and directivity.The TriplexBeamThe Triplex beam is a modifiedversion of the WBJK antennawhich uses f o 1 de d dipoles forthe half wave elements of the array. The useof folded dipoles results in higher radiationresistance of the array, and a high overall sys·tern performance. Three wire dipoles are usedfor the elements, and 300-ohm Twin-Lead isused for the two phasing sections. A recom·mended assembly for Triplex beams for 28 Me.,21 Me., and 14 Me. is shown in figure 24. Thegain of a Triplex beam is about 4. 5 db over adipole.23-8 Combination End-Fire andBroadside ArraysAny of the end-fire array s previous! y de·scribed may be stacked one above the other orplaced end to end (side bv side) to give greaterdirectivity gain while maintaining a bidirectionalcharacteristic. However, it must bekept in mind that to realize a worthwhile increasein directivity and gain while maintaininga bidirectional pattern the individual arraysmust be spaced sufficiendy to reduce themutual impedances to a negligible value.When two flat top beams, for instance, areplaced one above the other or end to end, acenter spacing on the order of one wavelengthis required in order to achieve a worthwhileincrease in gain, or approximately 3 db.
4 7 6 High Frequency Directive AntennasThus it is seen that, while maximum gainoccurs with two stacked dipoles at a spacingof about 0.7 wavelength and the space directivitygain is approximately 5 db over one elementunder these conditions; the case of twoflat top or parasitic arrays stacked one abovethe other is another story. Maximum gain willoccur at a greater spacing, and the gain overone array will not appreciably exceed 3 db.When two broadside curtains are placed oneahead of the other in end-fire relationship, theaggregate mutual impedance between the twocurt~ins ~s such that considerable spacing isrequued 1n order to realize a gain approaching3 db (the required spacing being a function ofthe size of the curtains). While it is true thata space directivity gain of approximately 4 dbcan be obtained by placing one, half-wave dipolean eighth wavelength ahead of anotherand feeding them 180 degrees out of phase, again of less than 1 db is obtained when thesame procedure is applied to two large broadsidecurtains. To obtain a gain of approximately3 db and retain a bidirectional pattern, aspacing of many wavelengths is required betweentwo large curtains placed one ahead ofthe other.A different situation exists, however, whenone driven curtain is placed ahead of an identicalone and the two are phased so as to givea unidirectional pattern. When a unidirectionalpattern is obtained, the gain over one curtainwill be approximately 3 db regardless of thespacing. For instance, two large curtainsplaced one a quarter wavelength ahead of theother may have a space directivity gain of only0.5 db over one curtain when the two are driven180 degrees out of phase to give a bidirectionalpattern (the type of pattern obtainedwith a single curtain). However, if they aredriven in phase quadrature (and with equal currents)the gain is approximately 3 db.The directivity gain of a composite arrayalso can be explained upon the basis of thedirectivity patterns of the component arraysalone, but it entails a rather complicated picture.It is sufficient for the purpose of thisdiscussion to generalize and simplify by sayingthat the greater the directivity of an endfirearray, the farther an identical array mustbe spaced from it in broadside relationship toobtain optimum performance; and the greaterthe directivity of a broadside array, the fartheran identical array must be spaced from it inend-fire relationship to obtain optimum performanceand retain the bidirectional characteristic.It is important to note that while a bidirectionalend-fire pattern is obtained with twodriven dipoles when spaced anything under ahalf wavelength, and while the proper phaserelationship is 180 degrees regardless of thespacing for all spacings not exceeding onehalf wavelength, the situation is different inthe case of two curtains placed in end-fire relationshipto give a bidirectional pattern. Formaximum gain at zero wave angle, the curtainsshould be spaced an odd multiple of one halfwavelength and driven so as to be 180 degreesout of phase, or spaced an even multiple ofone half wavelength and driven in the samephase. The optimum spacing and phase relationshipwill depend upon the directivity patternof the individual curtains used alone, andas previously noted the optimum spacing increaseswith the size and directivity of thecomponent arrays.A concrete example of a combination broadsideand end-fire array is two Lazy H arraysspaced along the direction of maximum radiationby a distance of four wavelengths and fedin phase. The space directivity gain of suchan arrangement is slightly less than 9 db. However,approximately the same gain can be obtainedby juxtaposing the two arrays side byside or one over the other in the same plane,so that the two combine to produce, in effect,one broadside curtain of twice the area. It isobvious that in most cases it will be more expedientto increase the area of a broadsidearray than to resort to a combination of endfireand broadside directivity. One exception,of course, is where two curtains are fed inphase quadrature to obtain a unidirectionalpattern and space directivity gain of approximately3 db with a spacing between curtainsas small as one quarter wavelength. Anotherexception is where very low angle radiation isdesired and the maximum pole height is strictlylimited. The two aforementioned Lazy Harrays when placed in end-fire relationshipwill have a considerably lower radiation anglethan when placed side by side if the array elevationis low, and therefore may under someconditions exhibit appreciably more practicalsignal gain.
CHAPTER TWENTY-FOURV-H-f and U-H-f AntennasThe very-high-frequency or v·h·f frequencyrange is defined as that range falling between30 and 300 Me. The ultra-high-frequency oru·h·f range is defined as falling between 300and 3000 Me. This chapter will be devoted tothe design and construction of antenna systemsfor operation on the amateur 50-Mc., 144-Mc., 235-Mc., and 420-Mc. bands. Although thebasic principles of antenna operation are thesame for all frequencies, the shorter physicallength of a wave in this frequency range andthe differing modes of signal propagation makeit possible and expedient to use antenna systemsdifferent in design from those used onthe range from 3 to 30 Me.24-1 Antenna RequirementsAny type of antenna system useable on thelower frequencies may be used in the v-h-f andu-h-f bands. In fact, simple non-directive halfwaveor quarter-wave vert i cal antennas arevery popular for general transmission and receptionfrom all directions, especially forshort-range work. But for serious v-h-f or u-h-fwork the use of some sort of directional antennaarray is a necessity. In the first place,when the transmitter power is concentrated intoa narrow beam the apparent transmitter powerat the receiving station is increased manytimes. A "billboard" array or a Sterba curtainhaving a gain of 16 db will make a 25-watttransmitter sound like a kilowatt at the other477station. Even a much simpler and smaller threeorfour-element parasitic array having a gainof 7 to 10 db will produce a marked improvementin the received signal at the other station.However, as all v-h·f and u-h-f workersknow, the most important contribution of ahigh-gain antenna array is in reception. If aremote station cannot be heard it obviously isimpossible to make contact. The limiting factorin v-h-f and u-h-f reception i s in almostevery case the noise generated within the receiveritself. Atmospheric noise is almost nonexistentand ignition interference can almostinvariably be reduced to a satisfactory levelthrough the use of an effective noise limiter.Even with a grounded-grid or neutralized triodefirst stage in the receiver the noise contributionof the first tuned circuit in the receiverwill be relatively large. Hence it is desirableto use an antenna system which will deliverthe greatest signal voltage to the first tunedcircuit for a given field strength at the receivinglocation.Since the field intensity being produced atthe receiving location by a remote transmittingstation may be assumed to be constant, the receivingantenna which intercepts the greatestamount of wave front, assuming that the polarizationand directivity of the receiving antennais proper, will be the antenna which gives thebest received signal-to-noise ratio. An antennawhich has two square wavelengths effectivearea will pick up twice as much signal poweras one which has one square wavelength area,assuming the same general type of antenna and
478 V-H-F and U-H-F Antennas<strong>THE</strong> R A 0 I 0that both are directed at the station being re·ceived. Many instances have been reportedwhere a frequency band sounded completelydead with a simple dipole receiving antennabut when the receiver was switched to a threeelementor larger array a considerable amountof activity from 80 to 160 miles distant washeard.Angle ofRadiationThe useful portion of the signalin the v-h-f and u-h-f range forshort or medium distance communicationis that which is radiated at a very lowangle with respect to the surface of the earth·essentially it is that signal which is radiatedparallel to the surface of the earth. A verticalantenna transmits a portion of its radiation ata very low angle and is effective for this reason;its radiation is not necessarily effectivesimply because it is vertically polarized. Asimple horizontal dipole radiates very littlelow-angle energy and hence is not a satisfactoryv-h-f or u-h-f radiator. Directive arrayswh1ch concentrate a major portion of the radiatedsignal at a low radiation angle will proveto be effective radiators whether their signalis horizontally or vertically polarized.In all cases, the radiating system for v-h-fand u-h-f work should be as high and in theclear as possible. Increasing the height of theantenna system will produce a very markedimprovement in the number and strength of thesignals heard, regardless of the actual typeof antenna used.TransmissionLinesTransmission lines to v-h-f andu-h-f antenna systems may beeither of the parallel-conductoror coaxial conductor type. Coaxial line is recommendedfor short runs and closely spacedopen-wire line for longer runs. Wave guidesmay be used under certain conditions for frequenciesgreater than perhaps 1500 Me. buttheir dimensions become excessively great forfrequencies much below this value. Non-resonanttransmission lines will be found to be considerablymore efficient on these frequenciesthan those of the resonant type. It is wise toto use the very minimum length of transmissionline possible since transmission line lossesat frequencies above about 100 Me. mount veryrapidly.Open lines s h o u I d preferably be spacedcloser than is common for longer wavelengths,as 6 inches is an appreciable fraction of awavelength at 2 meters. Radiation from theline will be greatly reduced if l-inch or I lizinchspacing is used, rather than the more common6-inch spacing.Ordinary TV-type 300-ohm ribbon may beused on the 2-meter band for feeder lengthsof about 50 feet or less. For longer runs, eitherthe u-h-f or v-h-f TV open-wire lines may beused with good overall efficiency. The v-h-fline is satisfactory for use on the amateur420-Mc. band.AntennaChangeoverIt is recommended that the sameantenna be used for transmittingand receiving in the v-h-f andu-h-f range. An ever-present problem in thisconnection, however, is the antenna changeoverrelay. Reflections at the antenna changeoverrelay become of increasing importanceas the frequency of transmission is increased.When coaxial cable is used as the antennatransmission line, satisfactory coaxial antennachangeover relays with low reflection canbe used. One type manufactured by AdvanceElectric & Relay Co., Los Angeles 26, Calif.,will give a satisfactorily low value of reflection.On the 235-Mc. and 420-Mc. amateur bands,the size of the antenna array becomes quitesmall, and it is practical to mount two identicalantennas side by side. One of these antennasis used for the transmitter, and theother antenna for the receiver. Separate transmissionlines are used, and the antenna relaymay be eliminated.Effect of Feed A vertical radiator forSystem on Radiation general coverage u-h-fAngle use s h o u I d be madeeither J4 or Yz wavelengthlong. Longer vertical antennas do not havetheir maximum radiation at right angles to theline of the radiator (unless co-phased), and,therefore, are not practicable for use wheregreatest possible radiation parallel to theearth is desired.Unfortunately, a feed system which is notperfectly balanced and does some radiating,not only robs the antenna itself of that muchpower, but distorts the radiation pattern of theantenna. As a result, the pattern of a verticalradiator may be so altered that the radiationis bent upwards slightly, and the amount ofpower leaving the antenna parallel to theearth is greatly reduced. A vertical half-waveradiator fed at the bottom by a quarter-wavestub is a good exam pIe of this; the slightradiation from the matching section decreasesthe power radiated parallel to the earth bynearly 10 db.The only cure is a feed system which doesnot disrurb the radiation pattern of the antennaitself. This means that if a 2-wire line is used,the current and voltages must be exactly thesame (though 180° out of phase) at any pointon the feed line. It means that if a concentricfeed line is used, there should be no currentflowing on the outside of the outer conductor.
<strong>HANDBOOK</strong>Antenna Polarization 479Radiator CrassSect ianThere is no point in usingcopper tubing for an antennaon the medium frequencies.The reason is that considerable tubing wouldbe required, and the cross section still wouldnot be a sufficiently large fraction of a wave·length to improve the antenna bandwidth char·acteristics. At very high and ultra high fre·quencies, however, the radiator length is soshort that the expense of large diameter con·ductor is relatively small, even though copperpipe of 1 inch cross section is used. With suchconductors, the antenna will tune much morebroadly, and often a broad resonance charac·teristic is desirable. This is particularly truewhen an antenna or array is to be used overan entire amateur band.It should be kept in m i n d that with suchlarge cross section radiators, the resonantlength of the radiator will be somewhat shorter,being only slightly greater than 0.90 of a halfwavelength for a dipole when heavy copperpipe is used above 100 Me.Insulation The matter of insulation is ofprime importance at very high fre·quencies. Many insulators that have very lowlosses as high as 30 Me. show up rather poor·ly at frequencies above 100 Me. Even the lowloss ceramics are none too good where the r·fvoltage is high. One of the best and most prac·tical insulators for use at this frequency ispolystyrene. It has one disadvantage, however,in that it is subject to fracture and to de forma·tion in the presence of heat.It is common practice to design v·h·f andu·h-f antenna systems so that the various rad·iators are supported only at points of relativelylow voltage; the best insulation, obviously, isair. The voltages on properly operated untunedfeed lines are not high, and the question ofinsulation is not quite so important, though in·sulation still should be of good grade.AntennaPolarizationCommercial broadcasting in theU.S.A. for both FM and tele·vision in the v·h·f range hasbeen standarized on horizontal polarization.One of the main reasons for this standardizationis the fact that ignition interference isreduced through the use of a horizontally po·larized receiving antenna. Amateur practice,however, is divided between horizontal andvertical polarization in the v·h·f and u·h·frange. Mobile stations are invariably verticalcallypolarized due to the physical limitationsimposed by the automobile antenna installa·tion. Most of the stations doing intermittentor occasional work on these frequencies use asimple ground-plane vertical antenna for bothtransmission and reception. However, thoseTABLE OF WAVELENGTHSfre- 1f4 Wave 1/4 Wavequency free An ..in Me. Space tenna50.0 59.1 55.550.5 58.5 55.051.0 57.9 54.451.5 57.4 53.952.0 56.8 53.452.5 56.3 52.853.054.055.754.752.4s 1.4144 20.5 19.2145 20.4 19.1146 20.2 18.9147 20.0 18.8148 19.9 18.6235 12.6 11.8236 12.5 11.8237 12.5 11.7238 12.4 11.7239 12.4 11.6240 12.3 11.6420 7.05 6.63425 6.95 6.55430 6.88 6.48lh WaveFreeSpace118.1116.9115.9114.7113.5112.5111.5109.541.040.840.440.039.925.225.125.024.924.824.614.113.913.81f2 WaveAn·tenna1.11.0109.9108.8107.8106.7105.7104.7102.838.538.338.037.637.223.623.523.523.423.323.213.2513.112.95All dimensions are in inches. Lengths have inmost cases been rounded off to three significantfigures. "VrWave Free-Space" column shownabove should be used with Lecher wires for frequencymeasurement.stations doing serious work and striving formaximum-range contacts on the 50-Mc. and144-Mc. bands almost invariably use horizon·tal polarization.Experience has shown that there is a greatattenuation in sign a 1 strength when usingcrossed polarization (transmitting antennawith one polarization and receiving antennawith the other) for all normal ground-wave con·tacts on these bands. When contacts are be·ing made through sporadic-£ reflection, however,the use of crossed polarization seems tomake no discernible d i f fer en c e in signalstrength. So the operator of a station doingv·h·f work (particularly on the 50-Mc. band)is faced with a problem: If contacts are to bemade with all stations doing work on the sameband, provision must be made for operation onboth horizontal and vertical polarization. Thisproblem has been solved in many cases throughthe construction of an antenna array that maybe revolved in the plane of polarization in ad·dition to being capable of .rotation in the azi·muth plane.An alternate solution to the problem whichinvolves less mechanical construction is sim·ply to install a good ground-plane vertical antennafor all vertically-polarized work, andthen to use a multi-element horizontally-polarizedarray for dx work.24-2 Simple HorizontallyPolarized AntennasAntenna systems which do not concentrate
480 V-H-F and U-H-F Antennas<strong>THE</strong> R AD I 0TO XMTRCOAXIAL LINETO TRANSMITTER®LOW ZTRANSMtSSJON LINE©Figure 1THREE NONDIRECTIONAL, HORIZONTALLY POLARIZED ANTENNASradiation at the very low elevation angles arenot recommended for v-h-f and u-h-f work. It isfor this reason that the horizontal dipole andhorizontally-disposed colinear arrays are generallyunsuitable for work on these frequencies.Arrays using broadside or end-fire elementsdo concentrate radiation at low elevationangles and are recommended for v-h-fwork. Arrays such as the lazy-H, Sterba curtain,flat-top beam, and arrays with parasiticallyexcited elements are recommended forthis work. Dimensions for the first three typesof arrays may be determined from the datagiven in the previous chapter, and referencemaybe made to the Table of Wavelengths givenin this chapter.Arrays using vertically-stacked horizontaldipoles, such as are used by commercial televisionand FM stations, are capable of givinghigh gain without a sharp horizontal radiationpattern. If sets of crossed dipoles, as shownin figure IA, are fed 90° out of phase the resultingsystem is called a turnstile antenna.The 90° phase difference between sets of dipolesmay be obtained by feeding one set ofdipoles with a feed line which is one-quarterwave longer than the feed line to the otherset of dipoles. The field strength broadside toone of the dipoles is equal to the field fromthat dipole alone. The field strength at a pointat any other angle is equal to the vector sumof the fields from the two dipoles at that angle.A nearly circular horizontal pattern isproduced by this antenna.A second antenna producing a uniform, horizontallypolarized pattern is shown in figurelB. This antenna employs three dipoles bentto form a circle. All dipoles are excited inphase, and are center fed. A bazooka is includedin the system to prevent unbalance inthe coaxial feed system.. A third nondirectional antenna is shown infigure IC. This simple antenna is made of twohalf-wave elements, of which the end quarterwavelengthof each ~s bent back 90 degrees.~he pattern from thts antenna is very muchhke that of the turnstile antenna. The fieldfrom the two quarter-wave sections that arebent back are additive because they are 180degrees out of phase and are a half wavelengthapart. The advantage of this antenna isthe simplicity of its feed system and construction.24-3 Simple Vertical-PolarizedAntennasFor general coverage with a single antennaa single vertical radiator is commonly em:ployed. A two-wire open transmission line isnot suitable for use with this type antennaand coaxial polyethylene feed line such a~RG-8/U is to be recommended. Three practicalmethods of feeding the radiator with concentricline, with a minimum of current inducedin the outside of the line, are shown in figure2. Antenna (A) is known as the sleeve antenna,the lower half of the radiator being a largepiece of pipe up through which the concentricfeed line is run. At (B) is shown the groundplanevertical, and at (C) a modification ofthis latter antenna.The radiation resistance of the groundplanevertical is approximately 30 ohms, whichis not a standard impedance for coaxial line.To obtain a good match, the first quarter wavelengthof feeder may be of 52 ohms surge impedance,and the remainder of the line of approximately75 ohms impedance. Thus, thefirst quarter-wave section of line is used as a
<strong>HANDBOOK</strong> Vertically Polarized Arrays 481® ©Figure 2THREE VERTICALLY-POLARIZEDLOW-ANGLE RADIATORSTShown at (A) is the" sleeve" or" hypodermic"type of radiator. At (B) is shown theground-plane vertical, and (C) shows a modi·fication of this antenna system which increasesthe feed-point impedance to a valuesuch that the system may be feel directlyfrom a coaxial line with no standing waveson the feed/ in e.matching transformer, and a good match isobtained.In actual practice the antenna would consistof a quarter-wave rod, mounted by meansof insulators atop a pole or pipe mast. Elaborateinsulation is not required, as the voltageat the lower end of the quarter-wave radiatoris very low. Self-supporting rods from 0.25 to0.28 wavelength would be extended out, as inthe illustration, and connected together. Asthe point of connection is effectively at groundpotential, no insulation is required; the horizontalrods may be bolted directly to the supportingpole or mast, even if of metal. The coaxialline should be of the low loss type especiallydesigned for v-h-f use. The outsideconnects to the junction of the radials, andthe inside to the bottom end of the verticalradiator. An antenna of this type is moderatelysimple to construct and will give a good accountof itself when fed at the lower end of theradiator directly by the 52-ohm RG-8/U coaxialcable. Theoretically the standing-waveratio will be approximately I. 5-to-1 but inpractice this moderate s-w-r produces nodeleterious effects, even on coaxial cable.The modification shown in figure 2C permitsmatching to a standard 50- or 70-ohm flexiblecoaxial cable without a linear transformer. Ifthe lower rods hug the lin.e and supporting mastrather closely, the feed-point impedance isabout 70 ohms. If they are bent out to form anangle of about 30° with the support pipe theimpedance is about 50 ohms.The number of radial legs used in a groundplaneantenna of either type has an importanteffect on the feed-point impedance and uponthe radiation characteristics of the antennasystem. Experiment has shown that three radialsis the minimum number that should beused, and that increasing the number of radialsabove six adds substantially nothing to theeffectiveness of the antenna and has no effecton the feed-point impedance. Experiment hasshown, however, that the radials should beslightly longer than one-quarter wave for bestre'Sults. A length of 0.28 wavelength has beenshown to be the optimum value. This meansthat the radials for a 50-Mc. ground-plane vertkalantenna should be 65" in length.Double SkeletonCone AntennaThe bandwidth of the amennaof figure 2C can be increasedconsiderably by substitutingseveral space-tapered rods for thesingle radiating element, so that the "radiator"and skirt are similar. If a sufficient numberof rods are used in the skeleton cones andthe angle of revolution is optimized for theparticular type of feed line used, this antennaexhibits a very low SWR over a 2 to I frequencyrange. Such an arrangement is illustratedschematically in figure 3.A NondirectionalVertical ArrayHalf-wave elements may bestacked in the vertical planeto provide a non-directionalpattern with good horizontal gain. An arraymade up of four half-wave vertical elementsis shown in figure 4A. This antenna providesa circular pattern with a gain of about 4.5 dbover a vertical dipole. It may be fed with300-ohm TV-type line. The feedline should beconducted in such a way that the vertical portionof the line is at least one-half wavelengthaway from the vertical antenna elements. Asuitable mechanical assembly is shown in figure4B for the 144-Mc. and 235-Mc. amateurbands.24-4 The Discone AntennaThe Discone antenna is a vertically polarizedomnidirectional radiator which has verybroad band characteristics and permits a simple,rugged structure. This antenna presents asubstantially uniform feed-point impedance,suitable for direct connection of a coaxialline, over a range of several octaves. Alsq,the vertical pattern is suitable for ground-wave
4 82 V-H-F and U-H-F AntennasTOP APEX CONNECTS TOINNER CONNECTORLOWER APEX CONNECTSTO OUTER CONDUCTORT -fALUMINUMTUBINIO38"TYP.1TS=••T~r.~2"X2"X18'SEMI-LOOSE FIT,~ALUMINUM CROSS-' BAR TIGHTENS IT UP.<strong>THE</strong>R AD I 010'··:j!20'Figure 3<strong>THE</strong> DOUBLE SKELETON CONEANTENNAA skeleton cone has been substituted for thesinqle element radiator of figure 2C. Thisgreatly Increases the bandwidth. If at leastJO elements are used for each skeleton coneand the angIe of revolution and elementlength are optimized, a low SWR can be obtainedover a frequency range of at least twooctaves. To obtain this order of bandwidth,the element length L should be approximately0.2 wavelength at the lower frequency endof the band, and the angle of revolution opti·mized for the lowest maximum VSWR withinthe frequency range to be covered. A greaterimprovement in the impeclonce-lrequencycharacteristic can be achieved by addingelements than by increasing the diameter ofthe element:s. With only 3 elements per"cone" one/ a much smaHer angle ol revo•I uti on a low SWR con be obtai nee/ over a fre•quency range of approximately 1.3 to 1.0when the element lengths are optimized.work over several octaves, the gain varyingonly slighdy over a very wide frequency range.Commercial versions of the Discone anten·na for various applications are manufacturedby the Federal Telephone and Radio Corporation.A Discone type antenna for amateur workcan be fabricated from inexpensive materialswith ordinary hand tools.A Discone antenna suitable for multi-bandamateur work in the v-h/u·h·f range is shownschematically in figure SA. The distance Dshould be made approximately equal to a freespacequarter wavelength at the lowest oper-GUYSFigure 4NONDIRECTIONAL ARRAYS FOR 144 MC.AND 235 MC.On right is shown two band installation. Thewhole system may easily be dissembled andcarried on a ski-rack atop a car for porta&/ euse.ating frequency. The antenna then will performwell over a frequency range of at least8 to 1. At certain frequencies within thisrange the vertical pattern will tend to "lift"slightly, causing a slight reduction in gain atzero angular elevation, but the reduction isvery slight.Below the frequency at w hi c h the slantheight of the conical skirt is equal to a freespacequarter wavelength the standing-waveratio starts to climb, and below a frequencyapproximately 20 per cent lower than this thestanding-wave ratio climbs very rapidly. Thisis termed the cut off frequency of the antenna.By making the slant height approximately equalto a free-space quarter wavelength at the lowestfrequency employed (refer to chart), a
<strong>HANDBOOK</strong>jQj0.7 DllIIIIII::DIIIIJlIII!Figure SA<strong>THE</strong> "DISCONE" BROAD-BANDRADIATORThis antenna system radiates o verticallypolarized w~e. ov,~r a very wide frequencyrange. The cltsc may be mode of solidmet a I s h ~,e t, a,group of radials, or wirescreen; the cone may best be constructedby forming o sheet of thin aluminum. A sin•gle antenna may be usee/ for operation on the50, 144, one/ 220 Me. amateur bonds. Thedimension D is determined by the lowest fre·quency to be employee/, one/ is given in thechart of figure 58.VSWR of less than 1. 5 will be obtained throughoutthe operating range of the antenna.The Discone antenna may be consideredas a cross between an electromagnetic hornand an inverted ground plane unipole antenna.It looks to the feed line like a properly terminatedhigh-pass filter.Construction Details The top disk and theconical skirt may befabricated either from sheet metal, screen (suchas "hardware cloth"), or 12 or more "spine"radials. If screen is used a supporting frameworkof rod or tubing will be necessary formechanical strength except at the higher fre·quencies .• If spines are used, they should beterminated on a s t i f f ring for mechanicalstrength except at the higher frequencies.The top disk is supported by means of threeinsulating pillars fastened to the skirt. Eitherpolystyrene or low-loss ceramic is suitable forthe purpose. The apex of the conical skirt isgrounded to the supporting mast and to theouter conductor of the coaxial line. The lineis run down through the supporting mast. Analternative arrangement, one suitable for certainmobile applications, is to fasten the base00.5 1.0 1.5 2 2.5 3 4Discone Antenna 483500400 ~. 300 ~ 160"' ~0w 140' ~a:"- 120tn 110-"" ~w 100-":;.: 9 0'3 80-"'0~0l'\,II'5DIN FEETDESIGNFigure SBCHART FOR <strong>THE</strong> "DISCONE"ANTENNAof the skirt direct! y to an effective groundplane such as the top of an automobile.24-5 Helical BeamAntennasMost v-h-f and u-h-f antennas are either verticallypolarized or horizontally polarized(plane polarization). However, circularly po·larized antennas have interesting characteristicswhich may be useful for certain applica·tions. The installation of such an antenna caneffectively solve the problem of horizontal vs.vertical polarization.A circularly polarized wave has its energydivided equally between a vertically polarizedcomponent and a horizontally polarized com•ponent, the two being 90 degrees out of phase.The circularly polarized wave may be either"left handed" or "right handed," dependingupon whether the vertically polarized componentleads or lags the horizontal component.A circularly polarized antenna will respondto any plane polarized wave whether horizontallypolarized, vertically polarized, or diagonallypolarized. Also, a circular polarizedwave can be received on a plane polarized antenna,regardless of the polarization of thelatter. When using circular! y polarized antennasat both ends of the circuit, however, bothmust be left handed or both must be righthanded. This offers some interesting possibilitieswith regard to reduction of QRM. At
484 V-H-F and U-H-F Antennas<strong>THE</strong> R AD I 0I..--ROUND OR SQUAREGROUND SCREENG1\COAX FEED POINT (RG-03/U)AT CENTER OFGROUND SCREEND=t s=-t- G=o.o>.CONDUCTOR OIA.-::: APPROX. 0.17 A)... =WAVELENGTH IN FREE SPACETRANSMITRECEIVEL-----------tL= 1.44 xFigure 6<strong>THE</strong> "HELICAL BEAM" ANTENNAThis type of clirectional antenna systemgives excellent perlormance over a frequencyrange of 1. 7 to 1.8 to 1. Its climensions aresuch that it orclinarily is not practicable,however, for use as a rotatable array on fre•quencies below about 100 Me. The centerconcluctor of the fee cl I in e shoulcl passthrough the grouncl screen for connection tothe feecl point. The outer concluctor of thecoaxial line shoulcl be grounclecl to theground screen.TD1the time of writing, there has been no standardizationof the "twist" for general amateurwork.Perhaps the simplest antenna configurationfor a directional beam antenna having circularpolarization is the helical beam popularizedby Dr. John Kraus, WBJK. The antenna consistssimply of a helix working against aground plane and fed with coaxial line. In theu-h-f and the upper v-h-f range the physicaldimensions are sufficiently small to permitconstruction of a rotatable structure withoutmuch difficulty.When the dimensions are optimized, thecharacteristics of the helical beam antennaare such as to qualify it as a broad band antenna.An optimized helical beam shows littlevariation in the pattern of the main lobe anda fairly uniform feed point impedance averagingapproximately 125 ohms over a frequencyrange of as much as 1. 7 to 1. The direction of"electrical twist" (right or left handed) de·pends upon the direction in which the helix iswound.A six-turn helical beam is shown schemati·cally in figure 6. The dimensions shown willgive good performance over a frequency rangeof plus or minus 20 per cent of the design frequency.This means that the dimensions arenot especially critical when the array is to beused at a single frequency or over a narrowband of frequencies, such as an amateur band.At the design frequency the beam width isabout 50 degrees and the power gain about 12db, referred to a non-directional circular! y polarizedantenna.The Ground Screen For the frequency range100 to 500 Me. a suitableground screen can be made from "chickenwire" poultry netting of l-inch mesh, fastenedto a round or square frame of either metal orwood. The netting should be of the type thatis galvanized after weaving. A small, sheetmetal ground plate of diameter equal to approximatelyD/2 should be centered on thescreen and soldered to it. Tin, galvanizediron, or sheet copper. is suitable. The outerconductor of the R!J-63/U (125 ohm) coax isconnected to this plate, and the inner conductorcontacts the helix through a hoi e in thecenter of the plate. The end of the coax shouldbe taped with Scotch electrical tape to keepwater out.The Helix It should be noted that the beamproper consists of six full turns.The start of the helix is spaced a distance ofS/2 from the ground screen, and the conductorgoes directly from the center of the groundscreen to the start of the helix.Aluminum tubing in the "SO" (soft) gradeis suitable for the helix. Alternatively, lengthsof the relatively soft aluminum electrical conduitmay be used. In the v-h-f range it will benecessary to support the helix on either twoor four wooden longerons in order to achievesufficient strength. The longerons should beof as small cross section as will provide suf·ficient rigidity, and should be given severalcoats of varnish. The ground plane buttsagainst the longerons and the whole assemblyis supported from the balance point if it is tobe rotated.Aluminum tubing in the larger diameters ordinarilyis not readily available in lengths greaterthan 12 feet. In this case several lengthscan be spliced by means of short telescopingsections and sheet metal screws.The tubing is close wound on a drum andthen spaced to give the specified pitch. Notethat the length of one com p 1 e t e turn whenspaced is somewhat greater than the circumferenceof a circle having the diameter D.Brood-Bond144 to 225 Me.Helical BeamA high! y useful v-h-f helicalbeam which will receive signalswith good gain over thecomplete frequency range from144 through 225 Me. may be constructed byusing the following dimensions (180 Me. designcenter):
<strong>HANDBOOK</strong>Helical BeamAntenna485D ............................ 22 in.S ......................... 16~ in.G ................•..........• 53 in.Tubing o.d ................. 1 in.The D and S dimensions are to the center ofthe rubing. These dimensions must be heldrather closely, since the range from 144 through225 Me. represents just about the practicallimit of coverage of this type of antenna sys·tem.High-BandTV CoverageNote that an array constructedwith the above dimensions willgive unusually good high-bandTV reception in addition to covering the 144-Mc. and 220-Mc. amateur bands and the taxiand police services.On the 144-Mc. band the beam width is ap·proximately 60 degrees to the half-powerpoints, while the power gain is approximately11 db over a non-directional circularly polarizedantenna. For high-band TV coverage thegain will be 12 to 14 db, with a beam widthof about 50 degrees, and on the 220-Mc. amateurband the beam width will be about 40 degreeswith a power gain of approximately 15 db.The antenna system will receive verticallypolarized or horizontally polarized signalswith equal gain over its entire frequency range.Conversely, it will transmit signals over thesame range, which then can be received withequal strength on either horizontally polarizedor vertically polarized receiving antennas.The standing-wave ratio will be very low overthe complete frequency range if RG-63/U coaxialfeed line is used.24-6 The Corner-Reflectorand Horn-Type AntennasThe corner-reflector antenna is a good directionalradiator for the v-h-f and u-h-f region.The antenna may be used with the radiatingelement vertical, in which case the directivityis in the horizontal or azimuth plane, or thesystem may be used with the driven elementDRIVEN DIPOLESUPPORT IN~MEMBERFigure 7CONSTRUCTION OF <strong>THE</strong> "CORNERREFLECTO~' ANTENNASuch an antenna is capable of giving highgain with a minimum of complexity in theradiating system. It may be usecl either withhorizontal or vertical polarization. Designclata for the antenna is given in the Corner-Reflector Design Table.horizontal in which case the radiation is horizontallypolarized and most of the directivityis in the vertical plane. With the antenna usedas a horizontally polarized radiating systemthe array is a very good low-angle beam arrayalthough the nose of the horizontal pattern isstill quite sharp. When the radiator is orientedvertically the corner reflector operates verysatisfactorily as a direction-finding antenna.Design data for the corner-reflector antennais given in figure 7 and in the chart CornerReflector Design Data. The planes which makeup the reflecting corner may be made of solidsheets of copper or aluminum for the u-h-fbands, although spaced wires with the endssoldered together at top and bottom may beused as the reflector on the lower frequencies.CORNER-REFLECTOR DESIGN DATACorner Freq. Feed Approx.Angle Band, Me. R H A G lmped. Gain, db90 so 110" 82" 140" 200" 230"' 18" 72 1060 so 110" 11S" 140" 230" 230" 18" 70 1260 144 38" 40" 48" 100" 100" S" 70 1260 220 24.5" 2S" 30" 72" 72" 3" 70 1260 420 13" 14" 18" 36" J6" screen 70 12NOTE: Refer to figure 7 for construction of corner-reflector antenna.
486 V-H-F and U-H-F Antennas<strong>THE</strong> R AD I 0@ UHF HORN ANTENNA~···"'" ··~-~D~ANGLE BETWEEN ~O'~, JSIDES OF HORN=&O"ljO450-QHM LINE0 ZA-A GAIN (DB)>.. 400 \A 420 TWO SIDES MADEOF WIRE MESH2X 390@ VHF HORIZONTALLY POLARIZED HORNFigure 8TWO TYPES OF HORN ANTENNASThe "two siclecJ horn" of Figure 88 may befeel by means of an open-wire transmissionline.Copper screen may also be used for the reflectingplanes.The values of spacing given in the cornerreflectorchart have been chosen such that thecenter impedance of the driven element wouldbe approximately 70 ohms. This means thatthe element may be fed directly with 70-ohmcoaxial line, or a quarter-wave matching transformersuch as a "Q" section may be used toprovide an impedance match between the center-impedanceof the element and a 460-ohmline constructed of no. 12 wire spaced 2 inches.In many v-h-f antenna systems, waveguidetransmission lines are terminated by pyramidalhorn antennas. These horn antennas (figureSA) will transmit and receive either horizontallyor vertically polarized waves. The use ofwaveguides at 144 Me. and 235 Me., however,is out of the question because of the relativelylarge dimensions needed for a waveguide operatingat these low frequencies.A modified type of horn antenna may still beused on these frequencies, since only one particularplane of polarization is of interest tothe amateur. In this case, the horn antennacan be simplified to two triangular sides ofthe pyramidal horn. When these two sides areinsulated from each other, direct excitation atthe apex of the born by a two-wire transmissionline is possible.In a normal pyramidal horn, all four triangularsides are covered with conducting material,but when horizontal polarization alone is ofinterest (as in amateur work) only the verticalareas of the horn need be used. If vertical polarizationis required, only the horizontal areasFigure 9<strong>THE</strong> 60° HORN ANTENNA FOR USE ONFREQUENCIES ABOVE 144 MC.of the horn are employed. In either case, thesystem is unidirectional, away from the apexof the horn. A typical born of this type is shownin figure 88. The two metallic sides of thehorn are insulated from each other, and thesides of the horn are made of small mesh"chicken wire" or copper window screening.A pyramidal horn is essentially a high-passdevice whose low frequency cut-off is reachedwhen a side of the horn is ~ wavelength. Itwill work up to infinitely high frequencies,the gain of the horn increasing by 6 db everytime the operating frequency is doubled. Thepower gain of such a horn compared to a ~wave dipole at frequencies higher than cutoffis:8.4 A 2Power gain (db) --- )1.2where A is the frontal area of the mouth of thehorn. For the 60 degree horn shown in figureBB the formula simplifies to:Power gain (db)= 8.4 D 2 , when Dis expressedin terms of wavelengthWhen D is equal to one wavelength, the powergain of the horn is approximately 9 db. Thegain and feed point impedance of the 60 degreehorn are shown in figure 9. A 450 ohmopen wire TV-type line may be used to feedthe horn.24-7 VHF HorizontalRhombic AntennaFor v-h-f transmission and reception in afixed direction, a horizontal rhombic permits
<strong>HANDBOOK</strong>VHF Rhombic 487w...J" z
488 V-H-F and U-H-F Antennas<strong>THE</strong>R AD I 0sistors in series. If 2-watt resistors are employed,this termination also is suitable fortransmitter outputs of 10 watts or less. Forhigher powers, however, resistors having greaterdissipation with negligible reactance in theupper v-h-f range are not readily available.For powers up to several hundred watts asuitable termination consists of a "lossy"line consisting of stainless steel wire (cottespondingto no. 24 or 26 B&S gauge) spaced 2inches, which in turn is terminated by two390-ohm 2-watt carbon resistors. The dissipativeline should be at least 6 wavelengthslong.24-8 Multi-Element V-H-FBeam AntennasFOR I:;)- METERS0•22"D-A=o•A-23f A-R-o•R= 25" ~~fts-t"FLATTEN ENOS OFTUBING AND DRILLFOR 6/32 SCREWSR- REFLECTOR40"L0NG75 HMTV LINEThe rotary multi-element beam is undoubtedlythe most popular type of v-h-f antenna inuse. In general, the design, assembly and tuningof these antennas follows a pattern similarto the larger types of rotary beam antennasused on the lower frequency amateur bands.The characteristics of these low frequencybeam antennas are discussed in the next chapterof this Handbook, and the information containedin that chapter applies in general to thev-h-f beam antennas discussed herewith.A Simple ThreeElement BeamAntennaThe simplest v-h-f beam forthe beginner is the three-elementY agi array illustrated infigure 12. Dimensions aregiven for Yagis cur for the 2-meter and 1 ~meter bands. The supporting boom for the Yagimay be made from a smoothed piece of 1" x 2 •wood. The wood should be reasonably dry andshould be painted to prevent warpage from exposureto sun and rain. The director and reflectorare cut from lengths of ~ 11 copper tubing,obtainable from any appliance store thatdoes service work on refrigerators. They shouldbe cut to length as noted in figure 12. The elementsshould then be given a coat of aluminumpaint. Two small holes are drilled at the centerof the reflector and director and these elementsare bolted to the wood boom by means of two1 11 wood screws. These screws should be ofthe plated, or rust-proof variety.The driven element is made of a 78" lengthof ~ 11 copper tubing, the ends bent back uponeach other to form a folded dipole. If the tubingis packed with fine sand and the bendingpoints heated over a torch, no trouble will behad in the bending process. If the tubing doescollapse when it is bent, the break may be repairedwith a heavy-duty soldering iron. TheFigure 12SIMPLE 3-ELEMENT BEAM FOR 2 AND1\4 METERSdriven element is next attached to the centerof the wood boom, mounted atop a small insulatingplate made of bakelite, micarta orsome other non-conducting material. It is heldin place in the same manner as the parasiticelements. The two free ends of the folded dipoleare hammered flat and drilled for a 6-32bolt. These bolts pass through both the insulatingblock and the boom, and hold the freetips of the element in place.A length of 75-ohm Twin-Lead TV-type lineshould be used with this beam antenna. It isconnected to each of the free ends of the foldeddipole. If the .antenna is mounted in the verticalplane, th~ 75-ohm line should be broughtaway from the antenna for a distance of fourto six feet before it drops down the tower tolessen interaction between the antenna elementsand the feed line. The complete antennais light enough to be turned by a TV rotator.A simple Yagi antenna of this type will providea gain of 7 db over the entire 2-meter or1 ~-meter band, and is highly recommended asan "easy-to-build" beam for the novice orbeginner.An 8-Eiement"Tippable" Arrayfar 144 Me.Figures 13 and 14 illustratean 8-element rotaryarray for use on the 144-Mc. amateur band. Thisarray is "tippable" to obtain either horizontalor vertical polarization. It is necessary thatthe transmitting and receiving station use thesame polarization for the ground-wave signalpropagation which is characteristic of this fre-
<strong>HANDBOOK</strong>VHFParasitic Arrays489r'""IRING BOLT!46" BOOM~~~: Fl LE END TO FITMAIN BOOMS- f' APPROX. 0.0.ELEMENTS- t"eon BOOM60"AS SHOWN, ANTENNA ISHORIZONTALLY POLARIZED ......___PULL TO SWJ NG MAIN BOOM 90°FOR VERTICAL POLARITY.CONTROL CORDSRG-8/U CABLETO RIG.Figure 13CONSTRUCTIONAL DRAWING OF AN EIGHT-ELEMENT TIPPABLE 144-MC. ARRAYquency range. Although polarization has beenloosely standardized in various areas of thecountry, exceptions are frequent enough sothat it is desirable that the polarization of an·tenna radiation be easily changeable from hori·zontal to vertical.The antenna illustrated has shown a signalgain of about 11 db, representing a power gainof about 13. Although the signal gain of theantenna is the same whether it is oriented forvertical or horizontal polarization, the hori·zontal beam width is smaller when the antennais oriented for vertical polarization. Conver·sely, the vertical pattern is the sharper whenthe antenna system is oriented for horizontalpolarization.The changeover from one polarization to theother is accomplished simply by pulling on the
490 V-H-F and U-H-F Antennas <strong>THE</strong> R AD I 0Figure 14<strong>THE</strong> EIGHT-ELEMENT 144-MC. ARRAY IN A HORIZONTAL POSITIONappropriate cord. Hence, the operation is basedon the offset head sketched in figure 13. AI·though a wood mast has been used, the samesystem may be used with a pipe mast.The 40-inch lengths of RG-59/U cable (elec·trically % wavelength) running from the centerof each folded dipole driven element to thecoaxial T·junction allow enough slack to per·mit free movement of the main boom whenchanging polarity. Type RG-8/U cable is runfrom the T·junction to the operating position.Measured standing-wave ratio was less than2:1 over the 144 to 148 Me. band, with thelengths and spacings given in figure 13.Construction ofthe ArrayMost of the constructionalaspects of the antenna arrayare self-evident from figure13. However, the pointers given in the follow·ing paragraphs will be of assistance to thosewishing to reproduce the array.The drilling of holes for the small elementsshould be done carefully on accurately markedcenters. A small angular error in the drillingof these holes will result in a considerablemisalignment of the elements after the array isassembled. The same consideration is true ofthe filing out of the rounded notches in theends of the main boom for the fitting of thetwo antenna booms.Short lengths of wood dowel are used freelyin the construction of the array. The ends ofthe small elements are plugged with an inchor so of dowel, and the ends of the antennabooms are similarly treated with larger discspressed into place.
<strong>HANDBOOK</strong>VHF Parasitic Arrays 4 91The ends of the folded dipoles are made inthe following manner: Drive a length of dowelinto the short connecting lengths of aluminumtubing. Then drill down the center of the dowelwith a clearance hole for the connecting screw.Then shape the ends of the connecting piecesto fit the sides of the element ends. After assemblythe junctions may be dressed with afile and sandpaper until a smooth fit is obtained.The mas~ used for supporting the array is a30-foot spliced 2 by 2. A large discarded ballbearing is used as the radial load bearing andguy-wire termination. Enough of the upper-mastcorners were removed with a draw-knife to permitsliding the ball bearing down about 9 feetfrom the top of the mast. The bearing then wasencircled by an assembly of three pieces ofdural ribbon to form a clamp, with ears fortightening screws and attachment of the guywires. The bearing then was greased and coveredwith a piece of auto inner tube to serveas protection from the weather. Another junkboxbearing was used at the bottom of the mastas a thrust bearing.The main boom s were made from %-inchaluminum electrical conduit. Any size of smalltubing will serve for making the elements.Note that the main boom is mounted at the balancecenter and not necessarily at the physicalcenter. The pivot bolt in the offset headshould be tightened sufficiently that there willbe adequate friction to hold the array in position.Then an additional nut should be placedon the pivot bolt as a lock.In connecting the phasing sections betweenthe T-junction and the centers of the foldeddipoles, it is important that the center conductorsof the phasing sections be connectedto the same side of the driven elements of theantennas. In other words, when the antenna isoriented for horizontal polarization and thecenter of the coaxial phasing section goes tothe left side of the top antenna, the centerconductor of the other coaxial phasing sectionshould go to the left side of the bottom antenna.The "Screen Beam" This highly effective rofor2 Meterstary array for the 144 Me.amateur band was designedby the staff of the Experimental PhysicsLaboratory, The Hague, Netherlands foruse at the 2 meter experimental station PE1PL.The array consists of 10 half wave radiatorsfed in phase, and arranged in two stacked rowsof five radiators. 0.2 wavelength behind thisplane of radiators is a reflector screen, measuringapproximately 15' x 9' in size. The antennaprovides a power gain of 15 db, and afront to back ratio of approximately 28 db.1 9"r3·r~" i6 POLY<strong>THE</strong>NEI t'3fj ~:•t"--j- lJ .--
492 V-H-F and U-H-F Antennas<strong>THE</strong> R AD I 0WOOD BLOCK~RADIATOR ==Figure 16<strong>THE</strong> MOUNTING BLOCK FOR EACH SETOF ELEMENTSThese tubes are welded onto the center tubeof each group of three horizontal bracing tubes,and are so located to support the horizontal dipoleat its exact center. The dipoles are attachedto the supporting rods by means ofsmall phenolic insulating blocks, as shown infigure 16. The radiators are therefore insulatedfrom the screen reflector. The inner tips ofthe radiators are held by small polystyreneblocks for rigidity, and are cross connected toeach other by a transposed length of TV-type400 ohm open wire line. The entire array isfed at the point A·A, illustrated in figure 15.The matching system for the beam is mountedbehind the reflector screen, and is shown infigure 17. A quarter-wave transformer (B) dropsthe relatively high impe.flance of the antennaarray to a suitable value for the low impedancebalun (D). An adjustable matching stub(C) and two variable capacitors (C, and C 2 )are employed for impedance matching. Thetwo varia b 1 e capacitors are mounted in aAPPROX,4 "Figure 17C1 &. C2=so.u1JF<strong>THE</strong> MATCHING UNIT IN DETAIL FOR<strong>THE</strong> PE1PL BEAM DESIGN, WHICH AL·LOWS <strong>THE</strong> USE OF 72-0HM COAXFigure 18HORIZONTAL RADIATION PATTERN OF<strong>THE</strong> PE1PL ARRAY. <strong>THE</strong> FRONT-TO·BACK RATIO IS ABOUT 28 db IN AMPLITUDE, AND <strong>THE</strong> FORWARD GAIN AP·PROXIMATELY 15 db.watertight box, with the balun and matchingstubs entering the bottom and top of the box,respectively.The matching procedure is carried out bythe use of a standing wave meter (SWR bridge).A few watts of power are f e d to the arraythrough the SWR meter, and the setting of theshorting stub on C and the setting of the twovariable capacitors are adjusted for lowestSWR at the chosen operating frequency. Thecapacity settings of the two variable capacitorsshould be equal. The final adjustment isto set the shorting stub of the balun (D) to removeany residual reactance that might appearon the transp!ission line. With proper adjustment,the VSWR of the array may be held toless than 1.5 to 1 over a 2 megacycle rangeof the 2-meter band.The horizontal radiation pattern of this arrayis shown in figure 18.Long Yogi For a given power gain, theAntennas Y agi antenna can be builtlighter, more compact, andwith less wind resistance than any other type.
<strong>HANDBOOK</strong>VHFParasitic Arrays493D11DRILL HOLES THROUGH BOOM ANDPASS ELEMENTS THROUGH HOLES.BOOM LEN(;TH:. 24 ', DIAM. t!-"'DRIVEN ELEMENTGAIN= 16.1 DBELEMENT DIMENSIONS2 METER BANDLENGTHSPACINGELEMENTFROM(OIAM, 1/8 ... ) 144MC. 145MC- 146MC. 147 MC, 01 POLEREFLECTOR 41"3"4047"401&3"4016 19"Of RECTORS 36 :t'' 36-t" 3"36. 36~· 01 = 7"I0.2= 14.5•DRIVEN ELEMENT 03= 22"04=- 38"36.5".A~CLEARANCE HOLE-I Ds= 70"BOOM!="OR SOL TDe= 102 ...··1t :uA /I07= 134•Ls WIRE FoR 3oo n. ~NSULAT!NG -LArreN De= tee,..MATCH. PLATE TUBING 09=t96".t:10WIRE FOR 450.0..AT ENOS.MATCH 010=230"011=242"Figure 19DESIGN DIMENSIONS FOR A 2-METER "LONG Y AGI" ANTENNAOn the other hand, if a Yagi array of the sameapproximate size and weight as another antennatype is built, it will provide a higherorder of power gain and directivity than thatof the other antenna.The power gain of a Yagi antenna increasesdirectly with the physical length of the array.The maximum practical length is entirely amechanical problem of physically supportingthe long series of director elements, althoughwhen the array exceeds a few wavelengths inlength the element lengths, spacings, andQ's become more and more critical. The effectivenessof the array depends upon a propercombination of the mutual coupling loopsbetween adjacent directors and between thefirst director and the driven element.Practically all work on Yagi antennas withmore than three or four elements has been onan experimental, cut-and-try basis. Figure 19provides dimensions for a typical Long Yagiantenna for the 2-meter VHF band. Note thatall directors have the same physical length.If the long Yagi is designed so that the directorsgradually decrease in length as theyprogress from the dipole bandwidth will beincreased, and both side lobes and forwardgain will be reduced. One advantage gainedfrom staggered director length is that thearray can be shortened and lengthened byadding or taking away directors without theneed for retuning the remaining group of parasiticelements. When all directors are thesame length, they must be all shortened enmasse as the array is lengthened, and viceversawhen the array is shortened.A full discussion of Long Yagi antennas,.including complete design and constructioninformation may be had in the VHF Handbook,available through Radio Publications, Inc.,Wilton., Conn.
CHAPTER TWENTY-FIVERotary BeamsThe rotatable antenna array has become almoststandard equipment for operation on the28-Mc. and 50-Mc. bands and is commonly usedon the 14-Mc. and 21-Mc. bands and on thosefrequencies above 144 Me. The rotatable arrayoffers many advantages for both military andamateur use. The directivity of the antennatypes commonly employed, particularly theunidirectional arrays, offers a worthwhile reductionin interference from undesired directions.Also, the increase in the ratio of lowangleradiation plus the theoretical gain ofsuch arrays results in a relatively large increasein both the transmitted signal and thesignal intensity from a station being received.A significant advantage of a rotatable antennaarray in the case of the normal station isthat a relatively small amount of space is requiredfor erection of the antenna system. Infact, one of the best types of installation usesa single telephone pole with the rotating structureholding the antenna mounted atop the pole.To obtain results in all azimuth directionsfrom fixed arrays comparable to the gain anddirectivity of a single rotatable three-elementparasitic beam would require several acres ofsurface.There are two normal configurations of radiatingelements which, when ~o~izontally polarizedwill contribute to obtammg a low angleof r~diation. These configurations are the endfirearray and the broadside array. The con-4 94ventional three- or four-element rotary beammay properly be called a unidirectional parasiticend-fire array, and is actually a type ofyagi array. The flat-top beam is a type of bidirectionalend-fire array. The broadside typeof array is also quite effective in obtaininglow-angle radiation, and although widely usedin FM and TV broadcasting has seen little useby amateur stations in rotatable arrays.25-1 UnidirectionalParasitic End-Fire Arrays(Yogi Type)If a single parasitic element is placed onone side of a driven dipole at a distance offrom 0.1 to 0. 25 wavelength the parasitic elementcan be tuned to make the array substantial!y unidirectional.This simple array is termed a two elementparasitic beam.25-2 The Two Element BeamThe two element parasitic beam providesthe greatest amount of gain per unit size ofany array commonly used by radio amateurs.
Parasitic Arrays ~954Cii 3e.z.)o.z•Figure 1GAIN VS ELEMENT SPACING FOR A TWO·ELEMENT CLOSE-SPACED PARASITICBEAM ANTENNA WITH PARASITIC ELE·MENT OPERATING AS A DIRECTOR ORREFLECTORFigure 2RADIATION RESISTANCE AS A FUNCTIONOF ELEMENT SPACING FOR A TWO-ELE·MENT PARASITIC ARRAYSuch an antenna is capable of a signal gainof 5 db over a dipole, with a front·to·back ratioof 7 db to 15 db, depending upon the adjust·ment of the parasitic element. The parasiticelement may be used either as a director oras a reflector.The optimum spacing for a reflector in atwo·element array is approximately 0.13 wave·length and with optimum adjustment of thelength of the reflector a gain of approximately5 db will be obtained, with a feed·point resist·ance of about 25 ohms.If the parasitic element is to be used as adirector the optimum spacing between it andthe driven element is 0.11 wavelength. Thegain will theoretically be slightly greater thanwith the optimum adjustment for a reflector(about 5. 5 db) and the radiation resistancewill be in the vicinity of 17 ohms.The general characteristics of a two-elementparasitic array may be seen in figures 1, 2 and3. The gain characteristics of a two-elementarray when the parasitic element is used as adirector or as a reflector are shown. It can beseen that the director provides a maximum of5.3 db gain at a spacing of slightly greaterthan 0.1 wavelength from the antenna. In theinterests of greatest power gain and size con·servation, therefore, the choice of a parasiticdirector would be wiser than the choice of aparasitic reflector, although the gain differ·ence between the two is small.Figure 2 shows the relationship betweenthe element spacing and the radiation resist·ance for the two element paras1t1c array forboth the reflector and the director case. Sincethe optimum antenna-director spacing for maxi·mum gain results in an antenna radiation re·sistance of about 17 ohms, and the optimumantenna-reflector spacing for maximum gainresults in an antenna radiation resistance ofabout 25 ohms, it may be of advantage in someinstances to choose the antenna with the high·er radiation resistance, assuming other fac·tors to be equal.Figure 3 shows the front·to·back ratio forthe two element parasitic array for both thereflector and director cases. To produce thesecurves, the elements were tuned for maximumgain of the array. Better front· to· back ratiosmay be obtained at the expense of array gain,if desired, but the general shape of the curvesremains the same. It can be readily observedthat operation of the parasitic element as areflector produces relatively poor front·to·back ratios except when the element spacingis greater than 0.15 wavelength. However, atthis element spacing, the gain of the array be·gins to suffer.Since a radiation resistance of 17 ohms isnot unduly hard to match, it can be argued thatthe best all-around performance may be ob·rained from a two element parasitic beam em·ploying 0.11 element spacing, with the para·sitic element tuned to operate as a director.This antenna will provide a forward gain of5.3 db, with a front·to·back ratio of 10 db, orslightly greater. Closer spacing than 0.11
4 96 Rotary Beams<strong>THE</strong> R AD I 0wavelength may be employed for greater front·to·back ratios, but the radiation resistance ofthe array becomes quite low, the bandwidthof the array becomes very narrow, and the tuningbecomes quite critical. Thus the Q of theantenna system will be increased as the spacingbetween the elements is decreased, andsmaller optimum f r e que n c y coverage willresult.40\\ .0.1'--. 'P........ "~~ ~-"_j),.c?~RElement Lengths When the parasitiC elementof a two-element array isused as a director, the following fonnulas maybe used to determine the lengths of the drivenelement and the parasitic director, assumingan element diameter-to-length ratio of 200 to400:476Driven element length (feet) = -FMc.450Director length (feet) = -*FMc.Element spacing (feet) =Me.Figure 4FIVE ELEMENT 28 MC BEAMANTENNA AT W6SAIAntenna boom is made of twenty footlength of Sears, Roebuck Co. threeinchaluminum irrigation pipe. Spacingbetween elements is five feet. Elementsare made of twelve foot lengthsof 7 /8-inch aluminum tubing, with extensiontips made of 3/4-inch tubing.Gamma matching device, elementclamps, and "Oxen Yoke" element-to•boom clamps are made by ContinentalElectronics & Sound Co., Dayton 27,Ohio. Beam dimensions are taken fromfigure S.00.1 0.1~ 0.2 O.Z!JELEMENT SPACING(:>-)(PARASITIC ELEMENT TUNED FOR MAXIMUM GAIN)Fi{lure 3FRONT-TO-BACK RATIO AS A FUNCTIONOF ELEMENT SPACING FOR A TWO-ELE·MENT PARASITIC ARRAYThe effective bandwidth taken between the1.5/1 standing wave points of an array cut tothe above dimensions is about 2. 5% of theoperating frequency. This means that an arraypre· cut to a frequency of 14,15 0 kilocycleswould have a bandwidth of 350 kilocycles (plusor minus 175 kilocycles of the center frequen·cy), and therefore would be effective over thewhole 20 meter band. In like fashion, a 15meter array should be pre-cut to 21,200 kilo·cycles.A beam designed for use on the 10-meterband would have an effective bandwidth ofsome 700 kilocycles. Since the 10-meter bandis 1700 kilocycles in width, the array shouldeither be cut to 28,500 kilocycles for operationin the low frequency portion of the band,or to 29,200 kilocycles for operation in thehigh frequency portion of the band. Operationof the antenna outside the effective bandwidthwill increase the SWR on the transmissionline, and noticeably degrade both the gain andfront-to- back ratio performance. The heightabove ground also influences the F /B ratio.25-3 The Three-Element ArrayThe three-element array using a director,driven element, and reflector will exhibit asmuch as 30 db front·to·back ratio and 20 dbfront-to-side ratio for low-angle radiation. Thetheoretical gain is about 9 db over a dipole infree space. In actual practice, the array willoften show 7 to 10 db apparent gain over ahorizontal dipole placed the same height aboveground (at 28 and 14 Me.).The use of more than three elements is desirablewhen the length of the supporting struc·ture is such that spacings of approximately
<strong>HANDBOOK</strong>Parasitic Arrays 4 970.2 wavelength between elements becomespossible. Four-element arrays are quite commonon the 28-Mc. and 50-Mc. bands, and fiveelements are sometimes used for increasedgain and discrimination. As the number of elementsis increased the gain and front-to-hackratio increases but the radiation resistance decreasesand the bandwidth or frequency rangeover which the antenna will operate withoutreduction in effectiveness is decreased.Material for While the elements may consistElements of wire supported on a woodframework, self-supporting elementsof tubing are much to be preferred. Thelatter type array is easier to construct, looksbetter, is no more expensive, and avoids theproblem of getting sufficiently good insulationat the ends of the e I em en t s. The voltagesreach such high values towards the ends ofthe elements that losses will be excessive,unless the insulation is excellent.The elements may be fabricated of thin·walled steel conduit, or hard drawn thin-walledcopper tubing, but dural tubing is much better.Or, if you prefer, you may purchase taperedcopper-plated steel tubing elements designedespecially for the purpose. Kits are availablecomplete with rotating mechanism and directionindicator, for those who desire to purchasethe whole system ready to put up.Element Spacing The optimum spacing for atwo-element array is, as hasbeen mentioned before, approximately 0.11wavelength for a director and 0.13 wavelengthfor a reflector. However, when both a directorand a reflector are combined with the drivenelement to make up a three-element array theoptimum spacing is established by the bandwidthwhich the antenna will be required tocover. Wide spacing (of the order of 0.25 wavelengthbetween elements) will result in greaterbandwidth for a specified maximum standing·wave ratio on the antenna transmission line.Smaller spacings may be used when boomlength is an important consideration, but for aspecified standing-wave ratio and forward gainthe frequency coverage will be smaller. Thusthe Q of the antenna system will be increasedas the spacing between the elements is decreased,resulting in smaller frequency coverage,and at the same time the feed-point impedanceof the driven e I em en t will be decreased.For broad-band coverage, such as the rangefrom 26.96 to 29.7 Me. or from 50 to 54 Me.,0.2 wavelength spacing from the driven elementto each of the parasitic elements is rec·ommended. For narrower bandwidth, such aswould be adequate for the 14.0 to 14.4 Me.band or the 144 to 148 Me. band, the radiatorto parasitic element spacing may be reducedto 0.12 wavelength, while still maintainingadequate array bandwidth for the amateur bandin question.Length of theParasitic ElementsExperience has shown thatit is practical to cut theprarslttc elements of athree-element parasitic array to a predeterminedlength before the installation of such an antenna.A pre-tuned antenna such as this willgive good signal gain, adequate front-to-hackratio, and good bandwidth factor. By carefullytuning the array after it is in position the gainmay be increased by a fraction of a db, andthe front-to-hack ratio by several db. Howeverthe slight improvement in performance is usuallynot worth the effort expended in tuningtime.The closer the lengths of the parasitic elementsare to the resonant length of the drivenelement, the lower will be the feed-point resistanceof the driven element, and the smallerwill be the bandwidth of the array. Hence, forwide frequency coverage the director shouldbe considerably shorter, and the reflector considerablylonger than the driven element. Forexample, the director should still be less thana resonant half wave at the upper frequencylimit of the range wherein the antenna is to beoperated, and the reflector should still be longenough to act as a reflector at the lower frequencylimit. Another way of stating the samething is to say, in the case of an array to covera wide frequency range such as the amateurrange from 26.96 to 29.7 Me. or the width of alow-band TV channel, that the director shouldbe cut for the up p e r end of the band andthe reflector for the lower end of the band. Inthe case of the 26.96 to 29.7 Me. range thismeans that the director should be about 8 percent shorter than the dtiven element and thereflector should be about 8 per cent longer.Such an antenna will show a relatively constantgain of about 6 db over its range of coverage,and the pattern will not reverse at anypoint in the range.Where the frequency range to be covered issomewhat less, such as a high-band TV channel,the 14.0 to 14.4 Me. amateur band, or thelower half of the amateur 28-Mc. phone band,the reflector should be about 5 per cent longerthan the driven element, and the director about5 per cent shorter. Such an antenna will performwell over its rated frequency band, willnot reverse its pattern over this band, and willshow a signal gain of 7 to 8 db. See figure 5for design figures for 3-element arrays.
4 98 Rotary Beams <strong>THE</strong> R AD I 0TYPE DRIVEN ELEI.AENT REFLECTOR tST DIRECTOR 2ND DIRECTOR 3RD DIRECTOR SPACIN' BET- APPROX.CAIN APPROX. RADIATIONLE-NGTH LENGTH LENGTH LENGTH LENGTH WEEN ELEMENTS DB RES I STANCE (ft)3-ELEMENT473 501 445-- -- .1!>-.1!> 7.5 20F(MC) F lMC) .f" (MC)3-ELEMENT ~ ~-nee) - -- .2!>-.25 8.5 35F(MC) F (MC)4-ELEMENT ~ ~ ~ ~-- . 2-.2-.2 ...F(MC) F(MC) F(MC) F(MC)5-ELEMENT J.U_ ~ ~ ~ ~ .2- .2-.2-.2 10.0 15F(MC) F(MC) F(MC) F(MC) F(MC)20FigureSDESIGN CHART FOR PARASITIC ARRAYS (DIMENSIONS GIVEN IN FEET)More ThanThree ElementsA small amount of additionalgain may be obtained throughuse of more than two parasiticelements, at the expense of reduced feed-pointimpedance and lessened bandwidth. One additionaldirector will add about 1 db, and a secondadditional director (making a total of fiveelements including the driven element) willadd slightly less than one db more. In thev-h-f range, where the additional elements maybe added without much difficulty, and whererequired bandwidths are small, the use of morethan two parasitic elements is quite practicable.Stocking ofYagi ArraysParasitic arrays (yagis) maybe stacked to provide additionalgain in the same manner thatdipoles may be stacked. Thus if an array ofsix dipoles would give a gain of 10 db, thesubstitution of yagi arrays for each of the dipoleswould add the gain of one yagi array tothe gain obtained with the dipoles. However,the yagi arrays must be more widely spacedthan the dipoles to obtain this theoretical improvement.As an example, if six 5-elementyagi arrays having a gain of about 10 db weresubstituted for the dipoles, with appropriateincrease in the spacing between the arrays,the gain of the whole system would approachthe sum of the two gains, or 20 db. A group ofarrays of yagi antennas, with recommendeds p acing and approximate gains, are illustratedin figure 6.25-4 Feed Systems forParasitic (Yogi) ArraysThe table of figure 5 gives, in addition toother information, the approximate radiationresistance referred to the center of the drivenelement of multi-element parasitic arrays. It isobvious, from these lo.., values of radiationresistance, that especial care must be takenin materials used and in the construction ofthe elements of the array to insure that ohmiclosses in the conductors will not be an appreciablepercentage of the radiation resistance.It is also obvious that some method of iqtpedancetransformation must be u s e d in manycases to match the low radiation resistanceof these antenna arrays to the normal range ofcharacteristic impedance u s e d for antennatransmission lines.A group of possible methods of impedancematching is shown in figures 7, 8, 9 and 10.All these methods have been used but certainof them o f fer advantages over some of theother methods. Generally speaking it is notmechanically desirable to break the center ofthe driven element of an array for feeding thesystem. Breaking the driven element rules outthe practicability of building an all-metal or"plumber's delight" type of array, and imposesmechanical limitations with any type ofconstruction. However, when continuous rotationis desired, an arrangement such as shownin figure 9D utilizing a broken driven elementwith a rotatable transformer for coupling fromthe antenna transmission line to the drivenelement has proven to be quite satisfactory.In fact the method shown in figure 9D is probablythe most practicable method of feedingthe driven element when continuous rotationof the antenna array is required.The feed systems shown in figure 7 will,under normal conditions, show the lowest lossesof any type of feed system since the currentsflowing in the matching network are thelowest of all the systems commonly used. The"Folded Element" match shown in figure 7 Aand the "Yoke" match shown in figure 7B arethe most satisfactory electrically of all standardfeed methods. However, both methods requirethe extension of an additional conductorout to the end of the driven element as a portionof the matching system. The folded-elementmatch is best on the 50-Mc. band and
<strong>HANDBOOK</strong>Stacked Vagi Arrays4 99!--o.2 >--+-o.n----jFEEDER LINEDIRECTIONALDIRECTIONAL® GAIN ABOUT 12 DBWITH 2 SECTIONSFEEDER LINE® GAIN ABOUT 1~ DBWITH 3 SECTIONS© GAl N ABOUT 17 DBFigure 6STACKED YAGI ARRAYSIt Is possible to attain a relatively large amount of gain over a limited bandwidth with stackedyogi arrays, The two-sect/on array at (A) will give a gain of about 12 db, while adding a thirdsection will bring the gain up to about 15 db. Adding two additional parasitic directors to eachsection, as at (C) will bring the goin up ta about 17 db.higher where the a.dditional section of tubingmay be supported below the main radiator elementwithout undue difficulty. The yoke-matchis more satisfactory mechanically on the 28-Me. and 14-Mc. bands since it is only necessaryto suspend a wire below the driven elementproper. The wire may be spaced belowthe self-supporting element by means of several
500 Rotary Beams <strong>THE</strong> R AD I 0sFOR D1=1"D2=.zs•~=16S= ,. .s FOR 0= 1"#12;1~E 3 u ~= 11®YOKE MATCHFOR 0~ 1•s = 2" B..Eu.c__=14#12.WIRE ~FOR 0= 1"S= 1.5•~=18#12 WIRE RRAD.FORD= 1•S= '. R~~·D~ =24#I WIREFORD= 1 11#12 ~7R~- ~ = 32RFEED= 9RRAD.R FEED = APPROX. 25RRAD.Figure 7DATA FORFOLDED-ELEMENTMATCHING SYSTEMSIn all normal applications ofthe clata given the main elementas shown is the drivenelement of a multi-elementparasitic array. Directors anclreflectors have not beenshown for the sake of clarity.small strips of polystyrene which have beendrilled for both the main element and the smallwire and threaded on the main element.The Folded-Element The calculation of theMatch Calculations operating conditions ofthe folded-elementmatching system and the yoke match, as shownin figures 7 A and 7B is relatively simple. Aselected group of operating. conditions hasbeen shown on the drawing of figure 7. In applyingthe system it is only necessary to multiplythe ratio of feed to radiation resistance(given in the figures to the right of the suggestedoperating dimensions in figure 7) bythe radiation resistance of the antenna systemto obtain the impedance of the cable to beused in feeding the array. Approximate valuesof radiation resistance for a number of commonlyused parasitic-element arrays are givenin figure 5.As an example, suppose a 3-element arraywith 0.15D-0.15R spacing between elements isto be fed by me an s of a 465-ohm line constructedof no. 12 wire spaced 2 inches. Theapproximate radiation resistance of such anantenna array will be 20 ohms. Hence we needa ratio of impedance step up of 23 to obtaina match between the characteristic impedanceof the transmission line and the radiation resistanceof the driven element of the antennaarray. Inspection of the ratios given in figure7 shows that the fourth set of dimensionsgiven under figure 7B will give a 24-to-1 stepup, which is sufficiently close. So it is merelynecessary to use a l-inch diameter driven ele·ment with a no. 8 wire spaced on 1 inch centers(~ inch below the outside wall of the l-inchtubing) below the l-inch element. The no. 8wire is broken and a 2-inch insulator placedin the center. The feed line then carries fromthis insulator down to the transmitter. Thecenter insulator should be supported rigidlyfrom the l-inch tube so that the spacing betweenthe piece of tubing and the no. 8 wirewill be accurately maintained.
<strong>HANDBOOK</strong> Matching Systems 501~----------------L----------------~r--1•'Vo L--1--'""' L--!@ DELTA MATCHDIMENSIONS SHOWN GIVEAPPftOX. MATCH TO 500.0.AIR-SPAC!:D L.INEFigure 8AVERAGE DIMENSIONSFOR <strong>THE</strong> DELTA AND"T" MATCH@ "T* MATCHD1=3DzIn many cases it will be desired to use thefolded-element or yoke matching system withdifferent sizes of conductors or different spacingsthan those shown in figure 7. Note, then,that the impedance transformation ratio ofthese types of matching systems is dependentboth upon the ratio of conductor diameters andupon their spacing. The following equationhas been given by Roberts (RCA Review, June,1947) for the determination of the impedancetransformation when using different diametersin the two sections of a folded element:z 2Transformation ratio = ( 1 + ~ )z.In this equation Z, is the characteristic impedanceof a line made up of the smaller ofthe two conductor diameters spaced the centerto-centerdistance of the two conductors in theantenna, and Z 2is the characteristic impedanceof a line made up of two conductors thesize of the larger of the two. This assumesthat the feed line will be connected in serieswith the smaller of the two conductors so thatan impedance step up of greater than four willbe obtained. If an impedance step up of lessthan four is desired, the feed line is connectedin series with the larger of the two conductorsand Z 1 in the above equation becomes the impedanceof a hypothetical line made up of thelarger of the two conductors and Z 2is madeup of the smaller. The folded v-h-f unipole isan example where the transmission line is connectedin series with the 1 a r g e r of the twoconductors.The conventional 3-wire match to give animpedance "multiplication of 9 and the 5-wirematch to give a ratio of approximately 25 areshown in figures 7C and 7D. The 4-wire match,not shown, will give an impedance transformationratio of approximately 16.The Delta Matchand T-MatchThe Delta match and theT-match are shown in figure8. The delta match has beenlargely superseded by the newer T-match, howeverboth these systems can be adjusted togive a low value of SWR on 50 to 600-ohm balancedtransmission lines. In the case of thesystems shown it will be necessary to makeadjustments in the tapping distance along thedriven radiator until minimum standing waveson the antenna transmission line are obtained.Since it is sometimes impracticable to eliminatecompletely the standing waves from theantenna transmission line when using thesematching systems, it is common practice tocut the feed line, after standing waves havebeen reduced to a minimum, to a length whichwill give satisfactory loading of the transmitterover the desired frequency range of operation.The inherent reactance of the T-match istuned out by the use of two identical resonatingcapacitors in series with each leg of theT-rod. These capacitors should each have amaximum capacity of 8 p.pfd. per meter of wavelength.Thus for 20 meters, each capacitorshould have a maximum capacity of at least160 p.p.fd. For power up to a kilowatt, 1000volt spacing of the capacitors is adequate.
502 Rotary Beams <strong>THE</strong> R AD I 0®DIRECT FEED WITHCOAXIAL CABLE:2.8 MC.- 4 TURNS 2" DIA., 2." LONGANT. TAPPED t TURN EACH SIDE14 MC.- 8 TURNS 2" DIA., 2" LONGANT. TAPPED 2 TURNS EACH SIDEFigure 9ALTERNATE FEEDMETHODS WHERE <strong>THE</strong>DRIVEN i:LEMENT MAYBE BROKEN IN <strong>THE</strong>CENTERCOIL SPACEDAPPROX. 0.~"COILS 10•DIAMETER@ROTARY LINKCOUPLING1 TURN LINKS ARE PARALLELC IS 200 JJJJFD VARIABLEThese capacitors should be tuned for minimumSWR on the transmission line. The adjustmentof these capacitors should be made at the sametime the correct setting of the T-match rods ismade as the two adjustments tend to be interlocking.The use of the standing wave meter(described in Test Equipment chapter) isrecommended for making these adjustments tothe T-match.Feed Systems Usinga Driven Elementwith Center FeedFour methods of excitingthe driven element of aparasitic array are shownin figure 9. The systemshown at (A) has proven to be quite satisfactoryin the case of an antenna-reflector twoelementarray or in the case of a three-elementarray with 0.2 to 0.25 wavelength spacing betweenthe elements of the antenna system. Thefeed-point impedance of the center of the drivenelement is close enough to the characteristicimpedance of the 52-ohm coaxial cable so thatthe standing-wave ratio on the 52-ohm coaxialcable is less than 2-to-1. (B) shows an arrangementfor feeding an array with a broken drivenelement from an open-wire line with the aid ofa quarter-wave matching transformer. With 465-ohm line from the transmitter to the antennathis system will give a close match to a 12-ohm impedance at the center of the driven element.(C) shows an arrangement which uses anuntuned transformer with lumped inductancefor matching the transmission line to the centerimpedance of the driven element.Rotary LinkCouplingIn many cases it is desirable tobe able to allow the antenna arrayto rotate continuously withoutregard to snarling of the feed line. If this is tobe done some sort of slip rings or rotary jointmust be made in the feed line. One relativelysimple methodof allowing unrestrained rotationof .the antenna is to use the method of rotarylink coupling shown in figure 9D. The two cou-
<strong>HANDBOOK</strong>The Gamma Match 503TO TRANSMITTER"'FLAT• LINESWR=1.oRESONANTSECTION~ANY ANTEN A Z.SIMPLE OR CO PLEXMATCHING STUBFigure 10<strong>THE</strong> GAMMA MATCHING SYSTEMSee text for details of resonating capacitorFigure IIIMPEDANCE MATCHING WITH A CLOSEDSTUB ON A TWO WIRE TRANSMISSIONLINEpiing rings are 10 inches in diameter and areusually constructed of ~-inch copper tubingsupported one from the rotating structure andone from the fixed structure by means of standoffinsulators. The capacitor C in figure 9D isadjusted, after the antenna has been tuned, forminimum standing-wave ratio on the antennatransmission line. The dimensions shown willallow operation with either 14-Mc. or 28-Mc.elements, with appropriate adjustment of thecapacitor C. The rings must of course be paralleland must lie in a plane normal to the axisof rotation of the rotating structure.The Gamma Match The use of coaxial cableto feed the driven elementof a yagi array is becoming increasingly popular.One reason for this increased popularitylies in the fact that the TVI-reduction problemis simplified when coaxial feed line is usedfrom the transmitter to the antenna system.Radiation from the feed line is minimized whencoaxial cable is used, since the outer conductorof the line may be grounded at severalpoints throughout its length and since the intensefield is entirely confined within the outerconductor of the coaxial cable. Other advantagesof coaxial cable as the antenna feedline lie in the fact that coaxial cable may berun within the structure of a building withoutdanger, or the cable may be run undergroundwithout disturbing its operation. Also, transmitting-typelow-pass filters for 52 ohm impedanceare more widely available and are lessexpensive than equivalent filters for two-wireline.The gamma-match is illustrated in figure 10,and may be looked upon as one-half of a Tmatch. One resonating capacitor is used,placed in series with the gamma rod. The capacitorshould have a capacity of 7 llllfd· permeter of wavelength. For 15-meter operationthe capacitor should have a maximum capacityof 105 llllfd. The length of the gamma rod determinesthe impedance transformation betweenthe transmission line and the driven elementof the array, and the gamma capacitor tunesout the inductance of the gamma rod. By adjustmentof the length of the gamma rod, andthe setting of the gamma capacitor, the SWRon the coaxial line may be brought to a verylow value at the chosen operating frequency.The use of an Antennascope, described in theTest Equipment chapter is recommended forprecise adjustment of the gamma match.The Matching Stub If an open-wire line isused to fe~d a low impedanceradiator, a section of the transmissionline may be employed as a matching stub asshown in figure 11. The matching stub cantransform any complex impedance to the characteristicimpedance of the transmission line.While it is possible to obtain a perfect matchand good performance with either an open stubor a shorted one by observing appropriate dimensions,a shorted stub is much more readilyadjusted. Therefore, the following discussionwill be confined to the pro b 1 em of using aclosed stub to match a low impedance load toa high impedance traqsmission line.If the transmission line is so elevated thatadjustment of a "fundamental" shorted stubcannot be accomplished easily from the ground,then the stub length may be increased by exactlyone or two electrical half wavelengths,without appreciably affecting its operation.While the correct position of the shortingbar and the point of attachment of the stub tothe line can be determined entirely by experimentalmethods, the fact that the two adjustmentsare interdependent or interlocking makessuch a cut-and-try procedure a tedious one.Much time can be saved by determining the approximateadjustments required by reference toa chart such as figure 12 and using them as astarter. Usually only a slight "touching up"will produce a perfect match and flat line.In order to utilize figure lZ, it is first neces·sary to locate accurately a voltage node orcurrent node on the line in the vicinity that
504 Rotary Beams<strong>THE</strong> R AD I 0•ne1138""7/311-1/311 '-'- .)/J'I"z5131.........4~"i)O...-"Nc~,.~~! ....'1'-1(,.,.,r;;;::;. ,.... .,,90Iao::Jc. w70~" waooso;:1-40~w>O.J.J2131tow~Ro to'}o20~a:I I 01-02 3 ..SWR~0 .JI 7 8 9 10 WFigure 12SHORTED STUB LENGTH AND POSITIONCHARTFrom the stonding wave ratio and current orvoltage null position it is possible to determinethe theoretically correct},length andposition of• a shorted stub. In ctual prac•tice a sTight discrepancy usu lly will befound between the theoretical and fhe ex•perlmentally optimized dimensions; thereforeIt may be necessary to "touch up" the c/;.mensions after using the above data as astarting point.DIRECTIONAL® GAIN ABOUT 6 DB FEED LINE®FEED LINEhas been decided upon for the stub, and alsoto determine the SWR.Stub adjustment becomes more critical asthe SWR increases, and under conditions ofhigh SWR the current and voltage nulls aremore sharply defined than the current and volt·age maxima, or loops. Therefore, it is best tolocate either a current null or voltage null, de·pending upon whether a current indicating de·vice or a voltage indicating device is used tocheck the standing wave pattern.The SWR is determined by means of a "di·rectional coupler," or by noting the ratio ofEmax to Emin or Imax to Imin as read on anindicating device.It is assumed that the characteristic imped·ance of the section of line used as a stub isthe same as that of the transmission line prop·er. It is preferable to have the stub sectionidentical to the line physically as well aselectrically.25-5 UnidirectionalDriven ArraysThree types of unidirectional driven arraysare illustrated in figure 13. The array shownin figure 13A is an end-fire system which mayFigure 13UNIDIRECTIONAL ALL-DRIVEN ARRAYSA unidirectional all-driven end·flre array isshown at (A), (B) shows an array with twohalf waves in phase with driven reflectors,A Lazy-H array with driven reflectors isshown at (C). Note that the directivity isthrough the elements with the greatest totalfeed· I ine length in arrays such as shown at(B) and (C),be used in place of a parasitic array of similardimensions when greater frequency coveragethan is available with theyagi type is desired.Figure 13B is a combination end-fire and co·
<strong>HANDBOOK</strong>Driven Arrays 505linear system which will give approximatelythe same gain as the system of figure 13A, butwhich requires less boom length and greatertotal element length. Figure 13C illustratesthe familiar lazy-H with driven reflectors (ordirectors, depending upon the point of view)in a combination which will show wide bandwidthwith a considerable amount of forwardgain and good front-to-hack ratio over the entirefrequency coverage.Unidirectional StackedBroadside ArraysThree p r act i c a b 1 etypes of unidirectionalstacked broadside arraysare shown in figure 14. The first type,shown at figure 14A, is the simple "lazy H"type of antenna with parasitic reflectors foreach element. (B) shows a simpler antenna arraywith a pair of folded dipoles spaced onehalfwave vertically, operating with reflectors.In figure 14C is shown a more complex arraywith six half waves and six reflectors whichwill give a very worthwhile amount of gain.In all three of the antenna arrays shown thespacing between the driven elements and thereflectors has been shown as one-quarter wavelength.This has been done to eliminate therequirement for tuning of the reflector, as aresult of the fact that a half-wave elementspaced exactly one-quarter wave from a drivenelement will make a unidirectional array whenboth elements are the same length. Using thisprocedure will give a gain of 3 db with the reflectorsover the gain without the reflectors,with only a moderate decrease in the radiationresistance of the driven element. Actually,the radiation resistance of a half-wave dipolegoes down from 73 ohms to 60 ohms when anidentical half-wave element is placed onequarterwave behind it.A very slight increase in gain for the entirearray (about 1 db) ma.r be obtained at the expenseof lowered radiation resistance, the necessityfor tuning the reflectors, and decreasedbandwidth by placing the reflectors 0.15 wavelengthbehind the driven elements and makingthem somewhat longer than the driven elements.The radiation resistance of each element willdrop approximately to one-half the value obtainedwith untunedhalf-wave reflectors spacedone-quarter wave behind the driven elements.Antenna arrays of the type shown in figure14 require the use of some sort of lattice workfor the supporting structure since the arraysoccupy appreciable distance in space in allthree planes.Feed Methods The requirements for the feedsystems for antenna arrays ofthe type shown in figure 14 are less criticalthan those for the close-spaced parasitic arraysshown in the previous section. This is anatural result of the fact that a larger numberof the radiating elements are directly fed withenergy, and of the fact that the effective radiationresistance of each of the driven elementsof the array is much higher than the feed-pointresistance of a parasitic array. As a consequenceof this fact, arrays of the type shownin figure 14 can be expected to cover a somewhatgreater frequency band for a specifiedvalue of standing-wave ratio than the parasitictype of array.In most cases a simple open-wire line maybe coupled to the feed point of the array withoutany matching system. The standing-waveratio with such a system of feed will often beless than 2-to-1. However, if a more accuratematch between the antenna transmission lineand the array is desired a conventional quarter-wavestub, or a quarter-wave matchingtransformer of appropriate impedance, may beused to obtain a low standing-wave ratio.25-6 Bi-DirectionalRotatable ArraysThe bi-directional type of array is sometimesused on the 28-Mc. and 50-Mc. bands wheresignals are likely to be coming from only onegeneral direction at a time. Hence the sacrificeof dis~rimination against signals arrivingfrom the opposite direction is likely to be oflittle disadvantage. Figure 15 shows two generaltypes of bi-directional arrays. The flattopbeam, which has been described in detailearlier, is well adapted to installation atop arotating structure. When self-supporting elementsare used in the flat-top beam the problemof losses due to insulators at the ends ofthe elements is somewhat reduced. With asingle-section flat-top beam a gain of approximately4 db can be expected, and with twosections a gain of approximately 6 db can beobtained.Another type of bi-directional array whichhas seen less use than it deserves is shownin figure 15B. This type of antenna system hasa relatively broad azimuth or horizontal beam,being capable of receiving signals with littlediminution in strength over approximately 40°,but it has a quite sharp elevation pattern sincesubstantially all radiation is concentrated atthe lower angles of radiation if more than atotal of four elements is used in the antennasystem. Figure 15B gives the approximate gainover a half-wave dipole at the height of thecenter of the array which can be expected. Alsoshown in this figure is a type of "rotatingmast" structure which is well suited to rotationof this type of array.
506 Rotary Beams <strong>THE</strong> R AD I 0"LAZY H" WITH REFLECTORCAIN APPROX. 9 DBt STUB®BROADSIDE HALF-WAVESWITH REFLECTORSGAIN APPROX. 7 DB©"TWO OVER TWO OVER TWO"WITH REFLECTORSGAIN APPROX. If.! DBFigure 14BROADSIDE ARRAYSWITH PARASITICREFLECTORSThe apparent gain of the ar•rays illustratecl will be great•er than the values given clueto concentration of the racli·atecl si,nal at the lower ele·vat/on angles.If six or more elements are used in the typeof array shown in figure 15B no matching sec·tion will be required between the antenna trans·mission line and the feed point of the antenna.When only four elements are used the antennais the familiar "lazy H" and a quarter-wavestub should be used for feeding from the an·tenna transmission line to the feed point ofthe antenna system.If desired, and if mechanical considerationspermit, the gain of the arrays shown in figure15B may be increased by 3 db by placing ahalf·wave reflector behind each of the ele·ments at a spacing of one-quarter wave. Thearray then becomes essentially the same asthat shown in figure 14C and the same con·siderations in regard to reflector spacing andtuning will apply. However, the factor that abi-directional array need be rotated through anangle of less than 180° should be consideredin this connection.25-7 Construction ofRotatable ArraysA considerable amount of ingenuity may beexercised in the construction of the supportingstructure for a rotatable array. Every personbas his own ideas as to the best method ofconstruction. Often the most practicable meth·od of construction will be dictated by the
<strong>HANDBOOK</strong>Bi-directional Arrays 507STUB0FLAT-TOP BEAM FORROTATABLE ARRAYGAIN 4 TO I 08®"TWO OVER TWO OVER TWO•TYPE OF ARRAYFigure 15TWO GENERAL TYPESOF BI-DIRECTIONALARRAYSAverage gain figures are giv•en for both the flat-top beamtype of array and for thebroadside-colinear array withdifferent numbers of elements.RAO!Al l-OADBEARINC. -TOTAL NUMBEROF ELEMENTS:>..toavailability of certain types of constructionalmaterials. But in any event be sure that soundmechanical engineering principles are used inthe design of the supporting structure. Thereare few things quite as discouraging as thepicking up of pieces, repairing of the roof, etc.,when a newly constructed rotary comes downin the first strong wind. If the principles ofmechanical engineering are understood it iswise to calculate the loads and torques whichwill exist in the various members of the structurewith the highest wind velocity which maybe expected in the locality of the installation.If this is not possible it will usually be worththe time and effort to look up a friend whounderstands these principles.RadiatingElementsOne thing more or less standardabout the construction of rotatableanteiUla arrays is the use of duraltubing for the self-supporting elements. Othermaterials may be used but an alloy known as24ST has proven over a period of time to bequite satisfactory. Copper tubing is too heavyfor a given strength, and steel tubing, unlesscopper plated, is likely to add an undesirablylarge loss resistance to the array. Also, steeltubing, even when plated, is not likely towithstand salt atmosphere such as encounteredalong the seashore for a satisfactory periodof time. Do not use a soft aluminum alloy forthe elements unless they will be quite short;24ST is a hard alloy and is best although thereare several other alloys ending in "ST" whichwill be found to be satisfactory. Do not usean alloy ending in "SO" or "S" in a positionin the array where structural strength is im·portant, since these letters designate a metalwhich has not been heat treated for strengthand rigidity. However, these softer alloys, andaluminum electrical conduit, may be used forshort radiating elements such as would beused for the 50-Mc. band or as interconnectingconductors in a stacked array.
5 08 Rotary Beams<strong>THE</strong> R AD I 0LINE OF /- - - -
<strong>HANDBOOK</strong>Tuning the Array 509driven onto a wooden dowel, as shown infigure 178. The element may then be mountedupon an aluminum support plate by means offour ceramic insulators. Metal based insulators,such as the Johnson 135-67 are recommended,since the all-ceramic types may break at themounting holes when the array is subjectedto heavy winds.25-8 Tuning the ArrayAlthough satisfactory results may be obtainedby pre-cutting the antenna array to thedimensions given earlier in this chapter, theoccasion might arise when it is desired tomake a check on the operation of the antennabefore calling the job complete.The process of tuning an array may fairlysatisfactorily be divided into two more or lessdistinct steps: the actual tuning of the arrayfor best front·to-back ratio or for maximum for·ward gain, and the project of obtaining thebest possible impedance match between theantenna transmission line and the feed pointof the array.Tuning the The actual tuning of the arrayArray Proper for best front-to·back ratio ormaximum forward gain may bestbe accomplished with the aid of a low-powertransmitter feeding a dipole antenna (polarizedthe same as the array being tuned) at leastfour or five wavelengths away from the antennabeing tuned and located at the same elevationas that of the antenna under test. A calibratedfield-strength meter of the remote-indicatingtype is then coupled to the feed pointof the antenna array being tuned. The transmissionsfrom the portable transmitter shouldbe made as short as possible and the call signof the station making the test should be transmittedat least every ten minutes.It is, of course, possible to tune an arraywith the receiver connected to it and with astation a mile or two away making transmis·sions on your request. But this method is morecumbersome and is not likely to give completesatisfaction. It is also possible to carry outthe tuning process with the transmitter connectedto the array and with the field-strengthmeter connected to the remote dipole antenna.In this event the indicating instrument of theremote-indicating field-strength meter shouidbe visible from the position where the elementsare being tuned. However, when the array isbeing tuned with the transmitter connected toit there is always the problem of making con·tinual adjustments to the transmitter so that aconstant amount of power will be fed to thearray under test. Also, if you use this system,use very low power (5 or 10 watts of power isusually sufficient) and make sure that the an·tenna transmission line is effectively groundedas far as d-e plate voltage is concerned. Theuse of the method described in the previousparagraph of course eliminates these problems.One satisfactory method for tuning the arrayproper, assuming that it is a system with severalparasitic elements, is to set the directorsto the dimensions given in figure 5 and thento adjust the reflector for maximum forwardsignal. Then the first director should be variedin length until maximum forward signal is obtained,and so on if additional directors areused. Then the array may be reversed in direc·tion and the reflector adjusted for best front·to-hack ratio. Subsequent small adjustmentsmay then be made in both the directors and thereflector for best forward signal with a reason·able ratio of front·to·back signal. The adjust·ments in the directors and the reflector willbe found to be interdependent to a certain de·gree, but if small adjustments are made afterthe preliminary tuning process a satisfactoryset of adjustments for maximum performancewill be obtained. It is usually best to makethe end sections of the elements smaller indiameter so that they will slip inside the largertubing sections. The smaller sliding sectionsmay be clamped inside the larger main sec·tions.In making the adjustments described, it isbest to have the rectifying element of the remote-indicatingfield-strength meter directlyat the feed point of the array, with a resistorat the feed point of the estimated value offeed-point impedance for the array.Matching to theAntenna Trans·mission LineThe problem of matching theimpedance of the antennatransmission line to the arrayis much simplified if the pro-DRIVEN ELEMENT~~RID DIP "ETER+-MOVEABLESHORTINGBARFigure 18ADJUSTMENT OF GAMMA MATCH BY USEOF ANTENNASCOPE AND GRID-DIPMETER
l~~7T08FT.MAST UP TOFULL HEIGHTIMPORTANT NOTEBE SURE THAT <strong>THE</strong> PI PES FITWITHOUT BINDING AS <strong>THE</strong>REIS SOME VARIATION INDIAMETERS OF TH£ SIZES USED.TWO 3/8" OR 1/2" MACHINESCREWS ABOUT 1.12'"' LONGTHROUGH BEARING PIECEAND TAPPED INT01'"' PIPE.~~~::;:=~r,ABOUT 2H OF 1 1/4" PIPE.THIS IS <strong>THE</strong> BEARING THATCARRIES <strong>THE</strong> LOAD.II,.1_II /'~~~~J~:!~~~~~:~JJ~1".ti:~GE.7-8FT.:::001>1'
<strong>HANDBOOK</strong>Tuning the Array 511cess of tuning the array is made a substantiallyseparate process as just described. Afterthe tuning operation is complete, the resonantfrequency of the driven element of the antennashould be checked, directly at the center ofthe driven element if practicable, with a griddipmeter. It is important that the resonant frequencyof the antenna be at the center of thefrequency band to be covered. If the resonantfrequency is found to be much different fromthe desired frequency, the length of the drivenelement of the array should be altered untilthis condition exists. A relatively small changein the length of the driven element will haveonly a second order effect on the tuning of theparasitic elements of the array. Hence, a moderatechange in the length of the driven elementmay be made without repeating the tuningprocess for the parasitic elements.When the resonant frequency of the antennasystem is correct, the antenna transmissionline, with impedance-matching device or networkbetween the line and antenna feed point,is then attached to the array and coupled to alow-power exciter unit or transmitter. Then,preferably, a standing-wave meter is connectedin series with the antenna transmission lineat a point relatively much more close to thetransmitter than to the antenna. However, forbest indication there should be 10 to 15 feetof line between the transmitter and the standing-wavemeter. If a standing-wave meter isnot available the standing-wave ratio may bechecked approximately by means of a neonlamp or a short fluorescent tube if twin transmissionline is being used, or it may be checkedwith a thermomilliammeter and a loop, aneon lamp, or an r-f ammeter and a pair ofclips spaced a fixed distance for clipping ontoone wire of a two-wire open line.If the standing-wave ratio is below 1. 5 to 1it is satisfactory to leave the installation asit is. If the ratio is greater than this range itwill be best when twin line or coaxial line isbeing used, and advisable with open-wire line,to attempt to decrease the s.w.r.It must be remembered that no adjustmentsmade at the transmitter end of the transmissionline will alter the SWR on the line. All adjustmentsto better the SWR must be made at theantenna endof the line and to the device whichperforms the impedance transformation necessaryto match the characteristic impedance ofthe antenna to that of the transmission line.Before any adjustments to the matching systemare made, the resonant frequency of thedriven element must be ascertained, as explainedpreviously. If all adjustments to correctimpedance mismatch are made at this frequency,the problem of reactance terminationof the transmission line is eliminated, greatlysimplifying the problem. The following stepsshould be taken to adjust the impedance transformation:1. The output impedance of the matchingdevice should be measured. An Antennascopeand a grid-dip oscillator are requiredfor this step. The Antennascopeis connected to the output terminals ofthe matching device. If the driven elementis a folded dipole, the Antennascopeconnects directly to the split section ofthe dipole. If a gamma match or T-matchare used, the Antennascope connects tothe transmission-line end of the device.If a Q-section is used, the Antennascopeconnects to the bottom end of the section.The grid-dip oscillator is coupledto the input terminals of the Antennascopeas shown in figure 18.2. The grid-dip oscillator is tuned to theresonant frequency of the antenna, whichFigure 19ALL-PIPE ROTATING MAST STRUCTURE FOR ROOF INSTALLATIONAn installation suitable for a building with a pitched roof is shown at (A). At (B) Is shown asimilar installation for a flat or shec/ roof. The arrangement as shown is strong enough to supporta lightweight 3-element 28-Mc. array one/ a I ight 3-element SO-Me. array above the 28-Mc.array on the enc/ of a 4-foot length of !-l-inch pipe.The lengths of pipe shown were chosen so that when the system is in the lowered position onecan stone/ on a household lac/c/er one/ put the beam in position atop the rotating pipe. The lengthsmay safely be revised upward somewhat if the array is of a particularly lightweight design withlow wind resistance.Just before the mast is installed it is a goocl iclea to give the rotating pipe a gooc/ smearing ofcup grease or waterproof pump grease. To get the lip of the top of the stationary section of 1 ~inch pipe to project above the flange plate, it will be necessary to hove a plumbing shop cut aslightly deeper thread Inside the flange plate, as well as cutting an unusually long thread onthe end of the 1 '14•inc:h pipe, It is relatively easy to waterproof thi:s assembly through the roofsince the 1 ~-inch pipe is stationary at all times. Be sure to use pipe compound on all the jointsancl then really tighten these joints with a pair of pipe wrenches.
512 Rotary Beams<strong>THE</strong> R A 0 I 0has been determined previously, and theAntennascope control is rurned for a nullreadingon the meter of the Antennascope.The impedance presented to the Antennascopeby the matching device may beread directly on the calibrated dial ofthe Antennascope.3. Adjustments should be made to thematching device to present the desiredimpedance transformation to the Antennascope.If a folded dipole is used as thedriven element, the transformation ratioof the dipole must be varied as explainedpreviously in this chapter to provide amore exact match. If a T·match or gammamatch system is used, the length of thematching rod may be changed to effecta proper match. If the Antennascope ohm·ic reading is lower than the desired reading,the length of the matching rod shouldbe increased. If the Antennascope readingis higher than the desired reading,the length of the matching rod should bedecreased. After each change in lengthof the matching rod, the series capacitorin the matching system should be reresonatedfor best null on the meter ofthe Antennascope.Raising andLoweringthe ArrayA practical problem always presentwhen tuning up and matchingan array is the physical locationof the structure. If the array isatop the mast it is inaccessible for adjustment,and if it is located on stepladders where it canbe adjusted easily it cannot be rotated. Oneencouraging factor in this situation is the factthat experience has shown that if the array isplaced 8 or 10 feet above ground on some stepladdersfor the preliminary tuning process, theraising of the system to its full height will notproduce a serious change in the adjustments.So it is usually possible to make preliminaryadjustments with the system located slightlygreater than head height above ground, andthen to raise the antenna to a position whereit may be rotated for final adjustments. If theposition of the sliding sections as determinednear the ground is marked so that the adjust·ments will not be lost, the array may be raisedto rotatable height and the fastening clampsleft loose enough so that the elements may beslid in by means of a long bamboo pole. Aftera series of trials a satisfactory set of lengthscan be obtained. But the end results usuallycome so close to the figures given in figure 5that a subsequent array is usually cut to thedimensions given and installed as-is.The matching process does not require rotation,but it does require that the antennaproper be located at as nearly its normal opec--Figure 20HEAVY DUTY ROTATOR SUITABLEFOR AMATEUR BEAMSThe new Corne/1-Dubi/ier type HAM-lrotor has extra heavy motor and gear ..ing system to withstand weight andInertia of amateur array under thebuffeting of heavy winds. Steel spurgears ancl rotor lock prevent "p/n ..wheellngn of antenna.
<strong>HANDBOOK</strong> Antenna Control Systems 513IIIIIICONTROL BOX---------,,.--_;S:.:;:O,;:;CK"iET 1 PLUGSOCKET!~..--~-------,&-CONTACT JONES PLUGS & SOCKETSANTENNA ROTATORII SYNCHRO. IL __-- _:E~R~O~jROTARY BEAM CONTROLD.P.O.T. TOGGLE SWITCHr---1IIIIIIro 115 v.A.c.1• ~~~~--+-~II TOGGLESWITCHSOCKETL--PLUGSOCKETI- _jFigure 21SCHEMATIC OF A COMPLETE ANTENMA CONTROL SYSTEMating pos1t10n as possible. However, on a particularinstallation the positions of the currentminimums on the transmission line near thetransmitter may be checked with the array inthe air, and then the array may be lowered toascertain whether or not the positions of thesepoints have moved. If they have not, and inmost cases if the feeder line is strung out backand forth well above ground as the antenna islowered they will not change, the positions ofthe last few toward the antenna itself may bedetermined. Then the calculation of the matchingquarter-wave section may be made, the sectioninstalled, the standing-wave ratio againchecked, and the antenna re-installed in itsfinal location.25-9 Antenna Rotation SystemsStructures for the rotation of antenna arraysmay be divided into two general classes: therotating mast and the rotating platform. Therotating mast is especially suitable where thetransmitting equipment is installed in the garageor some structure away from the mainhouse. Such an installation is shown in figure19. A very satisfactory rotation mechanism isobtained by the use of a large steering wheellocated on the bottom pipe of the rotating mast,with the thrust bearing for the structure locatedabove the roof.If the rotating mast is located a distancefrom the operating position, a system of pulleysand drive rope may be used to turn the antenna,or a slow speed electric motor may beemployed.The rotating platform system is best if atower or telephone pole is to be used for antennasupport. A number of excellent rotatingplatform devices are available on the marketfor varying prices. The larger and more expensiverotating devices are suitable for the rotaofa rather sizeable array for the 14-Mc. bandwhile the smaller structures, such as thosedesigned for rotating a TV antenna are design·ed for less load and should be used only witha 28-Mc. or 50-Mc. array. Most common practiceis to install the rotating device atop aplatform built at the top of a telephone poleor on the top of a lattice mast of sizeable crosssection so that the mast will be self-supportingand capable of withstanding the torque imposedupon it by the rotating platform.A heavy duty TV rotator may be employedfor rotation of 6 and 10-meter arrays. Fifteenand twenty meter arrays should use rotatorsdesigned for amateur use such as the CornellDubilier H AM-1 unit shown in figure 20.
514 Rotary Beams25-10 Indication of DirectionThe most satisfactory method for indicatingthe direction of transmission of a rotatablearray is that which uses Selsyns or synchrosfor the transmission of the data from the rotatingstructure to the indicating pointer at theoperating position. A number of synchros andSelsyns of various types are available on thesurplus market. Some of them are designed foroperation on 115 volts at 60 cycles, some aredesigned for operation on 60 cycles but at alowered voltage, and some are designed foroperation from 400-cycle or 800-cycle energy.This latter type of high-frequency synchro isthe most generally available type, and thehigh-frequency units are smaller and lighterthan the 60-cycle units. Since the indicatingsynchro must de I i v e r an almost negligibleamount of power to the pointer which it drives,the high-frequency types will operate quitesatisfactorily from 60-cycle power if the voltageon them is reduced to somewhere between6.3 and 20 volts. In the case of many of theunits available, a connection sheet is providedalong with a recommendation in regard to theoperating voltage when they are run on 60 cycles.In any event the operating voltage shouldbe held as low as it may be and still give satisfactorytransmission of data from the antennato the operating position. Certainly it shouldnot be necessary to run such a voltage on theunits that they become overheated.A suitable Selsyn indicating system is shownin figure 21.Systems using a potentiometer capable ofcontinuous rotation and a milliammeter, alongwith a battery or other source of direct current,may also be used for the indication of direction.A commercially-available potentiometer(Ohmite RB-2) may be used in conjunctionwith a 0-1 d-e milliammeter having a handcalibratedscale for direction indication.25-11 "Three-Band" BeamsA popular form of beam antenna introducedduring the past few years is the so-calledthree-band beam. An array of this type is designedto operate on three adjacent amateurbands, such as the ten, fifteen, and twentymeter group. The principle of operation ofthis form of antenna is to employ paralleltuned circuits placed at critical positions inthe elements of the beam which serve to electricallyconnect and disconnect the outer sectionsof the elements as the frequency of excitationof the an.tenna is changed. A typical"three-band" element is shown in figure 22.At the lowest operating frequency, the tunedtraps exert a minimum influence upon the elementand it resonates at a frequency determinedby the electrical length of the configuration,plus a slight degree of loading contributed bythe traps. At some higher frequency (generallyabout 1.5 times the lowest operating frequency)the outer set of traps are in a parallel resonantcondition, placing a high impedance betweenthe element and the tips beyond the traps.Thus, the element resonates at a frequency1.5 times higher than that determined by theoverall length of the element. As the frequencyof operation is raised to approximately 2.0times the lowest operating frequency, the innerset of traps become resonant, effectively disconnectinga larger portion of the element fromthe driven section. The length of the centersection is resonant at the highest frequencyof operation. The center section, plus the twoadjacent inner sections are resonant at theintermediate frequency of operation, and thecomplete element is resonant at the lowestfrequency of operation.The efficiency of such a system is determinedby the accuracy of tuning of both theelement sections and the isolating traps. Inaddition the combined dielectric losses of thetraps affect the overall antenna efficiency.As with all multi-purpose devices, some compromisebetween operating convenience andefficiency must be made with antennas designedto operate over more than one narrow band offrequencies. It is a tribute to the designers ofthe better multi -band beams that they performas well as they do, taking into account thetheoretical difficulties that must be overcome.,/\ISOLATIN~ TRAPS~~~~JLll ~RESiNANT_.fFEEDPOINTI ~jAT HI(;HES;FREQUENCYRESONANT t ATINTERMEDIATE FREQUENCYRESONANT t AT LOWEST FREQUENCYFigure 22TRAP-TYPE "THREE BAND"ELEMENTIsolating traps permit dipole to beself-resonant at three widely differentfrequencies.
CHAPTER TWENTY-SIXMobile EquipmentDesign and InstallationMobile operation is permitted on all amateurbands. Tremendous impetus to this phase ofthe hobby was given by the suitable design ofcompact mobile equipment. Complete mobileinstallations may be purchased as packagedunits, or the whole mobile station may be homebuilt, according to the whim of the operator.The problems involved in achieving a satisfactorytwo-way installation vary somewhat-with the band, but many of the problems arecommon to all bands. For instance, ignitionnoise is more troublesome on 10 meters thanon 75 meters, but on the other hand an efficientantenna system is much more easily accomplishedon 10 meters than on 75 meters.Also, obtaining a worthwhile amount of trans·mitter output without excessive battery drainis a problem on all bands.26-1 Mobile ReceptionWhen a broadcast receiver is in the car, themost practical receiving arrangement involvesa converter feeding into the auto set. The ad·vantages of good selectivity with good imagerejection obtainable from a double conversionsuperheterodyne are achieved in most caseswithout excessive "birdie" troubles, a com·515mon difficulty with a double conversion super·heterodyne constructed as an integral receiverin one cabinet. However, it is important thatthe b·c receiver employ an r-f stage in order toprovide adequate isolation between the con·verter and the high frequency oscillator in theb·c receiver. The r·f stage also is desirablefrom the standpoint of image rejection if theconverter does not employ a tuned output cir·cuit (tuned to the frequency of the auto set,usually about 1500 kc.). A few of the latemodel auto receivers, even in the better makes,do not employ an r·f stage.The usual procedure is to obtain converterplate voltage from the auto receiver. Experi·ence has shown that if the converter does notdraw more than about 15 or at most 20 rna. tot·al plate current no damage to the auto set orloss in performance will occur other than aslight reduction in vibrator life. The converterdrain can be minimized by avoiding a voltageregulator tube on the converter h·f oscillator.On 10 meters and lower frequencies it is pos·sible to design an oscillator with sufficientstability so that no voltage regulator is re·quired in the converter.With some cars satisfactory 75-meter opera·cion can be obtained without a noise clipperif resistor type spark plugs (such as thosemade by Autolite) are employed. However, anoise clipper is helpful if not absolutely neces·
516 Mobile Equipment<strong>THE</strong> R AD I 0sary, and it is recommended that a noise clipperbe installed without confirming the necessitytherefor. It has been found that quiet receptionsometimes may be obtained on 75 meterssimply by the use of resistor type plugs,but after a few thousand miles these plugsoften become less effective and no longer doa fully adequate job. Also, a noise clipper insuresagainst ignition noise from passingtrucks and "un-suppressed" cars. On 10 metersa noise clipper is a "must" in any case.Modifying theAuto ReceiverThere are certain things thatshould be done to the auto setwhen it is to be used with aconverter, and they might as well be done allat the same time, because "dropping" an autoreceiver and getting into the chassis to workon it takes quite a little time.First, however, check the circuit of the autoreceiver to see whether it is one of the fewreceivers which employ circuits which complicateconnection of a noise clipper or a converter.If the receiver is yet to be purchased,it is well to investigate these points ahead oftime.If the receiver uses a negative B resistorstrip for bias (as evidenced by the cathode ofthe audio output stage being grounded), thenthe additional plate current drain of the converterwill upset the bias voltages on the variousstages and probably cause trouble. Becausethe converter is not on all the time, itis not practical simply to alter the resistanceof the bias strip, and major modification of thereceiver probably will be required.The best type of receiver for attachment ofa converter and noise clipper uses an r-f stage;permeability tuning; single unit construction(except possibly for the speaker); push buttontuning rather than a tuning motor; a high vacuumrectifier such as a 6X4 (rather than an OZ4or a synchronous rectifier); a 6SQ7 (or miniatureor Loctal equivalent) with grounded cathodeas second detector, first audio, and a.v.c.;power supply negative grounded directly (nocommon bias strip); a PM speaker (to minimizebattery drain); and an internal r-f gain control(indicating plenty of built-in reserve gainwhich may be called upon if necessary). Manycurrent model auto radios have all of the foregoingfeatures, and numerous models have mostof them, something to keep in mind if the setis yet to be purchased.Noise Limiters A noise limiter either may bebuilt into the set or purchasedas a commercially manufactured unit for "outboard"connection via shielded wires. If thereceiver employs a 6SQ7 (or Loctal or miniatureequivalent) in a conventional circuit, it isa simple rna tter to build in a noise clipper byFigure ISERIES-GATE NOISE LIMITER FOR AUTORECEIVERAuto receivers using a 6SQ7, 786, 7X7, or6AT6 as seconcl c/etector one/ a.v.c, eon beconvertecl to the above circuit with but fewwiring changes. The circuit has the aclvantageof not requiring an aclclitional tube socketfor the limiter cliocle.substituting a 6S8 octal, 7X7 Loctal, or a 6T89-pin miniature as shown in figure 1. Whensubstituting a 6T8 for a 6AT6 or similar 7-pinminiature, the socket must be changed to a9-pin miniature type. This requires reamingthe socket hole slightly.If the receiver employs cathode bias on the6SQ7 (or equivalent), and perhaps delayeda. v.c., the circuit usually can be changed tothe grounded-cathode circuit of figure 1 withoutencountering trouble.Some receivers take the r-f excitation forthe a-v-e diode from the plate of the i-f stage.In this case, leave the a.v.c. alone and ignorethe a-v-e buss connection shown in figure 1(eliminating the 1-megohm decoupling resistor).If the set uses a separate a-v-e diode whichreceives r-f excitation via a small capacitorconnected to the detector diode, then simplychange the circuit to correspond to figure 1.In case anyone might be considering the useof a crystal diode as a noise limiter in conjunctionwith the tube already in the set, itmight be well to point out that crystal diodesperform quite poorly in series-gate noise clippersof the type shown.It will be observed that no tone control is
<strong>HANDBOOK</strong>Mobile Receivers 517shown. Multi-position tone controls tied inwith the second detector circuit often permitexcessive "leak through." Hence it is recommendedthat the tone control components becompletely removed unless they are confinedto the grid of the a-f output stage. If removed,the highs can be attenuated any desired amountby connecting a mica capacitor from plate toscreen on the output stage. Ordinarily from.005 to .01 p.fd. will provide a good compromisebetween fidelity and reduction of backgroundhiss on weak signals.Usually the switch SW will have to bemounted some distance from the noise limitercomponents. If the leads to the switch are overapproximately 1\1 inches long, a piece ofshield braid should be slipped over them andgrounded. The same applies to the "hot" leadsto the volume control if not already shielded.Closing the switch disables the limiter. Thismay be desirable for reducing distortion onbroadcast reception or when checking the intensityof ignition noise to determine the effectivenessof suppression measures taken onthe car. The switch also permits one to checkthe effectiveness of the noise clipper.The 22,000-ohm decoupling resistor at thebottom end of the i-f transformer secondary isnot critical, and if some other value alreadyis incorporated inside the shield can it may beleft a 1 one so long as it is not over 47,000ohms, a common value. Higher values must bereplaced with a lower value even if it requiresa can opener, because anything over 47,000ohms will result in excessive loss in gain.There is some loss in a-f gain inherent in thistype of limiter anyhow (slightly over 6 db),and it is important to minimize any additionalloss.It is important that the total amount of capacitancein the RC decoupling (r-f) filter notexceed about 100 p.p.fd. With a value muchgreater than this "pulse stretching" will occurand the effectiveness of the noise clipper willbe reduced. Excessive capacitance will reducethe amplitude and increase the duration of theignition pulses before they reach the clipper.The reduction in pulse amplitude accomplishesno good since the pulses are fed to the clipperanyhow, but the greater duration of the length·ened pulses increases the audibility and theblanking interval associated with each pulse.If a shielded wire to an external clipper is employed,the r-f by-pass on the "low" side ofthe RC filter may be eliminated since the capacitanceof a few feet of shielded wire willaccomplish the same result as the by-passcapacitor.The switch SW is connected in such a man·ner that there is practically no change in gainwith the limiter in or out. If the auto set doesnot have any reserve gain and more gain isneeded on weak broadcast signals, the switchcan be connected from the hot side of the volumecontrol to the junction of the 22,000,270,000 and 1 megohm resistors instead of asshown. This will provide approximately 6 dbmore gain when the clipper is switched out.Many late model receivers are provided withan internal r-f gain control in the cathode ofthe r-f and/ or i-f stage. This control shouldbe advanced full on to provide better noiselimiter action and make up for the loss in audiogain introduced by the noise clipper.Installation of the noise clipper often detunesthe secondary of the last i-f transformer.This should be repeaked before the set is per·manently replaced in the car unless the trimmeris accessible with the set mounted inplace.Additional clipper circuits will be found inthe receiver chapter of this Handbook.Selectivity While not of serious concern on10 meters, the lack of selectivityexhibited by a typical auto receiver will resultin QRM difficulty on 20 and 75 meters. A typicalauto set has only two i-f transformers ofrelatively low-Q design, and the second oneis loaded by the diode detector. The skirt sele.ctivityoften is so poor that a strong localwlll depress the a. v.c. when listening to aweak station as much as 15 kc. different infrequency.One solution is to add an outboard i-f stageemploying two good quality double-tuned transformers(not the midget variety) connected"back-to-back" through a small coupling capacitance.The amplifier tube (such as a 6BA6)should be biased to the point where the gainof the outboard unit is relatively small (1 or2), assuming that the receiver already has adequategain. If additional gain is needed, it maybe provided by the outboard unit. Low-capacitanceshielded cable should be used to coupleinto and out of the outboard unit, and the unititself should be thoroughly shielded.Such an outboard unit will shatpen the noseselectivity slightly and the skirt selectivitygreatly. Operation then will be comparable toa home-station communications receiver,though selectivity will not be as good as areceiver employing a 50-kc. or 85-kc. "Q5' er."Obtaining Power While the set is on thefor the Converter bench for installation ofthe noise clipper, provisionshould be made for obtaining filament andplate voltage for the converter, and for the exciterand speech amplifier of the transmitterif such an arrangement is to be used. To per:mit removal of either the converter or the autoset from the car without removing the other, aconnector should be provided. The best method
518 Mobile Equipment<strong>THE</strong> R A 0 I 0B+ TO RESTOF SET.(OPTIONAL)eyovcr------------,I 10HY. I1s+~POWERI, 181 REtE~e~R_L v "' - ...1. ~:tF.rfN~ 0 ~PEEcH1.UF er STAGESI ~ II -= IL ____ Jt!1TB ____ _jFigure 2B+ TOCONVERTERREGULAR""'-'-R-CFILTER~O.UFB+ 200 TO 250 V.TO XMTR.6 V. VIA CONTROLRELAY IN XMTR.USING <strong>THE</strong> RECEIVER PLATE SUPPLYFOR <strong>THE</strong> TRANSMITTERThis circuit silences the receiver on trans•mit, and in addition makes it possible to usethe receiver plate supply for feeding the ex•clter and speech amplifier stages In thetransmitter.is to mount a small receptacle on the receivercabinet or chassis, making connection via amatching plug. An Amphenol type 77-26 recep·tacle is compact enough to fit in a very smallspace and allows four connections (includingground for the shield braid). The matching plugis a type 70-26.To avoid the possibility of vibrator hashbeing fed into the converter via the heater andplate voltage supply leads, it is important thatthe heater and plate voltages be taken frompoints well removed from the power supply por·tion of the auto receiver. If a single-endedaudio output stage is employed, a safe placeto obtain these voltages is at this rube socket,the high voltage for the converter being takenfrom the screen. In the case of a push·pull out·pur stage, however, the screens sometimes arefed from the input side of the power supplyfilter. The ripple at this point, while sufficientlylow for a push-pull audio output stage,is nor adequate for a converter without addi·tional filtering. If the schematic shows thatthe screens of a push-pull stage are connectedto the input side instead of the output side ofthe power supply filter (usually two electroly·tics straddling a resistor in an R-C filter), thenfollow the output of the filter over into the r-fportion of the set and pick it up there at a convenientpoint, before it goes through any addi·tional series dropping or isolating resistors,as shown in figure 2.The voltage at the output of the filter usuallyruns from 200 to 250 volts wi rh typical con·verter drain and the motor not running. Thiswill increase perhaps 10 per cent when thegenerator is charging. The converter drain willdrop the B voltage slightly at the output of thefilter, perhaps 15 to 25 volts, bur this reduc·rion is nor enough to have a noticeable effectupon the operation of the receiver. If the Bvoltage is higher than desirable or necessaryfor proper operation of the converter, a 2-wattcarbon resistor of suitable resistance shouldbe inserted in series with the plate voltagelead to the power receptacle. Usually somethingbetween 2200 and 4700 ohms will befound about right.Receiver Disablingon TransmitWhen the battery drain ishigh on transmit, as is thecase when a PE·103A dynamotoris run at maximum rating and otherdrains such as the transmitter heaters and autoheadlights must be considered, it is desirableto disable the vibrator power supply in the re·ceiver during transmissions. The vibratorpower supply usually draws several amperes,and as the receiver must be disabled in somemanner anyhow during transmissions, openingthe 6-volr supply to the vibrator serves bothpurposes. It has the further advantage of intro·ducing a slight delay in the receiver recovery,due to the inertia of the power supply filter,thus avoiding the possibility of a feedback"yoop" when switching from transmit to receive.To avoid troubles from vibrator hash, it isbest to open the ground lead from the vibratorby means of a midget s.p.d.t. 6-volt relay andrhus isolate the vibrator circuit from the externalcontrol and switching circuit wires. Therelay is hooked up as shown in figure 3, Stand·ard 8-ampere contacts will be adequate forthis application.The relay should be mounted as close tothe vibrator as practicable. Ground one of thecoil terminals and run a shielded wire fromthe other coil terminal to one of the power re•ceptacle connections, grounding the shield atboth ends. By-pass each end of this wire toground with .01 11fd., using the shortest pos·sible leads. A lead is run from the correspond·ing terminal on the mating plug to the controlcircuits, to be discussed later.
<strong>HANDBOOK</strong> Mobile Receiver Installation 519Figure 3METHOD OF ELIMINATING <strong>THE</strong> BATTERYDRAIN OF <strong>THE</strong> RECEIVER VIBRATORPACK DURING TRANSMISSIONIf the reeelver ehassis has room for a mlcl·get s.p. c/. t. ref ay, the abave arrangement notonly s//enees the reeelver on transmit butsaves several amperes battery c/rain.If the normally open contact on the relay isconnected to the hot side of the voice coilwinding as shown in figure 3 (assuming oneside of the voice coil is grounded in accord·ance with usual practice), the receiver willbe killed instantly when switching from re·ceive to transmit, in spite of the fact that thepower supply filter in the receiver takes amoment to discharge. However, if a "slowstart" power supply (such as a dynamotor ora vibrator pack with a large filter) is used withthe transmitter, shorting the voice coil prob·ably will not be required.Using the ReeeiverPlate SupplyOn TransmitAn alternative and high·ly recommended proce·dure is to make use ofthe receiver B supply ontransmit, instead of disabling it. One dis ad·vantage of the popular PE-103A dynamotor isthe fact that its 450-500 volt output is too highfor the low power r·f and speech stages of thetransmitter. Dropping this voltage to a moresuitable value of approximately 250 volts bymeans of dropping resistors is wasteful ofpower, besides causing the plate voltage onthe oscillator and any buffer stages to varywidely with tuning. By means of a midget 6·volts.p.d.t. relay mounted in the receiver, con·nected as shown in figure 2, the B supply ofthe auto set is used to power the oscillatorand other low power stages (and possiblyscreen voltage on the modulator). On transmitthe B voltage is removed from the receiverand converter, automatically silencing the re·ceiver. When switching to receive the trans·miner oscillator is killed instantly, thus avoid·ing trouble from dynamotor "carry over."The efficiency of this arrangement is goodbecause the current drain on the main highvoltage supply for the modulated amplifier andmodulator plate (s) is reduced by the amountof current borrowed from the receiver. At least80 rna. can be drawn from practically all autosets, at least for a short period, without dam·age.It will be noted that with the arrangement offigure 2, plate voltage is supplied to the audiooutput stage at all times. However, when thescreen voltage is removed, the plate currentdrops practical! y to zero.The 200-ohm resistor in series with the normallyopen contact is to prevent excessivesparking when the contacts close. If the relayfeeds directly into a filter choke or large ca·pacitor there will be excessive sparking atthe contacts. Even with the arrangement shown,there will be considerable sparking at the con·tacts; but relay contacts can stand such spark·ing quite a while, even on d.c., before becom·ing worn or pitted enough to require attention.The 200-ohm resistor also serves to increasethe effectiveness of the .0 l·pfd. r·f by-passcapacitor.Auxiliary AntennaTrimmerOne other modification ofthe auto receiver which mayor may not be desirable de·pending upon the circumstances is the addi·tion of an auxiliary antenna trimmer capacitor.If the converter uses an untuned output circuitand the antenna trimmer on the auto set ispeaked with the converter cut in, then it isquite likely that the trimmer adjustment willnot be optimum for broadcast-band receptionwhen the converter is cut out. For receptionof strong broadcast band signals this usuallywill not be serious, but where reception ofweak broadcast signals is desired the loss ingain often cannot be tolerated, especially inview of the fact that the additional length ofantenna cable required for the converter in·stallation tends to reduce the strength ofbroadcast band signals.If the converter has considerable reservegain, it may be practicable to peak the antennatrimmer on the auto set for broadcast-band re·ception rather than resonating it to the con·verter output circuit. But oftentimes this re·suit!! in insufficient converter gain, excessiveimage troubles from loud local amateur sta·tions, or both.The difficulty can be circumvented by in·corporation of an auxiliary antenna trimmerconnected from the "hot" antenna lead on theauto r e c e i v e r to ground, with a switch inseries for cutting it in or out. This capacitorand switch can be connected across either the
520 Mobile Equipment<strong>THE</strong> R AD I 0converter end or the set end of the cable betweenthe converter and receiver. This auxiliarytrimmer should have a range of about 3to 50 p.p.fd., and may be of the inexpensivecompression mica type.With the trimmer cut out and the converterturned off (by-passed by the "in-out" switch),peak the regular antenna trimmer on the autoset at about 1400 kc. Then turn on the convert·er, with the receiver tuned to 1500 kc., switchin the auxiliary trimmer, and peak this trimmerfor maximum background noise. The auxiliarytrimmer then can be left switched in at alltimes except when receiving very weak broadcastband signals.Some auto sets, particularly certain GeneralMotors custom receivers, employ a high-Q highimpedanceinput circuit which is very criticalas to antenna capacitance. Unless the shuntcapacitance of the antenna (including cable)approximates that of the antenna installationfor which the set was designed, the antennatrimmer on the auto set cannot be made to hitresonance with the converter cut out. This isparticularly true when a long antenna cable isused to reach a whip mounted at the rear of thecar. Usually the condition can be corrected byunsoldering the internal connection to the antennaterminal connector on the auto set andinserting in series a 100-p.p.fd. mica capacitor.Alternatively an adjustable trimmer coveringat least 50 to 150 p.p.fd. may be substituted forthe 100-p.p.fd. fixed capacitor. Then the adjust·ment of this trimmer and that of the regular antennatrimmer can be juggled back and forthuntil a condition is achieved where the inputcircuit of the auto set is resonant with the convertereither in or out of the circuit. This willprovide maximum gain and image rejectionunder all conditions of use.Reducing BatteryDrain of theReceiverWhen the receiving installationis used frequently, andparticularly when the receiveris u s e d w i t h t h e carparked, it is desirable to keep the batterydrain of the receiver-converter installation atan absolute minimum. A substantial reductionin drain can be made in many receivers, withoutappreciably affecting their performance.The saving of course depends upon the designof the particular receiver and upon howmuch trouble and expense one is willing to goto. Some receivers normally draw (without theconverter connected) as much as 10 amperes.In many cases this can be cut to about 5 amperesby incorporating all practicable modifications.Each of the following modificationsis applicable to many auto receivers.If the receiver uses a speaker with a fieldcoil, replace the speaker with an equivalentP\1 type.Practically all 0.3-ampere r-f and a-f voltageamplifier tubes have 0.15-ampere equiva·lents. In many cases it is not even necessaryto change the socket wiring. However, whensubstituting i-f tubes it is recommended thatthe i-f trimmer adjustments be checked. Generallyspeaking it is not wise to attempt tosubstitute for the converter tube or a-f poweroutput tube.If the a-f output tube employs conventionalcathode bias, substitute a cathode resistor oftwice the value originally employed, or addan identical resistor in series with the onealready in the set. This will reduce the Bdrain of the receiver appreciably without ser·iously reducing the maximum undistorted output.Because the vibrator power supply is muchless than 100 per cent efficient, a saving ofone watt of B drain results in a saving of nearly2 watts of battery drain. This also mini·mizes the overload on the B supply when theconverter is switched in, assuming that theconverter uses B voltage from the auto set.If the receiver uses push-pull output and ifone is willing to accept a slight reduction inthe maximum volume obtainable without distortion,changing over to a single ended stageis simple if the receiver employs conventionalcathode bias. Just pull out one tube, doublethe value of cathode bias resistance, and adda 25-p.fd. by-pass capacitor across the cathoderesistor if not already by-passed. In somecases it may be possible to remove a phaseinverter tube along with one of the a-f outputtubes.If the receiver uses a motor driven stationselector with a control tube (d·c amplifier),usually the tube can be removed without upsettingthe operation of the receiver. One thenmust of course use manual tuning.While the changeover is somewhat expen·sive, the 0.6 ampere drawn by a 6X4 or 6X5rectifier can be eliminated by substituting sixUS-volt r·m·s 50-rna. selenium rectifiers (suchas Federal type 402D3200). Three in seriesare substituted for each half of the full-waverectifier tube. Be sure to observe the correctpolarity. The selenium rectifiers also make agood substitution for an OZ4 or OZ4·GT whichis causing hash difficulties when using theconverter.Offsetting the total cost of nearly $4.00 isthe fact that these rectifiers probably will lastfor the entire life of the auto set. Before pur·chasing the rectifiers, make sure that there isroom available for mounting them. While theseunits are small, most of the newer auto setsemploy very compact construction.Two-Meter Reception For reception on the 144-Mc. amateur band, andthose higher in frequency, the simple converter·
<strong>HANDBOOK</strong>One-Tube Converter 521auto-set combination has not proven very satisfactory.The primary reason for this is the factthat the relatively sharp i-f channel of the autoset imposes too severe a limitation on the stabilityof the high-frequency oscillator in theconverter. And if a crystal-controlled beatingoscillator is used in the converter, only a portionof the band may be covered by tuning theauto set.The most satisfactory arrangement has beenfound to consist of a separately mounted i.f.,audio, and power supply system, with the convertermounted near the steering column. Thei-f system should have a bandwidth of 30 to100 kc. and may have a center frequency of10.7 Me. if standard i-f transformers are to beused. The control head may include the 144-Mc. r-f, mixer, and oscillator sections, andsome times the first i-f stage. Alternatively,the control head may include only the h-f oscillator,with a broadband r-f unit includedwithin the main receiver assembly along withthe i.f. and audio system. Commercially manufacturedkits and complete units using thisgeneral lineup are available.An alternative arrangement is to build aconverter, 10. 7-Mc. i-f channel, and seconddetector unit, and then to operate this unit inconjunction with the auto-set power supply,audio system, and speaker. Such a systemmakes economical use of space and powerdrain, and can be switched to provide normalbroadcast-band auto reception or receptionthrough a converter for the h-f amateur bands.A recent development has been the VHFtransceiver, typified by the Gonset Communicator.Such a unit combines a crystal controlledtransmitter and a tunable VHF receivertogether with a common audio system andpower supply. The complete VHF station maybe packaged in a single cabinet. Variousforms of VHF transceivers are shown in theconstruction chapters of this Handbook.tion. A total transmitter power drain of about80 watts from the car battery (6 volts at 13amperes, or 12 volts at 7 amperes) is aboutthe maximum that can be allowed under theseconditions. For maximum power efficiency itis recommended that a vibrator type of supplybe used as opposed to a dynamotor supply,since the conversion efficiency of the vibratorunit is high compared to that of the dynamotor.A second school of thought states that themobile transmitter should be of relatively highpower to overcome the poor efficiency of theusual mobile whip antenna. In this case, themobile power should be drawn from a systemthat is independent from the electrical systemof the automobile. A belt driven high voltagegenerator is often coupled to the automobileengine in this type of installation.A variation of this idea is to employa complete secondary power system in thecar capable of providing 115 volts a.c. Shownin figure 4 is a Leece-N eville three phasealternator mounted atop the engine block, anddriven with a fan belt. The voltage regulatorand selenium rectifier for charging the carbattery from the a•c system replace the usuald-e generator. These new items are mountedin the front of the car radiator. The alternatorprovides a balanced delta output circuit whereinthe line voltage is equal to the coil voltage,but the line current is y3 times the coilcurrent. The coil voltage is a nominal 6-volts,RMS and three 6.3 volt 25 ampere filamenttransformers may be connected in delta onthe primary and secondary windings to stepthe 6-volts up to three-phase 115-volts. Ifdesired, a special 115-volt, 3-phase step-uptransformer may be wound which will occupyless space than the three f i 1 amen t transformers.Since the ripple frequency of a threephase d-e power supply will be quite high, asingle 10 mfd filter capacitor will suffice.26-2 Mobile TransmittersAs in the case of transmitters for fixed-stationoperation, there are many schools ofthought as to the type of transmitter which ismost suitable for mobile operation. One schoolstates that the mobile transmitter should havevery low power drain, so that no modificationof the electrical system of the automobile willbe required, and so that the equipment may beoperated without serious regard to dischargingthe battery when the car is stopped, or overloadingthe generator when the car is in mo-ALTERN A TOR IS ENGINE DRIVENBY AUXILIARY FAN BELT.
522 Mobile Equipment<strong>THE</strong> R AD I 010-i FTlFigure 65/16-WAVE WHIP RADIATOR FOR 10METERSIf a whip antenna is made slightly longerthan one-quarter wave it acts as a slightlybetter radiator than the usual quarter•wavewhip, and it can provide a better match tothe antenna transmission line If tht> react•ance Is tuned out by a series capacitor closeto the base of the antenna. Capacitor C 1 maybe a 100-J.LJ.Lfd. midget variable.Figure 5A CENTER LOADED 80-METER WHIPUSING AIR WOUND COIL MAY BE USEDWITH HIGH POWERED TRANSMITTERSAn anti-corona loop is placed at the topof the whip to reduce loss of power andburning of tip of antenna. Number of turnsin coil is critical and adjustable, high-Qcoil is refommended. Whip may be usedover frequency range of about 15 kilocycleswithout retuning.26-310-Meter MobileAntennasAntennas forMobile WorkThe most popular mobile antennafor 10-meter operationis a rear-mounted whip approx·imately 8 feet long, fed with coaxial line. Thisis a highly satisfactory antenna, but a fewremarks are in order on the subject of feedand coupling systems.The feed point resistance of a resonant quar·ter·wave rear-mounted whip is approximately20 to 25 ohms. While the standing-wave ratiowhen using 50-ohm coaxial line will not bemuch greater than 2 to 1, it is neverthelessdesirable to make the line to the transmitterexactly one quarter wavelength long electri·cally at the center of the band. This procedurewill minimize variations in loading over theband. The physical length of RG-8/U cable,from antenna base to antenna coupling coil,should be approximately 5 feet 3 inches. Theantenna changeover relay preferably should belocated either at the antenna end or the trans·mitter end of the line, but if it is more con·venient physically the line may be broken any·where for insertion of the relay.If the same rear-mounted whip is used forbroadcast-band reception, attenuation of broad·cast-band signals by the high shunt capaci·tance of the low impedance feed line can bereduced by locating the changeover relay rightat the antenna lead in, and by running 95-ohmcoax (instead of 50 or 75 ohm coax) from therelay to the converter. Ordinarily this w~ll pro·duce negligible effect upon the operation ofthe converter, but usually will make a worth·while improvement in the strength of broadcast·band signals.
<strong>HANDBOOK</strong>Mobile Antennas 523A more effective radiator and a better linerna tch rna y be obtained by making the whipapproximately 10 )i feet long and feeding itwith 75-ohm coax (such as RG-11/U) via aseries capacitor, as shown in figure 6. Therelay and series capacitor are mounted insidethe trunk, as close to the antenna feedthroughor base-mount insulator as possible. The 10 )ifootlength applies to the overall length fromthe tip of the whip to the point where the leadin passes through the car body. The leads insidethe car (connecting the coaxial cable,relay, series capacitor and antenna lead)should be as short as possible. The outer conductorof both coaxial cables should be groundedto the car body at the relay end with short,heavy conductors.A 100-11p.fd. midget variable capacitor issuitable for C 1• The optimum setting shouldbe determined experimentally at the center ofthe band. This setting then will be satisfactoryover the whole band.One suitable coupling arrangement for eithera Y.\-wave or 5/16-wave whip on 10 meters isto use a conventional tank circuit, inductivelycoupled to a "variable link" coupling loopwhich feeds the coaxial line. Alternatively, api-network output circuit may be used. If theinput impedance of the line is very low andthe tank circuit has a low C/L ratio, it maybe necessary to resonate the coupling loopwith series capacitance in order to obtain sufficientcoupling. This condition often is encounteredwith a Y.\-wave whip when the linelength approximates an electrical half wavelength.If an all-band center-loaded mobile antennais used, the loading coil at the center of theantenna may be shorted out for operation ofthe antenna on the 10-meter band. The usualtype of center-loaded mobile antenna will bebetween 9 and 11 feet long, including the center-loadinginductance which is shorted out.Hence such an antenna may be shortened toan electrical quarter-wave for the 10-meterband by using a series capacitor as just discussed.Alternatively, if a pi-network is usedin the plate circuit of the output stage of themobile transmitter, any reactance presentedat the antenna terminals of the transmitter bythe antenna may be tuned out with the pi-network.The All-BandCenter-LoadedMobile AntennaThe great majority of mobileoperation on the 14-Mc. bandand below is with centerloaded whip antennas. Theseantennas use an insulated bumper or bodymount, with provision for coaxial feed fromthe base of the antenna to the transmitter, asshown in figure 7.The center-loaded whip antenna must beCAR BODYUNSHIELDEDLOADING COILRG-58/U LINETO TRANSMITTER------COAXIAL LINEGROUNDED TOFRAME OF CARADJACENT TO BASEOF ANTENNAFigure 7<strong>THE</strong> CENTER-LOADED WHIPANTENNAThe center-loaclecl whip antenna, when provic/eelwith a tapped loacllng coil or a seriesof coils, may be usee/ over a wicle frequencyrange. The loacling coil may be shortecl foruse of the antenna on the 10-meter bane/.tuned to obtain optimum operation on the desiredfrequency of operation. These antennaswill operate at maximum efficiency over arange of perhaps 20 kc. on the 75-meter band,covering a somewhat wider range on the 40-meter band, and covering the whole 20-me terphone band. The procedure for tuning the antennasis discussed in the instruction sheetwhich is furnished with them, but basicallythe procedure is as follows:The antenna is installed, fully assembled,with a coaxial lead of RG-58/U from the baseof the antenna to the place where the transmitteris installed. The rear deck of the carshould be closed, and the car should be parkedin a location as clear as possible of trees,buildings, and overhead power lines. Objectswithin 15 or 20 feet of the antenna can exerta considerable detuning effect on the antennasystem due to its relatively high operating Q.The end of the coaxial cable which will pluginto the transmitter is terminated in a link of3 or 4 turns of wire. This link is then coupledto a grid-dip meter -and the resonant frequencyof the antenna determined by noting the frequencyat which the grid current fluctuates.The coils furnished with the antennas normallyare too large for the usual operating frequency,since it is much easier to removeturns than to add them. Turns then are removed,one at a time, until the antenna resonates atthe desired frequency. If too many turns havebeen removed, a length of wire may be splicedon and soldered. Then, with a length of insu·lating tubing slipped over the soldered joint,turns may be added to lower the resonant fre-
524 Mobile Equipment<strong>THE</strong> R AD I 0TOANT.26-4Construction andInstallation of MobileEquipmentFigure 8PI-NETWORK ANTENNA COUPLERThe pi-network antenna coupler is partlcu•larly satisfactory for mobile work since thecoupler affords some degree of harmonic reduction,provides a coupling variation tomeet varying load conditions caused by frequencychanges, and can cancel aut reactancepresentee/ to the transmitter at the endof the antenna transmission line.For use of the coupler on the 3.9-Mc. bane/C 1 should have a maximum capacitance ofabout 250 !1-f.Lfc/., L1 should be about 9 mlcrohenrys(30 turns 1" ella. by 2" long}, andC2 may include a fixed and a variable elementwith maximum capacitance of about 1400f.Lf.Lfc/. A 100-!1-f.Lfc/. variable capacitor will besuitable at C1 for the 14-Mc. and 28-Mc.bands, with a 350-!1-f.Lfc/. variable at c2. InductorL1 should have an Inductance of a•bout 2 mlcrohenrys (11 turns 1" ella. by 1 11long} for the 14-Mc. bane/, and about 0.8 mi•crohenry (6 turns 1" clio. by 1" long} for the28-Mc. band.quency. Or, if the tapped type of coil is used,taps are changed until the proper number ofturns for the desired operating frequency isfound. This procedure is repeated for the differentbands of operation.Feeding theCenter-LoadedAntennaAfter much experimenting ithas been found that the mostsatisfactory method for feed-ing the coax i a l line to thecenter-loaded antenna is with thebase of api-network coupler. Figure 8 shows the basicarrangement, with recommended circuit constants.It will be noted that relatively largevalues of capacitance are required for all bandsof operation, with values which seem particularlylarge for the 75-meter band. But referenceto the discussion of pi-network tank circuitsin Chapter 13 will show that the valuessuggested are normal for the values of impedance,impedance transformation, and operatingQ which are encountered in a mobile installationof the usual type.It is recommended that the following measuresbe taken when constructing mobile equipment,either transmitting or receiving, to ensuretrouble-free operation over long periods:Use only a stiff, heavy chassis unless thechassis is quite small.Use lock washers or lock nuts when mountingcomponents by means of screws.Use stranded hook-up wire except where r-fconsiderations make it inadvisable (such asfor instance the plate tank circuit leads in av-h-f amplifier). Lace and tie leads wherevernecessary to keep them from vibrating or floppingaround.Unless provided with gear drive, tuning capacitorsin the large sizes will require a rotorlock.Filamentary (quick heating) tubes shouldbe mounted only in a vertical position.The larger size carbon resistors and micacapacitors should not be supported from tubesocket pins, particularly from miniature sockets.Use tie points and keep the resistor andcapacitor "pigtails" short.Generally speaking, rubber shock mountsare unnecessary or even undesirable with passengercar installations, or at least with fullsize passenger cars. The springing is sufficiend y "soft" t h at well constructed radioequipment can be bolted directly to the vehiclewithout damage from shock or vibration. Unlessshock mounting is properly engineeredas to the stiffness and placement of the shockmounts, mechanical-resonance "amplification"effects may actually cause the equipment tobe shaken more than if the equipment werebolted directly to the vehicle.Surplus military equipment provided withshock or vibration mounts was intended foruse in aircraft, jeeps, tanks, gun-firing Navalcraft, small boats, and similar vehicles andcraft subject to severe shock and vibration.Also, the shock mounting of such equipmentis very carefully engineered in order to avoidharmful resonances.To facilitate servicing of mobile equipment,all interconnecting cables between unitsshould be provided with separable connectorson at least one end.Control Circuits The send-receive control circuitsof a mobile installationare dictated by the design of the equipment,and therefore will be left to the ingenuity ofthe reader. However, a few generalizationsand suggestions are in order.
<strong>HANDBOOK</strong>Control Circuits525Do not attempt to control too many relays,particularly heavy duty relays with large coils,by means of an ordinary push-to-talk switchon a microphone. These contacts are not designedfor heavy work, and the inductive kickwill cause more sparking than the contacts onthe microphone switch are designed to handle.It is better to actuate a single relay with thepush-to-talk switch and then control all otherrelays, including the heavy duty contactor forthe dynamotor or vibrator pack, with this relay.The procedure of operating only one relaydirecdy by the push-to-talk switch, with allother relays being controlled by this controlrelay, will eliminate the often-encountered difficultywhere the shutting down of one item ofequipment will close relays in other items asa result of the coils of relays being placed inseries with each other and with heater circuits.A recommended general control circuit, whereone side of the main control relay is connectedto the hot 6-volt circuit, but all other relayshave one side connected to ground, is illustratedin figure 9. An additional advantageof such a circuit is that only one control wireneed be run to the coil of each additional relay,the other side of the relay coils beinggrounded.The heavy-duty 6-volt solenoid-type contactorrelays such as provided on the PE-103Aand used for automobile starter relays usuallydraw from 1.5 to 2 amperes. While somewhatmore expensive, heavy-duty 6-volt relays ofconventional design, capable of breaking 30amperes at 6 v o 1 t s d. c., are available withcoils drawing less than 0.5 ampere.When purchasing relays keep in mind thatthe current rating of the contacts is not a fixedvalue, but depends upon (I) the voltage, (2)whether it is a.c. or d.c., and (3) whether thecircuit is purely resistive or is inductive. Ifin doubt, refer to the manufacturer's recommendations.Also keep in mind that a dynamotorpresents almost a dead short until the armaturestarts turning, and the starting relayshould be rated at considerably more than thenormal dynamotor current.Microphonesand CircuitsThe most generally used microphonefor mobile work is thesingle-button carbon. With ahigh-output-type microphone and a high-ratiomicrophone transformer, it is possible when"close talking" to drive even a pair of pushpull6L6's without resorting to a speech amplifier.However, there is a wide difference inthe output of the various type single buttonmicrophones, and a wide difference in the amount of step up obtained with different rypemicrophone transformers. So at least onespeech stage usually is desirable.One of the most satisfactory single buttonPUSH-TO-TALK PUSH-TO-TALKSWITCH ON MIKE RELAY~~·,g .. } ~ ~ALTERNATE MAIN POWER RECEIVER ANTENNA ANYCONTROLRELAY MUTING CHANGEOVER O<strong>THE</strong>RSWITCHRELAY RELAY RELAYSFigure 9RELAY CONTROL CIRCUITSimp/lfieel schematic of the recommeneleelrelay control circuit for mobile transmitters.The relatively small push-to-talk relay Iscontrol/eel by the button on the microphoneor the communications switch. Then one ofthe contacts on this relay controls the otherrelays of the transmitter; one s/ele of thecoil of all the aelelitional relays control/eelshoul el be grouneleel.microphones is the standard Western Electrictype F-1 unit (or Automatic Electric Co. equivalent).This microphone has very high outputwhen operated at 6 volts, and good fidelityon speech. When used without a speech amplifierstage the microphone transformer shouldhave a 50-ohm primary (rather than 200 or 500ohms) and a secondary of at 1 east 150,000ohms and preferably 250,000 ohms.The widely available surplus type T-17 microphonehas higher resistance (200 to 500ohms) and lower output, and usually wi 11 requirea stage of speech amplification exceptwhen used with a very low power modulatorstage.Unless an F -1 unit is used in a standardhousing, making contact to the button presentssomewhat of a problem. No serious damage willresult from soldering to the button if the connectionis made to one edge and the solderingis done very rapidly with but a small amount of solder, so as to avoid heating thewhole button.A sound-powered type microphone removedfrom one of the chest sets available in thesurplus market will deliver almost as muchvoltage to the grid of a modulator stage whenused with a high-ratio microphone transformeras will an F -1 unit, and has the advantage ofnot requiring button current or a "hash filter."This is simply a dynamic microphone designedfor high output rather than maximum fidelity.The standardized connections for a singlebuttoncarbon microphone provided with pushto-talkswitch are shown in figure 10. Practicallyall hand-held military-type single-button
526 Mobile Equipment<strong>THE</strong> R AD I 0~ ,------: ~:~G } OF MIKE~ PLUGPRESS-TO-TALKSWITCHf_f I I SHELL/ u (GROUND)Figure 10STANDARD CONNECTIONS FOR <strong>THE</strong>PUSH·TO·TALK SWITCH ON A HAND·HELD SINGLE-BUTTON CARBONMICROPHONEmicrophones on the surplus market use theseconnections.There is an increasing tendency among mo·bile operators toward the use of microphoneshaving better frequency and distortion char·acteristics than the standard single-buttontype. The high-impedance dynamic type isprobably the most popular, with the ceramic·crystal type next in popularity. The conven·tiona! crystal type is not suitable for mobileuse since the crystal unit will be destroyedby the high temperatures which can be reachedin a closed car parked in the sun in the sum·mer time.The use of low-level microphones in mobileservice requires careful attention to the elimi·nation of common·ground circuits in the micro·phone lead. The ground connection for theshielded cable which runs from the transmitterto the microphone should be made at only onepoint, preferably directly adjacent to the gridof the first tube in the speech amplifier. Theuse of a low-level microphone usually will re·quire the addition of two speech stages (a pen·tode and a triode), but these stages will takeonly a milliampere or two of plate current, and150 rna. per tube of heater current.PE-l03A Dyna·motor Power UnitBecause of its availabilityon the surplus market at alow price and its suitabilityfor use with about as powerful a mobile transmitteras can be employed in a passenger carwithout resorting to auxiliary batteries or aspecial generator, the PE·l03A is probablythe most widely used dynamotor for amateurwork. Therefore some useful information willbe given on this unit.The nominal rating of the unit is 500 voltsand 160 rna., but the output voltage will ofcourse vary with load and is slight! y higherwith the generator charging. Actually the 160rna. rating is conservative, and about 275 rna.can be drawn intermittently without overheat·ing, and without damage or excessive brushor commutator wear. At this current the unitshould not be run for more than 10 minutes ata time, and the average "on" time should notbe more than half the average 11 off" time.The output voltage vs. current drain isshown approximately in figure 11. The exactvoltage will depend somewhat upon the lossresistance of the primary connecting cableand whether or not the battery is on charge.The primary current drain of the dynamotorproper (excluding relays) is approximately 16amperes at 100 rna., 21 amperes at 160 rna.,26 amperes at 200 rna., and 31 amperes at 250rna.Only a few of the components in the baseare absolutely necessary in an amateur mobileinstallation, and some of them can just as wellbe made an integral part of the transmitter ifdesired. The base can be removed for salvagecomponents and hardware, or the dynamotormay be purchased without base.To remove the base proceed as follows:Loosen the four thumb screws on the baseplate and remove the cover. Remove the fourscrews h o 1 ding the dynamotor to the baseplate. Trace the four wires coming out of thedynamotor to their terminals and free the lugs.Then these four wires can be pulled throughthe two rubber grommets in the base plate whenthe dynamotor is separated from the base plate.It may be necessary to bend the eyelets in thelarge lugs in order to force them through thegrommets.Next remove the two end housings on thedynamotor. Each is held with two screws. Thehigh-voltage commutator is easily identifiedby its narrower segments and larger diameter.Next to it is the 12·volt commutator. The 6-volt commutator is at the other end of the armature.The 12-volt brushes should be removedwhen only 6-volt operation is planned, in orderto reduce the drag.If the dynamotor portion of the P E·lO 3Apower unit is a Pioneer type V5-25 or a Rus·sell type 530- (most of them are), the wires tothe 12-volt brush holder terminals can be crossconnected to the 6-volt brush holder terminalswith heavy jumper wires. One of the wires disconnectedfrom the 12-volt brush terminals isthe primary 12-volt pigtail and will come free.The other wire should be connected to the op·posite terminal to form one of the jumpers.With this arrangement it is necessary onlyto n;move the 6-volt brushes and replace the12-volt brushes in case the 6-volt commutatorbecomes excessively dirty or worn or startsthrowing solder. No difference in output volt·age will be noted, but as the 12-volt brushesare not as heavy as the 6-volt brushes it isnot permissible ~o draw more than about 150rna. except for emergency use until the 6-voltcommutator can be turned down or repaired.
<strong>HANDBOOK</strong>Noise Suppression 527w01-::>Q_1-::>0550500 ~450400100~........~~r-...PE-103A6 V. INPUT1--""150 200 250 300OUTPUT CURRENT, MA.Figure 11APPROXIMATE OUTPUT VOLTAGE VS.LOAD CURRENT FOR A PE-103ADYNAMOTORAt 150 rna. or less the 12-volt brushes willlast almost as long as the 6-volt brushes.The reason that these particular dynamotorscan be operated in this fashion is that thereare two 6-volt windings on the armature, andfor 12-volt operation the two are used in serieswith both commutators working. The arrange·ment described above simply substitutes forthe regular 6-v o It winding the winding andcommutator which ordinarily came into opera·tion only on 12-volt operation. Some operatorshave reported that the regulation of the P E·103A may be improved by operating both com·mutators in parallel with the 6-volt line.The three wires now coming out of the dy·namotor are identified as follows: The smallerwire is the positive high voltage. The heavywire leaving the same grommet is positive 6volts and negative high voltage. The singleheavy wire leaving the other grommet is nega·tive 6 volts. Whether the car is positive ornegative ground, negative high voltage can betaken as car-frame ground. With the negativeof the car battery grounded, the plate current· can return through the car battery and the ar·mature winding. This simply puts the 6 voltsin series with the 500 volts and gives 6 extravolts plate voltage.The trunk of a car gets very warm in sum·mer, and if the transmitter and dynamotor aremounted in the trunk it is recommended thatthe end housings be left off the dynamotor tofacilitate cooling. This is especially impor·tant in hot climates if the dynamotor is to beloaded to more than 200 rna.When replacing brushes on a PE·l03A checkto see if the brushes are marked negative andpositive. If so, be sure to install them accord·ingly, because they are not of the same mater·ial. The dynamotor will be marked to showwhich holder is negative.When using a PE·l03A, or any dynamotorfor that matter, it may be necessary to devoteone set of contacts on one of the control re·lays to breaking the plate or screen voltageto the transmitter oscillator, 1f these are sup·plied by the dynamotor, because the output ofa dynamotor takes a moment to fall to zerowhen the primary power is removed.26-5 Vehicular NoiseSuppressionSatisfactory reception on frequencies abovethe broadcast band usually requires greaterattention to noise suppression measures. Therequired measures vary with the particular ve·hicle and the frequency range involved.Most of the various types of noise that maybe present in a vehicle may be broken downinto the following main categories:( 1) Ignition noise.(2) Wheel static (tire static, brake static,and intermittent ground via front wheel bearings).(3) "Hash" from voltage regulator con·tacts.( 4) "Whine" from generator commutator seg·ment make and break.(5) Static from scraping connections betweenvarious parts of the car.There is no need to suppress ignition noisecompletely, because at the higher frequenciesignition noise from passing vehicles makesthe use of a noise limiter mandatory anyway.However, the limiter should not be given toomuch work to do, because at high enginespeeds a noisy ignition system will tend tomask weak signals, even though with the lim·iter working, ignition "pops" may appear tobe completely eliminated.Another reason for good ignition suppres·sion at the source is that strong ignitionpulses contain enough energy when integratedto block the a·v·c circuit of the receiver, caus·ing the gain to drop whenever the engine isspeeded up. Since the a-v·c circuits of thereceiver obtain no benefit from a noise clip·per, it is important that ignition noise be sup·pressed enough at the source that the a·v·ccircuits will not be affected even when theengine is running at high speed.Ignition Noise The following procedureshould be found adequate forreducing the ignition noise of practically anypassenger car to a level which the clipper canhandle satisfactorily at any engine speed atany frequency from 500 kc. to 148 Me. Some
528 Mobile Equipment<strong>THE</strong> R AD I 0of the measures may already have been takenwhen the auto receiver was installed.First either install a spark plug suppressoron each plug, or else substitute Autolite resistorplugs. The latter are more effective thansuppressors, and on some cars ignition noiseis reduced to a satisfactory level simply byinstalling them. However, they may not do anadequate job alone after they have been in usefor a while, and it is a good idea to take thefollowing additional measures.Check all high tension connections for gaps,particularly the "pinch fit" terminal connec•tors widely used. Replace old high tensionwiring that may have become leaky.Check to see if any of the high tension wiringis cabled with low tension wiring, or runin the same conduit. If so, reroute the low tensionwiring to provide as much separation aspracticable.By-pass to ground the 6-volt wire from theignition coil to the ignition switch at eachend with a 0.1-/Lfd. molded case paper capacitorin parallel with a .00 1-/Lfd. mica or ceramic,using the shortest possible leads.Check to see that the hood makes a goodground con t a c t to the car body at severalpoints. Special grounding contactors are availablefor attachment to the hood lacings on carsthat otherwise would present a groundingproblem.If the high-tension coil is mounted on thedash, it may be necessary to shield the hightension wire as far as the bulkhead, unlessit already is shielded with armored conduit.Wheel Static Wheel static is either staticelectricity generated by rotationof the tires and brake drums, or is noise generatedby poor contact bet we en the frontwheels and the axles (due to the grease in thebearings). The latter type of noise seldom iscaused by the rear wheels, but tire static mayof course be generated by all four tires.Wheel static can be eliminated by insertionof grounding springs under the front hub caps,and by inserting "tire powder" in all innertubes. Both items are available at radio partsstores and from most auto radio dealers.Voltage RegulatorHashCertain voltage regulatorsgenerate an objectionableamount of "hash" at thehigher frequencies, particularly in the v-h-frange. A large by-pass will affect the operationof the regulator and possibly damage thepoints. A small by-pass can be used, however,without causing trouble. At frequencies abovethe frequency at which the hash becomes objectionable(approximately 20 Me. or so) asmall by-pass is quite effective. A 0.001-/Lfd.mica capacitor placed from the field terminalof the regulator to ground with the shortestpossible leads often will produce sufficientimpcovement. If not, a choke consisting of about 60 turns of no. 18 d.c.c. or bell wirewound on a %-inch form can be added. Thisshould be placed right at the regulator terminal,and the 0.00 1-/Lfd. by-pass placed fromthe generator side of the choke to ground.Generator Whine Generator "whine" often canbe satisfactorily suppressedfrom 550 kc. to 148 Me. simply by by-passingthe armature terminal to ground with a special"auto radio" by-pass of 0.25 or 0.5 /Lfd. inparallel with a 0.001-/Lfd. mica or ceramic capacitor.The former usually is placed on thegenerator when an auto radio is installed, butmust be augmented by a mica or ceramic capacitorwith short leads in order to be effectiveat the higher frequencies as well as on thebroadcast band.When more drastic measures are required,special filters can be obtained which are designedfor the purpose. These are recommendedfor stubborn cases when a wide frequencyrange is involved. For reception only over acomparatively narrow band of frequencies,such as the 10-meter amateur band, a highlyeffective filter can be improvised by connectingbetween the previously described parallelby-pass capacitors and the generator armatureterminal a resonant choke. This may consistof no. 10 enamelled wire wound on a suitableform and shunted with an adjustable trimmercapacitor to permit resonating the combinationto the center of the frequency bandinvolved. For the 10-meter band 11 turns closewound on a one-inch form and shunted by a3-30 /L/Lfd. compression-type mica trimmer issuitable. The trimmer should be adjusted experimentallyat the center frequency.When generator whine shows up after oncebeing satisfactorily suppressed, the conditionof the brushes and corm~ utator should becheeked. U n 1 e s s a by-pass capacitor hasopened up, excessive whine usually indicatesthat the brushes or commutator are in need ofattention in order to prevent damage to thegenerator.Body Static Loose linkages or body or framejoints anywhere in the car arepotential static producers when the car is inmotion, particularly over a rough road. Locatingthe source of such noise is difficult, andthe simplest procedure is to give the car athorough tightening up in the hope that theoffending poor contacts will be caught by theprocedure. The use of braided bonding strapsbetween the various sections of the body ofthe car also may prove helpful.
<strong>HANDBOOK</strong>Noise Suppression 529Miscellaneous There are several other potentialnoise sources on a passengervehicle, but they do not necessarilygive trouble and therefore require attentiononly in some cases.The heat, oil pressure, and gas gauges cancause a rasping or scraping noise. The gasgauge is the most likely offender. It will causetrouble only when the car is rocked or is inmotion. The gauge units and panel indicatorsshould both be by-passed with the 0.1-fLfd.paper and 0.00 1-fLfd. mica or ceramic combinationpreviously described.At high car speeds under certain atmosphericconditions corona static may be encounteredunless means are taken to prevent it. The receiving-typeauto whips which employ a plasticball tip are so provided in order to minimizethis type of noise, which is simply a dischargeof the frictional static built up on the car. Awhip which ends in a relatively sharp metalpoint makes an ideal discharge point for thestatic charge, and will cause corona troubleat a much lower voltage than if the tip werehooded with insulation. A piece of Vinylitesleeving slipped over the top portion of thewhip and wrapped tightly with heavy threadwill prevent this type of static discharge underpractically all conditions. An alternativearrangement is to wrap the top portion of thewhip with S catch brand electrical tape.Generally speaking it is undesirable fromthe standpoint of engine performance to useboth spark-plug suppressors and a distributorsuppressor. Unless the distributor rotor clearanceis excessive, noise caused by sparkingof the distributor rotor will not be so bad butwhat it can be handled satisfactorily by anoise limiter. If not, it is preferable to shieldthe hot lead between ignition coil and distributorrather than use a distributor suppressor.In many cases the control rods, speedometercable, etc., will pick up high-tension noiseunder the hood and conduct it up under thedash where it causes trouble. If so, all controlrods and cables should be bonded to thefire wall (bulkhead) where they pass through,using a short piece of heavy flexible braid ofthe type used for shielding.In some cases it may be necessary to bondthe engine to the frame at each rubber enginemount in a similar manner. If a rear mountedwhip is employed the exhaust tail pipe alsoshould be bonded to the frame if supported byrubber mounts.LocatingNoise SourcesDetermining the source of certaintypes of noise is madedifficult when several thingsare contributing to the noise, because eliminationof one source often will make little orno apparent difference in t,he total noise. The_following procedure will help to isolate andidentify various types of noise.Ignition noise will be present only when theignition is on, even though the engine is turningover.Generator noise will be present when themotor is turning over, regardless of whetherthe ignition switch is on. Slipping the drivebelt off will kill it.Gauge noise usually will be present onlywhen the ignition switch is on or in the "left"position provided on some cars.Wheel static when present will persist whenthe car clutch is disengaged and the ignitionswitch turned off (or to the left position), withthe car coasting.Body noise will be noticeably worse on abumpy road than on a smooth road, particularlyat low speeds.
CHAPTER TWENTY-SEVENReceivers and TransceiversReceiver construction has just about becomea lost art. Excellent general coverage receiversare available on the market in many priceranges. However, even the most modest ofthese receivers is relatively expensive, and mostof the receivers are designed as a compromise-they must suit the majority of users, andthey must be designed with an eye to the price.It is a tribute to the receiver manufacturersthat they have done as well as they have. Evenso, the c-w man must often pay for a highfidelityaudio system and S-meter he neveruses, and the phone man must pay for the c-wman's crystal filter. For one amateur, the receiverhas too much bandspread; for the next,too little. For economy's sake and for easeof alignment, low-Q coils are often found inthe r-f circuits of commercial receivers, makingthe set a victim of cross-talk and overloadingfrom strong local signals. Rarely doesthe purchaser of a commercial receiver realizethat he could achieve the results he desiresin a home-built receiver if he left off the frillsand trivia which he does not need but whichhe must pay for when he buys a commercialproduct.The ardent experimenter, however, needsno such arguments. He builds his receivermerely for the love of the game, and the thrillof using a product of his own creation.It is hoped that the receiving equipment tobe described in this chapter will awaken theCAPACITORS:FIGURE 1COMPONENT NOMENCLATURERESISTORS•1- VALUES BELOW 999 JJJJFO ARE INDICATED IN UNITS. 1- RESISTANCE VALUES ARE STATED IN OHMS, THOUSANDSEXAMPLE: 150J.JJJFD OESICNATED AS 150. OF OHMS (K), AND MEGOHMS (M ).EXAMPLE! 270 OHMS= 2702- VALUES ABOVE 999 JJJJFO ARE INDICATED IN DECIMALS. 4700 OHMS= 4.7 KEXAMPLE: .005.UFD DES/(;NATED AS .00~.33.000 OHMS = 33 K100,000 OHMS= 100 K OR 0.1 M3- O<strong>THE</strong>R CAPACITOR VALUES ARE AS STATED. 33,000,000 OHMS= 33 MEXAMPLE: 10JJFD 1 O.SJJJJFD, ETC.2- ALL RESISTORS ARE 1-WATT COMPOSITION TYPE UNLESS4- iYPE OF CAPACITOR IS INDICATED BENEATH <strong>THE</strong> VALUE O<strong>THE</strong>RWISE NOTED. WATTAGE NOTATION IS THF.N INDICATEDDESIGNATION.BELOW RESISTANCE VALUE.SM =51 LVER MICA47 KC =CERAMICEXAMPLE:'if."SM= MICAP =PAPEREXAMPLE:250 .01 .001c;->p·~INDUCTORS:5- VOLTAGE RATING OF ELECTROLYTIC OR "FILTER• MtCROHENRI ES = JJHCAPACITOR' IS INDICATED BELOW CAPACITY DESIGNATION.MILLIHENRIES= MHHENRIES= H10 20 25EXAMPLE! .450 • 60o • 10""SCHEMATIC SYMBOLS•6- <strong>THE</strong> CURVED LINE IN CAPACITOR SYMBOL REPRESENTS OR<strong>THE</strong> OUTSIDE FOIL ''(;ROUND" OF PAPER CAPACITORS,<strong>THE</strong> NEGATIVE ELECTRODE OF ELECTROLYTIC CAPACITORS, + CONDUCTORS JOiNEDOR <strong>THE</strong> ROTOR OF VARIABLE CAPACITORS....l.-r-+ *CONDUCTORS CROSSINGBUT NOT JOINEDCHASSIS GROUND530
531experimenter's instinct, even in those individualsowning expensive commercial receivers.These lucky persons have the advantage ofcomparing their home-built product against thebest the commercial market has to offer. Sometimessuch a comparison is surprising.When the builder has finished the wiring ofa receiver it is suggested that he check hiswiring and connections carefully for possibleerrors before any voltages are applied to theSTANDARD COLOR CODE-RESISTORS AND CAPACITORSAXIAL LEAD RESISTOR INSULATED FIRST RING SECOND RING THIRD RING DISC CERAMIC RMA CODEUNI NSULATED !.COY COLOR END COLOR DOT COLORBROWN-INSULATED COLOR FIRST FIGURE SECOND FIGURE MULTIPLIER .5-DOT 3-DOTSLACK- NON-INSULATED BLACK 0 0 NONEGt"'";!;;BROWN 1 1 0RED 2 2 00 MUL TIPLlERORANGE1!\J!?Jfij3 3 ,ooo TOLERANCEYELLOW 4 4 o.ooo.l. .~.~~OLERANCE GREEN 5 5 00,000 TEMPERATUREMULTIPLIER BLUE 6 6 000,000COEFFICIENT1ST& 2ND SIGNIFICANT FIGS. VIOLET 7 7 0,000,000GRAY 6 6 00,000,000WIRE-WOUND RESISTORS HAVE 1STWHITE 000,000,000DIGIT BAND DOUBLE WIDTH.• •RADIAL LEAD DOT RESISTOR\5-DOfRADIAL LEAD CERAMIC CAPACITOR EXTENDED RANGE TC CERAMIC HICAPTEMP.COEFF.CAPACITYANCEMULTIPLIER_[fCAPA~TY~~~.d..---tB\1\ ~mm 1111It-TOLERANCELMuLTIPLIER'--TC MULTIPLIERRADIAL LEAD (BAND) RESISTORBY-PASS COUPLING CERAMIC CAPACIT'ORAXIAL LEAD CERAMIC CAPAC I TORMULTIPLIERTEMP. COEFf.l rrCAPACITYTOLERANCE1ST FIGURE2NDFIGURE~~~!MULTIPLIEJ LTOLERANCECURRENT STANDARD CODE:tH:~n """"'~~ """"'JAN &MOLDED MICA TYPE CAPACITORSRMA 3-DOT (oesocErE)RATED 500 V.D.C. ±2096 TOL.WHITE (RMA)BLACK(JAN) MULTIPLIER 1948RMACODE~MULrlPLIERCLASS TOLERANCE 0IST@RMA 5-DOT CODE (oesOLErE)SIG. FIGURE~o"~LERANCESIGNIFICANT FIG.RMA 6-DOT (oesocErEJ0~'}MULTIPLIER NO S1G FIGURERONT~RK.VOLr.1STMULTIPLIERNO SIG. FIG.~ULT1PL1ERULTIPLIERRANCEEAR WOR OLERANCE WORKING VOLTA
532 Receivers and Transceivers<strong>THE</strong> <strong>RADIO</strong>FIGURE 3COMPONENT COLOR CODINGPOWER TRANSFORMERSPRIMARY LEADS ------BLACKIF TAPPED:COMMON ------BLACKTAPENDHIGH VOLTAGE WINDING ---REOBLACK/YELLOWBLACK/ REOCENTER-TAP -----R£0/ YELLOWRECTIFIER FILAMENT WINDING-YELLOWCENTER-TAP -----YELLOW/BLUEFILAMENT WINDIN~ N° 1 ---t;REENCENTER-TAP-----GREEN/YELLOWFILAMENT WINDING N° 2 ---BROWNCENTER-TAP-----BROWN/YELLOWFILAMENT WINDING N° 3---SLATECENTER-TAP1-F TRANSFORMERSPLATE LEAD-------BLUEB+ LEAD REDGRID (ORO/ODE) LEAD---t;REENA-V-C (OR t;ROUNO) LEAD---BLACKAUDIO TRANSFORMERSSLATE /YELLOWPLATE LEAD (PRI.)----BLU£ OR BROWNB+ LEAD ( PRI.) REOGRID LEAD (SEC.)GRID RETURN (SEC.)GREEN OR YELLOWBLACKcircuits. If possible, the wiring should bechecked by a second party as a safety measure.Some tubes can be permanently damagedby having the wrong voltages applied to theirelectrodes. Electrolytic capacitors can beruined by hooking them up with the wrongvoltage polarity across the capacitor terminals.Transformer, choke and coil windings may beFigure 4TWO TRANSISTOR BROADCASTRECEIVER IS EXCELLENTBEGINNER'S PROJECTThis simple receivet works lrom a single 9-volt battery, providing many hours olpleasant listening. Two transistors and adiode in a rei/ex circuit provide ampleloudspeaker volume. Tuning control is atlelt, with audio control beneath it. Midgetspeaker is at right. Receiver is housed inplastic case.damaged by incorrect wiring of the high-voltageleads.The problem of meeting and overcomingsuch obstacles is just part of the game. A trueradio amateur (as opposed to an amateurbroadcaster) should have adequate knowledgeof the art of communication. He should knowquite a bit about his equipment (even if purchased)and, if circumstances permit, he shouldbuild a portion of his own equipment. Thoseamateurs that do such construction work areconvinced that half of the enjoyment of thehobby may be obtained from the satisfaction-susLSl5Ko.ssr susFigure 5SCHEMATIC OF TRANSISTOR BROADCAST RECEIVERB-9-volt transistor battery. RCA VS-300c,-365 !l!lld. Lafayette Radio Co. MS-214, or Allied Radio Co. 61H-009Lr--Transistor "Loopstick" coil. Lafayette Radio Co. MS-166LS-3" loudspeaker, 12-ohm voice coil. Lafayette Radio Co. SK-39T1-500 ohm pri., 12 ohm sec. Transistor transformer. Thordarson TR-18
<strong>HANDBOOK</strong>Circuitry 533of building and operating their own receivingand transmitting equipment.The Transceiver A popular item of equipmenton "five meters"during the late "thirties,"' the transceiver ismaking a comeback today complete with moderntubes and circuitry. In brief, the transceiveris a packaged radio station combining the elementsof the receiver and transmitter into asingle unit having a common power supplyand audio system. The present trend towardcompact equipment and the continued growthof single sideband techniques combine naturallywith the space-saving economies of thetransceiver. Various transceiver circuits for thehigher frequency amateur bands are shown inthis chapter. The experimenter can start fromthese simple circuits and using modern miniaturetubes and components can design andbuild his complete station in a cabinet no largerthan a pre-war receiver.27-1 Circuitry andComponentsIt is the practice of the editors of thisHandbook to place as much usable informationin the schematic illustration as possible. Inorder to simplify the drawing the componentnomenclature of figure 1 is used in all the followingconstruction chapters.The electrical value of many small circuitcomponents such as resistors and capacitors isoften indicated by a series of colored bandsor spots placed on the body of the component.Several color codes have been used in the past,and are being used in modified form in thepresent to indicate component values. Themost important of these color codes are illustratedin figure 2. Other radio componentssuch as power transformers, i-f transformers,chokes, etc. have their leads color-coded foreasy identification as tabulated in figure 3.27-2 A SimpleTransistorized PortableB-C ReceiverIllustrated in figures 4, 5, and 6 is an easyto construct two transistor portable broadcastreceiver that is an excellent circuit for thebeginner to build. The receiver covers therange of 500 kc. to 1500 kc. and needs noexternal antenna when used close to a highpower broadcasting station. An external antennamay be added for more distant reception.The receiver is powered from a single 9-voltminiature transistor battery and delivers goodspeaker volume, yet draws a minimum of currentpermitting good battery life.CircuitDescriptionOperation of the receiver maybe understood by referring tothe schematic diagram of figure5. The tuned circuit L-C resonates at thefrequency of the broadcasting station. A portionof the r-f energy is applied to the base ofthe 2Nll2 p-n-p type transistor. A tappedFigure 6INTERIOR VIEWOF TRANSISTORRECEIVERThe speaker and outputtransformer are mountedat the left of the Masonitechassis. Top, centeris the "loopstick" r:fcoil, and directly to tlieright is the 10 millihenryr-f choke in the collectorlead of the 2N112 transistor.Battery is at lowerright.
534 Receivers and Transceiverswinding is placed on coil L1 to achieve an impedancematch to the low base impedance ofthe transistor. Emitter bias is used on thisstage, and the amplified signal is capacitycoupledfrom the collector circuit to a 1N34diode rectifier. The rectifier audio signal isrecovered across the 2K diode load resistor,which takes the form of the audio volumecontrol of the receiver. The diode operates inan untuned circuit, the selectivity of the receiverbeing determined by the tuned circuitin the r-f amplifier stage.The audio signal taken from the arm of thevolume control ( R,) is applied to the base ofthe 2N112 r-f amplifier which functions simultaneouslyas an audio amplifier stage. Theamplified audio signal is recovered across the2K collector load resistor of the 2N112, andis capacitively coupled to the base of a 2N21 7p-n-p audio transistor. This stage is base andemitter biased, having the output transformerin the collector circuit. Optimum collector loadfor the 2N21 7 is approximately 500 ohms,and the 2N21 7 develops a maximum audiosignal of 75 milliwatts at this load impedance.Transformer T, matches the transistor circuitto the 12 ohm miniature loudspeaker. Thereceiver draws a maximum signal current of11 milliamperes from the 9-volt battery supply.<strong>THE</strong> <strong>RADIO</strong>Receiver The complete receiver isConstruction mounted upon a thin Masoniteboard measuring 7%" x4 :Vs" in size. The board fits within a smallplastic box having a hinged lid and handle.The dimensions of the box are 7 Y2" x 5" x1 Ys", exclusive of the handle. Placement of themajor components upon the board may be seenin figure 6. The "loopstick" antenna coil ismounted to the top center portion of the boardso that it fits directly under the handle of thebox. Slightly below and to the right of thecoil is the tuning capacitor, C. The midgetspeaker is mounted in a cut-out in the boardat the lower left corner with the output transformerT, directly above it. The volume controlR, is placed directly below the tuningcapacitor. The battery clip for the 9-volt cellis placed at the lower right corner of the board.A three terminal phenolic strip is attached tothe board above the battery clip. The leads ofthe 2N112 transistor are soldered to this strip.Before the components are mounted on theboard, it is placed within the plastic box andused as a drilling template for the box. Adrill-press and "fly-cutter" may be employedto cut the speaker hole in both the board andthe box. The plastic box should be drilledslowly to prevent excessive heat generationand consequent melting of the material.PLACE CARRIERHERE OR HERE"-\ vAVERA~ERECEIVERSELECTIVITYRANGEDESIREDSELECTIVITYCURVEMECHANICALFILTERSELECTIVITYCURVEBc-&008~---+~~~~~------------------~~~~~~------~~~~------~--------~1. ..,/ ,, ~ 3 KC I- I- e.~ Kc -.j1~ TO 30 KC WIDEAT -&0 DB DOWNFigure 7BANDWIDTH CURVES OF VARIOUS 1-F SYSTEMSBandwidth curves showing: A-Selectivity range ot most medium-priced single-conversion receivers withcrystal filter out ot circuit; 8-ldeo/ selectivity curve tor voke reception; and C-Selectivity curve ot a455 kc. mechanical filter with a 3.1 kilocycle bandwidth.
<strong>HANDBOOK</strong>Mechanical Filter535The 2N21 7 transistor is mounted upon asecond three terminal phenolic strip placed betweenthe tuning capacitor and the loud speaker.Use of a composition-type chassis precludesthe employment of the chassis as a commonground. A simple bare copper wire is thereforeused as a common "ground" return. This leadis actually the "B-plus Bus," shown in figure 5.The lead runs from the positive terminal ofthe battery clip, along the lower edge of thechassis then up to the "ground" terminal ofthe "loopstick" coil, L. All components thatnormally ground to the metal chassis in aconventional assembly are attached to this bus.Be sure to observe the polarity of the electrolyticcapacitors and the dry cell. Improper connectionof these components can damage thetransistors or other parts of the circuit.The transistors are the last items to be connectedinto the assembly. Since the transistorsare heat sensitive, it is necessary to protect themduring the soldering operation. Grasp the transistorlead to be soldered with a long-nosepliers between the body of the transistor andthe end of the lead. The thermal capacity ofthe metal pliers will act as a heat sink, preventingthe heat of the iron from damagingthe transistor. Be sure to hold the pliers inplace until the completed joint has had timeto cool. When all wiring is completed andchecked, the 9-volt battery can be inserted inthe clip (observe polarity! ) and the set canbe turned on by means of switch S,. A 0-100milliampere d-e meter placed temporarilyacross the open contacts of switch s, can beused to check the transistor operating current.The receiver should play immediately uponclosing S, as the transistors have no warm-uptime. If no external antenna is used, the receivershould be moved about to orient the"loopstick" coil L, for best pickup of eachindividual broadcast station. Adjacent channelinterference can often be eliminated by carefulrotation of the set to "null out" the offendingsignal. Ample loudspeaker volume will beobtained from local stations without the useof an external antenna.27-3A 455 Kc.MechanicalFilter AdapterEnjoy modern-style selectivity and pe~formancewith your present receiver by plugging inthis simple mechanical filter .adapter that replacesthe first i-f tube 1Many modern medium-priced and olderhigh priced communications receivers now ingeneral use are convenient to operate. The receivershave good frequency stability and adequatesensitivity, but lack sufficient "skirt"selectivity to reject strong signals that are onlya few kilocycles higher 'or lower in frequencyfrom a desired signal. The shaded area of curveA, figure 7 shows the typical selectivity characteristicof popular medium-priced communicationsreceivers. Although the "nose" of thiscurve is usually only a few kilocycles wide, the"skirt" selectivity 60 decibels down from thepeak may be from 15 to 30 kilocycles broad!Small wonder that strong local signals a fewkilocycles up the band from the station youare trying to copy may sometimes paralyzeyour receiver.It is possible to add a new order of selectivityto your present set by constructing thismechanical filter adapter unit, that is substi-Figure 8455 KC. MECHANICALFILTER ADAPTEREnjoy optimum selectivity and performancefrom your present receiver by plugging inthis simple adapter that replaces the firsti-f amplifier tube. The two 68J6 tubes areat opposite ends ol the chassis with themechanical filter between them. Adapterplug to fit i-f tube socket is shown inforeground.
536 Receivers and Transceivers<strong>THE</strong> <strong>RADIO</strong>ter such as the mechanical filter or a crystallattice filter. The second system utilizes a stringof high Q tuned circuits in the i-f amplifierto achieve the desired passband. This latter systemcan be space consuming, difficult to adjust,and expensive if quality components areemployed.Figure 9CONNECTIONS BETWEENMECHANICAL FILTER,ADAPTER AND RECEIVERAdapter makes connections to first i-fsocket of receiver. Power may also be obtainedfrom the receiver.tuted for the first 4 55 kc. intermediate frequencyamplifier tube, without making anyunder-chassis changes in the receiver. Simplyconnect the adapter to a power source (whichmay be your receiver), remove the first i-famplifier tube, and insert two short coaxialcables into the empty tube socket, as shownin figure 9. These cables carry the i-f signalto and from the adapter, which are then tuckedaway in an unoccupied corner of your receiveror speaker cabinet.Selectivity There are two systems generallyused to obtain aConsiderationsband pass characteristic thatapproaches the ideal communications selectivitycurve for voice modulated signals shown at Bin figure 7. One system utilizes a package fil-,III IOn the other hand, the mechanical filter isvery compact. It is readily available in a varietyof bandwidths, has an excellent selectivitycurve, and is roughly equivalent in cost toother systems having comparable selectivity.Curve C of figure 7 illustrates the selectivitycurve of the 3.1 kilocycle Collins mechanicalfilter which is suitable for A-M phone andsingle sideband reception.CircuitThe complete circuit of theDescription mechanical filter adapter isshown in figure 10. Two 6BJ6pentode tubes are employed in the ciircuit tocompensate for the insertion loss of the filter.The adapter picks up the 45 5 kc. signal fromthe control grid pin of the receiver's first i-famplifier tube socket through coupling capacitor,C. The signal passes through a coaxialline to the grid of the 6BJ6 amplifier rube,v,. The plate circuit of V, is capacity coupledto the input terminals of the mechanical filterto keep plate current from flowing throughthe filter coil. A much wider signal voltagerange can be handled by the filter without--,I IL _ _ _S!:f.l.E!d} _ __.._ __I_§H..l§L_Q __ ----1L--------------:ft-'C:'!5C-ih TO 1-F .r----Jf'C;,;o,_...,:'r------------'~ • ~TUBE SOCKET~ • ~GRID PINPLATE PINL---------'---...__TO GND. PIN, 1-F TUBE SOCKETFigure 10SCHEMATIC OF MECHANICAL FILTER ADAPTERc,, C 4-600 p,p,fd. ceramic (270- and 330 p,p,fd. in parallel)c,, c,-120 p,p,fd. ceramicCs, Ca-10 p,p,fd. tubular ceramic (Aerovox Cl-1)FL-455 kc. mechanical filter (3.1 kc. bandwidth) (Collins 455J-31)L,, Lt-200 microhenry slug-tuned "loop stick" coil (Superex VL, or J. W. Miller 6300)P,-Male octal plug with retaining ring (Amphenol 86-PM8)R,-18K, 10 watt. Place between pins 4 and 8 of P1 socket for plate supply greater than 130 volts.Use jumper for 130 volts or less (see text). R, is screen voltage dropping rtsistor.
<strong>HANDBOOK</strong>Mechanical Filter 537-- ---'TOPIN1 CsV1 (OBJO)-A--1!--~~Mt---+1SOCKET _:LTO GRID PIN1-F TUBE SOCKETlOOK .01--a:-.- ~cFigure 11OPTIONAL A-V-C CONNECTIONSFOR <strong>THE</strong> ADAPTERoverloading when no plate current flowsthrough the input coil. Both filter coils areruned to resonance at the intermediate frequencyby fixed capacitors c, and c3.The output terminals of the mechanical filterare connected directly to the input circuitof the second 6BJ6 tube. The output signalfrom this tube is again capacitively coupledback into the plate circuit of the first i-f stageof the receiver through a second coaxial lineand coupling capacitor Cr.. The 455 kc. tunedcircuits connected to the plate of V, and v,are composed of "loopstick"-type coils L andL,, shunted by fixed capacitors C and C.The input and output coaxial cables are 16-inch lengths of RG-58/U line. This cableforms a 40 1-'1-'fd. ground leg of a capacitorvoltage divider ( Cs being the other leg) thatreduces the signal voltage applied to V, toabout one-quarter of the voltage developedacross the secondary of the first i-f transformerof the receiver.The overall signal amplification of the adapterhas been held down to a few decibelsmore than the 10 db insertion loss of the filterthrough the use of small input and outputcoupling capacitors and large cathode bias resistorsin both amplifier stages. This is suitablefor receivers having two or more i-f amplifierstages, but additional gain from theadapter may be obtained by reducing the valueof one or both cathode resistors to 270 ohms.This may be desirable when the adapter is operatedwith a receiver having only one i-famplifier stage.Power is brought into the adapter unitthrough an eight-pin male plug. Connectionsto this plug are made so that the plug may beinserted into the NBFM adapter socket on certainNational receiver models. Many communicationsreceivers have an accessory power socketon the rear of the chassis from whichpower may be obtained. Adapter requirementsare 6.3 volts at 0.3 amperes, and 105-250volts at 10 milliamperes. This is little morethan the power requirements of the i-f tubereplaced by the adapter. A 250-volt supplysource requires that an 18K, 12-watt screenvoltage dropping resistor ( R,) be connectedbetween pins 4 and 8 of the power plug receptacle.When the d-e supply is 130-volts orless, a jumper may be placed directly betweenpins 4 and 8 (see figure 10).A method of applying automatic volumecontrol voltage from the receiver to the secondamplifier stage in the adapter is shown infigure 11. This modification is mainly usefulwhen the adapter is connected to a receiverthat has few a-v-e controlled stages. The automaticvolume control voltage is taken from thecontrol grid pin of the i-f amplifier tube socketthrough a simple R-C filter and is applied tothe control grid of v, through the output coilof the mechanical filter.--+--1-t --~--1-t ------+-+---1A+SHIELD BOLT3+-47e~------------------------~_L2fX 2-f' X 4" MIN/BOX (BUD CU-3003 OR EQUIVALENT)58II-~ -~OUTPUT CABLErt---1-- ___ B~Figure 12CHASSIS DRILLING TEMPLATE FOR ADAPTERA: #32 drill, B: 9/32" drill, C: 5/8" diameter socket punch, D: 3/4" diameter socket punchI
538 Receivers and Transceivers<strong>THE</strong> <strong>RADIO</strong>Constructionof the AdapterThe adapter is constructedon an aluminum box measuring2~" x 2~" x 4", agood compromise size that is compact, yet nottoo small for easy wiring. The primary designand construction consideration of this adapteris to completely isolate the input and outputcircuits. Any stray coupling can cause signalleakage around the filter unit, thus imparingits effectiveness. The construction shown allowsmaximum isolation, yet permits simplicityof construction. A drilling template forthe box is given in figure 12, and the placementof the major components may be seen infigures 8, 13, and 14. Note that the inputand output circuits are isolated within the boxby a shield passing across the center of thefilter socket.After drilling and punching all holes, thecube and mechanical filter sockets, power plug,and rubber grommets may be assembled. Placesolder lugs under all socket retaining bolts forground connections. Before any wiring is donea 3" x 3" piece of perforated aluminum(Reynold's "Do-it-yourself" aluminum, availableat large hardware stores) is formed intothe shield shown in figures 13 and 14. A %"flange is bent along all edges of this shieldexcept where it crosses the center of the 9-pinfilter socket. A small notch is cut in the shieldnext to the socket to pass the heater and platepower leads to the V, socket. The shield passesacross the filter socket between pins 3 and 4,and 8 and 9, then is bolted to a soldering lugthat has been soldered to pin 1 of the filtersocket. The flange is also bolted to the topof the box directly above coil L,, and two selftappingsheet metal screws are driven into theside flanges of the shield when the other halfof the box is assembled.CableAssemblyConstructing the two i-f tubesocket cables takes little time ifthe assembly shown in figure 15is followed. First, cut two lengths of RG-58/Ucoaxial cable 17 inches long and remove 1 Y2inches of the vinyl cover on one end of eachpiece. Slide the braided outer shield back overthe outer cover and trim the insulation andcenter conductor so that Y2 inch protrudes beyondthe shield. Next cut the insulation toexpose Y-4 inch of the center conductor andtrim one lead of the coupling capacitor C, solderingit to the center conductor with a ~ inchoverlap. Cut narrow strips of plastic insulatingtape and wrap them around this joint up tothe body diameter of the capacitor as shownin figure 15. Finally, slide the braided shieldover the capacitor, pull it tight and wrap ashort length of tinned copper wire twicearound the middle of the capacitor. Solder thewire to the shield and trim off the excessshielding. The tinned wires from each cableare then soldered to a pin taken from an octaltube base, making a plug-in ground connectionas shown in figure 8. Similar tube basepins are soldered to the other leads, and theexcess lead trimmed off. The exposed cableshield is then wrapped with plastic tape.If the receiver has a 7-pin miniature tubein the first i-f amplifier stage, short lengths of# 18 tinned wire may be used for the plug-inpins on the cables, or the capacitors andground lead may be soldered to a special 7-pinminiature male adapter plug (Vector P-7).AdapterWiringFor easy parts assembly, the filtershield may be temporarily removed,and replaced when wiringis completed. Heater, screen and plate powerwires are installed first, keeping all such leadsclose to the box wherever possible to minimizeFigure 13UNDERCHASSISVIEW OF <strong>THE</strong>ADAPTERSlug-tuned coil L: is atleft, below the secondstage 6 8 J 6 socket.Screen bypass capacitoris placed across socket.Capacitors Cz and C, maybe seen on each side ofthe center shield. Coil L1is at right of shield. Inputcable is at far right.Output c a b I e passesthrough rear wall ofchassis at left. Powerleads pass through centerof shield.
<strong>HANDBOOK</strong>Mechanical Filter 539stray signal pickup. The smaller componentsare now soldered in place, after which the coaxialcable input and output leads are connected.About Ys inch of the outer vinyl jacketis removed from the cables and the shieldbraid is twisted into a single conductor. Thecable ends are brought into the box throughrubber grommets. The twisted cable shield issoldered to a nearby ground lug and the centerconductor is soldered to the correct tubesocket pin. Finally coils L, and L, and theirassociated capacitors are placed in position andwired into the circuit.Alignmentof the AdopterThe adapter is connected toto the communications receiveras previously described,following a wiring and power check to ensurethat the correct voltages are applied to thevarious tube elements. The receiver shouldthen be tuned to the center of a strong, steadylocal signal. If the receiver has an S-meter,the a-v-e should be left on. The slugs of coilsL. and L, are now tuned for maximum carrierstrength reading on the S-meter. If the receiverhas no S-meter, a modulated signal from atest oscillator must be used in conjunction witha vacuum tube voltmeter placed across theaudio output circuit of the receiver. Tuningadjustments on the first and second i-f transformersin the receiver are also recommendedfor maximum output of the adapter.BASE PINFROMOCTALSOLDERReceiver TuningTechniquesFigure 1 SCROSS-SECTION VIEWOF SIGNAL CABLESPOLYETHYLEN~INNER INSULATIONA somewhat different tuningtechnique should beused for tuning A-M andsingle sideband signals on a receiver followinginstallation of the mechanical filter adapter.Modulated signals with carrier should be tunedin so that the carrier is placed on one edge,rather than in the center of the i-f passband(figure 7). Only one sideband of an A-Msignal will be heard at one time, and the receivertuning may be shifted so that the sidebandon which a heterodyne is present may be"pushed off the edge" of the bandpass curveof the i-f system.When receiving single sideband or c-w signalsthe beat frequency oscillator of the receiveris turned on and adjusted so that thefrequency of the b-f-o is near one edge of theFigure 14OBLIQUE VIEW OFADAPTER CHASSISCoil Lz is at left nearpower plug, with coil Lzmounted horizontally atright. Note c, and c,are placed directly acrosspins of mechanical crys ..tal socket.
Figure 16FRONT VIEW OF AMATEURBAND RECEIVER6-band receiver covers 80-6 meters, usingTV turret for "front end." Mechanical filtersprovide optimum selectivity.filter passband. The proper pitch control settingmay be determined by tuning the receiveracross a carrier while adjusting the pitchcontrol so that a beat note on only one sideof zero beat is heard. After marking this settingof the pitch control, again turn it so the test signalon only the other side of zero beat is heard.Mark this setting, then try tuning in a singlesideband signal. If intelligible speech cannotbe heard, shift the b-f-o pitch control to theother setting. Small adjustments of the pitchcontrol and the tuning of the receiver willproduce the correct voice tones. For best reception,the audio control of the receiver shouldbe fully advanced, and only enough r-f gainused to produce a readable signal. Volume istherefore controlled by the r-f gain control.For optimum c-w reception, the 0.5 kc. mechanicalfilter may be used. No changes need bemade to the adapter when filters are changed.27-4 A HighPerEormanceAmateur Band ReceiverThis receiver was designed for outstandingperformance on the 80, 40, 20, 15, 10, and 6meter amateur bands. It incorporates many upto-datefeatures such as mechanical filters foroptimum selectivity, double conversion forfreedom from images, and a novel r-f assemblyfor mechanical stability and good operating efficiencyon the higher frequencies. The receiveruses 12 tubes, plus a selenium rectifierpower supply. The complete r-f assembly is designedabout a television turret tuner havingpositions for 12 separate bands. The ruggedconstruction and excellence of the self-wipingcontacts of the TV tuner add greatly to theoverall stability of the receiver. Complete dialcalibration and an S-meter provide maximumtuning simplicity and logging ease.Because of the "snap-in" coil strips of thetuner it is possible to provide coils to tune anynarrow segment of the shortwave range between3.5 megacycles and 75 megacycles. Forexample, coils can be wound to cover the 13,19, 21, 31, and 49 meter international shortwavebroadcast bands if desired. Panel controlof the selectivity allows instant selection of theoptimum passband for either voice or c-w reception.The receiver is entirely self.containedexcept for an external speaker.CircuitDescriptionThe High Performance AmateurBand Receiver is a doubleconversion superhetrodyne (figure1 7). A modified TV turret tuner is employedto cover six amateur bands. The turretA-N-L AUDIO SPEAKER1!s v ...f\..J,.._P_o_w_ER--., ___________+_.,I~,..o_v_. ------------1+!v ~ SUPPLY r ~VOLTAGEREGULATORA-V·CBRIDGEFigure 17BLOCK DIAGRAM OF DE-LUXE AMATEUR COMMUNICATION RECEIVER
Amateur Band Receiver 541employs a 6BQ7 A cascade r-f amplifier anda 6]6 mixer-oscillator. All r-f tuned circuitsare changed as the tuner is switched fromchannel to channel. The intermediate frequencyoutput of the tuner is 1695 kc. A 6BA 7pentagrid mixer tube converts the first intermediatefrequency to the second i-f channel,which is 455 kc. To obtain a high order ofstability, the 6BA 7 conversion oscillator is crystalcontrolled at 2150 kc.Switch S, in the output circuit of the 6BA 7mixer stage permits the selection of one of thetwo mechanical filters. In this particular receiver,a 6 kc. and a 3.1 kc. filter were usedfor "broad" and "sharp" phone reception.Three stages of 4 55 kc. amplification followthe mechanical filter. The gain of these stagesis adjustable, permitting the overall receivergain to be set at a reasonable value. Too muchi-f gain. will make the receiver noisy and subjectto overloading effects on strong signals.Too little i-f gain will make the receiver seeminsensitive. The i-f gain is controlled by potentiometerR, and also by the value of the cathoderesistors in the second and third i-fstages. In order to prevent clipping of the excellentflat-top response of the 6 kc. mechanicalfilter, small coupling capacitors are placedacross the "top" of the i-f transformer coils towiden the "nose response" of the circuits. The"skirt" selectivity of the i-f strip is determinedby the passband of the mechanical filter andnot by the transformer response. To conservefilament power, 150 milliampere 6BJ6 tubesare employed in the i-f amplifier.A 6AL5 dual diode is used as a seconddetector and a-v-e rectifier. Automatic volumecontrol voltage is applied to the first two i-fstages, and a small portion of the a-v-e voltageis applied to the 6BQ7 A r-f amplifier and6BA 7 mixer. The a-v-e circuit is rendered inoperativefor c-w and SSB operation by segmentD of switch s,. A simple delayed a-v-esystem is used to allow best reception of weaksignals. Potentiometer Ra sets the delay voltageapplied to the cathode of the 6AL5 a-v-e diode.A second 6AL5 serves as a noise limiter. Thethreshold value of limiting is set by controlR,, and the noise limiter may be removed fromthe circuit by switch Sa. The audio output ofthe detector passes through volume control R,and is further amplified by one triode sectionof a 12A T7 followed by a 6AK6 power amplifierstage. The second section of the 12A T7functions as the beat frequency oscillator. Thefrequency of the b-f-o is controlled from thepanel by variable capacitor Ca.The S-meter circuit consists of a single vacuumtube voltmeter using a 12AU7 dual triodein a bridge circuit. The bridge is balancedby potentiometer R,, and the meter sensitivityis set by control R,. Automatic volume controlvoltage applied to one grid of the 12AU7 upsetsthe balance of the bridge and causes a~---I R-FCOILL1II"TV -TUNER"-,IIIL--8+ B+REG.Figure 18ACONVERTED TV-TUNER FORMS "FRONT-END" OF AMATEUR BAND RECEIVERSeveral forms of replacement TV -tuners may be purchased from large radio stores, Shown above is typical"cascode tuner" to be used in this receiver. Check your tuner for circuitry before you modify it. l·f outputfrequency is unimportant, since i-1 components of tuner are removed. Tuner trimmers permit minoradjustments to be made to alignment of r-f stages.A-Oscillator-mixer excitation measuring point on tunerC,A-B-C-15 p,p,fd. per section. All-Star Products Co., Defiance, Ohio. Model C3.C.-10 p,p,fd. ceramic. Centralab 8278c,-15 !LIL Bud LC-1641. (Mounted atop chassis after photographs were taken).L,-L,-See figure 21 for coil data
NOTES: 1-* SEE FILTER I NSf RUCTION SHEET FOR CAPACITOR SPECIFICATIONS.2-ALL RESISTORS 1-WATT UNLESS O<strong>THE</strong>RWISE NOTED.R1B+HI 14011.SEEFIG.18A2KR30::J0..._,.....0::Jen()(!)
<strong>HANDBOOK</strong>Amateur Band Receiver 543reading on the 0-1 d-e milliammeter connectedbetween the plates of the 12AU7 triodes.The selenium rectifier power supply delivers140 volts at the receiver current drain of 65milliamperes. The low plate voltage and smallpower consumption of the receiver result in avery low degree of thermal drift. If desired,the selenium supply may be replaced with amore conventional supply using a 5Y3-GTrectifier tube. Plate voltage to the receivershould be held below 160 volts. Plate voltageto the 6]6 oscillator section and the 12AU7S-meter tube is regulated by an OB2 gas regulator.Plate voltage is removed from the r-f andaudio stages when the receiver is in "standby"condition. All oscillators are left running continuouslyto reduce initial warmup drift.Receiver The receiver is built upon a platedLayout steel chassis measuring 8" x 15" x2Y2". Panel height is 8". Toachieve maximum rigidity, the panel is bracedto the chassis with two steel angle bracketsvisible in figure 19. The tuning dial is centeredon the panel, and the three gang tuningcapacitor is behind the dial, on the center-lineof the chassis. Directly to the left of the tuningcapacitor is a 3 Y2" x 5" cut-out in the chassis.The TV tuner is placed in this cut-out, heldin place by two 3" angle brackets cut fromaluminum stock. Height of the tuner abovethe chassis is determined by the shaft of thetuner which must pass through the bushingdrilled in the lower left corner of the dialescutcheon. Likewise, the placement of thethree gang tuning capacitor C is determined bythe shaft height of the main dial bushing. Cis therefore mounted above the chassis bymeans of long 6-32 bolts and steel spacers.Placement of the major components may beseen in figures 19 and 20. The two mechanicalfilters are mounted to the left of the tuner,directly behind the 6BA 7 mixer tube and the1695 kc. i-f transformer, T,. The filter selectionswitch S, is mounted below chassis so thatthe switch segments fall over the filter terminals.This switch is made up from a switchkit, and is mounted in place by means of twobrackets cut from soft aluminum. One bracketholds the front panel bushing of the switch,and the other is placed just forward of sectionS,B and holds the two fixed shafts of theswitch. A hole is cut in the center of thebracket to clear the rotary shaft of the switch.This bracket also serves as a shield isolatingthe output circuits of the mechanical filter fromthe input circuits. The particular switch usedis highly recommended, and it is suggested thatno substitution be made. It is imperative thatthe drive shaft of the switch be metal, and thatit is securely grounded. Some switches havefibre drive shafts, or metal shafts held in positionwith a cotter pin. Fibre drive shaftspermit signal leakage along the shaft, deterioratingfilter performance, and the cotter pintypeswitch does not insure the shaft is securelygrounded which is mandatory, when themetal shaft-type switch is used.The three intermediate frequency stages arespaced along the back of the chassis. At theright rear are the 6AK6 audio stage, the volt-Figure 19TOP VIEW OFAMATEURCOMMUNICATIONRECEIVERM.echani
544 Receivers and Transceivers<strong>THE</strong> <strong>RADIO</strong>age regulator tube, and the 12AU7 S-metertube. In front of the last i-f transformer canare the two 6AL5 tubes and potentiometersR, and Ra.At the right end of the chassis is the powertransformer, and between it and the panel isthe b-f-o transformer and the 12AT7 audio/b-f-otube. Between the power transformerand the three gang tuning capacitor are thepower supply filter capacitor and the auxiliaryfilament transformer. The i-f gain control potentiometerR, is placed between the filamenttransformer and the main tuning capacitor.Placement of the smaller components belowthe chassis may be seen in figure 20. TheTV turret and mechanical filter switch are atthe right of the chassis, with the audio outputtransformer mounted to the back wall of thechassis near the turret. The filter choke for thepower supply is mounted to the left side wallof the chassis, with the two selenium rectifiersin front of it. The 10 watt dropping resistorfor the regulator tube is mounted to the chassisbeween the filter choke and the rear tubesockets.ReceiverConstructionThe receiver should be builtin two parts to simplify construction.The power supply,audio and 455 kc. i-f sections are built first,and the receiver is placed in operating conditionup to the 1695 kc. i-f channel. The TVtuner is then modified, using the rest of thereceiver for alignment and adjustment tests.All major components should be marked inposition on the paper wrapper of the chassiswhich is then drilled and punched. All majorcomponents with the exception of the TVturret are mounted in place. The ground connectionsat each tube socket are made first,then the filament leads are wired in position.Use # 14 insulated wire for the filament leads,as the current drain is quite high. The powersupply should be wired first and tested. Itshould deliver approximately 140 volt! acrossa 2000 ohm, 10 watt resistor used as a temporarydummy load. When the power supplysection is completed, the audio stages, b-f-o,S-meter, second detector, and noise limiterstages can be wired. Operation of the audiosystem may be checked by injecting a smallaudio signal on pin # 2 of the 12A T7 stageand checking the volume of the tone in theloud speaker. ·The next step 1s to wire the three i-f amplifiersand the a-v-e circuit. Five terminalphenolic tie-point strips are mounted in frontof each i-f tube socket. The a-v-e, cathode bias,and screen dropping resistors are mounted betweenthe socket pins and the terminals of thestrip. The ceramic bypass capacitors are soldereddirectly between socket pins with theshortest possible leads. All interconnectingwiring is done between the terminal strips.After the i-f strip has been wired up to thearm of switch s,, segment B, the amplifiermay be tested. Pin # 1 of the first 6BJ6 i-ftube is temporarily returned to the a-v-e busby placing a 100 K resistor across the outputpins of the mechanical filter socket. Switch s,is adjusted to place the resistor in the circuit.Figure 20UNDER-CttASSISVIEW OF AMATEURBAND RECEIVERTo the /eft of the chassisare the selenium rectifiersand the filter choke.At the extreme right isthe mechanical filter selectorswitch mountedon two aluminum brackets.The controls alongthe panel are (left toright): BFO, AUDIOGAIN, MAN---AM---CW,tuning control, TV turretswitch, R-F GAIN,and filter selector switch.Output transformer ismounted to back ofchassis.
<strong>HANDBOOK</strong>A signal generator having a tone modulated455 kc. signal is attached to the input grid ofthe 6BJ6 tube, and an a-c voltmeter is connectedacross the speaker terminals. To insurethat the i-f system is aligned at the center ofthe filter passband it is imperative that thealignment frequency be very close to 455.0kc. The use of a BC-221 or equivalent signalgenerator is recommended. As the transformersare brought into alignment the input signalto the i-f strip should be decreased. A vacuumtube voltmeter may be employed to measurethe negative voltage developed on the a-v-e bus.A maximum voltage of about -s volts is developedwith a strong input signal to the i-fstrip. Overall gain is controlled by the settingof R, when switch S, is placed in the "A-M"position.After i-f alignment has been completed, the100 K input resistor should be removed fromthe filter socket and the 6BA 7 mixer stagewired. The capacitors connected between pins1 and 2 and ground of the 6BA 7 socket determinethe level of excitation to the 2150 kc.crystal. If difficulty is encountered in obtainingconsistent oscillation, the value of one orboth of these capacitors may be altered untilreliable oscillation is obtained. Operation ofthe crystal oscillator may be monitored with anearby receiver tuned to the crystal frequency.When the oscillator is working properly, a1695 kc. tone modulated signal should beAmateur Band Receiver 545coupled through a 0.01 ~-tfd. blocking capacitorto the "hot" end of the primary windingof transformer T. This transformer is adjustedfor maximum output signal. The receiver isnow ready for tuner installation.TunerModificationThe tuner used in this receiveris a Standard Coil Co.replacement type unit usinga 6BQ7 A cascode r-f amplifier and a 6]6 mixerstage. Several different variations of thistuner can be purchased at nominal pricesthrough large wholesale outlet stores. Thetuner has snap-in coil strips for 12 TV channels,and may have either a 21 Me. or 42 Me.i-f output range. Several simple changes mustbe made to the tuner for use in this receiver.The i-f output coil and the cascode neutralizingcircuit must be removed. The best thing to dois to remove all the clip-in coils and trace outthe tuner circuit, and then compare it with thecircuit shown in figure 18A. Cathode bias mustbe added to the first section of the cascode andseveral of the resistors in the tuner must bechanged. Rewire the tuner according to thecircuit of figure 18A. Note that regulated voltageis applied to the oscillator section of the6]6 and that a-v-e voltage is applied to thefirst section of the 6BQ7 A. The main tuningcapacitor C1A-B-C is connected to the appropriatepoints in the tuner by three short leads.A }~-inch coil is drilled in the rear of theFIGURE 21 -COl L TABLE FOR AMATEUR BAND RECEIVERBAND COIL WiNDING AND WIRE SIZ.E ;)c SHUNTY'II NATURALSPACING CAPACITY RESONANCE BAND COIL WINDING AND WIRE SIZE I •lc SHUNT 'I'JF NATURALSPACING CAPACITY RESONANCEL1ANTs 3 T. N° 2.2 E. e-w ANT: 7 T. N° 22. E. e-wL1GRfO:!>T, N°18E. WIRE CIA. !t8 J.WFD. 83MC. GRID: 13T. N°22.E. e-w 180!J1JFO. 1~.2 MC.eM PLATE, R-F; .5T. N°18E. WI REDIA. 821J1JFD 83 MC. 20M PLATE, R·F: 13T. N• 22 E. WIRE CIA. 200.UUFO 14.7~MC.L2L248.3- GRID, OSC.: !t T. N° 18 E. WIRE OIA. 02lWFD. 82 MC.1!t. 7- GRID,OSC.: 12.T.N°22.E. WIRE DIA. 2.001J.UFD. 17 t.AC.~2.3Me. PLATE, OSC. 1 5T. N° 18 E. e-w115.0~Me. PLATE. OSC.: 10 T. N° 30E. e-w10M15ML1ANT.: lOT. N°2.2E. e-w ANT.: I~T. N°30E. e-wL1GRID: lOT. N°22E. WIRE DIA. 75JJlJFD. 33t.AC. GRID: 39T. N° 30 E. e-w 100.U.UFD 7.9 MC.PLATE, R-F: lOT. N°22E. WIRE DIA. 62U.UFO. 32 MC. 40M PLATE, R·F: 38T. N° 30 E. e-w 110U.UF0. 7.8 MC.L2L226.3- GRID, OSC.: 10 T. N° 22 E. WIRE DIA. 68JJ.UFO. 32.~ MC. 8.7- GRID, OSC.: 25T. N°30E. e-w 1101J.1JFO. IOMC..28.0 •..Me. PLATE, OSC.: 9 T. N° 30 E. e-w Me. PLATE, OSC.,: 12T. N°30E. e-wL1ANT: 9 T. N° 22. E. e-w ANT: 10T. N° 30 E. e-wL1GRID: 9T. N° 2.2. E.15UJJFO,WIRE DIA. 150UJJFD. 22..3MC. GRID: ''ft8~'£VJiJf;'JJR~s5JJ.JJNtf: 10K 5.4t.AC.PLATE, R-F: 10T. N°2.2. E. WIRE DIA. 12.0JJUFD. 2.2.25 MC..L280MPLATE, R-F: SEE L1 ABOVE 12..U.UFD 5.7 MC.L"19.3- GRID,OSC..: lOT. N° 2.2. E, WIRE DIA. 2201J1JFO. 21.0 MC. 5.,- GRID,OSC.: SEE L1 ABOVE 39.1J1JFD 7.2.MC..19.75 5.7Me. PLATE,OSC.: lOT. N°30E. e-w Me. PLATE.OSC.: 2.0T.N°30E.I e-wNOTE 1 -GRID WINOIN. OF OSCILLATOR COIL L2 IS PLACED NEXT TO OSCILLATOR PLATE WINDIN •. R-F PLATE WINDIN.IS PLACED ABOUT 1/2-INCH AWAV' FROM OSCILLATOR-GRID WINDING.2-ANTENNA COIL SPACED ABOUT 1/te-INCH FROM GRID COIL.3-TOKIIES/STOII PLACED ACROSS 80M£TEIIII-F 'RID COIL.4-RESONANT FREQUENCY MEASURED WITH SHUNT CAPACITOR ACI?OSS COIL.
546 Receivers and Transceivers<strong>THE</strong> <strong>RADIO</strong>tuner case to permit the coaxial lead from theantenna receptacle to enter the tuner. Thetuner is now ready to be mounted in the receiver.Tuner CoilModificationThe tuner coils should beremoved from their snap-inholders. Notice that the r-fcoils have five terminals and that the mixeroscillatorcoils have six terminals. All coilforms and windings are removed from theclip-in assemblies and may be discarded. Thenew coils for all bands are to be wound onlengths of ~-inch diameter polystyrene ro~,cut to fit into the assembly clamps. For thisparticular tuner, the r-f coil forms are 1 %"long and the oscillator mixer coil forms are1 13/16" long. The ends of the forms aretapered slightly with a file so that they fitinto the slotted ends of the assembly clamp.After each set of coils is completed the frequencyband notation is marked on the unitfor future identification.Complete coil data is given in figure 21and a complete set of coils is shown in figure22. Start with the ten meter coils, as they havefew turns and are easy to wind. Make the r-fstage coil first. The windings may be tern-porarily held in place with ~ small piece ofcellophane tape until the coil form is cementedin the assembly clamp with Duco cement.The coil leads are then cleaned and solderedto the corresponding terminals. After the shuntcapacity is added, the natural resonance of thewinding should be checked against the valuegiven in figure 21. The coil should be suspendedin the clear and checked with a grid-dip oscillator. Turn spacing may be varied toproduce the correct resonance point. The mixer-oscillatorcoil is wound next. The two tunedwindings are placed at the ends of the formand the oscillator plate winding is placed betweenthem, positioned close to the grid windingas illustrated in figure 22. The coil windingsmay be held in position with a temporarypiece of cellophane tape. This coil is adjustedin the same manner as the r-f coil. When completed,the mixer-oscillator coil may be snappedinto the rener and the receiver turned on. Theoscillator operates on the low frequency sideof the signal for 10 meter reception, coveringthe range of 26.3 Me. to 28.0 Me. as thereceiver is tuned from 27,995 kc. to 29.7 Me.Listen for the oscillator in a nearby receiver.The frequency range of the oscillator can beset by varying the spacing of the turns of the6]6 oscillator coil, and also by adjustment ofthe oscillator trimmer capacitor c, in the tuner.When the correct frequency range is coveredthe winding may be permanently held in placewith a small application of Duco cement. Checkthe frequency range once again after you applythe cement as the frequency shifts slightlylower as the cement is applied. Make yourfinal correction before the cement dries. Theother coil windings are relatively non-criticaland require no adjustments after they arecompleted.Once the oscillator tuning range has beenset the signal generator may be used to calibr~tethe main dial of the receiver. If care isused in this process, the dial calibrations ca~be read to an accuracy of five kilocycles orless. The oscillator injection voltage may bechecked at terminal A on the tuner with av-t-v-m. The injection range should lie betweentwo to three volts on each band, and may becontrolled by varying the spacing between theFigure 22COIL SET FOR AMATEURBAND RECEIVERTuner
<strong>HANDBOOK</strong>"Hand ie-Talkie" 547oscillator grid winding and the untuned platewinding. These adjustments may be made withthe coils in the receiver by removing the bottomand side plates of the tuner.The conversion oscillator operates on thehigh frequency side of the signal on the 15,20, 40, and 80 meter bands, as tabulated infigure 21. The 80 coils are made from "loopstick"-type broadcast coils. Turns are removedfrom these coils until the natural resonancefalls at the desired frequency as shown in thetable. A small value of capacitance is usedacross these coils since the percentage-tuningrange is quite high.ReceiverAdjustmentAfter the first set of coils hasbeen completed, the receivermay be adjusted for optimumresults. A balance between r-f gain and i-fgain must be achieved. Under normal operatingconditions, the residual noise picked upby the antenna system should mask out the internalnoise level of the receiver. If the antennais removed from the receiver and theinternal noise level does not drop appreciablyit is usually a sign that the i-f gain of thereceiver is too high and the r-f gain is toolow. The value of i-f gain may be controlledby the setting of R,. Normally, there is noinstability or regeneration in the i-f strip. Thestability, however, is dependent to some extentupon the wiring technique you use; it maybe necessary to retard the i-f gain level toprevent instability. Receiver i-f gain may befurther decreased by increasing the value ofthe cathode resistor of the first 6BJ6 stage.I-f gain may be increased by decreasing thevalue of the cathode resistors of the secondtwo i-f stages.The a-v-e bias delay potentiometer Ra shouldbe set to provide about one volt of bias onpin # 1 of the 6AL5 a-v-e rectifier. This willretard a-v-e action on weak signals and improvethe signal-to-noise ratio of thresholdsignals.The level of beat oscillator injection can beset by varying the setting of coupling capadtorC.. Sufficient injection should be used toprovide good reception of single sideband signals.The injection level is not critical forc-w operation. Finally, the b-f-o capacitor C.should be set at mid-scale and the slug oftransformer T, adjusted until the b-f-o signalis tuned exactly to 455 kc.The response of the S-meter is controlledby the time constant in the grid of the 12AU7meter tube. If the grid capacitor in this circuitis increased in value from .05 ~-tfd. to 1.0 ~-tfd.the S-meter will respond much more slowlyto changes in signal strength. The value ofthis capacitor can be altered to suit the user'staste.Finally, various settings of the noise limiterthreshold control R, should be tried to findthe best clipping level for various types ofpulse-type interference.ReceiverOperationReceiver tuning is extremelysmooth, and the dial calibrationremains as good as the originalcalibration. The r-f input circuits are designedfor 50-100 ohm unbalanced transmission lineon all amateur bands, and r-f stage resonancemay be easily attained by means of the antennatrimming capacitor located within theset. For phone reception, switch s, is set toA-M and the audio level is controlled by theaudio potentiometer R.. For c-w or sidebandreception s, is set to C-W and audio controlR. is fully advanced. Receiver gain is nowcontrolled by the R-F gain potentiometer R..The B-F-0 pitch control Ca is adjusted to placethe oscillator on the edge of the filter passbandfor "single signal" reception of c-w signalsor single sideband reception of SSB transmissions.The S-meter is operative on the"A-M" position of s,; S-meter sensitivity iscontrolled by potentiometer R,.27-5 A "Handie-Talkie" for 144 Me.The 144 Me. band is the ideal frequencyrange for short-haul, portable equipment. Littlepower is required for local coverage, and theantenna size is moderate. Then too, this is justabout the highest frequency at which filamenttype battery tubes will operate.Shown in the accompanying illustrations isa battery powered, crystal controlled "handietalkie"designed for operation in the 2 meterregion. It is especially well suited for CivilDefense work or emergency service. This unitwas designed after the AN/URC-4 air-sea rescuetransmitter, and is completely self-contained,including batteries.Circuit The circuit of the "handie-Description talkie" is shown in figure 24.Six quick heating filament typetubes are used. Two tubes are used in thetransmitter portion of the transceiver, two inthe receiver portion, and the remaining twoin the common audio section.The transmitting portion of the unit employsa 3A5 dual triode and a 3A4 pentode.
548 Receivers and TransceiversThe first triode section of the 3A5 serves asan overtone crystal oscillator, delivering 24Me. energy from an 8 Me. crystal. The crystaloscillates in series mode and sufficient regenerationis introduced in the circuit by returningthe crystal circuit to a tap on the oscillatorplate coil. The oscillator may be adjusted bytuning the slug of coil L,. The 24 Me. platecircuit of the oscillator stage is capacitivelycoupled to a 3A4 pentode tripler stage whoseplate circuit is tuned to 72 Me. This stageis resonated by means of the slug of coil L •.The 3A4, in turn, is capacitively coupled tothe second section of the original 3A5 whichFigure 23BATTERY POWERED "HANDlE-TALKIE" FOR 2 METERSCompletely self-contained transceiver employscrystal controlled transmitter forgreatest stability. Send-receive switch is attop right with tuning control of receiverdirectly below.<strong>THE</strong> <strong>RADIO</strong>serves as a doubler from 72 Me. to 144 Me.Tuning of the plate circuit is accomplishedby the slug of coil L,. Inductive coupling isused between L, and the quarter-wave whipantenna.The receiver employs a 5678 sub-miniaturepentode r-f stage to provide a better signal-tonoiseratio and also to reduce spurious radiationfrom the super-regenerative detector. Ata distance of fifty feet from the transceiver,the detector radiation is inaudible to a sensitivereceiver. In the interest of economy, the5678 can be replaced with a 959 "acorn" tubeif miniaturization of the unit is not desired.The super-regenerative detector uses a specialquench circuit which provides smooth operationyet allows the usual variable quench controlto be eliminated. Values shown in theschematic are optimum, and care should betaken to keep all leads short. Capacitor C isthe receiver tuning control and has a rangeof 140 Me. to 150 Me. If it is desired to reducethe tuning range, one or two plates canbe removed from C and coil Ls readjustedaccordingly. The 5676 can be replaced by a95 7 "acorn" tube in the interest of economy.Audio output from the super-regenerativedetector is passed through a simple R-C filterto suppress the quench frequencies. The firstaudio stage is a 1S5 pentode using contact potentialgrid bias. Interstage coupling is accomplishedby a Centralab Coup/ate printedcircuit incorporating the six components requiredat this juncture. If this item is notavailable, the values of resistance and capacitanceshown can be substituted. The outputstage consists of a 3Q4 power pentode drivinga 2 Yz -inch speaker through the output transformerT,. For reception only one-half of the3Q4 filament is used to conserve "A" and "B"battery drain. During transmit periods theextra audio is needed and the other half ofthe filament is energized.The "send-receive" switch S1A-B-C-D-E isa miniature ceramic rotary switch. Section Aswitches the antenna from the receiver to thetransmitter. Secion B energizes the filamentsof the receiver r-f tubes on the "receive" position,and energizes the microphone and thesecond section of the 3Q4 filament on the"transmit" position. Section C turns off allfilaments in the "off" position, and sectionD opens the 67.5 volt battery circuit in the"off" position. Section E connects the loudspeakerto the audio system in the "receive"position of switch S,. The extreme counterclockwiseposition of s, is the "off' position.
<strong>HANDBOOK</strong>;'Handie-Talkie" 549Under normal operating conditions thetransceiver is held like a telephone and a shortwhip antenna is used. The operating rangemay be extended by removing the whip andusing a beam, connected to the unit by alength of 52-ohm coaxial transmission line.Three separate battery voltages are requiredby the "handie-talkie." 1 Y2-volts is required forthe filament circuit, 4.5 volts is required forgrid bias of the 3Q4 audio stage, and 67.5volts is required for the plate voltage. Allthese voltages may be obtained from miniaturebatteries placed within the case of the transceiver.TransceiverLayoutNo hard and fast assembly procedureis given for this unit,since each builder will want tocustom-fit the circuit into his particular case.The unit shown in the photographs was builtwithin a steel box measuring 9" long by 2%"wide by 2 Ys" deep. The width and depth dimensionswere determined by the size of thebattery pack. The "front" of the cabinet isone piece with flanged edges. The microphoneand speaker are attached to this portion ofthe box. The transceiver chassis is custom designedto fit the available components and isheld to the front of the box by two 6-32bolts through the rear of the speaker frame.The rear and sides of the cabinet serve merelyas a dust shield as no components are fastenedto them, except for the coaxial antenna receptacle.The portion of the cabinet directly overthe batteries is separately removable to permiteasy access to the battery pack. The remainingportion of the cabinet is affixed permanentlyto the front piece by means of sheet metalscrews. The crystal projects through a cut-outin this piece, and there are three %-inch holes+ 20JJFO.~"'i"5cl'--------,ROFF~XHT\~RNALANTENNAT0~-= S1AII(144MC) Lo-B+ 67.5 V.e cA4,7K .Q.Q.l-=--o.5 c~---------~c~og,r 1.s .-4.5 +67.5Figure 24SCHEMATIC, "HANDlE-TALKIE" FOR 2 METERSCoil data is given in Figure 26.C,-JI JLJLfd. "butterfly". Johnson 3MBIIS,A-B-C-D-E-5 pole, three position rotary switch. Centralab PA-2015T,-JOO ohms to grid. Triad A-IXTz-5 K pri., 6.7 K and 4 ohm sec. Triad M-4Zf ,-Telephone-type carbon microphone.:;~.~ +1. 5 } VOLTS-67.5
550 Receivers and Transceivers<strong>THE</strong> <strong>RADIO</strong>through which the transmitter tuning slugsmay be adjusted. These holes are dosed withsnap-in hole plugs after adjustments are completed.A general view of the transmitter chassismay be seen in figures 25, 27, and 28. Noticethat a small shield passes across the r-f stagetube socket, and pin 3 of the socket is groundedto the shield. The shield isolates coil L,from tuning capacitor C, which is placed directlybelow it. The two audio stages andtransformer T, mount at the end of the chassis.The tubes mount parallel to the front panelon a small aluminum angle bracket. Directlybelow T, is the 5676 detector tube, also placedin a horizontal position.The crystal socket, the transmitter coils, thetwo r-f tubes, and the "send-receive" switchS, are mounted at the far end of the chassis.A small shield plate is placed between coilsL, and L, to reduce interstage feedback. Outputtransformer T, is located between the 3A4plate coil ( L,) and the audio tubes.The r-f stage coil L, is wound upon a slugtunedform made from a concentric trimmercapacitor. The outer electrode of the capacitoris removed, exposing the polystyrene formupon which the coil may be wound. The innerslug of the capacitor serves as the tuning slugof the coil.FIGURE 26-COIL TABLE FOR "HANDlE-TALKIE"L1-5 TURNS N° 18 E. (SEE TEXT)l2-2 TURNS N°22. INSULATED WIRE COUPLEDTO GRID END QF COIL l3.L3-fiTURNS N°11 E •• 3/8-INCH DIAMETER,1/2.-INCH LONG, AIR WOUND.l4- 2.0 TURNS N• 2.8 E., 1/2.-INCH DIAMETER,1/2.-INCH LONt;. TAP 9 TURNS FROM GRID END.RESONATES IN CIRCUIT AT 2.4 MC.l5- 7 TURNS N° 18 E., 3/8-INCH DIAMETER,1/2-IN. LONG. RESONATES IN CiRCUIT AT 72 MG.L6- 5TURNS N°16 E., 3/8-INCH DIAMETER,1/2-IN. LONG. RESONATES IN CIRCUIT AT 144 MC.TransceiverWiringWiring techniques are extremelyimportant when working onVHF equipment that must becompacted into a small space. It is recommendedthat a "pencil-type" soldering iron be used,along with small gage wire whose insulationwill not melt if it is overheated. All socketgrounds and filament wiring are completedfirst. Coils are wound and put into position.After the coupling capacitors and other r-fcomponepts are mounted in place, the tubesare placed in the sockets and the tuned circuitsare adjusted to frequency. It may be necessaryto alter the number of turns on coils L,, L, L,and L due to slight variances in circuit componentsand tube inter-electrode capacities.Figure 25VIEW OF"WALKIE-TALKIE"CHASSIS SHOWINGMAJORCOMPONENTSOblique view of choJsislooking towards switchS1• Output coil '-• is inforeground with 3AStube behind it. To rightof 3AS Is 3A4 tripletstoge. Coils l..4 and l..s orebehind 3A4. To right ofswitch s, is receiver tuningcontrol c,. Shieldseparates C, from coil1.., directly above it. R-famplifier tube 5687mounts on vertical plotesupporting 1..,. Shield belowr-f tube isolotes Itfrom detector componentsbelow the chossls.Audio tubes ore at right,with T,
Figure 27UNDER-CHASSISVIEW OFWALKIE-TALKIESHOWING MAJORCOMPONENTSChassis of transceiver isspecially bent to fit componentsin smallest possiblespace. Oscillator coilL, is at left separatedfrom triplet coil Ls bysmall shield. Output coilL, is to rear. 5676 detectortube is at right rear,flanked by 20 p,fd. filtercapacitor. Audio tubesare under chassis parti·tion at right. Quench filtercapacitors are toright of coil Ls.These adjustments may be done quickly andeasily with the aid of a grid dip oscillator.After the r-f circuits are adjusted the rest ofthe transceiver may be wired. Short, directleads should be used in the r-f circuits and allwires should be routed in the shortest possiblepaths.TransceiverAlignmentAfter all wiring has been checked,the tubes should be removedfrom their sockets and the filamentbatteries connected to the unit. The filamentvoltage at each tube socket should bechecked in all three operating positions ofswitch St. Make sure the second filament sectionof the 3Q4 (pin # 7) is energized in the"transmit" position of St. The bias batteriesand B-plus batteries may now be attached tothe proper leads. Be sure of the correct batterypolarity before inserting the tubes in the sockets.Test the "handie-talkie" for proper receptionfirst. The slug of coil L, should be adjust-ed for best reception after the transceiver isassembled and the whip antenna placed in thecoaxial receptacle. However, it is possible toplace a short piece of wire in the antenna receptacleto pick up a test signal while thetransceiver is dissembled on the bench. Thecoupling between L, and La should be adjustedfor best super-regenerative action and sensitivity.After the receiver is operating, the transmittershould be energized. Tune the slug of coilL. for reliable crystal operation. A grid diposcillator held in the proximity of L, will serveas a r-f indicator. Remove the plate voltagefrom the grid dip oscillator for this test. Alternatively,oscillator operation may be monitoredin a nearby receiver tuned to the 24Me. overtone frequency of the crystal. The3A4 tripler stage may now be tuned for maximumoutput on 72 Me. as indicated on thegrid dip oscillator. Finally, the grid dip oscillator(or wavemeter) is held in close proximityFigure 28TRANSCEIVERCHASSIS BOLTEDTO PANEL OFWALKIE-TALKIEThe chassis is held inplace by bolts passedthrough frame of smallspeaker. Modulationtransformer T: is visiblein foreground with Centralab"Coup/ate" at rearof audio tube sockeb,Batteries occupy spaceto left above microphone.Crystal socket is at farright, which is "top" endof transceiver.
552 Receivers and Transceiversto the 144 Me. doubler plate coil, 16. Thiscoil is tuned for maximum output at 2 meters.After this preliminary alignment is completedthe chassis of the transceiver may be placed inthe case, along with the battery power pack.FinalAlignmentExperience with several of theseunits has shown that maximumoutput is obtained with a whipantenna slightly longer than quarter wavelength.The transmitter stages should be retunedfor maximum field strength in a nearbywavemeter and different length whip antennasshould be tried. The quarter wavelength whipis nineteen inches long, but whip lengths upto twenty-seven inches should be tried. Ingeneral, whip lengths of twenty-four inches orso seem to produce best results with this littletransceiver. Input to the final stage of the3A5 is approximately )f-watt under optimumloading conditions.The operating range between two of theseunits varies from a thousand feet or so overobstructed terrain to better than three milesover smooth ground. Communication betweenthe "handie-talkie" and a base station equippedwith a sensitive receiver and a high gain beamcan be maintained over greater range.27-6 Six MeterTransceiverfor Home or CarThe six meter band has become extremelypopular within the past few years. The pinnacleof the sunspot cycle, combined with theinflux of new amateurs has brought life toan otherwise neglected amateur band. Mostof the stations operating on this band employ<strong>THE</strong> <strong>RADIO</strong>relatively low power and the amateur havingmodest power suffers no handicap if he has agood antenna.The growth of mobile operation on the sixmeter band has been rapid and the need hasarisen for a compact transceiver that will workwell either in the car or at home. The unitdescribed in this section has been designed tomeet this need.This compact transceiver operates in the 50-54 Me. range. It employs a stable, superheterodynetype receiver and a 10 watt transmitter.The transmitter may be either crystal controlled,or driven with an external variablefrequency oscillator. A 5-watt audio systemoperates from a high impedance crystal microphoneand provides 100% modulation of thetransmitter. In the receiver mode, the audiosystem delivers sufficient power to drive anexternal speaker well above the noise levelof the automobile.Small enough to fit comfortably under thedash of today's car, the transceiver delivers awell modulated signal at a total plate powerdrain of 100 milliamperes at 250 volts potential.Voltage regulation of the plate and filamentvoltages of the receiver oscillator ensuresthe maximum stability required for mobilework.TransceiverCircuitA block diagram of the transceiveris shown in figure 30.Eleven tubes are employed, plusa filament voltage regulator. Change-overfrom receive to transmit is accomplished by apush-to-talk circuit that operates two d-e relaysfrom the microphone control button.The receiver employs a broad-band r-f stageand two i-f stages. The r-f stage uses a 6CB6television-type pentode with pre-tuned grid andplate circuits. This stage is capacity coupledto a second 6CB6 mixer operating with gridinjection. The variable frequency oscillator isa third 6CB6 in a Tri-tet configuration. Theoscillator circuit ( L,-C,-C.) tunes the 17.0-17. 7 Me. range. The plate circuit of thisFigure 29DE-LUXE SIX METER TRANSCEIVERFOR HOME OR CAROperating from a separate power package,this compact transceiver is well suited forthe serious six meter operator. A 10 wattcrystal controlled transmitter and superheterodynereceiver are combined with adual purpose audio system in this smallcabinet. Receiver controls are at left andtransmitter controls at right. Meter servesas $-meter and transmitter tuning indicator.External VFO may be used with transceiver.
<strong>HANDBOOK</strong>stage is gang-tuned to the oscillator and coversthe third harmonic range of 51-53 Me. Theharmonic voltage is capacitively coupled to theinput grid circuit of the mixer stage. Seriespadding capacitors in the oscillator and triplerstages permit close tracking over the tuningrange.An i-f channel of 1 Me. is employed in thistransceiver. When the oscillator is consideredto be on the high frequency side of the receivedsignal, the tuning range is 50-52 Me. Whenthe oscillator is considered to be on the lowfrequency side of the received signal, thetuning range is 52-54 Me. The broad band r-fcircuits of the receiver "front end" cover thecomplete range of 50-54 Me. so that signalson either side of the injection frequency maybe detected. This principle is sometimes referredto as super-imposition tuning, whereinthe receiver is tuned to two frequencies simultaneously.For example, if the local oscillatoris tuned to 51 Me., simultaneous reception ofsignals on 50 Me. and 52 Me. is possible.Since most amateur activity takes place in thelower one megacycle of the six meter band,the chance of two signals appearing at onepoint of the dial that in reality are two megacyclesapart is remote.Super-imposition tuning greatly simplifiesthe design of a VHF receiver. Double conversionis not required, and the local oscillatorneed tune only a relatively narrow frequencyrange. Saving, in parts and circuit complexity,IS obvious.6-Meter Transceiver 553Two stages of one megacycle i-f amplificationare used for maximum gain. Five i-ftransformers are used to narrow the "skirt"response of the i-f strip. A 12AL5 serves asthe detector, a-v-e rectifier, and automaticnoise limiter. Filament voltage to the 12AL5is reduced to 10.5 volts to provide better noiselimiting action.Two tubes are used in the audio system ofthe transceiver. A 12AX7 dual triode servesas a two stage resistance coupled speech amplifier.Only one section of the 12AX7 isused for reception purposes. The dual triodeis capacity coupled to a 12AQ5 audio outputstage. A tapped output transformer matches the12AQ5 to the r-f amplifier plate circuit of thetransmitter section, and a low impedance windingon the transformer drives an externalspeaker for reception. The screen voltage ofthe 12AQ5 is reduced for receiving operationto conserve power supply drain.The transmitter section of the transceiver usestwo pentode tubes. The oscillator is a 6CL6in a "hot cathode" circuit employing 25 Me.overtone crystals. Switch s, in the grid circuitof the oscillator permits selection of an externalvariable frequency oscillator. The platecircuit of the oscillator ( Cs-L,) is tuned to the50 Me. second harmonic of the crystal. Aswitch ( s,) in the oscillator screen circuitpermits the operator to turn on the oscillatorstage during reception for '"zero-beat" or frequencymarking purposes. The oscillator is inductivelycoupled to a 5763 r-f amplifierR-F AMP.(SOMC)MIXERI·F(fMC.)1-F(IMC.)DET.,A-N-L.I '~ I I II I I I IL - - - - -- - -b-- - - - --0-y-- - - - - -+ - - --- --'~OMC. ANT.OSC., MULT.(51-53MC.)IR!.¥ADJ.: @s-METERII AND PUSH-TO- I I---+1-:_J TALK SWITCH I Ir f F I ~K__ :___ -b-~Y~c- ___ ___________ --~1I I RY2.,t_ __ ..JIIIIIJ I II'-----b-_,_ I6RY1B t._ JBt 250 V.Figure 30BLOCK DIAGRAM OF 10 WATT SIX METER TRANSCEIVER
554 Receivers and Transceivers<strong>THE</strong> <strong>RADIO</strong>®ANTENNACHANGEOVERRY1E:® .---TO!1763 SCREENCONTROL-c-;;NTO TRANSMITTER OUTPUT CIRCUITTO J 1 (ANrENNA)TO RECEIVER INPUT~763 SCREEN Cl RCUITTO RY2 -SECTION c© O ~OPENSPEAKERTO SPEAKER CIRCUITCONTROL ~-c=VITO T1 SECONDARY®AUDIOCONTROL®UJ RY2.> TO GAIN CONTROL, R3TOGRIO, 12.AX7§£:a:TO VOLUME CONTROL, R 2.E:TO 12AQ6 SCREEN12AQ 5 SCREEN B+ 2.$0CONTROLTO 47 K RESISTOR© r--B+CONTROLc=:MJB+SJAM1~B+TO 6CL6. RY1 SECTION 82!»0B+ TO RECEIVER(SEE F!,URE 32)SJBFigure 31RELAY AND METER CIRCUITSRelays are shown in unenergized position.stage, which is plate modulated by the 12AQ5audio amplifier. The r-f amplifier is bridgeneutralized by capacitor Co for greatest stabilityat the operating frequency range. Theplate circuit of the 5763 is pi-coupled to a52- or 72-ohm external load.Transfer from reception to transmission isaccomplished by means of two d-e relays actuatedby a push-to-talk switch in the microphone.The six volt relay coils are connectedin series for twelve volt operation. Relaychange-over operation is shown in figure 31.Relay RY -1 actuates antenna changeover,screen control, and speaker control circuits.Relay RY-2 actuates audio control, 12AQ5screen control, and B-plus control circuits.Both relays are shown in "receive" position infigures 31 and 32.Proper tuning of the r-f amplifier is accomplishedby a simple crystal diode voltmeter( 1N38A) connected in the antenna circuit.MsThe stage is tuned for maximum reading ofthe voltmeter.A two pole, four position rotary switch ( s,)serves to place the 0-1 d-e milliammeter acrossthe appropriate circuits in the transceiver. Inposition 1 (M•) the meter serves as a 0-500volt meter measuring the transceiver platevoltage. Position 2 ( M,) converts the meterinto a 0-10 d-e milliammeter measuring poweramplifier grid current. Position 3 (M,) monitorsthe r-f voltmeter output, and position 4(M,, M,) places the meter in the a-v-e controlledB-plus line of the receiver where itserves as a signal strength meter.Under normal driving and charging conditions,the voltage of the standard "12 volt"battery can vary from 11 volts to 14.5 volts.This fluctuation can raise havoc with the receiveroscillator stability. It is annoying, tosay the least, to have the receiver calibrationchange with variations of car speed and generatorvoltage. To eliminate this effect, anA mperite ballast tube ( R,) is placed in serieswith the filament circuit of the 6CB6 receiveroscillator tube. This current sensitive devicecompensates for voltage changes in the filamentcircuit, holding the voltage at the filamentterminals of the oscillator tube withina few percent of a nominal 6.3 volts.The six volt tubes of the receiver are connectedin series parallel for operation acrossthe 12-volt power supply. A compensating resistoris required across the filament of the5 763 to equalize the current drain with thatof the 6CL6 oscillator.Transceiver Layoutand AssemblyFigures 29, 33, and 34provide a general layoutof the transceiver.The unit is built upon an aluminum chassismeasuring 9" x 7" x 1 Y2" in size. This assemblyfits within a steel wrap-around type cabinet4Y2" high, and approximately the samelength and depth. The cabinet is formed ofperforated metal to ensure proper ventilationwhile at the same time reducing spurious radiationto a minimum.Viewed from the top rear (figure 3 3), thereceiver portion of the unit occupies the righthalf of the chassis and the transmitter andmodulator occupy the left half. The externalplugs and receptacles are mounted on therear apron of the chassis as is the modulationgain control, R,. These components projectthrough a cut-out in the rear of the steel enclosure.All the ceramic variable capacitors thatserve as circuit adjustment trimmers are flush
RECEIVER(51-53MC.)6CL6 osc.A.v.c..ogs L7•I~1nl~..,._--.--1~ RY1A =X~z0m00"TUBEVOLTAGE CHARTPLATE SCREEN CATHODE~!" 112 J!~~ fc 4.7K 1N38Ao:T"M4 =ecee R-F tas 1358C86 MIX. 100 4S8CB6 OSC. 2.SO 10568J8 l-F 250 10012AX712.AQS100100250* - SEE Flt:URE .J52501751.01.012•RY1DUMMY LOADBASE VIEWSOLDER SIX 5.3 VOLT, !SOMA.BULBS TOGE<strong>THE</strong>R. JUMPERCENTER TERMINALS TOt:ETH£R.RY2Figure 32SCHEMATIC, SIX METER TRANSCEIVERCoil data and i-f transformer modification is given in figure 35. Relay and meter connections are given in figure 31.c,-C,-C,--6-30 p.p.fd. Centralab 827CRY,, RY.--3 pole, double throw with 6.3 volt d-e coil. Advance #MG-3C6VAC,-C,-S,-C,.......3-12 p.p.fd. Centralab 8278T,-T,-1500 kc. i-f transformer. J. W. Miller Co. 12-WJc,-C.--15 p.p.fd. per section. All-Star Products Co., Defiance, Ohio. Model C3(see figure 35 for modification)c,,.. ...... 7 p.p.fd. variable "piston-type". Erie 5328T.--5 K pri., 6.7 K sec., and 4 ohm winding. Triad M5-Z~ ·u ....... Un __,,.luntl HF-35 R,-Amperite Ballast Tube #3TF-4A----L-.11. __ .,. _..,_.,.,,.,.__r,..J:I......;,. Chnn.i§: Co. :t:tLTC-46412. VOLTS.01c2,36CB6 OSC.-l....0::::lVIn(!)
556 Receivers and Transceivers<strong>THE</strong> <strong>RADIO</strong>mounted to the chassis and may be tunedfrom the top of the unit, with the exceptionof the receiver oscillator trimmer C which ismounted below chassis to conserve space.Below the chassis, a small aluminum shieldmeasuring I )f" long and I~~" high is positionedbetween the oscillator output coil L, and thereceiver r-f amplifier plate coil L,. This is theonly interstage shield required in the transceiver.The two transfer relays RY-I and RY-2are mounted in the under-chassis area beneaththe modulation transformer T,. Betweenthe relays is placed an 8-terminal phenolicmounting strip. To the side of this strip isplaced the midget S-meter potentiometer R,.All sockets and trimming capacitors are mountedin place using 4-40 hardware with solderinglugs placed beneath the nuts in variousconvenient locations.TransceiverWiringThe wiring of the unit is quitesimple if done in the propersequence. The underchassis areacontains many small components but these neednot be crowded, provided proper care is takenin the layout and installation of parts. Be surethe i-f transformers have been modified accord-ing to the data of figure 3 5 before wiring isstarted.The majority of components are installedbetween the socket pins of the various tubesand five large phenolic terminal strips. One ofthese strips (previously mentioned) is placedbetween the two transfer relays. A four terminaland a six terminal tie-point strip aremounted in line running parallel to the i-ftube sockets of the receiver, towards the chassis-edgeside of the sockets. The i-f stage componentsmount between these tie-points andthe pins of the 6BJ6 and 12AL5 tube sockets.A fourth terminal strip ( 8 tie-points) is parallelto and directly beside the edge of thechassis adjacent to the r·f tube sockets of thereceiver. The smaller components of the r-fsection mount between this strip and the pinsof the 6CB6 sockets.The r-f coils of the receiver section (L, L,,L,, and L.) are mounted directly to the ceramictrimming capacitors associated with each coil.The interstage coupling coils ( L, and L) ofthe transmitrer section are mounted to theirrespective trimming capacitors and may beseen in the center of the chassis in figure 34.Figure 33REAR VIEW OFSIX MHERTRANSCEIVERCHASSISReceiver section of unitis at right with trimmercapacitors seen on chassisdeck. Center sectionaf three gang variablecapacitor is unused.Transmitter output controlsare located at upperleft. Filament volt·age regulator tube isat rear of chassis, justabove modulation levelcontrol potentiometer.5763 amplifier tube isbehind modulation trans•former, with shielded6CL6 tube to right. 1-fstrip runs along centerof chassis. Antenna andspeaker plugs and powerreceptacle are along rearlip of chassis. Receiveraudio control is at upperright of front panel.
<strong>HANDBOOK</strong>6-Meter Transceiver 557After all major components have beenmounted to the chassis the various socketground connections should be made, followedby the filament wiring. The speech amplifiershould be wired next, taking care to placesmall components directly between socketpins to conserve space. All the small componentsof the i-f system can be mounted abovethe tube sockets, and in the space between thesockets and the tie-point strips. The powerleads from the wired sections of the transceivershould now be run to the two relays.The next step is to complete the receiverwiring. Shielded leads are run to the volumecontrol R, mounted in the upper-left handcorner of the receiver panel (as viewed fromthe front) . The S-meter wiring circuit shouldbe completed as the final wiring step. Notethat the case of potentiometer R, is "hot" andshould be insulated from the chassis with fibrebushings.ReceiverCheck-outThe receiver may be tested beforethe transmitter wiring isdone. Relays RY-1 and RY-2are in an unenergized position during receptiO'!l.Examine both relays to make sure thatthe back contacts (normally closed) are ingood operating condition.FIGURE 35COIL TABLE FOR SIX METER TRANSCEIVERl1, l2., l4- 8 TURNS N°18 E., 3/8" DIAM., 1/2" LONG.l3-7TURNS N°16 E., 1/2" OlAM., 1/2" LONG.(SeW COIL STOCK )l5, l6-10 TURNS N'"22 E., 5/16" DIAM., 5/16" LONG.ADJUST SPACING BETWEEN COILS FOR MAXIMUM GRID DRIVE. COILS MOUNT END TO END.l7-6TURNS N° 16 E., 1/2" DIAM., 5/8" LONG. AIRWOUNO.T!,T5- REPLACE 100 lJlJFD PADDING CAPACITORSWITH 200JJJJFD CAPACITORS TO RESONATEWINDINGS TO 1 MC.A power supply capable of delivering 250volts at 100 milliamperes and 12.6 volts at3 amperes is required to test the transceiver.A 4-ohm loudspeaker is also required. Thereceiver is energized and the important voltagepoints are checked with a high resistance voltmeterand should comply closely with thevalues of figure 32. When the receiver sectionis determined to be in operational order, atone modulated 1 megacycle signal shouldbe loosely coupled to the plate circuit of the6CB6 mixer tube (pin 5) and the slug coresof all i-f transformers tuned for maximumaudio signal. As the stages are bought intoresonance, the input signal to the mixer tubeshould be reduced to prevent over-excitationof the stages.Figure 34UNDER-CtiASSISVIEW OFTRANSCEIVERChangeover relays arelocated at the left edgeof the chassis. The i-famplifier runs down thecenter of the chassis,with the receiver r-fstages to the right.Transmitter stages arelocated between front relayand i-f strip. Audiostages are placed alongrear of chassis. Smallcomponents are mountedbetween socket pins andadjacent phenolic terminalstrips.Note: filament circuit isdesigned for d.c. operation.If a.c. filament operationis desired, highresistance relay coils operatedfrom the highvoltage supply should beemployed,
558 Receivers and Transceivers<strong>THE</strong> <strong>RADIO</strong>The next step is to check the tuning rangeof the high frequency oscillator, which shouldbe 17.0-17.7 Me. The range may be checkedwith an accurately calibrated receiver or frequencymeter. The fixed trimming capacitordetermines the overall frequency span of theoscillator, and padding capacitor C determinesthe range setting. Padding capacitor C is adjustedso that 17.0 Me. occurs when the maintuning dial is set to five degrees, allowing aslight amount of "overlap" at the band edges.As the main tuning capacitor setting is decreasedthe oscillator will tune to 17.7 Me.If it is desired to widen or diminish the tuningrange, the value of the series capacitor may bechanged slightly. Once the proper oscillatorrange has been established, the tripler circuitshould be adjusted to track with it. The maintuning capacitor should be set to 17.7 Me. andpadding capacitor C, adjusted so that the tunedcircuit resonates at 53 Me., as checked with agrid dip oscillator. The main tuning dial isthen reset to place the oscillator on 17.0 Me.;trimming capacitor C, is adjusted so that thetripler circuit resonates at 51 Me. Once thetwo circuits have been brought into roughalignment they may be placed "on the nose"by adjusting C, and Co for constant grid currentin the mixer stage across the tuningrange. Temporarily unsolder the grounded terminalof the 0.1-megohm grid resistor of the6CB6 mixer tube (pin 1 ) and place a 0-100micro-ammeter in series with the resistor toground. Adjust C, at the high frequency endof the tuning dial and Co at the low frequencyend until the rectified grid current is relativelyconstant across the band. Under proper operatingconditions, the reading should be about15 microamperes.The last step is to align the tuned circuitsof the r-f stage. These are relatively broad bandand need only be peaked over the usual frequencyrange of operation. For most purposesthey should be resonated at 50.5 Me. for optimumoperation at the low frequency end ofthe band. Signals above 52 Me. will be slightlyattenuated by this adjustment. If operation atthe high frequency end of the six meter bandis desired, the circuits should be peaked at 53Me. with a consequent slight reduction of performancein the 50 Me. region.Completion of theTransmitter SectionOnce the receiver isjudged to be in goodoperating condition, attentionshould be turned to the transmitter.The lead from the XT AL-VFO switch s, tothe v-f-o input receptacle } on the rear of thechassis should be made of coaxial cable, asshould the lead from relay R Y -1 to the antennareceptacle J,. In addition, the lead fromthe receiver antenna input circuit to relay RY-1should be shielded.The majority of small components of boththe oscillator and amplifier stages may bemounted by their leads between socket pinsand a nearby 5-terminal phenolic tie-pointstrip. The plate coil L, of the 5763 amplifierstage is mounted above chassis, directly behindamplifier tuning capacitors Cn and C12.Small National Co. polystyrene feed throughbushings are used to support the coil. Neutralizingcapacitor Co is soldered to a rotor terminalof plate tuning capacitor Cn. The leadfrom Co to the "cold" end of L, passes througha }4-inch hole drilled in the chassis. To insurea good ground return, the rotors of Cn andC" are connected to the chassis with a short,heavy coppe~ strap.The final step is to complete the wiring tothe meter switch and the relays as outlined infigure 31. When this is completed, all thetransceiver wiring should be completely checkedand the chassis thoroughly cleaned of loosebits of solder, wire, etc.TransmitterAdjustmentIt is necessary to close both relaysto test the transmitter operation.For the time being,they may be wedged close with a bit of matchstickif a 12-volt d-e supply is not readilyavailable. A 25 Me. crystal should be pluggedin the panel holder and switch S, set to theXTAL position. Switch S, is set to OSC position,and the lead from the screen circuit ofthe 57 63 amplifier tube to the contact of relayRYl-B is temporarily opened, removingscreen voltage from the r-f amplifier. Filamentand plate voltage is applied to the transceiverand capacitor C, is tuned for oscillation of thecrystal. The meter of the transceiver is switchedto the GRID (M,) position and capacitor C,adjusted for maximum grid current. A readingof half-scale should be obtained. The couplingbetween L, and Lo can be varied slightly toobtain the proper reading.The dummy antenna (figure 32) is nextplugged into antenna receptacle ], and loadingcapacitor C,, set at full capacity. The plate circuitof the amplifier stage should be resonatedat 50 Me. with the aid of a grid dip oscillatorcoupled to coil L,. The turns of L, should beadjusted so resonance occurs at half-capacitysetting of tuning capacitor Cn. After this preliminaryadjustment has been made, capacitorell should be tuned through its range while
<strong>HANDBOOK</strong>28-Mc. Transceiver 559carefully noting the grid current of the amplifier.Unless neutralizing capacitor Co has beencorrectly set by a lucky accident, the grid currentreading will show an abrupt kick as capacitorCn is tuned through resonance.Thesetting of the neutralizing capacitor should beslowly varied so as to minimize the kick ofgrid current. After change of setting of Co,grid tuning capacitor C,, should be retuned formaximum grid current. An adjustment of Cowill soon be found that will reduce the gridcurrent variation to an imperceptible kick ofthe meter. When this point is found, the rotorof Co should be fastened in place with a dropof nail polish and the screen lead to R Y -1should be resoldered.Switch S, is now set to the transmit positionand power is again applied to the transmitter.The meter switch is placed in the OUTPUT (M,) position, and plate tuning capacitorCn adjusted for maximum meter reading.At the same time, the bulbs of the dummy antennashould glow. Capacitors Cn and Cu aretouched up for maximum indication of themeter.In order to test the modulator section of thetransmitter when using an a-c filament supply,it is necessary to disconnect the push-to-talkcircuit at pin # 2 of the microphone plug;otherwise there will be a loud hum on theaudio signal. When this is done, the microphonemay be plugged into the jack and thegain control ( R:,) advanced. The dummy antennashould increase in brilliance under propermodulation. If the coupling to the dummyload is too tight downward modulation willresult, and the bulb brilliance will drop. Thecapacity of loading capacitor c, should be increased,dropping the degree of loading.The transmitter is designed to operate intoa coaxial transmission line having an impedancevalue between 50 and 75 ohms. When ahigh value of standing wave ratio exists onthe line the impedance presented to the pinetworkamplifier may be of such magnitudeas to preclude the possibility of a propermatch. It is permissible to add a second micacapacitor in parallel with loading capacitor c,to extend the operating range of the pi-network.If more than 50 1-'1-'fd. has to be added,the SWR ratio should either be lowered byproper adjustments to the antenna, or thelength of the transmission line between thetransciver and antenna should be altered inorder to present a more reasonable load tothe amplifier circuit. Maximum carrier poweroutput when properly tuned is about 4Y2 watts.All r-f adjustments should be carried out withthe purpose of cbtaining maximum reading onthe output meter of the transmitter under agiven set of antenna conditions. Under theseconditions, maximum transceiver plate currentis 100 milliamperes at a plate potential of250 volts.The MeterCircuitA final word should be saidabout the metering circuit of thetransceiver. The particular meterused has a d-e resistance of 100 ohms, and afull scale reading of one milliampere. In orderto simplify the problem of obtaining metershunts, the meter is converted into a 0-1 voltmeterby adding a 900 ohm resistor in serieswith the meter in the grid current position( Ma) . The meter is placed across a 1000 ohmshunt in this position. Since the resistance ofthe meter circuit and the shunt are equal, onehalfthe grid current flows through the meter,and one-half through the shunt. Thus, whenthe meter reads full scale (one milliampere),two milliamperes of grid current are flowingin the grid return circuit.When the meter is switched to S-METERposition ( M,, M,) of switch Sa, the meter isplaced in a bridge circuit, the variable leg ofwhich is the internal resistance of the a-v-econtrolled, i-f amplifier tube. The bridge isbalanced by proper setting of R.; meter sensitivityis set by the value of the series resistor( 2.2K). Dropping the value of this resistancewill increase the sensitivity of the meter.When S., is set to the VOLTS position, themeter is placed in series with a 500 K resistor( M,) and is converted into a 0-500 volt meterto check the operating potential of the platesupply.27-7 A "Hot"Transceiverfor 28 MegacyclesThis little transceiver is vivid proof thatDX can be easily worked on the 10 meterband. During a test period of two weeks, usinga dipole antenna, over SO foreign phone contactswere made in four continents, includingthree contacts with India. Power input to thetransmitter section of the unit was under 10watts. Admittedly, such DX is not a dailyfeature with such low power, but it doesprove that a few watts into a well situated antennacan work wonders on the 28 megacycleband.The 28-10 transceiver (28 megacycles, 10watts) is designed to operate over the range
Figure 36TEN METER TRANSCEIVER HASSIMPLE CONTROL PANELUsed with a dipole antenna, this 10 watt28 Me. transceiver has made three contactswith India. Main tuning dial of receiver isat center with send-receive switch below it.At upper right is combination S-meter andtransmitter tuning indicator. Transmittertuning controls are at left with "push-tozero"switch near receiver tuning dial.Unit features self-contained power supplylor II 5 volt operation.of 28.0-29.7 Me. The receiver portion employsfive tubes in a single conversion circuit.Two tubes do double duty as the audiosection for reception and the modulator fortransmission. Four tubes are used in the r-fsection, making a total of eleven tubes plus avoltage regulator. A voltage doubler seleniumrectifier supplies plate voltage for the completetransceiver. The whole unit weighs less thaneleven pounds so it is readily transportable,even by air. The power source is 115 volts,50-60 cycles, a-c.TransceiverCircuitA block diagram of the transceiveris shown in figure 3 7.Change-over from receive totransmit is accomplished by a ceramic rotaryswitch, S,. The receiver employs one r-f stageand two i-f stages for optimum selectivity andsensitivity. The r-f stage employs a low drain6B]6 remote cut-off pentode with tuned gridand tuned plate circuits. The plate of the r-ftube is shunt fed through a 10 K compositionresistor and the tuned circuit is placed in thelow potential grid circuit of the next stage. A6U8 is employed here as a combined mixerand oscillator. Grid circuit injection is used.The triode section of the 6U8 serves as thelocal oscillator, tuning from 26.5-28.2 Me. for10 meter coverage. A three section variablecapacitor C1A-B-C simultaneously tunes theR.F. AMP.(28MC.)MIXER, OSC. I. F.(!.SMC.)I. F.(T.SMC.)DETECTOR,A·V·C, A-N-L,---~6BJ6)--------{ 6U6 }-------{ 6BJ6}-------{6BJ6 )---------1REGULATOR 6IL..-XMTR V·F·O(7.0-7.25 MG.)y__ .J_ ___ ,IIIr----,: VSlC__ .....:,_ ____...J_ _______ _jDOUBLER R·F AMPI I (28MC.)I I)--~: ~2~8~M~C·~:~--16AQ5)-------------~--{9006III L- _j I '---------,S1BIIStAR·FINDICATORI,---------~~------------------~----+---------------__1IIII : AUDIO AUDIOI J 3L-----::1 --{6BJ6)-------{6AQ5)--..!...-c-" 0I 1 IL..------------+-- _),....-I- __ JStE~-~ PHONES'15 v.""'Figure 37BLOCK DIAGRAM OF TEN METER TRANSCEIVER
<strong>HANDBOOK</strong>28-Mc. Transceiver 561r-f, mixer, and oscillator stages. Series paddingin the r-f and mixer circuits is employed forgood tracking across the band.An i-f channel of 1.5 Me. is used in thisunit. This particular frequency was chosen toeliminate the troublesome "image" problem,so prevalent on the 10 meter band when theusual 45 5 kc. i-f channel is employed. Selectivitysuffers a bit when the higher frequencychannel is used, but a total of eight tunedcircuits (four transformers) provide an acceptablepassband for voice reception. A 6AL5double diode serves as the detector, a-v-e rectifier,and automatic noise limiter. The a-n-1is left in the circuit at all times.The audio section of the 28-10 uses twotubes. A 6BJ6 serves as a resistance coupledpentode voltage amplifier which is coupled toa 6AQ5 power amplifier. A tapped outputtransformer matches the 6AQ5 to the r-f amplifierplate circuit of the transmitter section,and a low impedance winding on the transformermatches the audio system to externalearphones or speaker. The input circuit of the6BJ6 forms an R-C dividing network isolatingthe microphone input from the audio outputcircuit of the receiver. No switching is requiredin this circuit.The transceiver is designed to be used witha low impedance carbon microphone. Voltagefor microphone operation is taken from a tapon the cathode circuit of the 6AQ5 amplifierstage. This circuit is broken by switch SlDduring reception. At the same time, switch sectionS1E connects the earphone circuit to thelow impedance winding on the output transformerT •.Three tubes plus a gas-type voltage regulatorand a diode r-f indicator are used in thetransmitter portion of the transceiver. A 6BJ6serves as a "Tri-tet" oscillator covering therange of 7.0-7.2 5 Me. The plate circuit is slugtunedto the 14 Me. second harmonic. Aswitch ( S,) in the plate circuit of the oscillatorpermits the operator to turn on this stage for"zero-beat" or frequency marking purposes.The oscillator is capacitively coupled to a6AQ5 doubler stage whose plate circuit istuned to the 28 Me. region. Switch segmentS1C removes the plate voltage from this stageduring reception and applies it to the r-f sectionof the receiver. The plate circuit of the6AQ5 doubler stage is capacitively coupledto a second 6AQ5 serving as the modulatedamplifier. This latter stage is bridge neutralizedby capacitor C, for stability at the operatingfrequency. The plate circuit of the amplifieris a simple pi-network designed to match loadsof 50 to 7 5 ohms. The external antenna isswitched between the transmitter and the receiverby means of switch segment SlA. Aseparate section of the transfer switch (S1B)removes plate voltage from the 6AQ5 r.f.amplifier stage and applies an audio load to themodulator during reception. This auxiliaryloading prevents audio feedback when theheadphones are removed from the jack ;,, andpermits the use of headphones of any impedanceto be used without the danger of spuriousfeedback. The overall gain of the audio systemis quite high and it is built in a smallspace. As a result, it does not have the reserveof stability that would normally be expected.The switching system, too, tends to createsmall feedback loops that must be carefullycontrolled. The chief cause of audio instability(or feedback) in this case is due to the closeproximity of transformers T, and T, above thechassis. Feedback can be reduced or enhancedby reversing the polarity of the secondarywinding of T,. The additional audio filteringshown in the schematic of figure 3 8 reducedthis tendency to a minimum. As a final precaution,a small shield (cut from a segment oftin can) was soldered to the top of the coreof T,. The shield projected downwards overthe windings of the transformer, as seen inFigure 39. A more expensive solution wouldhave been to employ a shielded transformer inthis portion of the circuit. A similar shield issoldered to the core of the filament transformerto reduce hum pickup by the adjacent audiotransformers.The a-c operated power supply employs two"replacement type" selenium rectifiers in avoltage doubling circuit delivering 2 50 voltsat 100 milliamperes. A small filament transformer(T,) provides 6.3 volts for the tubesand pilot lamp. This particular voltage doublerhas the negative side of the high voltage incommon with one side of the primary line.As a result, there is a fifty-fifty chance thatthe chassis of the transceiver will be "aboveground" by the amount of the line voltage,which in this case is 115 volts. This can be alethal situation if permitted to exist. A practicalsolution is to remove the "ground" sideof the transmitter power cable from the usualtwo wire line, and connect it directly to theexternal ground, as shown in figure 41. Thisensures that the chassis of the transceiver is atground potential. The "hot" side of the linereturns to one pin of the line plug. If the plug
100K 220o:-5 0.5RSlAANTENNA6BJ6 100 4 6BJ6 6BJ6 6AL5-.;;--'i$fT3T4Jl:;f c -=B+AVC22K 30KMETER CIRCUIT""0:5t6usOSCILLATORSlF1.1.!5_0.51M 250K 250K 47Ko.s ~"'()."5 Q.5"'~...,;;o('1)()('1)
<strong>HANDBOOK</strong>28-Mc. Transceiver 563FIGURE 388COIL TABLE,10 METER TRANSCEIVERL 1- ANTENNA: 2. TURNS N° 18 INSULATED WIREAROUND \\COLD" END OF L1. GRID: 14TURNSN°18 E., 1/4" DIAM., 112."' LONG. J.W. MILLER 1447-FL2- 13 TURNS N° 18 E., SAME AS L 1L3-7 TURNS N° 18 £. 1 SAME AS L 1, TAP AT 3 TURNSFROM !;ROUND END.L4- 13TURNS N° 16 E •• 3/4• DIAMETER, 1/2" LONG..NATIONAL XR-~Z FORM.L5- J.W. MILLER FORM 1447-F, 114• OIAM., 1/2."LONG CLOSE WOUND WITH N° 2.8 E. FULL LENG.THOF SPACE.L6- 14 TURNS N°18 E., SAME AS L1L7- 1 TURNS N° 18 E., 5/8"' DIAM., 3/&"' LONG (8&WSTOCK)VOL TAG£ MEASUREMENTSTUBE PLATE SCREEN CATHODE8BJ6 R-F 180 1~0 2 .•8U8MIX. 280 18~osc. 1006BJ8 \·F'S 18~ 140 38AQ!)AUDIO280 280TX-12.!:»RX-17is inserted in a wall socket incorrectly nothingwill happen. The transceiver simply refuses tofunction, and no dangerous situation is created.Simply reversing the plug will energize thecircuit, with no danger to the operator. Thedual power lead connection, therefore, is asimple and foolproof system of protection andpermits the use of the inexpensive and lightweight selenium supply. The only other practicalalternative is to employ a power transformerwhich will add excessive weight andbulk to the package.Because this hazard exists with any radioequipment of the so-called ac-dc type, manyof the newer buildings are being wired withpolarized power receptacles having a separategrounding pin. The use of this new wiringtechnique will help to reduce the inherent"shock potential" of the practical and efficienthalf-wave rectifier systems, of which this is anexample.Transceiver Layoutand AssemblyThe component placementmay be seen infigures 39 and 40. The28-10 transceiver is built upon an aluminumchassis measuring 11" x 8Y2" x 2" in size,and fits within a steel wrap-around type cabinet5%" high. The cabinet is formed of perforatedmetal to ensure proper ventilation. Radiationof harmonics from the assembly is reducedto a minimum with this type of eoclosure.Layout of the major components above thechassis may be seen in figure 39. The receiverFigure 39REAR VIEW OFTRANSCEIVERCHASSISTuning capacitor of variablefrequency oscillatoris in foreground of photo,panel driven by an extensionshaft. To the leftof the capacitor is theadjustable slug core ofthe oscillator coli. Alongrear edge of the chassisare the selenium rectifiers,the filament transformerand the high voltagefilter capacitor.Transmitter tubes are atright of chassis, with amplifiertube next topanel. At far right Isvoltage reflulator tube.Audio section is at upperleft with 9006 dioderectifier in front of theinput audio transformer.1-f amplifier is at centerchassis, parallel to frontpanel.
564 Receivers and Transceiverssection of the unit occupies the center sectionof the chassis with the power supply directlybehind it. To the right is the transmitter sectionwith the variable frequency oscillator atthe rear. Panel control of the oscillator is accomplishedby an extension shaft and a flexiblecoupling. At the left of the chassis is theaudio system.Placement of the main tuning dial is dictatedby the height of the three-gang tuningcapacitor shaft above the chassis deck. In thiscase, it is necessary to cut a small notch in thesoft aluminum chassis to permit the dial driveas~embly to drop low enough to permit the dialshaft to align with the capacitor tuning shaft.No interstage shields are required belowthe chassis. Filter choke CHI mounts at therear of the chassis, directly below filamenttransformer T,. The 100 J.LJ.Lfd. filter capacitormounts to the left wall of the chassis belowand to the left of v-f-o tuning capacitor c,.The three slug-tuned coils of the receiver maybe seen in the photograph. Coil L, is locatedbehind change-over switch s,. Coil L, is towardsthe middle of the chassis, and adjacentto the 10 J.Lfd. audio decoupling capacitor. CoilL, is to the left of L, and between the r-f tubeand mixer tube sockets.<strong>THE</strong> <strong>RADIO</strong>The transmitter oscillator coil L is placed atthe left rear corner of the chassis. The adjustableslug of this coil is visible in figure 39between the selenium rectifiers and tuning capacitorC,.To facilitate wiring, four phenolic tie-pointterminal strips (six lug type) are placed beneaththe chassis. One is located parallel tothe edge, just to the left of the 6AQ5 buffertube socket. A second is placed parallel to thefirst, to the right of the same socket. The latterstrip is used for connections to the r-f circuitryof the receiver section. The third strip is placedparallel to the rear of the chassis just in frontof the i-f strip and is used for i-f connections.The last strip is located between the 6AL5 tubesocket and the main filter capacitor. The noiselimiter and a-v-e components are mounted tothis strip.To secure maximum stability it is necessaryto firmly mount the oscillator tuning capacityto the chassis. An aluminum block measuring1 Ys" x 1" was therefore cut from a section ofY2-inch aluminum stock and used as a supportblock for capacitor c,. The capacitor is boltedto the chassis by long 6-32 machine screwswhich pass through the block and are firmlybolted beneath the chassis.Figure 40UNDER-CHASSISVIEW OFTEN METERTRANSCEIVERThe transmitter componentsare along theleft side of the chassis.Receiver components areat the center, with r·fstage nearest the panel.F /Iter choke is mountedbelow chassis at rear.Audio stage componentsare grouped in upperright corner of chassis.See text for placementof the smaller components.
<strong>HANDBOOK</strong>28-Mc. Transceiver 5653-WIRE HOUSEHOLDWIRING CIRCUIT1~"HOT"115 0 115LEAD OF TRANSCEIVERPOWER SUPPLY"GROUND'' LEAD OF TRANS--- ---- -....r::-- CEIVER POWER SUPPl-Y-=Figure 41"GROUNDED NEUTRAL"WIRING SYSTEMMost dwellings have a three-wire groundedneutral system. 115 volts o-c con be obtainedfrom either side of the system toground. Separate ground lead o'! transceiveris always connected before /me plugis energ'zed. This eliminates usual 11 CJC·dc"TransceiverWiringshock hazard.The sequence of wiring followsmuch the same pattern as thatof the six meter transceiverdescribed in section 27-6. Small componentsare installed between tube socket pins, or betweensocket pins and adjacent terminal striplugs. Socket ground connections are made firstand then the filament wiring is completed. Thepower supply wiring can be done first.When the supply is completed it should betested. A 2500 ohm, 25-watt resistor shouldbe attached between the output of the supplyand ground. A full 250 volts should be developedacross this resistor with 115 volts a-capplied to the power supply.Next, the i-f and second detector section ofthe receiver should be wired. Short, directleads and the use of the adjacent terminalstrips will reduce crowding in this area. Theaudio stages should now be wired, along withswitch segments S1B, S1D, and S1E. Whenthis is completed, the i-f section and audio ofthe transceiver may be tested. A test speakershould be connected to jack Ja, and a 1.5 Me.tone modulated signal from a test oscillator isloosely coupled to the plate pin ( #6) of the6U8 socket. The top and bottom slugs of thei-f transformers are now adjusted for maximumoutput signal. As the stages approachaiignment, be sure to decouple the signal generatorto prevent overloading.After this portion of the transceiver hasbeen tested, the r-f circuits of the receivershould be wired. Coils L,, L,, and L are woundand the coil forms clipped in place. The frontsection of tuning capacitor C1A is employedfor the grid circuit of the r-f stage, the middlesection is for the interstage r-f circuit, and therear section is for the oscillator section of the6U8 tube. Connections between the coils, thetuning capacitor, and the tube sockets are madewith # 16 tinned solid wire. In particular, careshould be taken to prevent movement of theleads and components in the oscillator sectionas any minute vibration of this part of thecircuit would lead to receiver instability.Sufficient coupling exists between the wiringof the oscillator and mixer circuits to providethe proper level of injection, and nocoupling capacitor is required. The final wiringstep is to wind the antenna coil on formL. One end is attached to a nearby socketground lug and the other end goes to the"receive" contact of transfer switch S1A.When the receiver is completed, a 28.5 Me.may be injected into the antenna circuit andthe main tuning dial set near maximum capacity.The slug of receiver oscillator coil L,is now adjusted until the signal is received neara dial setting of thirty degrees. After this,coil L, and coil L, are peaked for maximumsignal strength. Dial calibration should waituntil the transmitter section is completed.Completion of theTransmitter SectionThe oscillator coil shouldbe wound and mountedin position. The leadfrom coil L to tuning capacitor C, is made of# 14 solid copper and passes through a ~-inchhole in the chassis. All oscillator componentsare firmly mounted to reduce vibration to aminimum. Coils L, and L are wound andsnapped into position as the buffer stage isbeing wired. The p-a plate coil ( L,) is mountedabove chassis between the 6AQ5 amplifiertube and the front panel. The plate (or "hot")end of the coil is supported on a l/2-inch ceramicinsulator and the coil lead passes througha ;4-inch hole in the chassis to the stator oftuning capacitor c,, mounted directly belowthe coil. The opposite end of coil L, is attachedto a polystyrene feed-through insulator. Neutralizingcapacitor c, is affixed at one end toa terminal of coil form Lo. The other end attachesto the plate r-f choke, as shown in theunder-chassis photograph. The output leadfrom transfer switch S1A to the antenna receptacleon the rear of the chassis is made froma short length of coaxial cable. When thewiring is completed all connections shouldbe checked and the chassis thoroughly cleanedof solder bits, pieces of wire, flux, etc.TransmitterAdjustmentThe 6BJ6 oscillator tube shouldbe placed in the socket, alongwith the OA2 voltage regulator.Transmitter frequency control capacitor C, is>et near full setting and the slug of oscillator
566 Receivers and Transceiverscoil L. is adjusted so as to place the oscillatorfrequency on 7.1 Me. The fourth harmonic ofthe oscillator will then be on 28.4 Me. Atminimum tuning capacitor setting the upperfrequency limit of the transmitter will beslightly above 29.7 Me. The 6AQ5 tubes arenow placed in their sockets, and a 0.5 d-e milliammeteris temporarily connected acrossmeter shunt R, in the grid circuit of the 6AQ5final amplifier stage. The screen lead (pin 6)of the 6AQ5 amplifier tube is temporarilyopened. Transfer switch s, is placed in the"transmit" position and the v-f-o tuning capacitorC, is set to 29 Me. The transmitter isenergized and the slugs of oscillator coil L,and buffer coil L, are adjusted for maximumgrid drive to the amplifier (about two milliamperes).Amplifier loading capacitor c, isset at full capacity, and tuning capacitor Ca istuned through its range while carefully notingthe grid current reading. The reading will showan abrupt kick as Ca is tuned through resonance.Next, the setting of neutralizing capacitorc, should be slowly varied by means of afibre-blade screwdriver so as to minimize thekick of grid current. After each movement ofCo, the slug of coil L, should be reset for maximumgrid current. A setting of Cs can readilybe found that will provide a minimum valueof grid current change as capacitor Ca is tunedthrough resonance. The last step is to resolderthe screen lead to pin 6 of the 6AQ5 amplifiertube socket.TransceiverOperationThe transmitter may now beattached to a suitable antennaor dummy load (see section27-6) for an operative check-out. Receivercalibration should be rechecked, and the dialmay be calibrated. The transfer switch shouldbe set to the transmitting position and thetransceiver loaded into the antenna system. Themeter now indicates the r-f voltage rectified bythe 9006 diode attached to the output of thepi-network system. Tuning ( Ca) and loading(C.) should be adjusted to provide a maximummeter reading, with Ca always being setlast for the final "touch up" adjustment. Maximumcapacity setting of C. provides minimumloading and vice-versa.It may be necessary when working into lowimpedance loads, or transmission lines havinga high value of SWR to parallel loading capacitorC. with an auxiliary 100 J.LJ.Lfd. micacapacitor to obtain optimum loading. This willhave to be determined by experiment.A mobile carbon microphone may now beplugged into jack],. An indication of the modulationlevel can be obtained by noting theincrease in brilliance of the lamps of the dummyantenna. Overloading will tend to causedownward modulation. The degree of modulationmay be varied by moving the microphoneaway from the lips. Using a telephone-typeF-1 unit, optimum modulation occurs whenspeaking in a normal tone about three inchesfrom the microphone. The transceiver outputis about six watts fully modulated with a poweramplifier input of ten watts.
CHAPTER TWENTY-EIGHTThe exciter is the "heart" of the amateurtransmitter. Various forms of amplifiers andpower supplies may be used in conjunctionwith basic exciters to form transmitters whichwill suit almost any need. Of great interesttoday are simple SSB exciters for fixed andmobile operation. These may be used "as-is"or with a small linear amplifier for mobilework, or may be combined with a high powerlinear amplifier for fixed station operation.Also occupying a position of importance isthe "package" VHF station capable of operationon one or more of the VHF amateurbands.Several different types of equipment designedto meet a wide range of needs are describedin this chapter. There are two differentsidband exciters for fixed/mobile service anda de-luxe VHF station capable of outstandingperformance on two VHF amateur bands. Alsoshown is a high stability variable frequencyoscillator unit designed for operation in thehigh frequency DX bands. To the amateurwho is interested in the construction phase ofhis hobby, these and other units shown in laterchapters should offer interesting ideas whichmight well fit in with the design of his basictransmitting equipment.As in the previous chapter, the componentnomenclature and color codes outlined in figures1, 2, and 3 of chapter 27 are used in thischapter unless otherwise noted.28-1 SSB Exciter forFixed or Mobile UseThe simplest and most economical methodof generating a single sideband signal is toemploy a phasing-type transmitter of the formoutlined in chapter 17 of this Handbook. Ifthe r-f phasing system operates on the carrierfrequency of the transmitter the complex frequencyconversion circuits may be omitted andthe complete exciter becomes inexpensive tobuild and simple to place in operation.A SSB exciter suitable for fixed or mobileoperation is shown in figures 1 and 4. Theexciter delivers a 3 watt peak power signal,567
568 Low Power Transmitters<strong>THE</strong> <strong>RADIO</strong>sufficient to drive a high power tetrode linearamplifier to a kilowatt level. Operation is confinedto the 80 meter band, although operationon the higher frequency bands is possible witha change of coils, phasing network, and crystal.The r-f phasing circuits are balanced at somefrequency within the amateur phone band andwill. retain a good degree of balance over afrequency range of plus or minus fifty kilocyclesof the adjustment frequency.CircuitDescriptionThe circuit of the single sidebandexciter is shown in figures2 and 3. A 12AU7 is employedas a crystal oscillator stage and buffer-amplifier.The first section of the double triode is usedin a Pierce oscillator circuit with the crystalconnected between the grid and plate of thetube. The oscillator operates directly on thechosen SSB frequency in the 80 meter band.The frequency of oscillation may be varied overa range of two hundred cycles or so by the 50p.p.fd trimming capacitor permitting the transmitterto be "zeroed in" on a particular SSBchannel. The second section of the 12AU7serves as an isolation amplifier with the platecircuit tuned to the operating frequency.Changes in the input impedance of the diodemodulator stage under operating conditionswould cause frequency shift of the oscillatorstage if direct coupling between these circuitswas used. The isolation afforded by the bufferstage effectively prevents frequency pulling ofthe oscillator stage during modulation.The output of the buffer stage is linkcoupled to a simple 90 o r-f phase shift networkwherein the audio signal from the audiophasing amplifier is combined with the r-fsignals. The network is of the R-C type made upof 50 ohm non-inductive resistors and 800p.p.fd. capacitors in a bridge configuration. Thereactance of the capacitors is very close to 5012AU7 4-1 N81 6CL6Figure 2BLOCK DIAGRAM OF PHASING-TYPESIDEBAND EXCITEROnly five tubes are required in this simplephasing-type sideband exciter. Audio systemhas su,icient gain to operate from crystalmicrophone.ohms in the center of the 80 meter phoneband. Bridge balance and carrier eliminationis achieved by adjustment of the variable potentiometers( R,, R., R,.) in the bridge circuit.Four selenium diodes are used in themodulator circuit. A 1N82 multiple diode assemblymay be employed, or four 1N81 diodeswhose front/back ratios are equal may be used.The output of the balanced modulator networkis coupled by a low impedance link tothe grid circuit of a neutralized 6CL6 linearamplifier. This stage operates class AB1 withcathode bias, delivering a 3 watt peak SSBsignal to a low impedance load circuit. Allcircuits are tuned by adjustable slug-tunedcoils ( L1, L,, and La ) .A cascade 12AU7 and a 6C4 comprise thespeech amplifier which may be driven from acrystal microphone. The output of the 6C4is transformer coupled into the audio phaseshift network PS-1. An interstage transformeris connected backwards for T1, providing arelatively low impedance secondary windingdelivering two equal audio voltages that are180 o out of phase. These voltages are appliedto the special audio network whose outputcircuits drive separate triode sections ofthe 12AU7 audio phasing amplifier. The12AU7 tube functions as a dual cathode fol-Figure 1<strong>THE</strong> GLOVE COMPARTMENTSIDEBAND EXCITERThis miniature phasing-type SSB exciter maybe mounted in the glove compartment of acar with room to sparef Using only four tubesthe exciter delivers a 3 watt peak sidebandsignal capable of driving a kilowatt tetrodeamplifier to full output. The l2AU7 and6CL6 r-f tubes are at the right end of thechassis, along with the crystal. Major tuningcontrols are on the right end of the chassis.Entire unit is bolted to the roof of the glovecomportment.
<strong>HANDBOOK</strong> S.S.B. Exciter 569lower which provides the necessary low impedancerequired to match the r-f phasing network.Sideband switching is accomplished by reversalof polarity of the audio channels byswitch S,. The diode modulator may be unbalancedto pass a carrier for tune-up purposesby throwing carrier insertion switch S, to theright n.nd biasing the diode modulators bymeans of carrier insertion control R,.Power requirements for the transmitter are300 volts at a current drain of 100 milliamperes,and 6.3 volts d-e at 1.85 amperes. Whenthe filaments are connected for 12.6 volt op-12AU7AUDIOPHASINGAMPLIFI~R1 Ko .•12.AU7esc.6CL6AMP12.AU7AUDIO6C4AUDIO12AU7AUDIO6VOLTFILAMENT CONNECTIONSNOTE*=MATCHED COMPONENTS ( 7 96 ORLESS), EXACT VALU£ CRITICALONLY IN THAT IT SHOULD MATCH<strong>THE</strong> MATING UNIT CLOSELY.TOP1Figure 3SCHEMATIC DIAGRAM OF FIXED/MOBILE SSB EXCITERC1, C:, c,, C1i-See figure 7Cs-10 JJ.JJ.fd variable ceramic, Centralab 8278.R1, R:, R,, R 4-See figure 7Rs-500K linear taper patentiometerR,, Rw--100 ohm potentiometerR,-25K linear taper potentiometerRs, R,--1 K patentiometerPS-I-Phase shift network package. See figure 7L,, L,, L,-See figure 8Tt-lnterstage audio transformer (1:3). Stancor A-53C used backwards.RFC-Miniature 2.5 mh. choke. Millen 1300-2500X-80 meter crystal (3800 - 4000 kc.)PC-3 turns # 18 e. wire around 50 ohm, 0.5 watt composition resistor.D,-D,-1N81 diodes; see text.
570 Low Power Transmitters<strong>THE</strong> <strong>RADIO</strong>IFigure 4TOP VIEW OF SSB EXCITERThe top plate is removable for easy access tointernal wiring and adjustments. Microphonejack and coaxial antenna receptacle are visibleat left of chassis. Four control potentiometersare along bottom of box (see text).eration current the current drain is 1.25 amperes.TransmiHerConstructionand WiringThe transmitter is constructedupon an aluminum box-chassismeasuring 10" x 2\12" x2 Y2" in size. Placement ofthe major components may be seen in figure 5.The 80 meter crystal and 12AU7 oscillator/buffertube are positioned at the left endof the chassis, and the 6CL6 linear amplifieris centered on the chassis. The balanced modulatoris mounted on a phenolic terminal boardon the opposite side of the box. Oscillator tuningcontrols are placed on the left side of thebox-chassis, along with the sideband selectorswitch s,, the carrier insertion switch S,, andthe audio gain control R,. Along the "bottom"of the chassis are placed (left to right) themodulator balance potentiometers Rs, Ro, andR10 and the audio balance control R •. The carrierinsertion potentiometer R, is mounted insidethe chassis. The three audio stages aremounted on the right-hand section of thechassis.The components of the audio phasing networkPS-I are mounted on a small phenolicterminal board that may be observed in thelower right corner of the chassis. This networkshould be wired and tested before it is placedin the transmitter.The AudioPhasing NetworkThe complete networkmay be purchased as afinished item (Millen75012) or it may easily be home built ifan audio oscillator and oscilloscope are availablefor testing purposes.The circuit of the audio network is shownFigure 5UNDER-CHASSIS VIEW OF SSBEXCITERPlacement of the main components may beseen in this inside view of the chassis. R-1circuitry is at the left with home-made audiophasing network at center, right. R-f modulatorpotentiometers and germanium diodesin lower left area of the chassis with audiosection at right. The 6CL6 linear amplifierplate coil is in the foreground of the centerof the chassis.in figure 7 A. The mounting base may be thinphenolic or any insulating material. The baseholds four precision resistors and four fixedmica capacitors padded with four adjustablemica trimmer capacitors. If desired, a seriesof fixed capacitors may be measured on acapacity bridge and hand picked units chosento replace the variable capacitors after finaladjustments have been completed. This wasdone with the network shown in the photographs.The two 100 K series resistors used in thenetwork are Continental Nobeloy 1% toleranceprecision resistors. The 133.3 K resistors, however,were made by taking two 150 K precisionContinental Nobeloy resistors and parallelingeach of them with a one-half watt 1.2megohm (plus or minus 10% tolerance) resistor.Careful selection of the 1.2 M units permitsclose adjustment to the desired targetvalue of 133.3 K. A convenient way to mountthe 1.2 M resistors is to slip them inside thehollow body of the precision resistors. Thedashed connections should be omitted initially,since the alignment procedure described belowpresumes that these connections will be madeat the proper time only.Adjustment of theAudio NetworkHowever, thealigned beforeIf a manufactured networkis employed no adjustmentsare required.home-made network must beit is placed in the transmitter
<strong>HANDBOOK</strong>S.S. B. Exciter 571chassis. The network resistors should bear theratio of 133.3 to 100.0, that is, 4 to 3 as closelyas can be determined. If in doubt as to theratio of the resistors you use, double-checktheir values on an accurate bridge. The adjustmentof the phase shift network now consistsonly of setting the four capacitors to theirproper values.An audio oscillator capable of operationfrom 225 to 2750 cycles per second (withgood waveform) is required, plus an oscilloscope.The oscillator should be carefully calibratedby the method described later. Connectthe output of the audio oscillator through astepdown transformer (T, will serve nicely)to a 1 K potentiometer (use R,) with the armgrounded.Adjust the arm position so that equal andopposite voltages appear on each half of thepotentiometer. A steady audio frequency signalof any convenient frequency may be used withthe oscilloscope acting as the voltmeter forthis job. Swing the vertical deflection lead ofthe 'scope from one end of the potentiometerto the other and adjust the arm to obtain equalvoltages. Hook up a temporary double cathodefollower using a 12AT7 with 500 ohms fromeach cathode to ground and connect as shownin figure 7B. (It will be convenient to provideleads M, N, and 1 and 2 with clips atthe ends to facilitate checking). Cathode pins8 and 3 of the 12AT7 should connect to theH and V deflection amplifiers in the oscilloscope,and the oscilloscope "ground" connectionshould be made to the common return ofthe cathode follower amplifier.First connect lead M to terminal A on PS-1and lead N to terminal A'. Connect leads 1and 2 to terminal M. (Note that the dashedconnections are missing at this stage of ad-PHASE SHIFT NETWORK PS-1FROM~~N1At12AU7 T1T1 (1:3)csrANCOR A53-CtsAL5:lf--!-..... --
572 Low Power Transmitters<strong>THE</strong> <strong>RADIO</strong>NATIONAL XR-50COl L FORM(112" or AM.)CONNECTIONSNOTE: TWO PARALLEL PIECES OF HOOkUP WIRE ARE WOUNDSIMULTANEOUSLY OVER PRIMARY WINDING TO MAKE8/F!LAR SECONDARY. BOTTOM END OF ONE WINDINGAND TOP END OF OPPOSITE WINDING ARE GROUNDEDTO MDUNT!N.G LUG ON COIL FORM.Figure 8BIFILAR COIL FOR SSBTRANSMITTER !L1lCoils L, and L3 have 4-turn link coils overbottom end instead of the bifilar winding.shift lead 1 from terminal A to terminal Bon the phase network PS-1. Adjust the trimmercapacitor C (figure 3) to obtain a circle onthe oscilloscope. It will be noted as this adjustmentis made the display will shift froman ellipse ''leaning" to one side through acircle or ellipse (with axes parallel to the deflectionaxes) to an ellipse which "leans" theother way. If desired, the appropriate gaincontrol on the oscilloscope may be varied sothat a circle instead of an "erect" ellipse isobtained at the point of correct adjustment.After checking the gain control on the 'scope,recheck and correct (if necessary) the phaseshift in the oscilloscope by moving lead 1back to terminal A, and then repeat the settingof capacitor C with lead 1 back on terminal B.In general, always make certain that the oscilloscopeis used in a phase-corrected manner.As a double check (if the deflection plates inthe oscilloscope are skewed, for instance) connectlead 2 to terminal A'. If the circle changesto a slanting ellipse, readjust C to produce anellipse "half-way" between the ellipse ( obtainedby switching lead 2) and a circle.Changing lead 2 from A' to A and backagain should give equal and opposite skewto the display when capacitor C is set correctly.Failure to get symmetrical ellipses (egg-shaped,or other display) is due to distortion in the'scope, the oscillator, the transformer, or thecathode follower. Conduct the test at the lowestsignal level possible to avoid distortion.Next, connect leads M and N to terminals Eand E', respectively. Connect leads 1 and 2 toE, set the oscillator frequency to 1960 c.p.s.,correct oscilloscope phase shift as before, andmove lead 1 to terminal G. Adjust capacitorC in PS-1 for a circle as was done for C,using the precautions outlined for that case.Now connect lead M to terminal D, andlead N to terminal F. Connect leads 1 and 2to terminal D, set the oscillator frequency at1307 c.p.s., correct oscilloscope phase shiftas before, and move lead 1 to the junction ofRa and Ca. Adjust capacitor C, for a circle onthe oscilloscope, as before.Repeat the above procedure for the remainingR-C pair, R, and C. Use terminals D andC at this time and set the oscillator to 326.7c.p.s. This comJ:>letes the tests except for a finalcheck to all adjustments. Connect A to A', Eto E', B to C, F to G, and A to E. The networkPS-1 is now completely wired.Checking theAudio NetworkIf the oscilloscope did notrequire changes in externalcompensation over the fourfrequencies used, an over-all frequency checkmay now be easily made on the phase shift network.To do this, connect lead 1 to point8-C, lead 2 to point F-G, lead M to point A-A'E-E', and lead N to point D. Now shift thearm of the potentiometer towards M until acircle appears on the oscilloscope screen at afrequency of 250 c.p.s. Then, as the oscillatorfrequency is varied from 250 c.p.s. to 2500c.p.s., this circle will wobble a little from oneside to the other, passing through a perfectcircular display at 440, 1225, and 2500 c.p.s.The audio band over which the wobble indicatesa plus or minus 1.3 degree deviationfrom true 90 degrees is 225 to 2750 c.p.s., or12 to 1 in range. This means that when othercircuits are properly adjusted, a sideband suppressionratio of 39 decibels is possible at theworst points within this range. The averagesuppression ratio will be about 45 decibels.Proper phase-shift network operation is necessaryto obtain this class of performance, sothe adjustment procedures have been outlinedin detail as an aid toward this goal. The phaseshift network should never require readjustment,so that when you are satisfied with theadjustment you may seal the trimmers withcement.Audio OscillatorCalibrationIt will be noted that thefrequency ratios are suchthat the 12th harmonicof 326.7 c.p.s., the 8th harmonic of 490c.p.s. and the 3rd harmonic of 1306.7 c.p.s.are all the same as the 2nd harmonic of
<strong>HANDBOOK</strong>S.S.B. Exciter 5731960 c.p.s .., namely 3920 c.p.s. Thus, if astable source of 3920 c.p.s. frequency (suchas a thoroughly warmed audio oscillator)is used as a reference, the frequency of thetest oscillator can be set very closely to onehalf,one-third, etc. of the reference frequencyif both oscillators feed an oscilloscope and theresulting Lissajous figures observed, as discussedin chapter 9 of this Handbook.Use of a calibrating frequency in this mannerassures that the frequency ratios used arecorrect, even though the exact frequencies usedare unknown. The frequency ratios (just asthe resistance ratio previously mentioned) arefar more important that the actual values offrequency (or resistance) used.TransmitterWiringThe r-f and audio sections ofthe unit should be wired beforethe phase network is placedwithin the chassis. Socket ground connectionsand filament wiring are done first. The oscillatorsection should be wired and tested first.The bifilar secondary winding of coil L, maybe made of two lengths of insulated hookupwire wound over the coil as shown in figure 8.The germanium diodes in the modulatorsection deserve special care in handling. Do notbend the leads close to the diode unit itself.The diodes are mounted by means of their leadsbetween the terminals of potentiometers Rs andRo and adjacent tie-point terminals. Protectthe diodes from excessive heat while solderingby holding the lead with pliers between thebody of the diode and the point where the solderingtakes place. Further, use only as muchheat as is necessary to make a good joint.The 50 ohm composition resistors used inthe modulator bridge circuit should be amatched pair, measured on an ohmmeter orWheatstone bridge. In the same fashion thebridge capacitors should be matched as closelyas possible to the target value of 800 1-'1-'fd.Both coils L, and L, of the linear amplifierare mounted beneath the chassis on either sideof the 6CL6 tube socket. To reduce couplingbetween the coils, a shield is placed over coilL,. The shield may be made from part of anold i-f transformer can measuring approximately1" x 1" x 2" in size.When the transmitter wiring is completed,the audio phase network PS-1 may be placedin the transmitter chassis and wired in place.TransmitterAdjustmentAfter the wiring has been completedand checked the transmitteris ready for adjustment.Insert the 12AU7 oscillator/buffer tube in thesocket, and plug in a crystal in the range of3.8- 4.0 Me. Apply plate power and adjustthe slug of coil L for maximum signalstrength of the oscillator in a nearby receiver.The oscillator should start promptly each timeplate voltage is applied to the unit. Frequencyof oscillation may be varied over a small rangeby changing the setting of the crystal tuningcapacitor.The next step is to neutralize the 6CL6linear amplifier. It is necessary to bypass thediode modulator to do this. Temporarily disconnectone lead of the bifilar link windingon L, and also disconnect the lead between potentiometerR10 and amplifier grid coil L,. Now,connect the free winding of the bifilar link ofL, to the ungrounded end of the link of L,.This will apply the oscillator signal directlyto the linear amplifier. Temporarily load the6CL6 stage with one or two flashlight bulbs( 6.3 volt, 150 ma. - brown bead) connectedin parallel at the output jack ]t. The slugs ofcoils L, and L, should be adjusted for maximumamplifier output. The 12AU7 oscillatortube should now be removed from the socket,and unless the setting of neutralizing capacitorC happens by chance to be correct, theflash lamp load will continue to give some indicationof output, showing that the 6CL6amplifier stage is oscillating. Neutralizing capacitorCo should be adjusted with the aid ofa fibre screwdriver until the bulb goes out.Readjust L, and L, for maximum bulb indicationas the setting of capacitor C is varied.A setting of C will be found at which oscillationwill not occur, regardless of the settingof the slugs of coils L, and Ls.When C is properly adjusted, the 12AU7tube should be replaced in its socket and coilsL, and Ls returned for maximum output. Thetemporary link connection between L, and L,should be removed and the diode modulatorreconnected into the circuit.Next, remove the r-f tubes and insert thethree audio tubes in their respective sockets.Apply a low level audio signal of 1225 c.p.s.to the microphone jack ], of the unit and connectthe horizontal deflection terminal of theoscilloscope to a cathode (pin 3) of the12AU7 audio phasing amplifier, and the verticaldeflection terminal to the other cathode(pin 8) after making certain that the 'scopeis phase-compensated at the test frequency of1225 c.p.s. Adjust audio balance control R,to produce a circle on the screen. Make thistest at as low an audio level as possible.Now, plug in the r-f tubes and connect thedummy load to the antenna receptacle, ]1.
574 Low Power Transmitters<strong>THE</strong> <strong>RADIO</strong>Connect the vertical plates of the oscilloscope(no ·scope amplifier used) to the terminalsof the dummy load so that the SO meter signalof the transmitter may be seen on the screen ofthe 'scope. Deliberately unbalance one of thediode modulators by setting potentiometer R,off-center. Adjust amplifier output coil formaximum pattern deflection at any convenientsweep speed. As the p::ttern grows when L, isresonated it may be necessary to reduce modulatorunbalance to keep from overloading theoutput stage or the 'scope. Remove all audioinput to the transmitter by adjusting audiogain control Rs to zero, then by successive adjustmentsof the modulator balance controlsad just for zero output as seen on the oscilloscope.It will be noted that as the correct pointis approached the adjustments will become pregressivelysharper. Potentiometers R, and R,should be juggled until the indicated carrieris a minimum. Slight capacity unbalances withinthe modulator bridge may prevent a perfectminimum from being obtained. It may be necessaryto place a small I 0 !L!Lfd variable ceramiccapacitor from either end of potentiometerRIO back to one side of the diode rectifierbank for best attenuation. Good carrier attenuationmay finally be achieved by jugglingthe settings of Rs, R,, and RIO, together withthe added padding capacitor.Next, apply a 1225 c.p.s. audio signal andadvance gain control Rs. An r-f envelopeshould be seen on the screen of the oscilloscope.The percentage of modulation (or ripple)noted on the pattern is an indication of thedegree of mis-adjustment of the balancingcontrols of the unit. The balance potentiometersshould now be touched up to produceminimum modulation on the observed carrier.Audio balance control R, should also be checkedfor minimum ripple pattern.It is a good idea to vary the audio levelinput to the transmitter to determine the pointof overload. This can be seen on he 'scopeas the point at which the modulation rippleon the pattern increases sharply as Rs is advanced.Always make sure that your audiolevel is set well below this overload point whenadjustments are being made. It should be possibleto produce a pattern having very littleripple on the observed carrier wave. This completesthe adjustments of the transmitter. Carrierinsertion for test purposes may be obtainedby reversing switch s, and adjusting carrierinsertion control R,.VOX CircuitsVoice control of the "sendreceive"circuits of thetransmitter are desirable for serious sidebandwork, and are doubly suited for mobile operation,since full attention may be given to roadproblems. A simple voice control (VOX) circuitsuitable for use with this transmitter isshown in figure 6. Connection to the speechamplifier is made at point A of figure 3. Thevoice signal is amplified by a triode stage andrectified by the 6AL5 diode. The resultingsignal is applied to the second half of thetriode tube which operates a sensitive relayconnected in the plate circuit. The relay contactsoperate the changeover circuits of thetransmitter. The time delay required to makethe transmitter "hang on" between syllablesof speech is determined by the R-C networkin the grid of the relay control tube. Increasingthe size of the capacitor will increase thedelay time and vice-versa.28-2 A MobileTransistorizedSSB ExciterPower consumption of radio equipment isof paramount importance in mobile systems.Without the addition of auxiliary expensiveFigure 9TEN THOUSAND MILES OF QSO'SHAVE BEEN MADE WITH THISTRANSISTORIZED FORTY METERSSB EXCITERThe reliability of transistorized sideband equipmenthas been proven with this miniaturephasing-type exciter. Overall size is only7Y4" X 4" X 2V2". Antenna, power, and microphoneconnectors are on end ol chassisand balancing potentiometers and buffet baseinductor (Lz) are mounted on side of chassis.
<strong>HANDBOOK</strong>Transistorized Exciter 575charging components, the usual automotiveelectrical system is capable of a relatively smalladditional power drain over and above thestandard equipment provided with the car.Single sideband affords an excellent opportunityfor the mobile enthusiast to obtainmore ''talk power" for a given degree of primarypower consumption. In addition, if thesideband equipment can be transistorized thepower drain will be reduced by a further importantamount.Described in this section is a 40 meter experimentaltransistorized sideband exciter deliveringsufficient output to excite a tetrodelinear amplifier to one kilowatt peak inputpower. The exciter operates directly from the12 volt ignition system of the car, using fiveinexpensive transistors, a printed circuit transistoramplifier, and a single vacuum tube.Until a power transistor is developed for r-fservice, it appears that the vacuum tube willstill be necessary as a buffer when high frequenciesare involved. Power requirements forthe exciter are 300 volts at 50 milliamperes,and 12.6 volts at 0.43 amperes.A new subminiature printed circuit amplifier(Centralab TA-ll) is used for the speechamplifier. Completely sealed in epoxy resinfor protection against mechanical abrasion andhumidity, this four stage transistor amplifierprovides a gain of over 70 decibels. The completeunit is only 1 J/,i inches long and lessthan one inch high.It is strongly recommended that spare transistorsbe on hand for substitution in the variouscircuits. This will give the builder confidenceand assurance that each stage is functioningas it should. Many times limitations on performanceare imposed by certain transistorswithout the operator being aware of the con-CH1SEC.~ ;-c£'";-~~~f4e- -~XT~L I 2 TRANSISTORMIC I AMPLIFIER~ b~:r ~e __ tjj' __ __• __ ~~J2 1 - C10 K- 0":""5- R6,10 KNOTES:AUDIO GAIN1. ALL COMPONENTS MARKED* ARE 196 TOLERANCE.EXACT VALUE CRITICAL ONLY IN THAT IT SHOULDMATCH <strong>THE</strong> MATING UNIT CLOSELY.TYPICAL O.C. VOLT.MEASUREMENTSPOINT VOLTAGE4Figure 10= ~~ 0 SCHEMATIC OF TRANSISTORIZED FORTY METER SSB EXCITERC,-25J1.J1.fd. variable ceramic capacitor. Centralab 823-DZL,-35 turns # 22 enamel, V2" diam., closewound on National XR-50 form. Start with two turn bifilarwinding (see figure 8, section 28-1 for bifilar detail). Mount coil assembly in small shield can.Lz, L·.-Same as Lt. Mount L2 in small shield can.L1-Same as Lt. Tap 18 turns from "cold 11 end. Mount in small shield can.RFC-2.5 mh. miniature r-f choke. Millen J300- 2500T 1-1 K pri., 60 ohm sec. transistor transformer. Chicago UM-111Tz-9~K $et., l~K pri. (u$e butkwurd$) IJTC-07CH ,-200K pri., 1 K sec. transistor transformer. Use primary only. Chicago UM-112X-7.2 - 7.3 Me crystalD,-D;,-Matched IN81 Diodes; see text.1.2~2.453.9NPN{;,£.
576 Low Power Transmittersclition. The concept of transistor action is newand novel to many amateurs; it is not easy torun to the corner radio store and test a transistorto make sure it is operating properly.ExciterCircuitryThe complete schematic of thetransistorized exciter is shown infigure 10. It is derived from thevacuum tube exciter circuit of figure 3, section1, and employs phasing technique on the frequencyof operation.A 2N112 transistor is used as a groundedemitter crystal oscillator employing crystals inthe 7.2- 7.3 Me. range. The r-f output iscoupled through a tuned circuit and a bifilarwinding to the crystal diode balanced modulator.The output in turn drives a 2N114 r-famplifier stage, operating in grounded emitterconfiguration. A base bias potentiometer R, isincluded in this stage which permits parameteradjustment for maximum output. The collectorof the transistor is tapped at a point on thegrid inductor of the linear amplifier stage providingmaximum drive without exceeding thetransistor ratings.A 12BY7 A is used as the linear amplifierstage. Neutralization is employed for greatestcircuit stability, and VHF parasitic oscillationsare eliminated by the use of a parasitic suppressorin the plate circuit.The subminiature transistorized audio amplifierused in this exciter has a rising frequencycharacteristic that is well suited forSSB work. However, the rising response carrieswell into the upper audio range whereinthe phase shift networks of the PS-1 unit<strong>THE</strong> <strong>RADIO</strong>cease to function properly. It is therefore necessaryto incorporate a simple low-pass filterthat will substantially eliminate all audio frequenciesabove 3000 cycles. Such a filter is incorporatedin the circuit between the microphoneinput jack ], and the transistorizedaudio amplifier. The filter consists of the primarywinding of a small audio transformer(CH,) and two 250 ppJd capacitors.The output signal from the encapsulatedtransistorized amplifier is transformer coupledto a 2N2 50 (or a 2N2 55 ) power transistorwhich serves as a driver for the phase shiftnetwork. The 2N250 transistor requires a"heat sink" for the collector to allow maximumcollector dissipation rating to be achieved. Thetransistor may therefore be mounted on anodizedaluminum discs or mica wafers (suppliedby the manufacturer of the transistor) ,thus insulating the case of the transistor (thecollector terminal) from the chassis of the exciter.The phase shift network driven by thistransistor is identical to that described in section28-1 and illustrated in figure 7.Two 2N170 NPN type transistors are basedriven by the phase shift network, and serveas emitter followers, providing a low impedancedriving source for the balanced modulator.Sideband selection is accomplished by switchs, whose purpose is to reverse the phase ofthe audio signals applied to the balanced modulator.Carrier injection is accomplished bypotentiometer Rs and switch S, which applyconducting bias to the modulator diodes.A voltage dropping network is employedFigure 11UNDER-CHASSISVIEW OFTRANSISTORIZEDSSB EXCITERThe audio and r-f networksare mounted on aphenolic sheet attachedto the bottom of thechassis. The 12BY7 A linearamplifier and shieldare mounted on a smallaluminum bracket at theleft of the assembly.Grid coil L, is beside thetube, and plate coil L,is behind the tube. CoilL, is to the right of L,with sideband switch S:behind it. The two r-fbalancing controls R, andRt are at the extremeright of the chssis panel.
<strong>HANDBOOK</strong>Transistorized Exciter 577to reduce the 12.6 volt primary supply to 9volts for the 2N1 70 stage, and to 4 volts and2 Y2 volts for the preamplifier stage. The exciteris designed for use with a primary supplyhaving the positive terminal of the batterygrounded.ExciterConstructionThe exciter is built withinan aluminum case measuring71,4" x 4" x 2V2" insize. Miniaturization of the audio phase shiftnetwork was accomplished by measuring aquantity of resistors and capacitors on a bridge.Final selection was made from those havingthe closest tolerances. These hand picked componentsare mounted on a phenolic terminalstrip. This same sheet is used to support thesmall r-f chokes as well as the balanced modulatorcomponents.It is a good idea when building equipmentto fit in a limited space to construct everythingin the form of sub-assemblies. This will permitthe components to be mounted in the open,facilitating wiring. In this particular unit, theaudio components and the r-f modulator aremounted on a phenolic board on one side ofthe box and the r-f components are attachedto the opposite wall of the box. The oscillatorcoil L, is placed within an old i-f transformercan as is the linear amplifier grid coil La. The12BY7 A tube is mounted on a small bracketat one end of the box, and has a tube shieldplaced over it. The plate coil L. of the amplifierstage is directly behind the tube (figure 11).Various voltage dropping resistors aremounted on a small terminal board at the centerof the chassis, and the audio transistorsare placed in the area between the lower terminalboard and the rear wall of the case.The r-f circuits are wired after the componentboards are mounted in place. Wiringis straightforward and simple. The two r-ftransistors are mounted by their leads to threeterminal phenolic tie-point strips. Be sure toprotect both the transistors and the modulatordiodes from excess soldering heat, as describedin section 28-1. The reader is referred to thissection concerning the adjustment and testprocedure for the home-made phase shift network.Testing TheExciterThe transistorized SSB excitermay be bench tested by runningthe transistors from a groupof 1 Y2 volt batteries. It is wise to conductpreliminary tests at an operating potential ofabout 9 volts. The voltage may be boosted tothe operating value after tuning and alignmentadjustments have been completed. Volt-age should be first applied to the transistoroscillator. R-f output may be measured at theterminals of the bifilar coil with the aid of avacuum tube voltmeter. The slug of coil L,is adjusted for maximum reading. Under diodeload, about 4 volts r.m.s. may be measuredfrom either side of the winding to ground.The bifilar winding is relatively critical. Toomany turns will load the oscillator to a pointof instability, and too few turns deliver insufficientdrive to the modulator. Two turns,loosely coupled to the primary winding, seemto be about correct. When the bridge is unbalanced,approximately 0.2 volt r.m.s. may bemeasured from either arm of output potentiometerRa to ground. This voltage is steppedup by means of a pi-network circuit so thatmaximum excitation voltage is applied to thebase of the 2N114 r-f transistor.Preliminary adjustments, in general, followthose described for the vacuum tube exciterdescribed in section 28-1. Potentiometers R,,R,, and R, are adjusted for minimum carrieroutput in the collector circuit of the 2N114.It may be necessary to add a 10 p,p,fd variableceramic capacitor from one side of the modulatorbridge to ground to achieve maximumcarrier suppression. The final adjustment is toapply an audio signal and touch up the audiophasing potentiometer R, for maximum rejectionof the unwanted sideband. ·When these preliminary adjustments havebeen made, the 12BY7 A tube may be insertedin the socket, and the shield slipped over thetube. Grid inductor La is resonated for maximumgrid excitation with the aid of grid diposcillator, and the plate inductor L can betuned in the same manner. A dummy loadconsisting of two 6.3 volt, 150 rna. flashlampbulbs (brown bead) connected in parallel isattached to output jack J•. Plate voltage andexcitation are now applied to the linear amplifierstage and La and L. adjusted for maximumoutput. When audio excitation is removedfrom the exciter the r-f amplifier will probablybreak into oscillation and the antennabulbs will continue to glow. Neutralizing capacitorC should now be adjusted with afibre screwdriver until oscillation stops. Afterthe exciter is operating properly, full batteryvoltage may be applied to the transistor stageswith a consequent increase in grid drive andpower output of the linear amplifier. Approximately2 volts r.m.s. grid drive may be measuredwith the vacuum tube voltmeter at thegrid pin of the 12BY7 A linear amplifier stageunder conditions of maximum excitation.
578 Low Power Transmitters28-3 A VHF Transceiverof Advanced DesignModern tubes and circuit techniques permitsensitivity and stability levels to be achievedin the VHF region that were possible only atthe lower frequencies a few years ago. Thetransceiver described in this section makes fulluse of these advanced circuits to produce superbresults in the amateur 50 Me. and 144Me. bands.The transceiver (pictured in figure 12)comprises a complete VHF station and powersupply in one small, streamlined cabinet. Thetransmitter section is crystal controlled on anyone of eleven frequencies chosen at randomin the two bands, and runs eighteen watts inputfully modulated. The receiver section employscascode-type r-f amplifiers resulting in anoise figure better than 5 decibels on bothbands. Optimum communications selectivity isobtained by the use of a two-section crystallattice filter in the intermediate frequency amplifier.Maximum frequency stability is achievedby the use of crystal controlled VHF conversionoscillators and a highly stable low frequencytunable oscillator. A signal strengthmeter is incorporated in the receiver, and atuning meter is used to obtain optimum transmitteradjustment. The power supply operatesfrom 115 volts a.c., and either 6- or 12-voltsd.c.Styling for the transceiver permits its useeither as a fixed, home station or as a mobileunit. The major tuning controls are placed atthe left of the panel to permit operation whilethe automobile is being driven. A push-to-talksystem is employed for rapid break-in operation,and band changing takes place in a few<strong>THE</strong> <strong>RADIO</strong>seconds. Twenty tubes including a voltage regulatorare employed in the unit, and the totalprimary power consumption is approximately105 watts.CircuitDescriptionThis high performance transceiveris a complex device, andthe wiring diagram has beenbroken down into several schematic drawings(figures 14, 15, 17, 18, and 19). A block diagramof the complete circuit is given in figure13.The Receiver Section. The receiver portionof the unit employs thirteen tubes in a doubleconversion superheterodyne circuit. Separate r-f""front ends" are employed for each band. Each"front end" employs a 6BZ7 cascade r-f amplifierand a 6]6 converter/oscillator tube. Thedesired section is activated by energizing thefilaments and connecting the output circuitof the mixer stage to the first i-f amplifier.Switch S, accomplishes this action. The 144Me. cascode amplifier is neutralized for optimumsignal-to-noise ratio, while the neutralizingcircuit is omitted from the 50 Me. amplifiersince a satisfactory noise figure is obtainedwithout the extra coil. Overtone crystals operatingin the VHF region are used in a regenerativeoscillator circuit to provide mixingvoltage for the 6]6 triode converter. The completeschematic of the r-f section is given infigure 14.The tunable intermediate frequency amplifiercovers the range of 30- 34 Me. ""Low side"'oscillator operation is used for two meter operationwith the conversion crystal at 114 Me."High side" operation is used for six meterservice with the crystal at 84 Me. The fourmegacycle i-f tuning range is covered by aFigure 12VHF Transceiver for Six andTwo Meters is Designed forFixed or Mobile OperationThis transmitter-receiver providesthe highest order of re•liable communication in the twomost populor VHF bands. Receivertuning controls areplaced at the left of the unitto permit ease of tuning inmobile installation. Double conversionreceiver and crystal lat ...tice i-1 filter provide maximumstability and selectivity lor reception.Crystal controlled transmitterdelivers a ten watt car ..rier. Band change takes only afew seconds. Left hand meter issignal strength indicator for receiver,and meter at right ismulti-purpose mil/ammeter fortransmitter. The transmitter tuningcontrols are at right. Perfo ..rated metal cabinet insuresmaximum ventilation 'or continuousoperation.
<strong>HANDBOOK</strong>Advanced Transceiver 579TV-type rotary induction tuner, padded to thecorrect operating range. The use of this rypeof tuned circuit provides exceptional stabilitycombined with small size and has proven tobe a very satisfactory choice. A 6BJ6 remotecutoff pentode is used as the first i-f amplifierstage ( V,) and a 6BE6 is used as the secondmixer ( Vo). The tunable oscillator stage operates2009 kilocycles lower than the variableintermediate frequency and employs a singlesection of a 6]6 tube (V,). A small inductance( L11) is placed in series with the rotary coilof the oscillator section to ensure proper trackingover the 4 Me. tuning range.The greater portion of the receiver gain andselectivity is achieved in the second intermediatefrquency amplifier which operates ata center frequency of 2009 kc. A double latticecrystal filter is employed to ensure maximumselectivity at this frequency. The filteris made in two sections and has a single 6BJ6amplifier tube ( Vs) between the sections tocompensate for the insertion loss of the filter.The first section of the filter is placed immediatelyafter the 6BE6 converter tube toachieve maximum i-f amplifier protection fromstrong signals adjacent to the desired one. Two144MC,V•eBZ7V26J6crystals are used, one at a frequency of 2007kc. and the other at a frequency of 2011 kc.,providing a "flat top" to the passband of about4 kilocycles. The crystals used in this filter arespecially cut for filter work, and the substitutionof ordinary "transmitter type" crystals willdegrade the performance of the filter. The unitsspecified in figure 15, however, are not expensive.The i-f transformers are standard 1500kc. units with the mica tuning capacitorschanged as shown in figure 15 to permit operationat 2 Me. The lattice filter may be removedfrom the circuit by means of the "sharpbroad"switch, S,. All four crystals are shortedout in the "broad" position, and one crystal ineach filter section is disconnected from the associatedi-f transformer. In addition, anothersection of the switch ( S,C) drops the gainof the amplifier tube (Vs) to compensate forthe rise in stage gain that occurs when thecrystals are removed from the circuit. Twoadditional stages of i-f amplification ( V,, v,.)follow the crystal filter, providing an abundanceof gain for proper operation of the a-v-eand audio squelch circuits.The detector and associated audio componentsof the receiver are shown in figure 17. A~:.,. J31I PHONES!I II IIII I-------1---JZEROr--aI ' 'L---------..0...- ---
580 Low Power Transmitters6AL5 (V11) is employed as detector, a-v-e,and a-n-1 tube. The noise limiter is of the lowdistortion type and is left in the circuit at alltimes. One section of the 6AL5 serves as thea-v-e rectifier, and the resulting voltage is appliedto four i-f tubes for smooth a-v-e actionon strong signals. The last i-f stage (V•o) isleft off the a-v-e line, and the S-meter is connectedbetween the plate supply return of thistube and an adjacent controlled tube. A changein the plate current of the controlled tube(V,) upsets the balance of the bridge circuit,and a meter reading is produced that is proportionalto the a-v-e voltage. This voltage isalso applied to one section of a double triodeaudio tube ( v12) which functions as a squekhtube. This circuit renders the audio section ofthe receiver inoperative until the receiver a-v-ecircuit is activated, thus eliminating a largeportion of the background noise commonly associatedwith VHF reception. The squelch circuitmay be disabled by switch s,. Two audioV16BZ7ALz<strong>THE</strong> <strong>RADIO</strong>stages (V, and V") deliver sufficient audiopower to fully drive a loudspeaker and to overcomethe motor and wind noises of a car inmotion. An auxiliary jack (}a) provides earphonereception if desired. The receiver is inactivatedduring transmission periods by relaycontacts RY2B which remove the high voltagefrom the plates and screens of the receivertubes.The Transmitter Section. Shown in figure 17is the transmitter section of the transceiver.Six tubes are used, three for the audio portionof the unit and three for the r-f portion.Crystals in the 24- 25 Me. region are employedfor both 6- and 2-meter operation. Any oneof eleven crystals may be selected by switchs,. The oscillator is a 6CL6 (V,.) in a "hotcathode"doubler circuit, delivering energy inthe 48 - 50 Me. region. A split-stator tuningcapacitor is employed in the plate circuit toprovide a balanced output configuration. Theoscillator stage may be turned on during revz6J650MC.VJ6BZ7Ar--f1--+--+--- S 7 AIII(SEE FIG. 18 FORSECTIONS 8& C )NOTES'1, SEE FIG. 18 FOR FILAMENT WIRING.2.. C = .001 DISC CERAMIC3. ALL RESISTORS 1/2 WATTIIIII1-F ourSEE FIGURE 15Figure 14SCHEMATIC, R-F PORTION OF RECEIVER SECTIONC-.001 p,fd, disc ceramic capacitor. Centralab DD-102c,, C.-20 p,p,fd. Johnson 20M IIJ,, J,-BNC-type coaxial receptacleL,-L,_5ee figure 21 for coil informationS7 A, D-Part of S7 (Centralab Wfer PA-9) See figure 18.RY,A, RY,A-See figure 19X 1-ll4 Me. overtone crystal. Precision Crystal Lab., Santa Monica, Calif.X,-84 Me. crystal (see X,)
V106BJ6%>z0.,00;ill;SEE F/~. 14R 3 ~---1---'1>'1'-----
582 Low Power Transmittersception by depressing the "zero" switch S",which applies plate voltage to the oscillatorby way of the receiver circuits.The balanced output circuit of the oscillatorstage is inductively coupled to a push-pullminiature beam amplifier stage employing a6360 ( V"). The grid circuit of this stage isbroadly resonant to the 50 Me. region by virtueof the interelectrode capacitances of thetube. For six meter operation, the 6360 functionsas a class-C amplifier with the plate circuit(Ls-C) tuned to 50 Me. Bias and screenvoltages are set for optimum operating valuesby segments of switch S,. In addition, the platereturn of the stage is routed through the secondaryof modulation transformer T,. The r-foutput circuit is inductively coupled to the 50Me. antenna coaxial receptacle through relaycontacts R Y2A. ,When switch s, is moved to the I44 Me.position, the aforementioned 6360 stage is convertedinto a high efficiency tripler stage, providingoutput in the I40 - I 50 Me. region.Switch segment S,D raises the operating biason the stage, segment S,E drops the screenvoltage, and segments SaA-B select the properplate tank coil, L".A separate amplifier stage employing another6360 dual beam power tube (Vn) isused for 2-meter operation. The tube functionsas a class-C amplifier and is driven by thepush-pull tripler stage (V;o). The grid circuitof the I44 Me. amplifier is inductively coupled<strong>THE</strong> <strong>RADIO</strong>to the tripler, and the plate circuit ( Cu-L,) isinductively coupled to the output circuit. Aseparate antenna receptable (]1) is used forI44 Me. operation. Plate voltage is applied tothis tube by switch section s,p,The use of inductive coupling between thethree stages of the transmitter reduces spuriousemissions to a minimum. Channel 2 televisioninterference, so common to six meter transmittersis absent when this transmitter is used,thanks to the combination of interstage inductivecoupling and the employment of 25 Me.crystals. Power input to the modulated stageon both bands is close to IS watts, permittinga carrier power of over I 0 watts. ,Transmitter adjustment is greatly simplifiedby the use of r-f volt-meters connected to theoutput circuits of each amplifier stage. Smallgermanium diodes (D1 and D,) are used torectify a minute portion of the output voltagewhich is read on the multi-meter M1 of thetransmitter. All tuning adjustments of thetransmitter are conducted so as to enhance thereading of the r-f voltmeter under a given setof antenna conditions.The transmitter is plate modulated by two6AQ5 tubes ( V,, V,.) operating class AB1.Plate voltage may be removed from these tubesand from the modulated amplifier by the "tuneoperate"switch S,. A single I2AX7 serves asa two stage resistance coupled audio amplifier,having sufficient gain for the use of a crystalor dynamic microphone.Figure 16REAR VIEW OFVHF TRANSCEIVERAt the left rear of the chassis isthe dual purpose power supplywith the vibrator and filter capacitorsmounted in front of thepower transformer. At the frontof the chassis are the two 6360tubes and the tuning capacitors ofthe transmitter. Note that thestages are isolated by shieldsplaced between the tuning capacitorsand tubes.The modulator and power supplyfilter choke occupy the c
V136AQ5 T4 NOTES1- SEE FIG. 18 FOR FILAMENT WIRING2-C;;.007.l/F DISC CERAMIC3- ALL RESISTORS 0.5 W. UNLESS O<strong>THE</strong>RWISE NOTED.4-53 IN 144 MC. POSITIONV2.o6AQ5:X>zcta00"~+-------~S~+~B~u~s ________________________________________________ +---~~RYzsMETER CIRCUIT58A-BM 1 ~ " L ,p-.,-------jl•M2---o ~ o--' -··M 3 --- o------.rw- M 41 KM1=0-10 MA.M2-= 0-10 MA.M3,M4:::0-t00MA.10 K~ ,--c055TUNEOPERATE1~K1N.O. N.C.56"ZERO"C,, C,, C,--15/15 JJ-JJ-fd. Hammarlund HFD-15XC,, Cw-30 JJ-JJ-fd. Johnson 25JI2D1, D,-Germanium diode INBILu-LJS-See figure 21 for coil informationRFC-,-7.0 JJ-h. Ohmite Z-50RFC:-1.8 JJ-h. Ohmite Z-144RFC,-0.15 JJ-h. J. W. Miller 4644...,.. 001~1KVFigure 17SCHEMATIC, TRANSMITTER AND AUDIO SECTION OF TRANSCEIVERB+ XMTRS,A-S,F-Same asS,, figure 15. Space decks so coils L,. and L,. fit between the decks.S,-SPDT "push-button" switchS,--2 pole, 3 position non-shorting switch. Mallory 3223-GRY,, RY:-See figure 19M:-0-1 d.c. milliammeter. De-jur # JJ2Tz-25 watt modulation transformer, IOK pri., 5 K sec. Stancor A-3845T,-tnterstage transformer, 1:3. Stancor A-53CX,--24-26 Me. overtone crystal. Precision Crystal Co., Sonto Monica, Calif.)>0....
584 Low Power Transmitters<strong>THE</strong> <strong>RADIO</strong>The Power Supply Section. The completetransceiver requires a plate supply of 250 voltsat 150 milliamperes, and either 6.3 volts at7.8 amperes or 12.6 volts at 3.9 amperes. Thesevoltages may be obtained from the power supplywhose schematic is given in figure 18. Thehigh voltage portion of the supply employs avoltage doubler selenium-type rectifier systemoperating from a 117 volt half-wave windingof the power transformer. This transformer hastwo auxiliary windings. One winding may beused for 115 volt, 50-60 cycle operation, andthe other one is intended for vibrator operaarionfrom either a 6- or 12-volt d.c. primarysource. The choice of a.c. or d.c. operation maybe made by inserting the correct terminal plugin receptacle SO,, mounted on the rear of thetransceiver chassis.8-12 V. O·CPL1(SEE FIL, SCHEMATIC)T1(SEE rEXT)P-8 158MODIFIEDGND "HOT 11115 V.V"B+HVGND~--~~------~,~~------~------~®V2V1C =.001 CERAMIC CAPACITOR115 VOLT"'\.. WIRING3 510 5 7'HOT" r-1 r(J r:J r:~ ~ 89GND -·-r:::r~-.-"'"11• 11 13106-12 VOLT 0-C WIRINGFigure 18 p,SCHEMATIC, POWER SUPPLY OF TRANSCEIVERCH,-4 h. at 175 mo. Stancor C-1410RFC,-4 Jih., 6 amp. J. W. Miller 5221RFC4-2.5 mh., 200 mo. J. W. Miller 5222Sw-DPST toggle switch, 12 amp. H & H 80600-268SO ,-21 terminal receptacle. Cinch-Jones P321-SBSR1, SRz-Se/enium rectifier, 156 volt, 250 mo. International Rectifier Corp. 6RS250PL 1 , PL 1-21 terminal plug. Cinch-Jones S321-CCT7 1 -117 volts at 200 mo., three 6.3 volt windings. Stancor P-8158 (modify as per text).V/8,-Radiart vibrator type 5503-6 (6 volt) or 5503-12 (12 volt)P,-6.3 volt, ISO mo. Pilot lamp (brown bead)S 7-4 pole, 2 position. Centralab PA9 deck with PA-300 hardware.VIB1FOR 6-VOLT OPERATIONA-JUMPER PINS 2-9IN PL1B-JUMPER PINS 7-20IN PLI.FOR 12-VOLT OPERATIONA-JUMPER PINS 10-20IN PL1.
<strong>HANDBOOK</strong>Advanced Transceiver585The power transformer is made from a TVreplacementtype, with a new vibrator windingreplacing the usual 6.3 volt filament windings.Information covering the conversion of thetransformer will be found later in this section.The special winding may be arranged for either6-volt d.c. or 12- volt d.c. operation, butnot both. Thus, the choice of d.c. supply voltagemust be determined before the transformeris modified.The filaments of the transceiver are arrangedin a series parallel circuit so that theymay be operated from either 6- or 12-voltsd.c., or 6.3 volts a.c. with no change in circuitry.The appropriate filament connections tothe power supply are automatically made whenthe proper terminal plug is inserted in SO,. Asimplified drawing of the primary power wiringof the transceiver is shown in figure 18.Relay and Switching Circuits. The controlcircuits of the de-luxe transceiver are shown infigure 19. Two relays control change-over operation.The relay coils are of the d.c. type,and are connected in series with the push-totalkbutton of the microphone and the filamentvoltage supply. Closing the microphone switchgrounds one end of the relay coil circuit andactuates both relays. Relay RY, controls the144 Me. antenna changeover, and switches theS-meter to the r-f output meter circuit of thetransmitter section. Relay R y, conrols the 50Me. antenna changeover, and switches the Bplus from the receiver to the transmitter.Meter M, serves as a multi-meter for thetransmitter portion (figure 17). The basicmeter has a 0-10 d.c. millammeter movementand is used as-is for measuring the grid currentof V" and V". Meter switching is accomplishedby means of Ss. The third position of the switchconnects the meter across the shunt in the platecircuit of the modulated amplifier stage. Theresistance of the shunt is chosen so that fullscale meter deflection is 100 milliamperes.Transceiver Layoutand AssemblyThe transceiver is housedin a special cabinet measuring14" long, 10"deep, and 5 Y2" high. The cabinet is formedfrom "cane metal" stock and has a removablefront panel. The back, which is also made ofsolid material is welded to the wrap-aroundcane-metal cover. The chassis of the unit ismade of sheet steel which is copper platedafter all major holes have been drilled. Chassisheight is 2 inches. Placement of the majorcomponents may be seen in figure 16. Thereceiver section occupies the right-hand por-RxBlRELAY SWITCHING CIRCUITR y,(SHOWN VNENERGIZEO)RY2.c:: 6>. R-F COILA..c::: 2.A R-FCOIL~:~~~ANT. COl L..t::::: i~ ~ 9 5-METER.c:: TO RECEIVERB TO B+~ TOXMTRA ~ ~~;R 2 ANT.COILLo---il•Txc .c:: ~~~~-METER c E}UNUSEDp~TO R-F DIODESTO CONTROL CIRCUITIN MICROPHONEI[RTO FILAMENTVOLTAGE SUPPLYTRANSCEIVER CONTROLS~ POTENTIOMETERS51-SELECTIVITY52-XMTR CRYSTAL$3-BANOSWITCH, XMTR54-FILAMENT$So- TUNE -OPERATE$6- "ZERO~" TUNE57- BAND SWITCH, RECEIVER$8- METER SELECTOR, XMTR59-SQUELCH$10- POWER (BACK OF CHASSIS)R1-RECEIVER VOLUMER2-XMTR AUDIO GAINR3-$-METER ADJ.Figure 19RELAY CONTROL WIRING ANDLIST OF TRANSCEIVER CONTROLSRY,, RY:--3 pole, DT. Advance MF3C/6VD.Hook coils in parallel for 6 volt operation.tion of the chassis, with the power supply tothe rear left, and the transmitter at the upperleft of the photograph. The variable TV-typetuner is panel driven by the main tuning dialthrough a simple speed reduction gear (Crowe).To the side of the tuner are tubes V,, V,, andV 7 • To the rear of the tuner are the two separate"front end" sections of the receiver. The twocascade stages are in the corner, with the 6]6mixer/oscillator tubes to the left. The variouscoil slugs and tuning capacitors are mountedto the chassis deck and are adjustable from thetop of the unit. Near the center of the chassisand running from the front to the back is the2009 kc. i-f strip. The four crystals of the i-ffilter are seen near the front of the chassis directlybehind i-f transformer T,. To the leftof the i-f strip arranged in a line are the voltageregulator tube v,., the first lOJ.Lfd. filtercapacitor for the speech amplifier, the speechamplifier tube V", the second 10J.Lfd. filtercapacitor, and the receiver audio amplifier/squelch tube V12. To the left of these componentsare the two 6AQ5 modulator tubesand transformers T, and Ta. In the far left
586 Low Power Transmitterscorner of the chassis are placed the power transformerT,, the vibrator VIB,, and the twopower supply filter capacitors. Note that thecase of one of the capacitors is "hot" to groundand the unit is mounted on a fibre base ring.The transmitter components occupy the upperleft chassis area of the transceiver. The6CL6 oscillator stage is at the center of thechassis with tuning capacitor C6 to the left. Asmall aluminum shield isolates the oscillatorcomponents from the 6360 buffer/amplifiertube. A second shield plate is placed betweenthe second stage tuning capacitor C and the6360 144 Me. amplifier.Space is provided on the chassis to mountfive miniature crystal holders for the transmitteroscillator circuit. A small aluminumclamp holds five additional crystal sockets tothe rear of crystal selector switch s,, as seenin the under-chassis photograph of figure 20.The power supply area beneath the chassisis separated from the remainder of the circuitryby a 2 inch high copper shield that enclosesthe selenium rectifiers and other smallsupply components. Leads passing out of thiscompartment go through chassis mounting typeminiature feedthrough capacitors mounted inholes in the shield ( Centralab FT-1000).The "broad-sharp" switch S, may be seen inthe upper left area of the chassis. Each segmentof the switch is placed over a group of i-fcrystal sockets so that the leads from the crystalsto the switch contacts are extremely short.To the left of this switch is the S-meter "zero"potentiometer R,. The receiver bandswitch s,is located adjacent to the cascode amplifier<strong>THE</strong> <strong>RADIO</strong>stages at the rear of the chassis and is actuatedfrom the panel by an extension shaft.The r-f section of the transmitter is visiblein the upper right area of the chassis, runningparallel to the front panel. A small brassshield is passed across the center of each of the6360 tube sockets. The shield passes betweenpins 1 and 9, and 3 and 4. Transmitter bandchangeswitch Sa is located in the center compartmentand the adjoining shields preventcoupling between the coils of this circuit andeither the 144 Me. amplifier coil or the 50Me. oscillator coil.TransceiverWiringThe transceiver is a complexpiece of equipment and shouldbe wired with care. Space is ata premium and wiring errors are difficult tofind when working in a small area. It is thereforesuggested that the unit be wired in sections.The receiver should be wired first, andplaced in operation with an auxiliary powersupply. The transmitter may be wired nextand tested for operation with the receiver.Finally, the power transformer should be convertedand the power supply can be wired andtested within the transceiver. The final checkis to make sure that all sections operate properlyas a complete system.Wiring the Receiver Section. The receiversection should be wired first. It is imperativethat all r-f wiring and ground leads be shortand direct. As many components as possibleare mounted between the pins of the tubesockets and the socket ground lugs. Miniaturedisc ceramic capacitors are used for bypass pur-Figure 20UNDER-CHASSIS VIEW OFTRANSCEIVERGeneral layout of the components below thechassis may be seen in this view. At the lowerright are the power supply components, separatedfram the rest of the circuitry by acopper shield. RY1, RY,, and the audio drivertransformer are just above this shield. Acrossthe front of the chassis is the transmittersection with the inter-stage shields mountedacross the tube sockets. At the center is thecrystal switch with five crystals mounted tothe wafer of the switch. Directly behind thisswitch is the i-f section of the receiver andthe audio output transformer. In the lowerleft corner are the cascade r-f stages and thebandswitch for the receiver which is mountedon a small bracket and panel driven withan extension shaft. In front of this switchare the components of the tuned first conversionstage. The trimmers used with the TV"inductuner" are mounted on the tuner terminals.Sections of !;4-inch coaxial cables connect thevarious antenna circuits to the change-overrelays and the coaxial antenna receptacleson the rear of the chassis,
<strong>HANDBOOK</strong>Advanced Transceiver 587FIGURE 21COl L TABLE FOR 2 AND 6 METER TRANSCEIVERRECEIVER SECTION, FIGURES 14 AND 15l 1- 6T. # 20 E., 3/6" 0\A., 1/2" LON C.. TAP 2 1/2. T.FROM GFWUNO END.L2-12 T. ** 20 E, 3/16" 0\A, 7/18• LONG.. MOUNT BETWEENPIN 3 AND PIN 6 OF SOCKET VLL3,L4-5T.#18E. 114" 0\A., SPACED WIRE CIA. ON CTCIRON CORE FORM PLS6. CENTER-TO-CENTER DISTANCE BETWEEN L3 AND l4 IS 3/4".L5- PLATE WINDING: 6T. # 16 E, 5/16" 0\A. 1/2." LONG.~RID WINDING; 1 T. PLASTIC HOOKUP WIRE BETWEENTWO "COLD" END TURNS OF PLATE WINDING.LB-10 T. -tt= 20 E, 3/8'" Dl A., 1/2" LONG. TAP 4 T. FROM C. NO. ENDL7,L8-16T.$$ 2.0E,CLOSEWOUND ON CTC IRON CORE FORMPL$6. CENTER-TO-CENTER DISTANCE BETWEEN L 1AND LB IS 3/4".L9- PLATE WINDING: 9 T. 41' 18 E, 5/18" 0\A. 5/8" LON(;GRID WINDING: 1 T. PLASTIC HOOKUP WIRE BETWEENTWO "COLD'' END TURNS OF PLATE WINDING.L10A, B, C -MALLORY INDUCTUNER, VHF TV TYPE, MODEL 8303L11- 0,33lJH. J.W. MILLER 4586 R-F CHOKE.TRANSMITTER SECTION, FI
588 Low Power Transmittersnearby receiver tuned to 84 Me. A signal inthe 50 - 54 Me. range is now injected in theantenna receptacle (],) of the receiver section,and the adjustable slugs of coils 1, and 1stuned for greatest signal response. The laststep is to adjust the slug of antenna coil L.Switch s, should be set to the 144- 148Me. range and tubes V, and v, are placed intheir sockets. Oscillator operation is checkedat 114 Me., and coils 1,, L, and 1, are adjustedfor maximum signal response in the 2 meterband. When the receiver is operating properlyon both bands, the transmitter section of thetransceiver may be wired and tested.Wiring the Transmitter Section. The r-f sectionof the transmitter is wired first. Couplingcoil L, between the oscillator and buffer ismade of two sections of coil stock that aretelescoping, and the plate coil is placed withinthe grid coil. Coils 1" and 1 " are mountedbetween the wafer sections of Switch Sa. The50 Me. antenna coil is placed between the twowindings of 1,,, and the 144 Me. grid coil ofthe 2 meter amplifier is placed between thedual windings of coil L,,. The various resistorsassociated with the bandswitch are mountedbetween the terminals of the switch. The 2meter plate coil L, is fastened to the terminalsof tuning capacitor Co, which project throughY2-inch clearance holes cut in the chassis. The2 meter antenna coil is supported on two miniatureceramic insulator posts mounted adjacentto Ls. When the r-f wiring is completed,the various coils may be adjusted to operatingfrequency range by varying the spacing betweenthe turns.The audio section of the transmitter shouldbe wired next. This is a straightforward operationand comparatively simple. Small shieldedleads are run from the second section of the12AX7 speech amplifier to the audio gaincontrol R, mounted on the rear apron of thechassis near the microphone jack. A shieldedlead runs from the microphone jack to thefirst section of the 12AX7. The coils of relaysR y, and R y, are connected in series with thecontrol pin of ],, and are returned to the filamentcircuit of the transmitter.Testing the Transmitter Section. The transmitteris now temporarily connected to theauxiliary power supply and the 6C16 oscillatortube and an appropriate crystal are inserted intheir respective sockets. A test lamp shouldbe made from a 6.3 volt, 150 rna. (brownbead) pilot lamp attached to a small loop ofwire. This test lamp is coupled loosely to os-<strong>THE</strong> <strong>RADIO</strong>cillator coil 112 and will light brightly when theoscillator stage is functioning. Capacitor C. isadjusted for steady operation of the oscillator.Bandswitch Sa is now set to 50 Me. and switchs, is set to the "tune" position removing plateand screen voltage from the amplifier tube.The 6360 is inserted in the buffer socket andmeter switch S, is set to read the grid currentof this stage. A reading of about 3.5 milliamperescan be obtained by proper adjustment ofcapacitor C.. The plate circuit of the 50 Me.stage should be set to approximate resonantfrequency with a grid-dip osilclator and adummy load or antenna is attached to the 50Me. antenna receptacle ],. Switch s, is now setto the operate position and the push-to-talkrelay circuit actuated. Amplifier tuning capacitorc, is adjusted for maximum meter readingwhen the meter is switched to the r-f voltmeterposition. Plate current of the 6306 tubeunder this condition is approximately 70 milliamperes.The actual value of plate current maybe checked by setting meter switch S, to the"amplifier" position ( 0 - 100 milliamperes).The next step is to check the modulator. Inserttubes v,., v,, and v20 in their respectivesockets. Advance gain control R, while speakinginto the microphone in a normal tone.When checked in a nearby receiver the modulationshould be clear and crisp.After the transmitter section is operatingproperly on 50 Me., attention should be givento the 2 meter section. A 144 Me. antenna isattached to receptacle ],, and the 6360 amplifiertube (V11) is inserted in its socket. Switchs, is placed in the "tune" position, and thecoupling between the grid coil of this stage andbuffer plate coil L, adjusted for 3.5 milliamperesof grid current to V,. Switch Sa automaticallyactivates tube vl1 when the switchis placed in the 144 Me. position. Finally,switch S, is placed in the "operate" positionand the tuning ( Ca) and loading (Co) controlsare adjusted for maximum indication on ther-f voltmeter. Plate current to the 2 meterstage is held at 70 milliamperes as measuredon the amplifier position of meter switch S,.The bandchange operation will now merelyconsist of selecting the proper transmittingcrystal, setting Sa to the proper position, andadjusting the transmitter controls for maximumreading of the r-f voltmeter. The completebandchanging operation takes about fifteenseconds.The PowerSupplyWhen the transceiver is in workingorder, attention should beturned to the power supply sec-
<strong>HANDBOOK</strong> Miniature S.S.B. Transmitter 589tion (figure 18) . At the time of construction,no power transformer was available that wouldprovide the desired voltages and at the sametime be capable of operation from both a 115volt a.c. and a d.c. vibrator primary source. Itis necessary, therefore, to find a power trans·former that has suitable high voltage windingsand to remove the filament windings, replacingthem with a special multi-purpose windingthat will fill the bill. The Stancor P-8158 wasused in this transceiver, and does the job nicely.This transformer has a 115 volt primary winding,a 11 7 volt secondary winding designedfor voltage doubler service, and three 6.3 voltfilament windings. The three filament windingsare on the outside of the transformer and maybe easily removed.Power Transformer Modification. The fourtransformer bolts should be removed, exposingthe copper shorting band of the transformer,which is taken off. Take off the layers of cambrictape to expose the uppermost filamentwinding. Now, remove the unused filamentwindings, being careful not to damage theother windings of the transformer. A newwinding made of enameled wire should nowbe placed over the remaining windings. Thiswinding will be for either six or twelve voltoperation, and will have a voltage tap for filamentoperation when the supply is run fromalternating current. During battery operationthe filaments of the transceiver will be rundirectly from the d.c. primary source.Decide whether you want a six or twelvevolt primary winding. The six volt windingwill be made of # 14 wire, whereas the twelvevolt winding will be made of # 16 wire. Eachwinding will have a tap at the proper point toprovide voltage for the tube filaments. Thisnew winding is shown on transformer T,,figure 18.The new winding will be put on in twohalves, since two separate windings are requiredfor proper operation of the "full-wave"vibrator. The six volt winding consists of twoseparate windings of 14 turns of #14 wire.The windings are wound on in series, with thetwo adjacent leads brought out as a center tap(see terminal B, T,, figure 18). The 6.3 voltfilament tap (terminal D, T,) is placed tenturns from the junction of the two windings(point B). The windings should be carefullytaped in place and the transformer reassembled.If twelve volt operation is desired, two separatewindings of 28 turns of # 16 wire are woundin series on the transformer. The filament tapfor a-c operation of the filaments (D) isplaced at the 12.6 volt spot on the winding,or twenty turns from point B. Thus, the filamentswork as a "six volt string" for 61 115volt operation and as a "twelve volt string"for 121115 volt operation. The two differentarrangements of plug PL-2 necessary to accomplishthis are shown in figure 18.Testing the Power Supply. After transformerT, has been modified, the power supply shouldbe wired in accord with figure 18. Before it isused with the transceiver, it should be testedwith both a.c. and d.c. input circuits. The highvoltage system should develop approximately250 volts across a 1600 ohm 50 watt resistor,and a full 6.3 volts should be measured acrossa 1 ohm, 10 watt filament load resistor. The12 volt configuration should develop the samehigh voltage under a similar load, and shoulddeliver 12.6 volts across a 2 ohm, 10 watt resistor.If it is found that the voltages developedduring operation from the d-e source are lowerthan the corresponding voltages derived froma-c operation, it is an indication that the modifiedprimary winding has too many turns. Thetransformer should be removed from the installationand one turn removed from each halfof the primary winding. This will boost thed.c. plate voltage about 10%. After the voltagehas been set, the power supply wiring andswitching circuits of the transceiver may becompleted.28-4 A MiniaturizedSSB Transmitterfor 14 Me.This 40 watt p.e.p. sideband transmitter wasdesigned to provide the greatest amount of"talk power" for a given size and weight. Itwas built by W6A VA expressly for the purposeof permitting amateurs in far-off countriesto have the opportunity to experimentwith SSB transmission. The completely selfcontainedtransmitter has been shipped aroundthe world by air freight, and has been operatedin such locations as Canton Island, Fiji, Brunei,Singapore, and other exotic spots.The transmitter and power supply is containedwithin an aluminum case measuring9" x 6" x 5" and weighs less than sevenpounds. When encased in a strong cardboardbox it is completely transportable by air express.Primary power source can be either 115-or 230-volts, 50- 60 cycles. Air tested by manyhours of operation under conditions of extremeheat and humidity, this small sideband transmitterhas proven to be an effective design
590 Low Power Transmittersthat may provide inspiration for a more elaborateassembly. Tuning adjustments are simpleand fool proof, and the transmitter remains inalignment without time consuming balancingof r-f and audio phase networks.TransmitterCircuitryA block diagram of the circuitis shown in figure 24. Fourteentubes are used including twovoltage regulators. The filter method of sidebandgeneration is employed using a 500 kc.Collins mechanical filter having a passband of3.1 kc. The sideband signal is heterodyned tothe 14 Me. operating frequency by two mixerstages. A voltage doubler type silicon rectifierpower supply provides high voltage for transmitteroperation.The complete transmitter schematic is givenin figures 26 and 29. Sub-miniature tubes areused wherever possible to conserve space. Theaudio and sideband generation section of thetransmitter is illustrated in figure 26. The heartof this section is the Crosby triple triode modulatorstage (V,A, V:,A, V,B). The firsttriode serves as a cathode follower audio stageand is driven by a sub-miniature 5 702 resistancecoupled audio amplifier v,. The secondtriode is also a cathode follower ( V:,A) andis excited by the mixing oscillator stage V,A,B. The cathode follower output voltage is developedacross a common cathode resistor andis injected into the cathode of a grounded gridmixer tube V,B. A double sideband A-Msignal is developed in the plate circuit of themixer tube. The phase of the signal at theplate of mixer V,B is opposite to that of asignal injected on the control grid of the tube.Cancellation of the carrier is therefore possibleby injectin~ the r-f carrier on this grid via<strong>THE</strong> <strong>RADIO</strong>the "carrier null" potentiometer R,. For maximumcarrier elimination capacitor C is adjustedto neutralize the capacity feed-throughwhich may occur between V,A and V,B.With proper adjustment of C and R, a carrierreduction of greater than 40 decibels belowmaximum SSB signal may be obtainedwith stability.The mixing carrier is generated at approximately498 kc., the frequency of the "20db attenuation point" of the mechanical filter.The carrier is coupled into the Crosby modulatorthrough cathode follower V,B. A singlesideband suppressed carrier signal appears atthe output terminals of the mechanical filterand is amplified by a single 5 702 pentodestage ( V,), which drives a cathode coupled6BG 7 mixer ( y,) . A crystal oscillator ( V10)operating on 4002 kc. provides mixing voltageto deliver a 4.5 Me. sideband signal in theoutput circuit of the mixer tube. Two doubletuned transformers (T, and T,) provide adequatesupression of the mixer voltage. Furthersignal amplification is obtained at 4.5 Me. byvirtue of a 6BG 7 cascode r-f amplifier stage,y,_At this point the signal is mixed in a secondcathode coupled mixer stage (Vs) and heterodynedto the 14 Me. amateur band. The mixingoscillator operates in the 9.7- 9-8 Me. regionand uses a 57 44 triode ( Vn) in a Piercecircuit. The 14 Me. SSB signal is developedacross a tuned circuit in the plate of the mixertube. The signal level at this point is quitelow, and a second cascade r-f amplifier ( y,)is employed to boost the signal to a level highenough to drive a class AB, tetrode linear amplifier.The RCA 6524 dual beam amplifier·rube requires only 33 volts peak drive signalto develop 50 watts p.e.p., so it was chosen forthe final r-f amplifier stage. With the two sectionsof the tube connected in parallel and 4 7 5volts applied to the plates approximately 40watts p.e.p. can be obtained from the tube.For maximum stability and freedom from parasiticoscillations, suppressors are placed in thegrid and plate leads of the tube and the stageis neutralized.A simple pi-section output circuit is em-.1 ··~:::~~~Figure 23SINGLE SIDEBAND IN A SMALL PACKAGE!A complete filter-type 14 Me. single sideband transmittercapable of 40 watts p.e.p. and less than one-halfcubic loot in size is described in this section. Com ..plete/y self contained, this seven pound sideband packageis well suited for portable or mobile operation. Primarypower source is 115/230 volts, 50-60 cycles. Siliconrectifiers are used in high voltage supply and a6524 is employed as high /eve/linear amplifier.
<strong>HANDBOOK</strong>Miniature S.S.B. Transmitter 5914.!1MC.AMPLIFIERV7FIRST 14MC.AMPLIFIERVo11~/ 230V,50-60""LAYOUT, CHASSIS#1 1 SIDE :I: 110\C10 --:: 'I0V1V4AND0 -,3 ANDSHIELDRSHIELD 1 3-..: ~' a T1LAYOUT,CHASSIS#1 1 FRONTLAYOUT, CHASSIS# 1, 51 DE# 2.0 -, 0V6ANO _,SHIELD ·~ ~V10~~~tENL~'O.t' - v 0 V110-~,v.I )AND\:_:;--SHIELDFigure 24BLOCK DIAGRAM AND PARTS LAYOUT OF 14 MC. SIDEBAND TRANSMITTERFourteen tubes (including two voltage regulators) are used in this compact transmitter. A "Crosby"modulator and 500 kc. Collins Mechanical Filter are employed for sideband generation. First frequencyconversion is to 4 Me., and second frequency conversion is to J 4 Me.Layout of major components on both sides and front of special chassis is shown in the illustration.Figure 25OBLIQUE VIEW OFAMPLIFIER ANDEXCITER CHASSISThe power amplifierchassis is at the left. Theplate coil, plate tuningcapacitor and loadingswitch are in the loreground.Side # r of theexciter is shown. The inter-chassisarea is coveredwith perforatedmetal. The tip jacks lorthe receiver disablingcircuit are on the rightedge of the chassis. Cornersof the mountingbase are rounded to permitclose lit with roundedcorners of drawnaluminum case.
592 Low Power Transmittersplayed with the linear amplifier stage. In orderto conserve space, the usual variable capacitoris omitted from the output side of the networkand a rotary switch and selection of fixed capacitorsis employed in its place. Proper choiceof capacitors permits the transmitter to bematched to 52- or 72-ohm transmission lineshaving s.w.r. values as great as 2.5/1.The problem of obtaining the necessary d-epower from a supply that would fit withinthe small cabinet space was solved by employinga bridge rectifier using the new Sarkes-T arzian silicon rectifiers. The forward voltagedrop of these rectifiers is of the order of only1.5 volts and the back resistance is extremelyhigh. Because of the very small size, a greatnumber of rectifiers may be placed in a smallarea (figure 3 3 ) . Care must be taken withthese rectifiers not to overload them with anaccidental short circuit. Peak currents are criticalbecause the small mass of the rectifierelement will heat instantaneously and couldconceivably reach failure temperature withina time lapse of a few microseconds. The use ofV15702.001v~5702ATOCHASSIS#2.,Ve~c+12.5 FROMSUPPLY!:!.QiliB (CONTROL)TO FIG. 2.91- CAPAC!rDRS MARKED* ARE 240 1J1JF FOR TYPE5008-31 MODEL FILTER.2- ALL RESISTORS 1/2-WATT UNLESS O<strong>THE</strong>RWISE NOTED.3-C=.Ot.UF DISC CERAMIC CAPACITOR.4- SUBMINIATURE TUBE SOCio---+----4>-------- +475 v.r----------~~---~-----~~----0.3V.(AT EACH TUBE SOCirET)Figure 26SCHEMATIC, LOW FREQUENCY SECTION AND POWER SUPPLY OF SSB TRANSMITTERDL 1-Time delay relay, 45 seconds. Amperite 6N045-T. Normally open.T 1-455 kc. miniature i-1 transformer. J. W. Miller 12-C9. Remove turns from windings to resonate at500 kc.T 4 -150 volt, 125 ma., 500 volt, 125 ma., 6.3 volt, 4 amp. Two 115 volt, 50-60 cycle primary windings.Walgren 3266. Walgren Electric Mfg. Co., Pasadena, Calif.CH,, CHz- 2 henry at 130 ma. Stancor C-2303SR 1-Four silicon rectifiers. Max. inverse volts = 280. Sarkes-Tarzian M-500 or 1N1084SR.-Eight silicon rectifiers, two in series lor each leg. Max. inverse volts = 280. Sarkes-Tarzian M-500or 1N1084MF-Col/ings Mechanical Filter, 3.1 kc. bandwidth, 500 kc. center frequency. Type 5008-31X 1 -Approximately 498 kc. frequency chosen to place carrier oscillator at "20 db. down" point on filtercurve (see filter data sheet).RFC-2.5 mh. miniature r-1 choke. Millen J300-2500.s,-see figure 29
Figure 27INTERIOR OF LOW FREQUENCYPORTION OF FILTER TYPE SSBTRANSMITTER500 kc. filter and low frequency componentsare on the bottom deck of the chassis. Smallcomponents are mounted between socket pinsand miniature tie-point terminals. Interiorshield isolates low frequency section fromconversion oscillators and 4 Me. amplifierstages. The "twenty meter 11 conversion crys ..tal X, projects through the front panel ofthe transmitter (left). Note that transformersare mounted below chassis level to bringoverall height even with that of miniaturetubes. 6BK7 A amplifier tube is also submounted.Power leads that pass through Inter-stageshielding are routed through feed-thru typeceramic insulators (Centralab type FT-1000).an auxiliary supply for tuning operations,therefore, is highly recommended.Transmitter Layoutand AssemblyThe transmitter is builtin three sections whichare held together by acommon front panel. The exciter (figure 27),the linear amplifier (figure 31), and the powersupply (figure 33) make up these three sections.When the three sections are placed inposltlon they appear as in figure 32. A rearview of the complete transmitter (minus thecase) is shown in figure 28. The sections arebolted together and the complete assembly isfastened to the front panel. Because space is ata premium in such a configuration, each chassisis individually designed and shaped to fitthe unusual layout. A sketch of the exciterchassis is shown in figure 24. It is formedfrom two sides and a base. The base mountsin a vertical position in the completed assembly(see figure 28). Side # 1 of the exciterchassis contains the low frequency portionof the exciter and can be seen in figure 26, andthe actual placement of major components isshown in figures 25 and 27. The mechanicalfilter occupies the upper right portion of thedeck. Input and output circuits of the filterare isolated by a shield partition that passesacross the midsection of the filter. This shieldalso braces side # 1 to the interior fulllengthshield seen in figure 27. Power leadsfrom this portion of the exciter pass through.001 fLfLfd. Centralab type FT ceramic feedthroughcapacitors. These capacitors are employedwherever power leads pass through aninterstage shield. Note that coupling transformerT, is submounted to bring its heightin line with that of the sub-miniature tubes.All components are mounted on this side ofthe chassis and it is wired before it is attachedto the base.Side # 2 of the exciter chassis is attachedto the base, and is further braced to side # 1by an end plate and an interstage shield. The6BK7 A socket, and transformers T, and Taare submounted to bring their height down tothat of the sub-miniature tubes. The compon-Figure 28REAR VIEW OF ASSEMBLED SSBTRANSMITTERThe three sub-assemblies of the transmitterare fastened together and bolted to the frontpanel. At the left is the power supply section.The a.c. power receptacle is at the bottomof the assembly with the two filter chokesat the left. Above the chokes is the bank ofsilicon rectifiers with the two voltage regulatortubes and dropping resistor at the top ofthe supply.The linear amplifier comprises the center sectionof the assembly. Antenna and receivercoaxial receptacles are mounted on a smallbracket bolted between the outer sections.Plate coil is visible at the top of the chassiswith adjustable loading switch to the right.The right section of the assembly is the exciter,shown in detail in figure 27,
594 Low Power Transmitters <strong>THE</strong> RA D I 0vo6B'G7SIDE *I< 2 OF CHASSIS *I< IV7V868G7 6111Vg6BK7AA8KONTROL).-------------------------~------------------------------------------_JF/t;.26PCCHASSIS* 2TOL2~SSB StDJ4ANTENNARFCCW DISABLEBr:=-:} RECEIVER220 Js5 022of-----~_r:;:>-----~c--'_ _.~~s v.::=:: 0-+ H•____ fSHtI-ALL RESISTORS 1/2 WATT UNLESS O<strong>THE</strong>RWISE NOTED.2.- C = .OI.J.JF CERAMIC CAPAC/ TOR3- S££ FIGURE 26 FOR SOCkET CONNECTIONS OFSUBMINIATURE TUBES.~ 0-10+300V. +475 v.Figure 29T:, T.--4.5 Me. interstage transformer. J. W. Miller 6204L,--20 turns #22 e., Y4" diam .. ¥8" long on ceramic form with iron slug. J. W. Miller 4504L:, L.-15 turns # 18, %" diam, %" long on polystyrene form. Link = 2 turns hookup wire.L,--12 turns#l6, ll/4" diam., IY2" longc,--50 1111.fd. ceramic trimmer. Centralab 822-ANCz--50 1111fd. Johnson SOKIOC,-9 1111.fd. Johnson 9M11c,-so 1111fd. Johnson SOLISCs--75 1111.fd. fixed capacitor. Switch Sz adds seven 33 1111.fd. capacitors in succession.CM-19 or Centralab TCZ-33).s,A-E--5 pole, 3 position. Centralab PA-2015S:--Centralab P-121 Index Assembly with PIS progressive shorting deck.X:--4002 kc. Precision Crystal Lab., Santa Monica, Calif.X.-Frequency = 20 meter frequency minus 4500 kc.PC-Parasitic choke. 52 ohm, I watt resistor wound with 6 turns # 16 wire.RFC--2Y 2 mh. miniature r-f choke. Millen J300-2500M-0-10 d.c. milliammeter.B,-37.5 volts. Burgess XX22 plus two type Z flashlight cellsSH,-150 ma. shunt. 100 ohmsRFCr-Ohmite Z-14 r-f choke(EI-Menco typeents are mounted and wired before the side isattached to the base. The base contains no wiring,except for two leads to the receiver dis·abling jacks ], and ]s.Amplifier chassis #2 occupies the centerportion of the transmitter assembly, and canbe seen in figures 23, 25, and 30. The tubesocket and major parts are mounted on a flatplate, with the grid circuit components enclosedin a "step" shaped box. This shield is
<strong>HANDBOOK</strong> Miniature S.S.B. Transmitter 595Figure 30LINEAR AMPLIFIERAND EXCITERSECTIONS OF SSBTRANSMITTERLeft: Linear amplifier Isconstructed upon analuminum sheet withgrid circuit enclosed bysmall L-shaped shieldbox. Low impedance linkleads from exciter passthrough grommet in box.The coaxial output cablepasses through chassishole to changeoverswitch. Space to the rightof grid enclosure is occupiedby meter and timedelay relay.Right: Front view of assembledexciter chassisshows main panel con ..trois. Chassis mounts invertical position in finalassembly. "Twenty me•tet 11 conversion crystalsocket is at top, left, offront of assembly.clearly visible in figure 32. The clearance spaceprovided by the "step" is occupied by the"SSB- standby-CW" switch s, which is mountedto the panel in close proximity to the gridcircuit components of the amplifier stage. Theimportant leads to the control switch pass outof the grid compartment through ceramic feedthrough capacitors mounted on the wall of the"step" shield. The coaxial r-f output lead andthe grid bias lead also pass through this shieldand may be seen in figure 32.Figure 31 provides a close-up of the interiorsection of the amplifier chassis. Gridcoil L, is affixed to a polystyrene insulator andis mounted to the side of the chassis-box. Neutralizingcapacitor C, is placed on a small polystyreneplate adjacent to the 6524 amplifiertube. Plate r-f choke is next to C, and the5KV plate blocking capacitor is supported betweenthe top end of the choke and the statorof tuning capacitor C. The _j)late inductor Loccupies the far end of the chassis. Pi-networkswitch S, is placed next to the tuning capacitor,and the various padding capacitors aremounted directly on the back of the switchsection.Power supply chassis #3 is attached to thefront panel, and the weight of the supply issupported by a sheet metal screw run throughthe rear of the case into the supply chassisafter final assembly. The power supply componentsare mounted upon a phenolic boardwhich in turn is fastened to the aluminumchassis frame. The silicon rectifiers are held inposition by fuse clips mounted on the boardsand are placed so that they obtain the maximumpossible ventilation. The primary powerplug is placed on the rear of the supply chassisand projects through a hole cut in the caseof the transmitter. The time delay relay DL-1is mounted on the side of the amplifier chassisand may be seen in figure 31. The small biasbattery for the amplifier stage fits below thisspace.TransmitterWiringIt is important that no lowfrequency energy pass aroundthe mechanical filter as spuriousFigure 31INTERIOR VIEW OF LINEARAMPLIFIER GRID COMPARTMENTTime delay relay is mounted on side of gridbox, occupying space between the panel meterand plate tank assembly. The grid coil andtuning capacitor can be seen below the tubesocket. Neutralizing capacitor is made fromJohnson 30MB with every third plate removed,but capacitor listed in ports list of figure 29is satisfactory.
Figure 32COMPLETE R-F ASSEMBLY OF 20METER SSB TRANSMITTERThe two r-f units are bolted together in thisview. The power supply chassis mounts on theright-hand edge of this assembly. Bias batterieslit in the lower area ol the center unit,while plate meter fits into upper space infront of time delay relay. Compare this viewwith figure 23, This photo is taken from theright front, looking upward at the assembly.leakage will deteriorate the sideband and carriersuppression to a great degree. Power leadsare therefore bypassed with feed through typeceramic capacitors at each partition. Severalcircuits (V1B, V,, Va, V,, and V,) are ofthe "hot cathode'' type in which r-f voltageappears on the cathode of the tube. It is verynecessary, therefore, to bypass the filamentlead of these tubes to prevent a signal leakagepath along the common filament wiring.Bypass capacitors are placed directly at thetube socket pins to conserve space and allwiring is short and direct. Coil L ( 14 Me.mixer plate coil) is mounted within the chassisassembly on a small bracket and a short extensionshaft is soldered to the slug to permitcircuit adjustment from the front panel. Allwiring must be checked for opens, shorts, transpositions,and accidental grounds before poweris applied to the transmitter.Testing theTransmitterThe transmitter should be testedin sections before it is assembledin the case. The useof an auxiliary power supply is recommended.The exciter section should be tested first. Aone kilocycle audio signal of low harmoniccontent is applied to the microphone jack ],."Carrier insert" potentiometer R, is set at zero(ground end) and the antenna of a receivercapable of tuning to 500 kc. is attached topoint A, figure 26. Before the audio signal isapplied, "carrier null" potentiometer R:, andneutralizing capacitor C are adjusted for minimumsignal at the crystal frequency of 498kc. When the audio level is advanced an unmodulatedcarrier should be heard in the receiver.The frequency of the carrier will bethe crystal frequency plus the frequency ofthe audio signal. (i.e.; a 2 kc. audio signalwill produce a carrier frequency of 500 kc.).The carrier may be observed on an oscilloscopeas described in chapter 17 and audio signallevel and null adjustments are varied to reducethe residual "ripple" modulation of the carrier.Conversion crystal X, and tubes V,, v,, Vs,and V10 are inserted in the proper sockets andthe receiver is tuned to 4.5 Me. and coupledto the secondary winding of T,. TransformerT1, T,, and Ta are adjusted for maximum sig-Figure 33COMPACT SILICON POWER SUPPLYRUNS ENTIRE TRANSMITTERMiniature silicon rectifiers permit the completepower supply to be built in extremelysmall space. Rectifiers are mounted in fuseclips bolted to phenolic board. Power transformeroccupies bottom area of assembly andis fastened to aluminum assembly plate bythe four bolts in the foreground. Primarypower plug can be seen at top ol unit, withthe two filter chokes mounted back-to-hac/
Figure 34"DUPLEX" TRANSMITTER-RECEIVERFOR 220 MC. <strong>RADIO</strong> LINKThe width of the amateur 220 Me. band permitssimultaneous duplex transmission betweentwo remote points. This compact VHF packagecontains a complete 220 Me. "Radio Link." Acrystal controlled transmitter operates on224.6 Me. The receiver operates at a frequencyof 220.1 Me. Signal from the transmitteracts as local oscillator for the i-f sig·nal of 4.5 Me.The complete station is housed in single steelcabinet. Transmitter section is at left withcrystal mounted on panel. Receiver section isat right with power supply occupying /owerportion of cabinet. Transmitter may be tonemodulated for i.c.w. transmission.nallevel at 4.5 Me. If a broadband oscilloscope,such as the Heathkit 0-11 is at hand, the signalmay be observed visually and the audio levelmay be set for minimum carrier modulation.When monitored in the receiver the 4.5 Me.signal should be a pure carrier with little orno tone modulation. When audio gain controlR, is retarded, the signal should graduallyweaken and disappear. After satisfactory operationis obtained at this frequency, crystal x,and tubes V, and Vn are placed in their sockets.Capacitor C (figure 29) is tuned for maximumSSB signal in the 14 Me. band. The interstagecoupling transformers are peaked formaximum sideband signal and the receivermay be used for monitoring purposes. Voicemodulation of the transmitter should be sharpand clean. The carrier may be inserted fortest purposes by advancing potentiometer R,.The next step is to check the operation of,r----4.5 MC. ~2.2.0.1 MC.---z--Rx22.4.6 MC.--=--2.2.0.1 MC.TxMODULATORV\3SPEECHY14AUDIO OSC.V1sCARBONMICROPHONE &le+300V. AT 2.10 MA.r---- -----,IIRECT. I 5Y3 5Y3 l RECi.Vte I1 Vt7L.. ___ J,___ _j1t5V. '\..VHF "<strong>RADIO</strong> LINK"Figure 35BLOCK DIAGRAM OF 220 MC. TRANSMITTER-RECEIVERSeventeen tubes are employed in the VHF station. The receiver has two grounded grid r-f stages formaximum sensitivity. Local oscillator injection is ptovided from the transmitter. Three 4.5 Me i-f stagesprovide excellent gain and adequate selectivity. Transmitter is crystal controlled from 25 Me. crystaland is plate modulated. Power supply provides 300 volts at 210 milliamperes. Transmitter draws 80milliamperes, receiver draws 75 milliamperes, and modulator draws apptoximately 45 milliamperes underI OOo/o modulation. Speech system is designated to be used with high gain mobile-type carbon microphone.
598 Low Power TransmittersFigure 36REMOVAL OF FRONT PANEL SHOWSPLACEMENT OF R-F CHASSISThe receiver and transmitter sections are boJt ..ed together to form o single unit which sitsneor the top edges of the power transformerand modulator chassis. Power supply chassis11is bolted to cabinet on edge 11 along lefthand side, and modulator chassis is attachedin the same fashion along the right hand sideof the cabinet. Ventilation holes are drilledalong upper edge of rear portion of cabinet.the linear amplifier stage. Bias voltage shouldbe applied to the 6524 and meter switch Saset to the grid position. With full carrier insertion,one or two milliamperes of grid currentwill flow. The stage is neutralized byadjusting capacitor c, for minimum grid currentfluctuation as the plate circuit is tunedthrough resonance. A dummy load is then attachedto the antenna receptacle ( Ja) and theamplifier stage tuned in the usual manner.Make sttre that plate voltage is never removedwhen excitation is applied to the tube as thescreen current will increase to such a valueas to endanger the tube. Always remove thescreen lead when you remove the plate voltage.When the carrier is nulled out, the restingplate current of the linear amplifier will beabout 20 m:~., rioing t0 about 100 rna. undervoice peaks. No grid current should be indicatedunder these conditions.<strong>THE</strong> <strong>RADIO</strong>28-S A DuplexTransmitter-Receiverfor 220 Me.Duplex operation is permitted in the 220Me. amateur band and opens interesting possibilitiesfor unusual and novel forms of equipment.This class of operation consists of twoone-way communication links separated in frequencyfrom each other sufficiently to permitinterference-free operation. The width of the220 Me. band permits placing one link at thelow frequency end of the band and the otherlink near the high frequency end of the bandwithout the danger of excessive interferencebetween the links.Shown in figure 3 5 is the block diagram ofa transmitter-receiver unit designed to serveas one end of a typical duplex communicationlink. Transmission takes place on 224.6 Me.,and reception takes place on 220.1 Me. Thefrequency separation between the two links is4.5 Me. At the opposite end of the link, transmissiontakes place on 220.1 Me., and receptiontakes place on 224.6 Me.The complete duplex station is built withina single cabinet and employs a common powersupply. Separate antennas are used for transmissionand reception, and communication ismaintained in the same manner as in the caseof a "land-line," that is, simultaneous receptionand transmission are possible.Transmitter-ReceiverCircuitryThe schematic of thetransmitter - receiveris given in figure 3 7and figure 38. The receiver section employseight tubes in a superheterodyne circuit havingtwo grounded grid r-f amplifier stages (figure3 5 ) . The intermediate frequency of the receiveris 4.5 Me. and three stages of i-f amplificationare used. Since the frequency separationof the two links is 4.5 Me. it is feasible toemploy the signal from the transmitter portionof the unit as the injection frequency for thefirst mixer of the receiver. Thus with the useof two properly chosen transmitter crystals(one at each end of the duplex link) the twotransmitter-receivers are locked on frequencyand tuning of the receivers is unnecessary. Thetransmission frequency at each end of the circuitcontrols the frequency of reception of thereceiver portion of the transmitter-receiver.Two 6AJ4 tubes are used in the receiver asgrounded grid r-f amplifier st:lges. The gainper stage is quite low but is sufficient to overcomethe noise level of the mixer tube (V:,).R-f energy from the transmitter section pro-
Vs6AB4(ZSMC.)f-+--------'~--~(>----~H·1 K,001TW-c(22SMC.)PIN!, PL1RFC2.:c,...z0m00;t:::TRANSMITTER COIL TABLEL 1- 26 T., 1/2.* DIA. 1 3/4" LONG.L2- 7 T., 1/2." DIA., 1/2" LONG.L3- 2. T., 1/2."' DlA. 1 1/4"' LONG.L4- 2T. 1 3/8" DIA., OVER L3.LS- 2."' LONG, 1" WI DE, •10 WIREl.6- 1 f. #12, 1 1/2" OIA. OVER L5.L 7- 15 T. #1 6, 1/8"0\A., 3/4'' LONG.NOTES1-ALL RESISTORS 1/Z WATT UNLESS O<strong>THE</strong>RWISENOTED.RECEIVER COIL TABLE(ALL COILS 1/4" DIA.)L8-3 T. # 16, 1/2." L., ANT. TAP 1 112. 1 CATH.1T.L9- 3 T. # 16, 3/8" LONG, TAP 1 1/2. T.L!0-3T.41:16, 3/8" LONG, TAP 2.114 T.L11-2.T.#16, 3/&"LONG.NN0I~()B+RECEIVERFigure 37SCHEMATIC, R-F SECTION OF TRANSMITTER-RECEIVERTRANSMITTER SECTIONRECEIVER SECTIONCt, c,, c,, C,-10-10 p,p,fd. Johnson IIMBIICs, Cs, Cto, C11-4 p,p,fd. Erie 3139-EC 4 , C,-4 p,p,fd. Erie 3139-ERFC,-Ohmite Z-235 r-f chokeC,-10 p,p,fd. Johnson 9M IIRFCt--Ohmite Z-28 r-f chokeT,-T.-4.5 Me. interstage i-f transformer. J. W. Miller 1466RFCz--Ohmite Z-235 r-f chokePL,-6 contact receptacle. Cinch-Jones P-306-ABXt-24,944 kc. crystal. Precision Crystal Lab., Santa Monica, Calif.PIN 5 1PL1lOOK·"c' T(AUDIO)PIN 6,PL1CJc"0C1)X
600 Low Power TransmittersV1412AU7T6V1312AX7T7CARBONMIC.MODULATOR SECTION._---~+300V. TO PINS2. & s, PLt.01cKEYPOWER SUPPLY SECTIONAUDIO TONE OSCILLATORe~----~+--P-1L_o_,~~.3V. TO P1N3, PL 1FROM RECEIVER(PINeS, PLt)T5!-ALL RESISTORS 1/2. WATT UNLESSO<strong>THE</strong>RWISE NOTED.RECEIVER AUDIO STAGEFigure 38SCHEMATIC, AUDIO AND POWER SUPPLY SECTIONS OF TRANSMITTER-RECEIVERT.,, Ts-10K pri., 4 ohm sec. Stancor A-3879T.-10K pri., Pri. to V2 sec. = 2:1. Stancor A-4713T,-10 pri., 8K sec. Stancor A-3845T tr-360-0-360 volts at 200 ma., 5 v. at 6 amp., 6.3 v. at 9 amp. Stancor P-8351CH,-2 henry at 200 ma. Stancor C-2325B-Miniature blower motor and fan, 115 voltsFigure 39REAR VIEW, R.F DECKS OFTRANSMITTER-RECEIVERReceiver deck is at the left andtransmitter deck is at the right.The two chassis are bolted togetherto form one unit. Acrossthe back lip (left to right) are;Receiver antenna receptacle,pwoer plug PL,, meter terminals,transmitter antenna receptacle,and transmitter loading capacitor.Rows of (,!," holes are drilled inthe transmitter chassis to improveventilation of the rectfierarea under the chassis.
Figure 40UNDER-CHASSIS VIEW, R-F DECKSOF TRANSMITTER-RECEIVERSmall components of both sections are mountedbetween tube socket pins and phenolicterminal strips mounted at center of chassis.Small copper shields are placed across centerof 6AJ4 r-f tube sockets of the receiver.Shield plate made of perforated aluminumsheet is placed over bottom of transmitterchassis to reduce injection voltage to receivermixer stage. Neutralization capacitors oftransmitter amplifier stage are mounted directlyacross socket on either side of thegrid coil.vides the proper mixing voltage for this stage.Three stages of 6BA6 amplifiers at 4.5 Me.comprise the i-f section of the receiver. A-v-eis applied to each stage. A 6T8 multi-purposetube (V,) is employed as diode detector, a-v-erectifier, noise limiter, and first audio stage.The transmitter section is mounted upon aseparate chassis. A 6AB4 triode oscillator ( Vs)operates on 24.994 Me. The plate circuit ofthis stage is split to provide balanced driveto a push-pull 12AT7 tripler, which in turnFigure 41TRANSMITTER-RECEIVER CABINET,SHOWING POWER SUPPLYAND MODULATORThe power supply and modulator chassis aremounted vertically against the side walls ofthe cabinet. The tubes are mounted on thetop edge of the chassis. A small blower motoris placed at the rear of the chassis for continuousduty operation. R-f section sits uponpower transformer and is held in position bypanel bolts and sheet metal screws passedthrough rear of cabinet.drives a second 12AT7 tripler stage to 224.6Me. All stages employ capacity coupling. Athird 12AT7 (Vn) is used as a push-pull neutralizedamplifier at 224.6 Me. The plate circuitof this stage is a "hair pin" inductance( L.,) tuned by a miniature butterfly capacitorC. Inductive coupling is employed between thedriver stage ( V,o) and the amplifier, and betweenthe amplifier and the antenna circuit. A1N34 germanium diode serves as a r-f voltmeterin the antenna circuit for transmitt~rtuning purposes.The transmitter modulator, audio tone oscillator,and receiver audio amplifier stage aremounted upon another chassis (figure 3 8).The modulator uses two tubes. A 12AU7 ( V")serves as a two stage resistance coupled speechamplifier, and a 12AX7 (V,") is used as aclass B modulator. The speech amplifier is designedto be used with a mobile-type carbonmicrophone. M-e-w transmission is permissiblein the 220 Me. band and a 12AU7 (V,) audiooscillator has been added for this purpose. Theoscillator may be keyed for code transmissions.The power supply occupies the remainingchassis. It uses two 5 Y3 rectifier tubes to provide300 volts at a current drain of 210 milliamperes.No standby circuits are requiredsince both transmitter and receiver operatesimultaneously.Assemblyand WiringThe complete assembly is housedin a steel cabinet measuring7" x 12" x 8" (figures 34 and36). The receiver and transmitter portions arebuilt upon two aluminum chassis measuring5" x 8" x 1 ". These chassis are bolted togetheras seen in figures 39 and 40 and occupy thecenter section of the cabinet. The power supplyand modulator are built upon two similar
602 Low Power Transmitterschassis that are mounted against the side wallsof the cabinet as seen in figure 41. The cornersof these chassis are rounded to permit mountingthem snugly against the walls of the cabinet.The various tubes are mounted in a verticalposition on the upper side of the chassiswhile the transformers mount on the verticalsurface. To the rear of the cabinet is a smallblower motor to provide adequate ventilationfor protracted operation of the equipment. Thevarious manual controls fasten directly to themain panel of the cabinet and flexible leadsare run from them to their respective circuitsin the equipment. The r-f deck sits atop thepower transformer and the modulation transformerand is held in position by sheet metalscrews that pass through the rear wall of thecabinet into the chassis.The Transmitter Chassis. The placement ofmajor parts on the transmitter chassis may beseen in figures 39 and 40. The crystal oscillatorstage is at the front of the chassis with thecrystal mounted in a horizontal position, projectingthrough the front panel of the cabinet.The multiplier stages fall in a line, with thefinal amplifier stage at the rear of the chassis.The coaxial antenna receptacle J, and antennatuning capacitor C are mounted to the rear.wall of the chassis.Common v.h.f. wiring techniques are employedin the assembly. Short, direct leads inthe r-f section are mandatory. The sockets ofthe two tripler tubes (V, and V10) and theamplifier tube (Vn) are positioned at anangle of 45 degrees so that a line drawnthrough pins 1 and 6 is parallel with the frontedge of the chassis. The midget butterfly capacitorsare mounted in line with the tubesockets. permitting very short leads to be runfrom the plate pins of the tubes to the statorsof the capacitors.The nine-pin tube sockets are the type thatmount from beneath the chassis and have ametal ring encasing the phenolic portion of thesocket. The various pins of each socket thatmust be grounded are bent down and soldereddirectly to this ring. The grid resistors of V"and Vw are installed directly between the gridpins and the grounded cathode pins of the tubesocket. The coupling capacitors between thestages are placed between the grid pins andthe stator rods of the butterfly tuning capacitors.Filament pins 4 and 5 of the nine-pinsockets are grounded to the socket ring and acommon filament lead runs between the number9 pins.Inductors L and L, are made of manufac-<strong>THE</strong> <strong>RADIO</strong>tured coil stock and are mounted to the statorrods of the tuning capacitors. Coils L, and L,may be wound from enameled wire. The variouspower leads, decoupling resistors and capacitorsare placed clear of the r-f circuitryand the components are mounted upon phenolictie point strips placed adjacent to the tubesockets. The power leads and meter leadsthat leave the chassis pass through ceramicfeed through capacitors (Centralab FT-1000)mounted on the walls of the chassis.Preliminary Transmitter Adjustments.Transmitteroperation may be checked using thepower supply shown in figure 38, or any auxiliarysupply capable of providing about 200volts at 80 milliamperes. If the permanent supplyis used for tune-up purposes, a 5000 ohm,10-watt dropping resistor should be placed inseries with the B-plus lead to the exciter toprotect the tubes in case of misadjustments.Tuned circuit L-C may be set to 25 Me. withthe aid of a grid dip oscillator and the 6AB4oscillator tube and crystal x, plugged in theirsockets. Proper operation of this and succeedingstages may be checked with a 2 volt, 60 milliampere(pink bead) flash light bulb attachedto a small loop of wire which is held in proximityto the coil of the stage being adjusted.When oscillation is obtained, the 12AT7 triplertube is plugged in its socket and circuitL,-C, set to 75 Me. with the grid dipper. Platevoltage is again applied and the circuits touchedup for maximum bulb brilliance when thepickup loop is held near L,. The second triplertube vlO is plugged in the socket and circuitLa-Ca adjusted for maximum output at 225Me. using the indicator lamp. Spacing of theturns of La may require adjustment to permitresonance. Plate current drawn by these threetubes should be about 70 milliamperes whenthe dropping resistor is shorted out.The next step is to neutralize the amplifierstage. This may be done by observing the gridcurrent of Vn while capacitors C. and c, areadjusted. The leads of a high resistance voltmeterare temporarily clipped across the 2.7Kgrid resistor of the amplifier stage, and theexciter circuits are repeaked for maximum gridvoltage reading. Over twenty volts should beobtained. Plate voltage to the amplifier stage isremoved for this test. Neutralizing capacitorsC. and C are now set to minimum capacityand coil L is varied in position with respectto L, to obtain maximum grid voltage. Theamplifier plate tuning capacitor Cc is tunedthrough resonance while grid voltage is observed.If a change in voltage occurs during
<strong>HANDBOOK</strong>220-Mc. Duplex 603the tuning process the settings of capacitorsC and C are advanced in unison until nochange of grid voltage is noticeable. As thecapacity settings are increased (keeping thetwo settings approximately equal) the "kick"of grid voltage will gradually decrease, untilthere is either no movement of the meter asCo is varied, or else merely a gradual rise ofa volt or so as C is swung through its range.A 6.3 volt, 150 milliampere (brown bead)lamp is now connected to antenna receptacle]1 and plate voltage is applied to the amplifierstage and exciter. Capacitors Co and c, aretuned for maximum bulb brilliance. The platecurrent of the amplifier stage will be close to30 milliamperes under conditions of maximumoutput.Once initial adjustments have been made,a 0-1 d.c. milliammeter may be connected tothe r-f voltmeter terminals and the transmittermay be completely tuned by adjusting all resonantcircuits for maximum indication of themeter. It is wise, however, to make the preliminaryadjustments stage by stage as describedabove to acquaint the operator withproper transmitter operation.The Recei~·er Chassis. The placement of themajor components of the receiver chassis maybe seen in figures 39 and 40. The line-up ofstages forms a "U" on the chassis, passingdown one side, across the front and back alongthe other side. Two six terminal phenolic tiepointstrips are mounted along the center lineof the chassis, supporting various decouplingand voltage dropping resistors. Layout of ther-f stages may be seen in the under-chassisview ot figure 40. A small copper shield plate1" high and 1 Y2 inches long passes across thecenter of each socket and is grounded at eachend by soldering to the socket retaining screws.The center stud of the socket, and pins 1, 3,4, 6, and 9 are all soldered to this plate. Tuningcapacitors C, C., Co, and Cu are mountedclose to the tube sockets, and the VHF-typebypass capacitor at the "cold" end of each coil(Centralab type ZA) is mounted next to thetuning capacitor. Coils L, L., L10, and Lu aremounted between the terminals of the respectivetuning ca1oacitor and the adjacent VHFbypass capacitor. Care must be taken to keepthe coil leads as short as possible-less than1;4-inch long in any case.One filament choke of each r-f amplifierstage is grounded to the shield plate and theocher choke is bypassed by a disc ceramic capacitorand returns to the common filamentlead. Wiring of the i-f stages is straightforward,VERTICAL ARRAYFOR TRANSMISSIONRAD/ELEMENT LENGTHSAND SPACINGREFLECTOR(R) 2!>. 5"ANTENNA (A) 23 . .5"01 RECTOR (0) 22.2.5*SPACING (5) 10"WOODENSUPPORTHORIZONTAL ARRAYFOR RECEPTIONFEED SYSTEM FOR EACH ANTENNA1.----- 2~. 5 00 -----1/JE:::~'h~I··-t" OIA. ALUMINUM:COAXIAL BALUNTOTAL LENGTH= 17 1/4"MAD£ OF 7SJJ COAXIAL LINE II75 !l. COAXIAL LINE ITO TRANSMITTER-RECEIVER)#SWIRE14-300 .n TV Ll NEI (RANDOM1 LENGTH)Figure 42DUAL ANTENNA SYSTEM FOR 220MC. TRANSMITTER-RECEIVERThe same polarization must be used on eachlink. This means that polarization of trans·mission and reception is different at each station.for the companion link to the one illustratedin this drawing, the vertical arrayshould be used for reception, and the horizontalatray for transmission.components being mounted between the socketand i-f transformer terminals and the phenolictie-point strip. Power leads are terminated atthe six prong plug mounted on the rear apronof the chassis.No connection need be made between ther-f circuit of the transmitter and the mixerstage of the receiver as ample proximity couplingexists. In fact, a shield plate is placed overthe bottom of the transmitter chassis to reducethe mixer injecion to a tolerable level.Preliminary Receiver Adjustment. After allwiring is completed, the i-f section of the receivermay be aligned. A 4.5 Me. tone modulatedsignal is loosely coupled to the plate pin(# 5 ) of mixer tube V, and earphones or ana-c meter connected to the output circuit ofaudio tube V, (Pins 5 and 6 of the powerplug). The slug cores of transformers T,, T,,T:,, and T, are adjusted for maximum signal.The receiver and transmitter sections shouldbe bolted together and the transmitter tuned
Figure 43COMPACT VFO PROVIDES HIGHSTABILITY SIGNAL FOR HIGHFREQUENCY OX BANDSThis small v.t.o. employs latest techniques toachieve stable, drift-tree signal. Only 7Y2" X9" X 7" in size, the unit uses a high-C 1.75Me. oscillator to/lowed by a cathode followerto achieve a maximum degree of circuit isolation.Buffer tuning and control switch are atlett of panel, with large, easily read frequencycontrol dial at center.up wih a dummy load. This will provide injecionsignal for proper receiver operation at220.1 Me. A remote signal (preferably thatof the companion unit) should be used as asignal source and the tuned circuits of the r-famplifiers adjusted for best signal reception.It may be necessary to alter the spacing of coilsLs - Ln to provide proper circuit resonancewhich should occur at about the middle of thetuning range of the associated capacitor.Modulator and Power Supply Construction.The layout of the modulator and power supplycan be seen in figure 41. The units are wiredand the control leads are brought out throughinsulated grommets placed in the end of thechassis. A common ground lead is run fromeach chassis to the switches and controls mountedupon the front panel. The small blowermotor is mounted at the rear of the cabinetand a row of Ys" ventilation holes is drilledaround the top edge of the cabinet as shownin the photograph. The power supply andmodulator should be tested before they aremounted in the cabinet, and the whole assemblyshould be in operating condition beforeit is placed within the cabinet. The tone ofthe audio oscillator (V,) may be varied byplacing a capacitor acwss the primary windingof the oscillator transformer T s.AntennaInstallationTwo separate antennas are requiredfor duplex operation. Toreduce coupling, one link ishorizontally polarized and the other link isvertically polarized as illustrated in figure 42.Care must be taken that identical polarizationis employed at both ends of each link. Threeelementbeam antennas are used for each link.A folded dipole and balun system are employedto provide a good match to the coaxial transmissionline. After the links have been set up,transmitter and receiver tuning adjustmentsshould be peaked to maximize the signals. Itwill be found advantageous to experiment withthe position of the antennas to provide maximumsignal pickup with a minimum of interferencefrom the local transmitter. In general,the receiver antenna should be oriented formaximum signal from the distant station, andthe transmitter antenna should be oriented forminimum local signal pickup. For general use,the two antennas may be attached to the samesupporting structure, separated from each otherby four or five feet. As with any VHF equipment,time spent in placing the antennas willpay big dividends in system operation.28-6 A High StabilityV.F.O. For theDX OperatorStability and freedom from drift are theprime requisites of a high quality variable frequencyoscillator. To meet the requirementsof the discriminating operator the v.f.o. mustbe stable with respect to warm-up, ambienttemperature variations, line voltage shift, vibration,and the presence of a strong r-f field.To achieve these parameters in a unit simpleenough for home construction is a large order.The variable frequency oscillator shown infigures 43, 46, and 47 is a successful attemptto meet these requirements and is recommendedto those operators who desire a v.f.o. havinga high order of stability and resetability.FrequencyStabilityThe perfect variable frequencyoscillator has the frequency determiningelements completelyVOLTAGE MEASUREMENTS, V.F.O.SCREEN (PINJ) 6U8-7QCATHODE (PIN B) 6U8- 10.5R-FVOLTS, GRID (PIN9) 6U6-10.3R-FVOLTS 1 CATHODE (PIN B) 6U8- 9.5R- F VOLTS, J1. ACROSS 47 OHM RESISTOR- 6. 2Figure 44VOLTAGE CHART FOR V.F.O.
<strong>HANDBOOK</strong>High-Stability V.F.O.605(160>)L1teue-teue6AQ5( BO>)L2RFC1NOlES'l-ALL MICA CAPACITORS EL-MENCOSILVER MICA OR EQUIVALENT.2-ALL CERAMIC CAPACITORS DISC-TYPECENTRALAB ODOR EQUIVALENT.eue6AQ.54- WIRE CABLETO SUPPLY~----------------t--------------1---i~ P13 4(6.3V.) (GND.)1( 8~300)2(CONTROL)Figure 45C,-.002 JLfd. Centralab high accuracy capacitor 950-202.Cz-300 JLJLfd. Bud MC-1860C,-25 JLJLfd. Bud LC-1642L1-Vt31 :~:::: f:o2:/. :;~ 0 3~;; :~d~·· approx. I" long. (See text) Wind on National ceramic form XR-72. TapLz-4,5 Me. lnterstage "TV" transformer. J. W. Miller 1466. Remove secondary winding and replacewith B turns #22 d.c.c. wound next to primary winding.RFC,-2.5 mh. National R-100RFCz-VHF choke, 500 ma. National R-60isolated both electrically and physically fromthe rest of the transmitting equipment. Thisgoal cannot be achieved in practice since avacuum tube or transistor must be coupled tothese circuit elements to maintain oscillation.The coupling of such items deteriorates theelectrical isolation of the frequency determiningcircuit. By the use of proper coupling circuitsthe effects of the tube or transistor canbe minimized.The frequency determining circuit is also.subject to temperature variations, changes inthe relative humidity of the surrounding air,and absorption of heat from nearby objects. Byemploying temperature and humidity resistivematerials and removing as many heat generatingcomponents from the vicinity of the frequencydetermining circuits, a good degree ofstability in the v.f.o. may be obtained.Finally, steps must be taken to ensure thatthe oscillator of the v.f.o. unit is completelyfree of parasitics, and that the output waveformFigure 46REAR VIEW OF V.F.O. CHASSIS,SHOWING PLACEMENT OF PARTSOscillator inductor L, is at the left of chassis,separated from the heat producing electrontubes. To right of L, is the precision paddingcapacitor C, mounted to the chassis. The maintuning capacitor is firmly affixed to a heavydural mounting plate bolted to the gear boxof the dial drive. The rear of the tuning capacitoris attached to the chassis by a mountingstud. To the right are the three tubes andthe output transfermer. Directly under thetuning capacitor is the polystyrene chassisplate holding the feed' through bushings forthe oscillator leads to the under-chassis area.Control switch s, is at upper right.
606 Low Power Transmitters<strong>THE</strong> <strong>RADIO</strong>Figure 47UNDER-CHASSIS VIEW OFV.F.O. UNITPower leod filters are at lower right of chassis,with r-f circuit components at left. Allports are firmly mounted to terminals andtie points to reduce vibration. Power leadsare laced. Buffer tuning capacitor C, is atupper left.is low in harmonic content. The variable frequencyexciter to be described in this sectionmeets these fundamental requirements.The V.F.O. Circuit The circuit diagram ofthe high stability v.f.o.is shown in figure 45, and the complete physicallayout may be seen in figures 46 and 47.The pentode section of a miniature 6U8tube is used as the oscillator. The oscillatorcircuit is tuned to the 160 meter region andconsists of a high-Q oscillator coil resonatedto the working frequency by a precision ceramiccapacitor in parallel with a variable aircapacitor. The ceramic capacitor (Centralabtype 950! has a measured temperature coefficientof less than plus or minus ten parts permillion over the temperature range of -40degrees to plus 60 degrees Centigrade. Theuse of ordinary ceramic or silver mica capacitorsas a substitute for this unit is not recommended,since the temperature coefficient ofsuch units cannot be closely controlled inquantity production.The oscillator coil, L,, is wound upon a ceramiccoil form that has a very low coefficientof expansion. The coil turns are spaced to ensurelow inter-turn capacitance. The wire iswound upon the form under tension and at arelatively high temperature. As the wire cools,it shrinks and tightly clasps the form so thatfor all practical purposes the wire and theform have the same coefficient of expansionwith regard to temperature changes. This styleof winding reduces the possibility of the oscillatordeveloping random frequency jumps withchanges in temperature and humidity.A 56 ohm resistor is inserted in series withthe grid lead of the 6U8 oscillator t.ube. Thissuppressor eliminates any tendency toward parasiticoscillation that can result in this typeof oscillator circuit. The parastic tends to makefundamenal oscillation unstable at certain settingsof the oscillator tuning capacitor. Thisinstability may be caused by variations of theinherent inductance of the tuning capacitor atthe frequency of parasitic oscillation.All components of the oscillator circuit arespaced clear of the oscillator tube to preventheat transfer from the tube to the frequencydetermining circuit.The pentode oscillator is R-C coupled tothe triode section of the 6U8 which serves asa cathode follower. The input impedance ofthe cathode follower is extremely high andprovides excellent isolation between the oscillatorcircuit and the output stage. Plate andscreen voltage of the oscillator and plate voltageof the cathode follower are regulated by anOA2 gas regulator to provide maximum isolationfrom power supply fluctuations.The r-f output of the triode section of the6U8 is taken from the low impedance cathodecircuit and is capacity coupled to a 6AQ5 miniaturepentode tube which serves as a frequencydoubler to the 80 meter region. The v.f.o.covers the frequency range of 1750 kc. to1850 kc., and the plate circuit of the 6AQ5doubler tunes 3500 kc. to 3700 kc. The 80meter output is 0.8 watt which is more thansufficient to drive a tetrode buffer such as an807 or 6146. If more output is desired, a6CL6 may be substituted for the 6AQ5 (withappropriate socket wiring changes) to deliverapproximately 2 watts.One filament terminal of each tube isgrounded and the free terminals are bypassedto ground with .005 fLfd. ceramic capacitors.The filament leads then pass through an r-fchoke to ensure maximum lead isolation.Switch S, disables the v.f.o. by removing theplate voltage. A second section of the switchopens an auxiliary circuit that can control thepower relays of the transmitter. A third positionof switch S, permits the v.f.o. to beturned on by itself for zero-beat operation.Transmitter keying for c-w operation is donein the stages following the v.f.o.-exciter toachieve maximum frequency stability and freedomfrom frequency shift during keyinp.
<strong>HANDBOOK</strong>High-Stability V.F.O. 607Mechanical Designof the V.F.O.The mechanical designof the variable frequencyoscillator is equallyimportant as the electrical design if maximumstability and reliability are to be achieved.Provision must be made for dissipation of theheat generated by the vacuum tubes, and thev.f.o. components should be mounted on asheet of conducting material which will act asa heat "sink" and will tend to resist rapidchanges in the temperature of the componentsbolted thereto.The v.f.o. is built upon a cadmium platedsteel chassis measuring 7 Yz" x 9" x 1 Y2 ". Thegear box of the National HRO-type dial isbolted to the chassis and the main tuning capacitorc, is affixed by its front bearing to a3" x 3" x 0.125" dural plate held to thecapacitorgear box by three long machine screwsand three 1 Ys" metal spacers. The tuning capacitoris driven through a high quality flexiblecoupling. This coupling should be free ofback-lash, and should not permit end pressureon the shaft of the capacitor. A ] ohnson 104-250 coupling is recommended. Make sure thatthe shaft of the gear box and that of the capacitorare perfectly in line as misalignmenttends to force a degree of back-lash in thesystem.The rear mounting foot of the capacitor isfastened to the steel chassis by means of a long6/32 bolt and a l-inch metal spacer. This assemblyprovides a very rugged arrangementwith a minimum of flexing between the capacitorand the chassis.Oscillator coil L and the precision ceramiccapacitor C are mounted to the chassis to theside of the tuning capacitor as seen in the rearview photograph. The slug of coil L is removedand discarded before the coil is wound.Moveable oscillator slugs are not conduciveto any great degree of oscillator stability andthis potential source of instability should betaken from the circuit before it can do anydamage.The leads from the oscillator circuit to the6US pentode tube pass through a 1 VI" holecut in the steel chassis. A Ys "-thick square ofpolystyrene is mounted under this hole and twominiature feedthrough bushings are mountedin the insulating plate. The residual capacityto grounds of these important leads is therebyheld to an absolute minimum value.The vacuum tubes and auxiliary circuitsare placed on the opposite side of the tuningcapacitor from the sensitive tuned circuit L-C.In front of the oscillator tube is the SO meteroutput coil L,, the 6AQ5 socket and the OA2socket. Switch S, is mounted to the front paneldirectly in front of the 6AQ5. The front panelis held to the chassis by means of four 6/32bolts placed in the extreme corners of thefront lip of the chassis. The panel is spacedslightly away from the chassis by an extra setof nuts slipped over the bolts before the panelis affixed to the chassis.All control leads to the v .f .o. unit passthrough a four wire cable which enters the underchassis area via a rubber grommet on theback lip of the chassis. Each lead is bypassedwith a .005 /Lfd. ceramic capacitor and is filteredwith a low resistance r-f choke. The coaxialoutput receptacle for the v .f.o. is alsomounted on the rear lip of the chassis, directlybehind the 6US oscillator tube.Wiring the V.F.O. The under-chassis area ofthe v.f.o. should be wiredfirst. Common ground connections are madeto the three sockets by means of soldering lugsand lock washers placed beneath one socket retainingnut. The filament leads are wired next.The bypass capacitors for pins 1, 3, and 4 ofthe 6US socket are placed between the socketpins and the grounded center stud of the socketwith the shortest possible leads. All componentsof the 6US stage should be mounted solidlyin place between adjacent tube pins or tonearby phenolic tie-point terminals. The gridresistor and capacitor of the oscillator sectionare mounted between pin # 2 of the 6U8socket and the inter-chassis feedthrough insulatorin the polystyrene block.The plate coil of the 6AQ5 doubler stage ismade from a 4.5 Me. sound TV i-f transformer.The secondary winding and tuning capacitorare removed and a link winding of 8 turns iswound on the form, closely spaced to the primarywinding. The transformer is then replacedin the shield can and mounted atop thechassis. The r-f choke in the plate circuit ofthe doubler stage is mounted between two terminalsof a phenolic tie-point strip bolted toone of the transformer lugs.The power supply r-f filter circuits areplaced on two four terminal tie-point stripsmounted in the far corner of the chassis. Thebypass capacitors are mounted between the individualterminal lugs and the adjacent groundconnections of the strip and the r-f chokes aremounted between the strips.After the under-chassis wiring is completedthe wiring atop the chassis may be done. Theleads of the tuned circuit are made of # 12tinned wire. The coil is wound with #20
608 Low Power Transmitters<strong>THE</strong> <strong>RADIO</strong>enameled wire space-wound to a length of approximatelyone inch. The easiest way to obtainthe correct spacing is to wind two separatewindings on the coil. One winding is the desiredone of # 20 wire, the second is composedof #24 enameled wire and is used only forspacing. The two windings are put on simultaneouslyand the # 24 wire is removed afterthe other winding has been properly terminatedat both ends. To obtain the best temperaturecoefficient of the coil the wire should bewound on when it is warm. The easiest way toaccomplish this is to place rolls of the twowire sizes in the open and warm them slowlyuntil they are almost too hot to touch. Thewires are then removed from the oven andwound on the coil form before they have achance to cool to room temperature. The wireis kept under tension during the winding processand may be reheated with a soldering ironor heat lamp if it begins to grow cold. Whenthe coil is completed and the wire cools, itwill be found to be tightly wound around thecoil.After the coil has been completed the fourthturn (cathode tap) should be cleaned carefullywith a small, sharp knife blade and ashort length of # 22 wire is soldered to theturn. The opposite end of this lead is attachedto a lug terminal at the base of the coil. Allthe unused lugs may then be removed from thecoil form. A separate ground lead is run fromthe ground lug of the coil form to the groundlug of C and then to the rotor terminal oftuning capacitor C, to prevent any intermittentground paths through the chassis and the capacitormounting assembly.V.F.O. PowerSupplyThe schematic of the v.f.o.power supply is shown infigure 48. The power supplyis constructed as a separate unit since it is desiredto keep the heat and vibration of thetransformers and chokes away from the v.f.o.circuits. The power construction is straightforward,with all leads bypassed to prevent r-fpickup from the transmitter. The supply isbuilt upon a 7 Y:z" x 5 Y:z" miniature amplifierfoundation chassis and provides 300 volts at50 milliamperes for the v.f.o. circuits.V.F.O. Alignmentand AdjustmentAfter the v.f.o. wmnghas been checked the6U8 and OA2 tubesshould be plugged in their sockets. Switch S,is set to the "off" position and the power supplyis turned on. The filaments of the tubesshould light, and when s, is set to either the115 V.'VT15V4-GFigure 48SCHEMATIC, POWER SUPPLY FORVFOTt-325-0-325 volts, 55 ma. Stancor PC-8407CH,, CH.-7 henry at 50 ma. Stancor C-1707"zero-beat" or "transmit" position the v.f.o.signal may be heard in a nearby receiver tunedto the 160 meter region. The oscillator is nowrun for a few hours before it is adjusted tothe correct frequency range. The next step is toplug the 6AQ5 in its socket, and a 6.3 volt,150 rna. (brown bead) pilot lamp is placedacross the terminals of the coaxial output jack]t, serving as a dummy load. The v.f.o. isturned on and the setting of c, and the slugof coil L, are varied to provide an indicationin the bulb.The next step is to calibrate the oscillator. ABC-221 frequency meter or calibrated receivershould be used. The dial of the v.f.o. shouldbe set so the bandspread scale reads "0" whencapacitor C, is fully meshed. When the dialis tuned to a reading of "10" the capacitor isnearly fully meshed, and the operating frequencyof the oscillator should be 3500.00 kc.Unless you are extremely lucky, this will notbe the case. Since there are no adjustable paddingcapacitors in the frequency determiningcircuitcoil L, must be adjusted a bit at a timeuntil the correct frequency falls at the designateddial reading. The turns of L may bevaried in position with the aid of a pen-knifeblade, and minute adjustments to the top fewturns may be made until the correct calibrationis reached. Once the calibration of theoscillator has been set, the turns of L may bepermanently fixed in place by means of a fewdrops of colorless nail polish. Use the minimumamount of polish possible. Placing thepolish on the coil will vary the operating frequencyslightly, so a final slight adjustmentmust be made after the last spot of polish hasbeen placed on the coil. Full coverage of allthe amateur bands except 80 meters will beobtained with this value of tuning capacitor.Decreasing the size of c, will provide greaterbandspread on the high frequency bands, and
<strong>HANDBOOK</strong>High-Stability Y.F.O. 609increasing the capacitance of C will providefull coverage of the 80 meter band. In eithercase, it will be necessary to juggle the turns onL to obtain the correct tuning range.Operating Checkof the V.F.O.When completed the v.f.o.should be placed in theoperating position and runfor a period of several hours. During this time,the frequency of the v.f.o. should be comparedagainst a known standard such as a 100 kc.crystal, or Standard Frequency Station WWV.Under normal conditions the v.f.o. will havea small positive initial warm-up drift of lessthan 100 cycles on 80 meters. It should settledown after a period of time and remain relativelystable if the temperature of the room isconstant.Temperature stabilization may be accomplishedby the addition of a small value ofnegative coefficient capacitance to the tuningcircuit. This is a cut-and-try process and is notrecommended unless the builder has plenty oftime and previous experience with the task.It is not really required unless the oscillator isoperated under extremes of room temperature.The last step is to place the v.f.o. in themetal cabinet. The unit should not be operatedout of the cabinet as the enclosure affordssome degree of shielding from the r-f field ofthe transmitter. The rear of the chassis isfastened to the bottom of the cabinet by twosheet metal screws passed from the under sideof the bottom of the cabinet up into the rearlip of the chassis. To prevent frequency changesduring operation the lid of the cabinet shouldbe fastened shut by means of two sheet metalscrews run through the front corners of thelid into the cabinet.
CHAPTER TWENTY-NINEThe trend in design of transmitters for operationon the high frequency bands is towardthe use of a single high-level stage. The mostcommon and most flexible arrangement includesa compact bandswitching exciter unit,with 15 to 100 watts output on all the highfrequencybands, followed by a single poweramplifier stage. In many cases the exciter unitis placed upon the operating table, with a coaxialcable feeding the drive to the power amplifier,although some operators prefer to havethe exciter unit included in the main transmitterhousing.This trend is a natural outgrowth of the increasingimportance of v-f-o operation on theamateur bands. It is not practical to make aquick change in the operating frequency of atransmitter when a whole succession of stagesmust be returned to resonance following thefrequency change. Another significant factor inimplementing the trend has been the wide acceptanceof commercially produced 75 and150-watt transmitters. These units provide r-f610excitation and audio driving power for highlevelamplifiers running up to the 1000-wattpower limit. The amplifiers shown in thischapter may be easily driven by such exciters.29-1 Power AmplifierDesignChoice ofTubesEither tetrode or triode tubes maybe used in high-frequency poweramplifiers. The choice is usuallydependent upon the amount of driving powerthat is available for the power amplifier. Iia transmitter-exciter of 100-watt power capabilityis at hand (such as the Heath TX-1)it would be wise to employ a power amplifierwhose grid dPiving requirements fall innhe same range as the output power of theexciter. Triode tubes running !-kilowatt input(plate modulated) generally require some58 to 80 watts of grid driving power. Sucha requirement is easily met by the outputlevel of the 100-wat,t transmitter which should
Design611TO ANTENNACIRCUITTO ANTENNACIRCUIT+UNBALANCED COAXIALFEED SYSTEM+BALANCED TWIN- LINEFEED SYSTEMFigure 1LINK COUPLED OUTPUT CIRCUITSFOR PUSH-PULL AMPLIFIERSbe employed as the exciter. Tetrode tubes(such as the 4-250A) require only 10 to15 watts of actual drive from the exciter forproper operation of the amplifier stage at 1-kilowatt input. This means that the outputfrom the 100-watt transmitter has to be cutdown ~o the 15 watt driving level. This is anuisance, as it requires the addition of swampingresistors to the output circuit of the transmitter-exciter.The triode tubes, therefore,would lend themselves to a much more convenientdriving arrangement than would thetetrode tubes, simply because their grid driverequirements fall within the power outputrange of the exciter unit.On the other hand, if the transmitter-exciteroutput level is of the order of 15 - 40 watts(the ] ohnson Ranger, for example) sufficientdrive for triode tubes running 1-kilowatt inputwould be lacking. Tetrode tubes requiring lowgrid driving power would have to be employedin a high-level stage, or smaller triode tubes requiringmodest grid drive and running 250watts or so would have to be used.Power AmplifierDesign-Choiceof CircuitsEither push-pull or singleended circuits may be employedin the power amplifier.Using modern tubesand properly designed circuits, either type iscapable of high efJiciency operation and lowharmonic output. Push-pull circuits, whetherusing triode or tetrode tubes usually employlink coupling between the amplifier stage andthe feed line running to the antenna or the antennatuner.It is possible to use the link circuit in eitheran unbalanced or balanced configuration, asshown in figure 1, using unbalanced coaxialline, or balanced twin-line.BIAS 110 v. +Figure 2CONVENTIONAL PUSH-PULLAMPLIFIER CIRCUITThe mechanical layout should be symmetricaland the output coupling provision must beevenly balanced with respect to the plate coilc,-Approx. 1.5 JLJLfd. per meter of wavelengthper sectionC,-Refer to plate tank capacitor design inChapter IIc,-May be 500 JLJLfd., 10,000-volt type ceramiccapacitorNC-Max. usable capacitance should be greater,and min. capacitance less than ratedgrid-plate capacity of tubes in amplifier.50o/o greater air gap than c,.R,-100 ohms, 20 watts. This resistor servesas low Q r-f choke.RFC ,-All-band r-f choke suitable for platecurrent of tubesM,-M,-Suitable meters for d-e grid and platecurrentsAll low voltage .001 JLfd. and .01 JLfd. by-passcapacitors are ceramic disc units (CentralabDD or equiv.)L,-50-watt plug-in coil, center linkL,-Piug-in coil, center link, of suitable powerrating.Common technique is to employ plug-inpla-te coils with the push-pull amplifier stage.This necessitates some kind of opening for coilchanging purposes in the "electrically tight'"enclosure surrounding the amplifier stage. Caremus·t be used in the design and constructionof the door for this opening or leakage of harmonicsthrough the opening will result, withthe attendant TVI problems.Single ended amplifiers may also employlink-coupled output devices, although the trendis to use pi-network circuits in conjunctionwith single ended tetrode stages. A tapped orotherwise variable nank coil may be used whichis adjustable from the front panel, eliminatingthe necessity of plug-in coils and openingsinto the shielded enclosure of the amplifier.Pi-network circuits are becoming increasinglypopular as coaxial feed systems are cominginto use to couple the output circuits of transmittersdirectly to the antenna.
612H.F. Power Amplifiers<strong>THE</strong> <strong>RADIO</strong>29-2Push-Pull TriodeAmplifiersFigure 2 shows a basic push-pull triodeamplifier circuit. While variations in the methodof applying plate and filament voltages andbias are sometimes found, the basic circuitremains the same in all amplifiers.Filament Supply The amplifier filament transformershould be placedright on the amplifier chassis in close proximityto the rubes. Short filament leads are necessaryto prevent excessive voltage drop in the connectingleads, and also to prevent r·f pickupin the filament circuit. Long filament leadscan often induce instability in an otherwisestable amplifier circuit, especially if the leadsare exposed to the radiated field of the platecircuit of the amplifier stage. The filamentvoltage should be the correct value specif,iedby the tube manufacturer when measured at thetube sockets. A filament transformer having atapped primary often will be found useful inadjusting the filament voltage. When there isa choice of having the filament voltage slightlyhigher or slightly lower than normal, thehigher voltage is preferable. If the amplifier isto be overloaded, a filament voltage slightlyhigher than the rated value will give greatertube life.Filament bypass capacitors should be low internalinductance units of approximately .01!Lfd. A separate capacitor should be used foreach socket terminal. Lower values of capacitanceshould be avoided to prevent spuriousresonances in the internal filament structure ofthe tube. Use heavy, shielded filament leads forlow voltage drop and maximum circuit isolation.Plate Feed The series plate voltage feedshown in figure 2 is the mostsatisfactory method for push-pull stages. Thismethod of feed puts high voltage on the platetank coil, but since the r-f voltage on the coilis in itself sufficient reason for protecting thecoil from accidental bodily contact, no additionalp11otective arrangements are made necessaryby the use of series feed.The insulation in the plate supply circuitshould be adequate for the voltages encountered.In geneml, the insulation should berated to withstand at least four times the maximumd-e plate voltage. For safety, the platemeter should be placed in the cathode returnlead, since there is danger of voltage breakdownbetween a metal panel and the metermovement at plate voltages much higher thanone thousand.Grid Bias The recommended method of obtainingbias for c-w or plate modulatedtelephony is to use just sufficient fixedbias to protect the tubes in the event of excitationfailure, and to obtain the rest by thevoltage drop caused by flow of rectified gridcurrent through a grid resistor. If desired, thebias supply may be omitted for telephony if anoverload relay is incorporated in the plate circuitof the amplifier, the relay being adjustedto trip immediately when excitation is removedfrom the stage.The grid resistor R, serves effectively asan r-f ~choke in the grid circuit btx:ause the impressedr-f voltage is low, and the Q of theresistor is poor. No r-f choke need be used inthe grid bias return lead of the amplifier, otherthan those necessary for harmonic suppression.The bias supply may be built upon the amplifierchassis if care is taken to prevent r-ffrom finding its way into the supply. Ampleshielding and lead filtering must be employedfor sufficient isolation.The Grid Circuit As the power in the gridcircuit is much lower thanin the plate circuit, it is customary to use aclose-spaced split-stator grid capacitor withsufficient capacitance for operation on thelowest frequency band. A physically small capacitorhas a greater ratio of maximum tominimum capacitance, and it is possible to obtaina unit that will be satisfactory on all bandsfrom 10 to 80 meters without the need for auxiiiarypadding capacitors. The rotor of the gridcapacitor is grounded, simplifying mounting ofthe capacitor and providing circuit balance andelectrical symmetry. Grounding the rotor alsohelps to retard v-h-f parasitics by by-passingthem to ground in ,the grid circuit. The L/Cratio in the grid circuit should be fairly low,and care should be taken that circuit resonanceis not reached with the grid capacitor atminimum capacitance. That is a direct invitationfor instability and parasitic oscillationsin the stage. The grid coil may be wound ofno. 14 wire for driving powers of up to 100watts. To restrict the field and thus aid inneutralizing, the grid coil should be physicallyno larger than absolutely necessary.Circuit Layout The most important considerationin constructing apush-pull amplifier is to maintain electricalsymmetry on both sides of the balanced cir-
<strong>HANDBOOK</strong>Design 613cuit. Of utmost importance in maintaining electricalbalance is the control of stray capacitancebetween each side of the circuit andground.Large masses of metal placed near one sideof the grid or plate circuits can cause seriousunbalance, especially at the higher frequencies,where the tank capacitance between oneside of the tuned circuit and ground is oftenquite small in itself. Capacitive unbalancemost often occurs when a plate or grid coil islocated with one of its ends close to a metalpanel. The solution to this difficulty is tomount the coil parallel to the panel to makethe capacitance to ground equal from each endof the coil, or to place a grounded piece ofmetal opposite the "free" end of the coil toaccomplish a capacity balance.Whenever possible, the grid and plate coilssrhould be mounted at right angles to eachother, and should be separated far enoughapart to reduce coupling between them to aminimum. Coupling between the grid and platecoils will tend to make neutralization frequencysensitive, and it will be necessary to readjustthe neutralizing capacitors of the stage whenchanging bands.All r-f leads should be made as short anddirect as possible. The leads from the tubegrids or plates should be connected directlyto their respective tank capacitors, and theleads between the tank capacitors and coilsshould be as heavy as the wire that is usedin the coils themselves. Plate and grid leadsto the tubes may be made of flexible tinnedbraid or flat copper strip. Neutralizing leadsshould run directly to the tube grids and platesand should be separate from the grid and plateleads to the tank circuits. HltVing a portion ofthe plate or grid connections to their tank circuitsserve as part of a neutralizing lead canoften result in amplifier instability at certainoperating frequencies.ExcitationRequirementsIn general it may be statedthat the overall power requirementfor grid circuit excitationto a push-pull triode amplifier is approximately10 per cent of the amount of the power outputof the stage. Tetrodes require about 1 per centto 3 percent excitation, referred to the poweroutput of the stage. Excessive excitation topentodes or tetrodes will often result i11 reducedpower output and efficiency.Push-PullAmplifierConstructionSymmetry is the secret of suecessfulamplifier design. Shownin figure 3 is the top v,iew ofa 350 watt push-pull all bandFigure 3LAYOUT OF 350-WATT PUSH-PULLTRIODE AMPLIFIERTwo 81 1 -A tubes are employed in this circuit.Plate tuning capacitor is at left of chassis,with swinging-link type plug-in coil assemblymounted above it. Rotor of split-stator capacitormay be insulated from ground to increasevoltage breakdown rating of capacitor. Notethat pickup link is series-tuned to reduce circuitreactance. One corner af rotor plate ofseries capacitor is bent so that capacitorshorts itself out at maximum capacitance.Grid circuit coil and capacitor are at right.Center-linked plug-in coil is employed. Parasiticchokes are placed in grid leads adjacentto the tube sockets, and tube filaments arebypassed to ground with .01 !'fd. ceramiccapacitors. Complete area above the chassisis enclosed with perforated screen to reduceradiation of r.f. energy.amplifier employing 811-A tubes. The circuitcorresponds to that shown in figure 2 exceptthat the 811-A's are zero bias tubes. The biasterminals of the circuit are therefore jumperedtogether and no external bias supply is requiredat plate potentials less than 1300 volts.All r-f components are mounted abovedeck. The plate circuit tuning capacitor andswinging link tank coil are to the left, withthe two disc-type neutralizing capacitors betweenthe tank circuit and the tubes. At theright of the chassis is the grid tank circuit.Small parasitic chokes may be seen betweenthe tube sockets and the grid circuit. Plateand grid meters are placed in the under-chassisarea where they are shielded from the r-f fieldof the amplifier.Larger triode tubes such as the 810 and8000 make excellent r-f amplifiers at the kilowattlevel, but care must be taken in amplifierlayout as the inter-electrode capacitance ofthese tubes is quite high. One tube and oneneutralizing capacitor is placed on each sideof the tank circuit (figures 4 and 5) to permitvery short interconnecting leads. The relativeposition of the tubes and capacitors is trans-
614 H.F. Power Amplifiers<strong>THE</strong> <strong>RADIO</strong>Figure 4UNIQUE CHASSIS LAYOUT PERMITSSHORT LEADS IN KILOWATTAMPLIFIERLarge size components required for high levelamplifier often complicate amplifier layout.In this design, the plate tank capacitor sitsastride small chassis running lengthwise onmain chassis. Inductor is mounted to phenolicplate atop capacitor. Variable link is paneldriven through right-angle gear drive. Platecircuit is grounded by safety arm when paneldoor is opened. Note that plate capacitor ismounted on four TV -type capacitors whichserve to bypass unit, and also act as supports.A small parasitic choke is visible next to thegrid terminal of the 810 tube.posed on each side of the cha5sis, as shown inthe illustrations. The plate tank coil is mountedparallel to the front panel of the amplifieron a phenolic plate supported by the tuningcapacitor which sits atop a small chassis-typebox. The grid circuit tuning capacitor is locatedwithin this box, as seen in figure 6. An externalbias supply is required for proper amplifieroperation. Operating voltages may be determinedfrom the instruction sheets for theparticular tube to be employed.Whenever the amplifier enclosure requiresa panel door for coil changing access it is wiseto place a power interlock on the door thatwill turn off the high voltage supply wheneverthe door is open!Figure 5LEFT-HAND VIEW OF KILOWATTAMPLIFIER OF FIGURE 4Above shielded meter box is the protective11micro-switch" which opens the primary powercircuit when the panel door is not closed. Tubesockets are recessed in the chassis so thattop of tube socket shells are about 1;2-inchabove chassis level. On right side of amplifier(facing it from the rear) the tube socket isnearest the panel, with the neutralizing capacitorbehind it. On the opposite side, thecapacitor is nearest the panel with the tubedirectly behind it. This layout transpositionproduces very short neutralizing leads, sinceconnections may be made through the statorof plate tuning capacitor.29-3Push-PullTetrode AmplifiersTetrode tubes may be employed in push-pullamplifiers, although the modern trend is toparallel operation of these tubes. A typicalcircuit for push-pull operation is shown infigure 7. The remarks concerning the filamentsupply, plate feed, and grid bias in Section29-2 apply equally to tetrode stages. Becauseof the high circuit gain of the tetrode amplifier,extreme care must be taken to limitinterstage feedback to an absolute minimum.Many amateurs have had bad luck with retrodetubes and have b~en plagued with parasiticsand spurious oscillations. It must be rememberedwith high gain tubes of this type
<strong>HANDBOOK</strong>P-P Tetrode Amplifier615Figure 6UNDER CHASSIS VIEW OF1-KILOWATT TRIODE AMPLIFIERThe grid circuit tuning capacitor and platecircuit r-f choke are contained in the belowchassis enclosure formed by a small chassismounted at right angles to the front panel.The bandswitch coil assembly lor the gridcircuit is mounted on two brackets above thiscutout. A metal screen attached to the bottomol the amplifier completes the TVI-proolenclosure.that almost full output can be obtained withpractically zero grid excitation. Any minuteamount of energy fed back from the plate circuitto the grid circuit can cause instability oroscillation. Unless suitable precautions are incorporatedin the electrical and mechanical designof the amplifier, this energy feedback willinevitably occur.Fortunately these precautions are simple.The grid and filament circuits must be isolatedfrom the plate circuit. This is done by placingthese circuits in an "electrically tight"" box.All leads departing from this box are by-passedand filtered so that no r..£ energy can passalong the leads into the box. This restricts theenergy leakage path between the plate and gridcircuits to the residual plate-to-grid capacityof the tetrode tubes. This capacity is of theorder of 0.25 flfLfd. per tube, and under normalconditions is sufficient to produce a highly regenerativecondition in the amplifier. Whetheror not the amplifier will actually break into oscillationis dependent upon circuit losses andresidual lead inductance of the stage. Sufficeto say that unless the tubes are actually neutralizeda condition exists that will lead tocircuit instability and oscillation under certainoperating conditions. With luck, and aFigure 7CONVENTIONAL PUSH-PULLTETRODE AMPLIFIER CIRCUITPush-pull amplifier uses many ol the samecomponents required by triode tubes (seefigure 2). Screen supply is also required.B-Biower lor filament seals ol tubes.C~,-Low internal inductance capacitor, .001JIId., SKY. Centralab type 8585- 1000.NC-See text and figure 8.PC-Parasitic choke. 50 ohm, 2-watt compositionresistor wound with 3 turns # 12 e.wire.Note: Strap multiple screen terminals togetherat socket with %" copper ribbon. AttachPC to center ol strap.heavily loaded plate circuit, one might be ableto use an un-neutralized push-pull tetrode amplifierstage and suffer no ill effects from theresidual grid-plate feedback of the tubes. Infact, a minute amount of external feedback inthe power leads to the amplifier may just (bychance) cancel out the inherent feedback ofthe amplifier circuit. Such a condition, however,results in an amplifier that is not "reproduceable."There is no guarantee that a duplicateamplifier will perform in the same, stablemanner. This is the one, great reason thatmany amateurs having built a tetrode amplifierthat "looks just like the one in the book" findout to their sorrow that it does not "work likethe one in the book."This borderline situation can easily be overcomeby the simple process of neutralizing thehigh-gain tetrode tubes. Once this is done, andthe amplifier is tested for parasitic oscillations(and the oscillations eliminated if theyoccur) the tetrode amplifier will perform in anex;cellent manner on all bands. In a word, itwill be "reproduceable."
616 H.F. Power Amplifiers<strong>THE</strong> <strong>RADIO</strong>Figure 8REAR VIEW OF PUSH PULL4-250A AMPLIFIERThe neutralizing rods are mounted on ceramicfeedthrough insulators adjacent to each tubesocket. Low voltage power leads leave thegrid circuit compartment via Hypass capacitorslocated on the lower left corner of thechassis. A screen plate covers the rear of theamplifier during operation. This plate wasremoved for the photograph.As a summation, three requirements mustbe met for proper operation of tetrode tubeswhetherin a push-pull or parallel mode:1. Complete isolation must be achieved betweenthe grid and plate circuits.2. The tubes must be neutralized.3. The circuit must be parasitic-free.AmplifierConstructionThe push-pull tetrode amplifiershould be built aroundtwo "r-f tight" boxes for thegrid and plate circuits. A typical layout that hasproven very satisfactory is shown in figures 8and 9. The amplifier is designed around a Barker& Williamson "butterfly" tuning capacitor.The 4-250A tetrode tubes are mounted at therear of the chassis on each side of the capacitor.The base shells of the tubes are groundedby spring clips, and short adjustable rods projectup beside each tube to act as neutralizingcapacitors. The leads to these rods are crossconnectedbeneath the chassis and the rodsprovide a small value of capacitance to theplates of the tubes. This neutralization is necessarywhen the tube is operated with highFigure 9UNDER CHASSIS VIEW OF4-250A AMPLIFIERThe bias supply for the amplifier is mountedat the front of the chassis between the twocontrol shafts. A blower motor is mountedbeneath each tube socket. A screened plateis placed on the bottom of the chassis to cQmpletethe under-chassis shielding.power gain and high screen voltage. As theoperating frequency of the tube is increased,the induotance of the internal screen supportlead of the tube becomes an important part ofthe screen ground return circuit. At some criticalfrequency (about 45 Me. for the 4-250Atube) the screen lead inductance causes aseries resonant condition and the tube is saidto be "self-neutralized" at this frequency.Above this frequency the screen of the tetrodetube cannot be held at ground potential by theusual screen by-pass capacitors. With normalcircuitry, the tetrode tube will have a tendencyto self-oscillate somewhere in the 120 Me. to160 Me. region. Low capacity tetrodes that canoperate efficiently at such a high frequency arecapable of generating robust parasitic oscillationsin this region while the operator is vainlytrying to get them operating at some lowerfrequency. The solution is to introduce enoughloss in the circuit at the frequency of theparasitic so as to render oscillation impossible.This procedure has been followed in thisamplifier.During a long series of experiments designedto stabilize large tetrode tubes, it wasfound that suppression circuits were most effectivewhen inserted in the screen lead of the
<strong>HANDBOOK</strong>Pi-Network Amplifiers 617tetrode. The screen, it seemed, would have r-fpotentials measuring into the thousands of voltsupon it during a period of parasitic oscillation.By-passing the screen to ground with copperstrap connections and multiple by-pass capacitorsdid little to decrease the amplitude of theoscillation. Excellent parasitic suppression wasbrought about by strapping the screen leadsof the 4·250A socket together (figure 7) andinserting a parasitic choke between the screenterminal of the socket and the screen by-passcapacitor.After this was done, a very minor tendencytowards self-oscillation was noted at extremelyhigh plate voltages. A small parasitic chokein each grid lead of the 4-250A tubes eliminatedthis completely.The neutralizing rods are mounted upon twofeedthrough insulators and cross-connected tothe 4-250A control grids beneath the chassis.These rods are threaded so that they may berun up and down the insulator bolt for neutralizingadjustment.Because of the compact size of many tetrodesit is necessary to cool the filament seals of thetube with a blast of air. A smaU blower can bemounted beneath the chassis to project coolingair directly at the socket of the tube as shownin figure 9.Inductive Tuning ofPush-Pull AmplifiersThe plate tank circuitof the push-pull amplifiermust have a lowimpedance to ground M harmonic frequenciesto provide adequate harmonic suppression. Theusual split-stator tank capacitor, however, hasan uncommonly high impedance in the VHFregion wherein the interference-causing harmonicslie. A push-pull vacuum-type capacitormay be used as these units have very low internalinductance, but the cost of such a capacitoris quite high.A novel solution to this problem is to employa split stator capacitor made up of twoinexpensive fixed vacuum capacitors. Amplifieradjustment can then beSit be accomplishedby inductive tuning of the plate tank coil asseen in figure 10. Two fixed vacuum capacitorsare mounted vertically upon the chassisand the upper terminals are attached to theplates of the amplifier tubes by means of lowimpedance straps. Resonance is established byrotation of a shorted copper loop located withinthe amplifier tank coil. This loop is madeof a %" long section of copper water pipe,two inches in diameter. Approximate resonanceis established by varying the spacing betweenthe turns of the copper tubing tank coil. Inductivecoupling is used between the tank coiland the antenna circuit in the usual manner.Sufficient range to enable the operator to covera complete high frequency band may be hadwith this intere~ting tuning method.29-4 T etrode Pi-Network AmplifiersThe most popular amplifier today for bothcommercial and amateur use is the pi-networkconfiguration shown in figure 11. This circuitis especially suited to tetrode tubes, althoughtriode tubes may be used under certain circumstances.A common form of pi-network amplifier isshown in figure 11A. The pi circuit forms thematching system between the plate of the amplifiertube and the low impedance, unbalancedantenna circuit. The coil and input capacitorFigure 10INDUCTIVE TUNING MAY BEEMPLOYED IN HIGH POWERAMPLIFIERTwo fixed vacuum capacitors form split-statorcapacitance, providing very low inductanceground path for plate circuit harmonics. Tuningis accomplished by means of shorted,single-turn link placed in center of tankcoil. Shorted link is made from %·inch sectioncut from copper water pipe. Larger link outsideof tank coil is antenna pick-up coil.
618 H.F. Power Amplifiers <strong>THE</strong> RA D I 00~Bl AS + 115 v.rv,£"EXCITATION! T 9RFC1@.QQlc-V1 C2. L2.~52.~BIAS + 1 tSV.'\..h! 'l":M,C4p =RFC1'"""''' r:.o'cR1V1 C2. L2.LOW ZOUTPUr-=- .a 1c.oatcrw1- BIAS + t15v.rvFigure 11TYPICAL PI-NETWORK CONFIGURATIONSA-Split grid circuit provides out-of-phase voltage for grid neutralization of tetrode tube. Rotary coilis employed in plate circuit, with small, fixed auxiliary coil for 28 Me. Multiple tuning grid tank T 1covers 3.5- 30 Me. without switching.8-Tapped grid and plate inductors are used with "bridge type" neutralizing circuit for tetrode amplifierstage. Vacuum tuning capacitor is used in input section of pi-network.C-Untuned input circuit (resistance loaded) and plate inductor ganged with tuning capacitor comprisesimple amplifier configuration.PC1, PC:-57 ohm, 2 watt composition resistor, wound with 3 turns # 18 c. wire.
<strong>HANDBOOK</strong>Pi-Network Amplifiers 619of the pi may be varied to tune the circuit overa 10 to 1 frequency range (usually 3.0 - 30Me.). Operation over the 20 - 30 Me. rangetakes place when the variable slider on coil L,is adjusted to short this coil out of the circuit.Coil L therefore comprises the tank inductancefor the highest portion of the operating range.This coil has no taps or sliders and is constructedfor the highest possible Q at the highfrequency end of the range. The adjustablecoil (because of the variable tap and physicalconstruction) usually has a lower Q than thatof the fixed coil.The degree of loading is controlled by capacitorsC and C. The amount of circuit capacityrequired at this point is inversely proportionalto the operating frequency and tothe impedance of the antenna circuit. A loadingcapacitor range of 100 flflfd. to 2500flflfd. is normally ample to cover the 3.5 - 30Me. range.The pi circuit is usually shunt-fed to removethe d.c. plate voltage from the coils and capacitors.The components are held at ground potentialby completing the circuit groundthrough the choke RFC. Great stress is placedupon the plate circuit choke RFC,. This componentmust be specially designed for thismode of operation, having low inter-turn capacityand no spurious internal resonancesthroughout the operating range of the amplifier.Parasitic suppression is accomplished bymeans of chokes PC-1 and PC-2 in the screenand grid leads of the tetrode. Suitable valuesfor these chokes are given in the parts list offigure 11. Effective parasitic suppression is dependentto a large degree upon the choice ofscreen bypass capacitor C. This componentmust have extremely low inductance throughoutthe operating range of the amplifier andwell up into the VHF parasitic range. The capacitormust have a voltage rating equal to atleast twice the screen potential (four times thescreen potential for plate modulation). Thereare practically no capacitors available that willperform this difficult task. One satisfactory solutionis to allow the amplifier chassis to formone plate of the screen capacitor. A '"sandwich"is built upon the chassis with a sheet of insulatingmaterial of high dielectric constant anda matching metal sheet which forms the screenside of the capacitance. A capacitor of thistype has very low internal inductance but isvery bulky and takes up valuable space beneaththe chas:sis. One suitable capacitor forthis position is the Centralab type 858S-1000,~EXCITER --·----AMPLIFIER------!807+ -BIASFigure 12CAPACITIVE COUPLING CIRCUITBETWEEN EXCITER AND FINALAMPLIFIER STAGE MAKES USEOF ONE COMMON TANK CIRCUITGrid circuit of amplifier stage serves as platecircuit of exciter In this simplified circuit.Leads between the tubes must be kept shortand direct.rated at 1000 flflfd. at 5000 volts. This compactceramic capacitor has relatively low internalinductance and may be mounted to thechassis by a 6-32 bolt. It is shown in variousamplifiers described in this chapter. Furtherscreen isolation may be provided by a shieldedpower lead, isolated from the screen by a .001l'fd. ceramic capacitor and a 100 ohm carbonresistor.Various forms of the basic pi-network amplifierare shown in figure 11. The A configurationemploys the so-called ""all-band"' gridtank circuit and a rotary pi-network coil in theplate circuit. The B circuit uses coil switchingin the grid circuit, bridge neutralization, anda tapped pi-network coil with a vacuum tuningcapacitor. Figure 11C shows an interestingcircuit that is becoming more popular forclass AB 1 linear operation. A tetrode tube operatingunder class AB1 conditions draws nogrid current and requires no grid drivingpower. Only r-f voltage is required for properoperation. It is possible therefore to dispensewith the usual tuned grid circuit and neutral.izingcapacitor and in their place employ a simpleload resistor in the grid circuit acrosswhich the required excitation voltage may bedeveloped. This resistor can be of the order of50- 300 ohms, depending upon circuit requirements.Considerable power must be dissipatedin the resistor to develop sufficiwtgrid swing, but driving power is often cheaperto obtain than the cost of the usual grid circuitcomponents. In addition, the low impedancegrid return removes the tendency towardsinstability that is so common to thecircuits of figure 11A and 11B. Neutralizationis not required of the circuit of figure 11 C,and in many cases parasitic suppression maybe omitted. The price that must be paid is the
620 H.F. Power Amplifiers<strong>THE</strong> <strong>RADIO</strong>Figure 13MIDGET LINEAR AMPLIFIER FORMOBILE SIDEBAND OPERATIONFEATURES WATER COOLED4W-300B TETRODE TUBESMiniature amplifier is comparable in size withsideband exciter. Antenna, plate, and screencurrent are monitored by panel /amps used inplace ol larger and more expensive meters.Inductor slug of grid coil is located belowplate circuit tuning capacitor.additional excitation that is required to developoperating voltage across grid resistor R '·The pi-network circuit of figure 11C isinteresting in that the rotary coil L, and theplate tuning capacitor C are ganged togetherby a gear train, enabling the circuit to be tunedto resonance with one panel control instead ofthe two required by the circuit of figure 11A.Careful design of the rotary inductor will per·mit the elimination of the auxiliary high frequencycoil L, reducing the cost and complexityof the circuit.A single tuned circuit may be employed tocouple the power amplifier grid circuit to thebuffer stage as shown in figure 12. Care mustbe taken to ensure that a good ground existsbetween the two circuits.29-5 A CompactLinear Amplifier forMobile SSBSingle sideband transmission is ideal formobile operation because it provides the maximumamount of "talk power" for a givenamount of primary power drain. The new threephase alternator systems available for automotiveuse are capable of supplying over onethousand watts of primary power, making theproposition of a kilowatt sideband transmittera reality.The compact kilowatt linear amplifier describedin this section is designed for remotecontrol operation on any one amateur band. Itmay be driven by any exciter capable of oneor two watts p.e.p. output.AmplifierCircuitThe circuit of the mobile linearamplifier is shown in figure 1-LDesigned around two Eimac4W -300B water cooled tetrode tubes, the amplifiermay nevertheless be used with 4X-250Bor 4CX-300A tubes if air cooling is employed.For purely mobile operation the water cooledtubes are recommended as air blowers are bulky,noisy, and draw a high current from theautomobile primary power system. Water cooling,although unfamiliar to many amateurs, isrelatively simple and inexpensive. A StewartWarner fuel pump is used to pump distilledwater from a reservoir through polyethylenetubes to the anodes of the 4W-300B tetrodeand back through other tubes to the reservoir.This container is a surplus one gallon G.I.gasoline can (figure 1 7). The temperature ofthe water remains under 100°F. even when thetransmitter is operated for extended periods oftime. The water jackets of the tubes are connectedin series as shown in the drawing. Ithas been found by experiment that it is notnecessary to have any other form of coolingmedium to maintain the tubes at a safe temperature.The water pump draws a current of0.2 amperes at maximum pumping speed. Itis mounted in a corner of the turtle-back ofthe automobile, along with the G.I. water can.The pump motor is connected so that it startsas soon as filament voltage is applied to thetwo 4W-300B tubes.The two tetrode tubes are connected in parallelwith a parallel resonant plate circuit anda simple "pi-type"' grid circuit. The plate circuitconsists of a minature Jennings GSLA-120variable vacuum capacitor (C) in parallelwith a silver plated copper tubing tank coil.Plate voltage is fed to the tubes through a r-fchoke and the anodes of the tubes are coupledro the tank circuit through a 500 ,u,ufd., 5 KVceramic capacitor. The low impedance antennasystem is directly tapped at a low impedancepoint on the tank coil. A small 6 volt automobileheadlamp is placed in series with theantenna lead to provide a simple indicator ofr-f current. Antenna transfer is accomplishedby means of a minature Jennings RB-1 s.p.d.t.vacuum relay.Special low impedance sockets for thesetubes are supplied by the Eimac Co. and theiruse is recommended. The sockets have a built-
<strong>HANDBOOK</strong>Mobile SSB Amplifier62182Figure 14SCHEMATIC, MOBILE LINEAR AMPLIFIERc,-120 JLJLid., SKY variable vacuum capacitor. Jennings GSLA-120L,-(20 meters): J5 turns #I Be., o/a" diam., I" long. Adjust to resonance with J,shorted and tubes in sockets.L,-(20 meters); 6 turns, '14-inch silver plated copper tubing, 2'/ 2 " i.d., 2" long.Adjust antenna tap lor proper loading.RFC,-2.5 mh. National R-100RFC,-225 turns #28 manganin, o/a" dia., 2" long, in series with National R-100choke. Or, heavy-duty Ray par RL- 102 choke may be substituted lor these twoitems.RY,-SPDT relay, d.c. coil to match battery voltage. Jennings RB-I vacuum relayemployedB,, 8:, 8,-/ndicator pilot lamps. 81 should have 500 ma. rating; Bz, 60 ma. rating;and 8, may be automobile headlight lamp heavy enough to carry antennaeurrent.in screen capacitor which provides an extremelylow impedance screen ground return circuit,reducing the problem of parasitics to a minimum.A simple "series" tuned input circuit isemployed in the grid circuit. The internal inputcapacitance of the tubes is effectively inparallel with the grid coil resonating it to theoperating frequency. Since no grid current isdrawn during class ABl operation a small batterymay be used to provide the correct valueof operating bias. Flashlight bulbs are placedin the screen and negative high voltage leadsinstead of meters to conserve space.AmplifierConstructionClass ABl operation plus thecomplete absence of parasiticsresults in a very minimumamount of harmonic generation. The use of alow inductance variable vacuum capacitor providesan effective return path to ground forhigh frequency harmonics appearing in theFigure 15SMALL SIZE OF LINEAR AMPLIFIERIS DUE TO USE OF MINIATURE300 WATT, WATER COOLEDTETRODE TUBESParallel operation of the 4W-300B tubes isemployed. Plate tank inductor is at right,with miniature variable vacuum capacitor behindit. Plate choke is in the foreground,made up of a 2.5 mh. choke in series with ahome-made unit. Commercial choke may beused, as indicated in parts list of figure J 4.Special low inductance Eimac sockets are employedwith the tubes. Vacuum-type antennarelay is at top, center.
622 H.F. Power Amplifiers<strong>THE</strong> <strong>RADIO</strong>plate circuit of the amplifier. As a result, a veryminimum of shielding and lead filtering :srequired with this amplifier even when 28 Me.operation is desired. Placing the amplifierwithin the turtle-back of the automobile andthe insertion of a low pass TV-filter in thecoaxial antenna line reduces TVI-producingharmonics to such a low value that no interferenceis caused to a nearby television receivertuned to a channel 2 signal of moderatestrength.The amplifier is built upon a shallow metaldeck just large enough to hold the variouscomponents (figure 15). The two 4W-300Btube sockets are placed in the left area of thechassis with the variable vacuum capacitormounted at the right corner of the front paneLThe silver plated copper tubing tank coil isbolted between the rear terminal of the capacitorand one of the capacitor mounting bolts.Make sure that a good connection exists betweenthe capacitor, the panel, and the chassisas this ground return is part of the plate tankcircuit.Centered on the front panel are the threepilot lamp indicator sockets, the antennachangeover relay and the coaxial antenna receptacles.At the rear of the chassis is the plater-f choke. Two r-f chokes in series were usedin the original model, but have since been replacedwith one of the new Raypar miniaturechokes (see parts list, figure 14) .The grid circuit components are mountedin the under-chassis area of the amplifier (figure16). A shield plate is placed over thebottom of the amplifier to completely isolatethe components from the field of the platetank circuit. Grid coil L is mounted to thepanel and slug-tuned to resonance. The gridterminals of the tube sockets are connectedwith a short length of w-inch wide copperstrap. A wire interconnecting lead will Jnvanblylead to VHF parasitic oscillation of thestage.AmplifierOperationThe amplifier must be completelytested before it is installed in theautomobile. Grid, plate, and screenmeters should be inserted in the power leadsto the amplifier and preliminary tune-up intoa dummy load is done at reduct>d screen andplate potentials. A small driving signal is appliedto the amplifier and grid coil L resonated.The screen and plate currents will increasesharply in value when the grid circuitis tuned to the operating frequency. The degre:!of plate loading is determinoo by the positionof the antenna tap on the plate tank coil.Screen current is a very sensitive indicat::>rof the degree of amplifier antenna loading. Atplate potentials of 2000 or so the total screencurrent at one kilowatt peak input is about 10milliamperes, varying slightly with individualtubes. As the plate potential is raised abovethis value the screen current gradually dropsuntil at 3000 volts plate potential the screencurrent is near zero. In general, overcouplingof the amplifier will be indicated by lowscreen current and undercoupling will resultin high values of screen current. At 3000 voltsplate potential, an overcoupled condition isindicated by negative screen current, 'lnd undercouplingwill produce positive screen current.Only a slight deviation from zero currentduring voice operation indicates proper operarion.Maximum peak input under these conditionsis about 1500 watts ( 3000 volts at 500rna.).The Power Supply This amplifier is designedto work from a threephasealternator system of 1500 watts capacity.A three-phase rectifier supply is employed toFigure 16UNDER-CHASSIS VIEW OF MOBILESSB LINEAR AMPLIFIER SHOWSSIMPLICITY OF CIRCUITInput circuit is at the left, with grid terminalsof socket connected with short copper strap.Screen series resistors are at center, with gridchoke and power plug at right. Under-chassisarea is enclosed with aluminum bottom plateheld in place with self-tapping sheet metalscrews.
<strong>HANDBOOK</strong>Mobile SSB Amplifier623provide 2500 volts at 400 milliamperes (figure18). Since the frequency of the alternatoris 120 cycles or higher and the ripple outputof the supply is less than 5% before filtering,a small capacitor is sufficient to produce pured.c. and to provide good regulation for peakpower surges.In the interest of power conservation andcompactness the new Sarkes-Tarzian siliconrectifiers are used in the high and low voltagepower supplies. These rectifiers can be operatedin series with no special selection. It ispossible therefore to provide compact and inexpensivehigh voltage rectifiers by connectinga number of the standard type M-500 rectifiersin series to obtain the required peak inversevoltage rating. The higher cost of thebank of silicon rectifiers as compared to vacuumtube rectifiers is offset by the eliminationof the tubes and a special three-phase filamenttransformer. If desired, the specialSarkes-T arzian 280-SM rectifier stack may besubstituted for the individual units. Since theto----- ·r·DIA. HIGH-VACUUM HOSES6V. D.C.0.2.A.(TURNS ON WITHK-531 FILS)Figure 17WATER CIRCULATION SYSTEMFOR X-531 (4W-300Bl TUBESalternator will not supply much more than1500 watts, the primary voltage drops rapidlyT1 (SEC. :11. 1)50;owt----.w--..--~-{.; + 32.5 V. (SCREEN)AT tOO MA10.UF +450100 Ksw-T• (SEC. #Z)Figure 18SCHEMATIC, THREE-PHASE MOBILE POWER SUPPLYDI-D;-500 mo. silicon rectifier. Sorkes-Torzion M-500D,-Diz-2.5 KV, 130 mo. rectifier stock. Sorkes-Torzion 280-SM silicon rectifier. See text forsubstituteT ~-Special three-phase power transformer to deliver 2500 volts d.c. at 400 mo., and 325volts d.c. at I 00 mo. Design data and core may be obtained from Arnold Engineering Co.,Marengo, Ill. Core number: ATA-1573 with 12-mil Hypersil laminations for 120 c11cleoperation.Primary winding: (12 volts)-Each leg, 15 turns #Be. wire( 6 volts)-Each leg, 7 turns # Se. wireLow Voltage secondary, Each leg, 265 turns #30e. wireHigh Voltage secondary: Each leg, 2400 turns #28e. wire7.5 KV insulation employed. High voltage winding placed next to core, low voltagesecondary next, and primary winding outside, with 2KV insulation. Vacuum impregnatewindings. When finished, polarize windings by placing 6 volt, 50 c.p. lamps in serieswith each primary lead. Bulbs should only glow red when primary windings are correctlypolarized. Correct polarization of secondor11 windiNis will deliver desired ouU>ut voltaaes.
Figure 19WATER COOLED LINEARAMPLIFIER PACKAGED FORFIXED STATION USEEithet 4W-3008 (watet cooled) tubes ot 4X-250B (ait cooled) tubes may be employed inthis compact lineat amplifiet. Input and outputcapacitots of pi-netwotk ate vacuumunits, and powet supply (thtee phase) is containedwithin the cabinet (tight). Wide bandplate t.f. choke is at left of cabinet, tunningftom ftont to back. Amplifiet opetates atplate potential of 3000 volts, with p.e.p. of1500 watts.under overload conditions and the silicon rectifierscannot be damaged, even with a directshort circuit across the output terminals of thehigh voltage power supply.This unusual equipment is described forothers who may wish to make models basedupon ideas presented in the design. Althoughsome of the components shown in the amplifieror power package are unique or are notreadily available, it is hoped that the readercan make use of his facilities and abilities touse many of the ideas shown herewith.Variations Uponthe Basic CircuitIt is possible to increasethe power level of thisamplifier by merely addingmore tubes connected in parallel with theoriginal two. Three tubes will permit a p.e.p.of two kilowatts at 3000 volts plate potential.• •Figure 20MULTI-BAND LINEAR AMPLIFIERFOR MOBILE SERVICE FEATURESTHREE SEPARATE TANK CIRCUITSSWITCHED BY CONTROL RELAYSThree vacuum tonk capacitors are located attop left cotnet of panel. To the tight is thetmocoupler.l. ammeter in output circuit. Zerocentetscteen metet and 0-500 d.c. ma. platemetet ate at lower tight.Four tubes in parallel will provide a p.e.p. oftwo kilowatts at 2000 volts plate potential. Anexperimental amplifier having five tubes inparallel has run 3500 watts p.e.p. into dummyloads. The five tube amplifier employed neutralizationand parasitic suppress.ion and performedin normal fashion with no signs of instability.Shown in figure 19 is a fixed station versionof this amplifier. Employing a pi-section platetank circuit and a three-phase power supply,the complete unit is built within a standard12" x 16" x 19" cabinet. Of unusual interestis the r-f choke employed in the plate circuitof the linear amplifier stage. This choke isusually a critical item as the full value of r-fplate potential is developed across it. For maximumamplifier efficiency and minimum lossthe plate choke must present an impedance ofseveral thousand ohms at any frequency ofoperation. Special r-f chokes are available thatwill perform this exacting task, and the onedescribed equals the best of the commercialitems at a fraction of the cost. The choke coilconsists of 2 50 turns of # 22 double cottoncovered manganin wire close-wound upon aY:2-inch diameter wooden dowel rod. The smalld.c. resistance of the winding effectively destroysundes,ired resonance effects within thechoke, providing a high and uniform impedanceacross all amateur bands from 80- to 10-meters. The choke will work well up to powerlevels of the order of 2000 watts, p.e.p., or1000 watts, plate modulated.29-6 A Multi-bandMobile Linear AmplifierThe kilowatt linear amplifier described inthis section is designed for remote control mobileoperation on any three amateur bands.The unit shown in the photographs operateson the 80, 40, and 20 meter bands, but operationon 15 or 10 meters is possible by alteringthe constants of the plate tank circuit.
625INPUT 01j, cLDFC1~~ ~•IH300 ~BW cJ3RECEIVERNOTE *=SCREEN BYPASS IN TUBE SOCKETFigure 21SCHEMATIC, MULTI-BAND MOBILE LINEAR AMPLIFIERc,, c,, Cs-Jennings variable vacuum capacitor, type GSLA. Use 250 JlJlld. unit lor 80 meters and 120J.LJ.lld. units for other bands. Adjust coils to resonate circuit with capacitors set near maximum value.L,-16 turns, 7;4-inch silver plated copper tubing, 2Y2" i.d., 3Y2" long. Antenna tap approximately 2turns from ground end.Lz-8 turns, same as L1o Antenna tap approximately 2 turns from ground end.L,-5y2 turns, same as L, Antenna tap approximately % turn from ground end.RFC,-2.5 mh. National R-100RFCz-Wide band choke, 500 ma. rating. Raypar RL-102RY,, ,, s-DPDT relay, d.c. coil to match battery voltage. Jennings type RB vacuum relayRY,-SPST relay, d.c. coil to match battery voltage, Jennings type RB-I vacuum relay (Jennings RadioCo., San Jose, Calif.)M,-0- 500 ma. d.c. meterM,-500- 0- 500 d.c. micro-ammeter, zero center movement, shunted to 50- 0- 50 ma.Ms-0- 5 ampere, r.l. ammeterAmplifier The amplifier employs two ex-Circuit ternal anode tetrodes such as the4W-300B, 4X-250B, or 4CX-300A. The water cooled tubes are recommendedfor mobile operation, while the air cooledtubes may be used for fixed operation. Theschematic of the amplifier is shown in figure21 and is a version of the untuned input circuitof figure 11 C.Three separate parallel tuned plate circuitsare employed in the linear amplifier. Eachcircuit is made up of a silver plated coppertubing coil paralleled with a miniature variablevacuum tuning capacitor. These new Jenningscapacitors are extremely small in sizeand are tuned by means of a slotted sh:1fr insteadof the more cumbersome and expensivecounter dial arrangement. The low impedancemobile antenna system is tapped directly tOexperimentally determined points on the tJ.nkcoils. The proper tuned circuit is switched tothe linear amplifier by means of miniaturevacuum-sealed d.p.d.t. relays. Three relays arerequired, one for each tank circuit. The tankcircuits are relatively low impedance devicesas they are tuned with capacitors having a largevalue of maximum capacitance, permittinggood efficiency in matching the tubes to thevery low load impedance presented by t~emobile antenna system.Although the tubes operate with "zero"driving power the resistive input system demandsthat approximately 9 watts of excitationbe employed to develop 50 volts (peak)across the 300 ohm circuit impedance. Themobile SSB exciters ot Chapter 28 used inconjunction with a 2E26 linear buffer stagewill supply an abundance of excitation forthis kilowatt amplifier. Alternatively, a highimpedance grid ci~cuit such as shown in figure
626 H.F. Power Amplifiers<strong>THE</strong> <strong>RADIO</strong>llB may be used, dropping the grid drivingpower level to a watt or so.Selection of the proper tank circuit is accomplishedby means of a rotary switch ( S,)located at the driver's position in the automobile.One of the three miniature plate circuitreLays at a r:me may be operated by this switch.The antenna change-over relay is actuated bythe push-to-talk circuit in the microphone.AmplifierConstructionThe amplifier is built upona shallow metal deck (figure22). The two Eimac low inductancesockets for the tetrode tubes areplaced in the rear corner of the chassis withthe 80 meter tank coil to their left. One endof the coil is bolted to the chas,sis and theother end is attached to the rear terminal ofthe variable vacuum capacitor which is mountedto the metal panel. The 40 and 20 metertank coils are mounted directly to their respectivetuning capacitors.The front area of the chas,sis is taken upwith the plate circuit switching relays RY,-,-a.These relays are bolted to the chass,is andconnections are made to the various terminalsby means of flexible 1,4-inch silver plated copperstrap. The three meters are mounted in theremaining panel space. Under-chassis wiringis extremely simple, as seen in figure 23. Thesocket grid terminals are connected togetherwith a short length of copper strap. A shieldis not required over the bottom of the chas.sissince the grid circuit is untuned.AmplifierOperationThe amplifier should be benchtestedbefore it is placed withinthe automobile. Operating conditionsare similar to those described in Section29-5. A center-reading milliammeter is usedin the screen circuit to observe screen currentduring loading adjustments. Linear operat1ionmay be checked with the aid of envelope detectors,as discussed in an earlier chapter. Normaloperation takes place at 2500 volts at 400milliamperes, peak current. Screen current isless than 5 milliamperes under these conditions.Grid current is zero.29-7 An InexpensiveCathode DrivenKilowatt AmplifierAn objectionable feature of triode tubes isthat they must be neutralized in conventionalgrid driven circuits. Terrade tubes may oftendispense with the neutralizing circuit, but theyrequire a screen power supply whose cost andcomplexity dissipate the saving afforded bythe elimination of the neutralizing circuit. Theuse of a cathode driven amplifier employingzero bias tubes overcomes these two disadvantages.The grids act as an excellent screen betweenthe plate and cathode so neutralizationis not usually required. The very small plateto cathode capacitance permits a minimum ofintercoupling on frequencies below 30 Me.when inexpensive tetrode or pentode tubes areused. No screen or bias supplies are required.Figure 22REAR VIEW OFMULTI-BANDMOBILE LINEARAMPLIFIERThe three amplifier tank
<strong>HANDBOOK</strong>Multi-Band Amplifier 627FeedthroughPowerA large portion of the excitingpower appears in the platecircuit of the cathode drivenstage and is termed the feedthrough power.In any amplifier of this type, whether it betriode or tetrode, it is desirable to have a fairlylarge ratio of feedthrough power to peak griddriving power. The feedthrough power acts asa swamping resistor across the driving circuitto reduce the effect of grid loading. The ratioof feedthrough power to driving power shouldbe at least 10 to 1 for best stage linearity.Class AB 1 CathodeDriven AmplifiersThe only way that theadvantages can be takenof the inherent feedbackin a cathode driven amplifier is to driveit with a source having a high internal impedanceand to operate the stage class AB 1wirhout grid or screen current. It is a characteristicof class AB 1 cathode driven (socalled"grounded grid") amplifiers that a variationin load impedance on the stage reflectsback as a nearly proportional change at theinput became the input and output circuitsare essentiaHy in series. In the same manner, achange in gain (such as may be caused by anonlinearity of gain with respect to signallevel) of a cathode driven stage reflects backas a change in the input impedance of thatstage. If the cathode driven stage is fed by asource having high internal impedance suchas a tetrode amplifier, changes in gain or nonlinearityin the cathode driven stage will becompensated for. For example, if the gain ofthe cathode driven stage drops slightly, the inputimpedance of the stage will increase andthe load impedance on the driver stage willrise. If the driver has high internal impedancethe output voltage will also rise when its loadimpedance rises. The increased output voltagewill raise the output voltage of the cathodedriven stage so thM the output is nearly up towhere it should be even though the gain ofthe cathode driven stage is low. The equivalentcircuit for this action is shown in figure 24."True" class AB1 cathode driven amplifiers,however, have rather low stage gain and theuse of low-mu triodes is required for reasonableefficiency. Such low gain circuits requiremore stages to reach a given power level. Highdriving voltage is required which is difficultto achieve in the cathode circuit. Since i'l s•~emsimpractical to opera·te most available tubesclass AB 1 in a cathode driven circuit (i.e., nogrid current and no screen current) the possibilityof taking advantage of the inherent feedbackin the circuit is lost because a low driverFigure 23UNDER-CHASSIS VIEW OF MULTI-BAND LINEAR AMPLIFIERCoaxial input receptacle is at left, with composition-typegrid resistors. Simplicity of circuitrequires relatively few components beneaththe chassis. Grid terminals of socketsare connected with wide copper strap.!;Duree impedance is then required to supplygrid driving power for class AB2 operation.The latter class of operation offers stage gainfigures of 10 to 25 with moderate values ofcathode excitation voltage. The linearity, however,is approximately the same as for conventionalgrid driven circuits having lowsource impedance, and about 30 decibels signal-to-distortionratio is the maximum thatcan be achieved.®LOADRFigure 24EQUIVALENT CIRCUIT OF CASCADECLASS ABl CATHODE DRIVENAMPLIFIERSA-Cathode driven amplifiers may be arrangedin cascade to provide high outputlevels. Any variation in load impedance(R) reflects back as nearly proportionalchange at the input, as input and outputcircuits are essentially in series as shownin figure B.a-Equivalent circuit of figure A. R, has avery high resistance, and R2 and R3 havevery low resistances compared to theload R.
628 H.F. Power Amplifierswith high frequency tubes such as the 813and 7094, the use of filament chokes for operationabove 7 Me. is recommended.This type of amplifier has a tendency towardparasitic oscillations in the 100 Me. regionat which point the inherent inductance ofthe ground paths begins to affect operation.Variations in grounding technique can eliminatethe parasitic, but the addition of parasiticsuppressors in the plate leads of the tubesmakes the circuit relatively insens,itive to variationsin the ground return path.Figure 25CATHODE DRIVEN KILOWATTAMPLIFIER FOR 40 AND 80 METERSCathode driven amplifier presents ultimate insimplicity and stability. Designed around two803 tetrodes, this kilowatt amplifier for 40and 80 meter operation has only plate tanktuning control and plate cu11ent meter onpanel. Amplifier is completely enclosed forreduction o# spurious radiation.A Practical Cathode Shown in figures 25,Driven Amplifier 27, and 28 is a cathodedriven amplifier designedfor 40 and 80 meter operation. It iscapable of 1000 watts power input and maybe employed for SSB, c.w., or a.m. phone.Combined grid driving and feedthrough poweris approximately 110 watts for 1 kilowatt input.Actual grid driving power is of the orderof 9 to 10 watts, so the ratio of feedthrough todrive power is about 10 to 1.The schematic of the amplifier is shown infigure 26. Two 803 tetrode tubes are used inparallel in cathode driven configuration. Thesetubes are available on the surplus market for adollar or so and perform very well up to approximately10 Me. All three grids of thetubes are returned to ground and the tubes operatein near zero bias condition. The internalscreening of the tubes begins to deterioratein the 15 Me. region and amplifier instabilitymay be encountered above 10 Me. unless precautionsare taken to ensure that the grids remainclose to r-f ground potential.When operating in the 3.5 - 7 Me. regionit is entirely possible to apply the r-f drivingvoltage directly to the filament circuit withoutthe necessity of adding filament r-f chokes toisolate the driving voltage from the filamenttransformer. This makes for circuit simplicityand lower initial cost. If this circuit is usedAmplifierConstructionThe amplifier is built upon a10" x 19" x 4" aluminumchassis. The plate tuning capacitorC is centered on the chassis with thetwo 803 tubes mounted to one side. The tubesockets are recessed below the chassis to bringthe top of the tube base level wi
Figure 27REAR VIEW, CATHODE DRIVEN803 AMPLIFIERPlate tuning capacitor is at the center, withtank coil to left. The 803 tubes (right) aremounted with sockets below deck, so thattops of bases are level with chassis. Plater.f. choke is placed between tubes, with parasiticchokes (PC) supported from top terminalof choke.supporting the sockets. The complete amplifieris enclosed in a shield made of aluminum.Holes are drilled above the 803 tubes in thetop sheet (figure 25) and in the rear plate topermit circulation of air about the tubes.AmplifierOperationThe resting plate current of theamplifier under quiescent condi-'tion is between 30 rna. and 75rna. depending upon the plate voltage, climbingto 450 rna. or so under full excitation at aplate potential of 2500 volts. Care should betaken in tune-up and full excitation shouldnever be applied unless the amplifier is operatingat rated plate voltage with maximumantenna loading. Applying full drive to thetubes with no plate voltage will surely causedamage to the grid structure of the tubes.The meter indicates plate and screen current;actual plate current is about 40 milliampereslower than the meter reading. At 1000watts p.e.p., the power output measures approximately600 watts. Tuning of the platecircuit is facilitated by means of a thermocoupler-f meter placed in the coaxial lead tothe antenna coaxial receptacle.Figure 28UNDER-CHASSIS VIEW OF CATHODEDRIVEN AMPLIFIERInput coaxial connector is at the right onrear of chassis. Mica filament bypass capacitorsare placed between the tube sockets,with RFC, beneath them.29-8 A Low DistortionSideband Linear AmplifierA properly designed grounded grid linearamplifier is a very satisfactory high level stagefor a sideband transmitter, even though thepo~sibility of taking advantage of the inherentfeedback is lost because of the reasons explainedin Section 29-7 of this chapter. With care,a signal-to-distortion ratio of 30 decibels caneasily be achieved and the circuit is unsurpassedfor stability and ease of operation.Data given in the tube manuals and instructionsheets are usually derived for maximumtube output rather than the condition of minimumdistortion. It is therefore necessary toadjust circuit operating voltages to arrive at asatisfactory power output level consistent witha pre-set value of maximum allowable distortion.The maximum distortion level for thisamplifier was set at 30 db S/D. At this figure350 watts output p.e.p. can be obtained at 3000volts plate potential, and 680 watts outputp.e.p. can be obtained at 3500 volts.The GroundedScreenConfigurationFor maximum shielding withinthe tube, it is necessary tooperate both grid and screenat r-f ground potential. At thesame time, the control grid has a negative operatingbias applied to it, and the screen has apositive operating potential. Both elements ofthe tube, therefore, must be bypassed togl'ound over a frequency range that includesthe operating spectrum and the region of possibleVHF parasitic oscillations. This is quitea large order. The inherent inductance of theusual byp1ss capacitors is sufficient to introducesufficient regeneration into the circuit todegrade the linearity of the amplifier at highsignal levels even though the instability is notgreat enough to cause parasitic oscillation. The
630 H.F. Power Amplifiers<strong>THE</strong> <strong>RADIO</strong>Figure 29GROUNDED SCREEN GRIDCONFIGURATION PROVIDES HIGHORDER OF ISOLATION IN TETRODEAMPLIFIER STAGEA-Typical amplifier circuit has cathode returnat ground potential. All circuits returnto cathode.B-All circuits return to cathode, but groundpoint has been shifted to screen terminalof tube. Operation of the circuit remainsthe same, as potential differences betweenelements of the tube are the same as incircuit A.C-Practical grounded screen circuit. ~~commonminus" lead returns to negative ofplate supply, which cannot be grounded.Switch S; removes screen voltage for tuneuppurposes.solution to this problem is to eliminate thescreen bypass capacitors by actuaUy groundingthe screen terminals of the amplifier to thechassis by means of a low inductance strap. Itis possible to do this by juggling the groundreturn of the amplifier, as shown in figure 29.Circuit A shows the conventional d.c. returnpaths wherein all power supplies are returnedto the grounded cathode of the stage. Metersare placed in the common return leads, andeach meter reads only the current flowing inthe singular circuit. The d.c. ground lead isremoved from the cathode in circuit B andplaced at the screen terminal of the tube. Circuitoperation is still the same, as all powersupply returns are still made to the cathode ofthe tube. The only change is that the screenr.f. ground and d.c. ground are now one andthe same. The cathode circuit is now at anegative potential with respect to the chassisground by an amount equal to the screenvoltage. In addition, the return of the highvoltage plate supply is now negative with respectto ground by an amount equal to thescreen voltage supply.A practical version of this circuit is shownin figure 29C. The grid bias and screen suppliesare incorporated in the amplifier andseparate terminals are provided for the positiveand negative leads of the high voltagesupply. A "tune-opera:te" switch S, is added tothe circuit which removes the screen potentialfor tune-up purposes. The negative of thehigh voltage supply "floats below ground" atthe value of the screen voltage.This circuit is unique in that it requires amental orientation of the user to accept the factthat d.c. ground is no longer "ground." However,the circuit is no more complex than theusual grounded cathode d.c. return circuit, andthe advantage of having the screen at r.f.ground potential far outweighs the unusualaspects of the circuit. Operation of the circuitis normal in all respects, and it may be appliedto any form of tetrode amplifier with goodresults.The AmplifierCircuitThe circuit of an operationalcathode driven, groundedscreen amplifier is shown infigure 30. A 4-250A or 4-400A tetrode tubeis used with an untuned cathode circuit and api-network plate circuit. Bias and screen suppliesare included in the amplifier. The importantoperating parameters are also shownin the diagram. P.e.p. excitation is 12 watts(including losses) for 3000 volt operationand about 22 watts (including losses) for3500 volt operation.The resonant plate circuit is capaC'itivelycoupled to the amplifier tube and consists ofa 300 ,.,,.,fd. variable vacuum capacitor, a kilowatttapped-coil turret, and a 1500 ,.,,.,fd. variableair-type loading capacitor. D.c. groundis established by returning the output sectionof the network to ground by means of chokeRFC-3. Three .001 ,.,fd., 5 KV ceramic capacitorsin parallel form the plate blocking capacitor.The high voltage is applied to the tubevia a Raypar heavy duty r.f. choke in serieswith two air wound VHF choke coils. Escape
<strong>HANDBOOK</strong> SSB Linear Amplifier 631INPUTJtALCOUTT~NOTES:I. ALL RESISTORS 1 WATT UNLESSO<strong>THE</strong>RWISE SPECIFIEDCLASS ABl OPERATING CONDITIONS,4- 250A /4-400AD.C. PLATE VOLTAGE 3000 3500D.C. SCREEN VOLTAGE 300 300STATIC PLATE CURRENT (MA.) 60 60D.C. GRID BIAS VOLTAGE -60 -60MAX. SIG. GRID CURRENT (MA.) 3.6 10MAX. SlG. SCREEN CURRENT (MA.) 4.5 2.0MAX. SIG. PLATE CURRENT (MA.) 195 265MAX. SIG. PLATE DlSSIP. (WATTS) 235 235STATIC PLATE DISSIPATION(WATTS) 180 2.10GRID DRIVING POWER (WATTS) 0.63 3.4FEED THRU POWER (WATTS) 6.6 15.8POWER OUTPUT (WATTS) 350 690EFFICIENCY (o/o) 60 70RELAYCONTROL\tSV.'\.1Figure 30SCHEMATIC, LOW DISTORTION LINEAR AMPLIFIERc,-300 ;1Jlld., JOKY variable vacuum capacitor. Jennings UCS-300C:-1500 !l!lld., variable capacitor. Cardwell 8013C,-JOO 1111ld. variable ceramicCJ,-C,-0.1 11ld., 600 volt bulkhead-type capacitor. Sprague "Hypass"L,-5,-Barker & Williamson #850 pi-network inductor. Inductances: 80 meters, 13.511h.; 40 meters, 6.5 11h.; 20 meters, 1.75 11h.; 15 meters, I 11h.; 10 meters,0.8 /lh.Lz-Dual winding filament choke. B&W FC-15, 15 amperes capacityT,-5 volts at 10.5 amperes. Stancor P-6135T:-335- 0-335 volts, 75 ma. Chicago PSC-70CH,-8 henry, 85 ma. Stancor C-1709D,-D,-500 ma. silicon rectifier. Sarkes-Tarzian M-500D.:-Diode, type IN67 or equivalentM,-0- 800 ma. d.c. meter. Marion type MM3M,-0- 50 ma. d.c. meter. Marion type MM3M,-0- 50 ma. d.c. meter. Marion type MM3RFC,-Heavy duty wide band r.l. choke, 800 mo. Raypar RL-100RFC2-20 turns #20e, airwound, !;' 2 " diam, J" longRFC,-2.5 mh. National R-100
Figure 31KtLOWA TT CATHODE-DRIVENAMPLIFIERCareful arrangement of components permitssymmetrical panel layout. Grid, screen, andplate current meters are spaced across topof panel, with amplifier loading control (C,)at left, and plate bandswitch (SI) at right.Counter dial at center drives variable vacuumcapacitor C1. Tune-operate switch Sz is atlower right, balanced on left by pilot light.Amplifier is completely screened to reducespurious energy radiation. Screened area ontop of shield allows circulation of air aroundtube.of r.f. energy through this lead 1s thus heldto an absolute minimum value.Driving power is coupled to the filamentcircuit of the tube through two .001 fLfd. ceramiccapacitors. The filament transformer isisolated from the exciting voltage by means ofa bifilar wound r.f. choke coil, capable of carryingthe filament current of the tube ( 10.5amperes).A dual purpose power supply delivering375 volts serves as screen and bias source. Theoutput voltage of the supply is held constantby three regulator tubes in series. The "commonminus" lead (see figures 29 and 30) isat the juncture of the regulated -7 5 volts andthe regulated 300 volts. The supply thus delivers-75 volts for the bias circuit and plus300 volts for the screen circuit. The exact value•of bias voltage may be set by adjusting potentiometerR,.A simple automatic load control circuit isincorporated in the input circuit of the amplifier.A diode samples the incoming signal andproduces a d.c. signal proportional to the inputsignal. This control voltage may be applied toan amplifier tube in the sideband exciter insuch a way as w limit the incoming signal toa predetermined value which may be set byadjustment of capacitor C and potentiometerRz. This circuit is analogous to the automaticvolume control circuit in a receiver. The timeconstant of the a.l.c. circuit is determined bythe 0.1 /Lfd. capacitor and 3 3K resistor acrossthe output jack, }a.Amplifier The amplifier is built upon anConstruction aluminum chassis measuring13" x 17" x 4". The frontpanel is 13%" high, and the above-chassisFigure 32TOP VIEW OFKILOWATTAMPLIFIER USING4-400A TETRODEPlate circuit inductor isplaced parallel to paneland driven by right-angledrive unit. Loading ca·pacitor is at right, withvariable vacuum capacitormounted vertically atrear of chassis. 4-400Atube and air socket areat rear right corner ofchassis, with bias transformerin lelt corner.Plate blocking capacitorsare placed betweenaluminum plates andmounted atop vacuumcapacitor. Plate r-f chokeis hidden between tubeand variable vacuum capacitor.At base of capacitorare the 8-plusVHF filter chokes andbypass capacitors.Meters are isolated byaluminum shield runningfull length of chassis.Meter leads pass downconduit (left corner I tounder-chassis area. Shieldplate covers entire enclosure.
Figure 33UNDER-CIIASSISVIEW OFCATHODE-DRIVENAMPLIFIERBias and screen supply,and input circuits areplaced in the under•chassis area. Filamenttransformer and filamentchoke are at left, withtube socket next to thechoke. Screen terminalsof socket are groundedwith wide aluminumstrap passing across ter ..minals to socket shelland chassis. Grid terminalis bypassed to centerof strap. Input capacitorsmount between filamentterminals and co•axial input receptacle onrear of chassis.At right are power supplycompartment (seefigure 34) and voltageregulator tubes. Vacuumcapacitor gear drive andoutput circuit voltmeterare at center of chassis.Blower motor is next tofilament transformer.Under-chassis area ispressurized so that airescapes through tubesocket vent. After bottomplate is in place,seam around plate issealed with cellophanetape to provide air-tightenclosure.shield is 9 Ys" high. This shield is made of onepiece of aluminum, bent into a U-shape, havingY2 -inch flanges on all sides. The openside of the "U" is bolted to the front panel asshown in the photographs. Elastic stop-nutsare affixed to the shield every 1 1;2" along theflange, or it may be drilled and tapped for6-32 machine screws. A shield of this typecan be homemade, or it may be fabricated inany large sheet metal shop for a small sum.The three indicating meters are mountedon the top front of the panel and are shieldedfrom the intense r.f. field of the amplifier bya half-box section of shielding bent to fitaround the rear of the meter cases. The metershield measures 2 1;2" deep, 4JA," high, and islong enough to make a close fit with the outershield at each end. Meter leads pass from theunder-chassis area through the upper deck viaa short secnion of \12-inch pipe, threaded ateach end and bolted between the chas\\LS deckand the meter shield.The major components are mounted withan eye towards panel symmetry and short leads.The vacuum tube socket and variable vacuumcapacitor C are placed to the rear of the chassis.The capacitor is mounted in a vertical position,gear driven from the under-chassis area(figure 3 3). Plate choke RFC-1 is placed betweenthe air socket and the capacitor. Direct-SSB Linear Amplifier 633ly in front of the tube is the panel driven 1500fLfLfd. loading capacitor C, mounted on twoheavy aluminum brackets, bringing its shaftin line with that of the variable inductor, L.Plate inductor L, is affixed to the chassisparallel to the front panel, and is driventhrough a miniature right angle drive, as seenin figure 32, making the control knobs of L,and C occupy symmetrical positions on thepanel.The variable vacuum capacitor is centeredbetween the sides of the chassis and is paneldriven by a counter dial and a right angledrive unit. To preserve the panel appearance,the counter dial is mounted with its shaftabout 2 Ys" above the lower edge of the panel.The center-line of the right angle drive unit,however, falls about 1 Ys" above the paneledge. Two 114" brass gears are used to couplethe shafts, as seen in figure 34. The gears areheld in position by an angle plate cut from asoft aluminum sheet.The Tube Socket An Eimac air cooled socketis used for the amplifiertube. The small 115 volt a.c. blower motor ismounted near the socket, and the entire underchassisarea is sealed air-tight with a bottomplate. A ');,\" wide aluminum strap passesacross the center of the socket, extending down
634 H.F. Power Amplifiers<strong>THE</strong> <strong>RADIO</strong>Figure 34CLOSE-UP OF POWER SUPPLYCOMPARTMENT AND GEARDRIVE SYSTEMOff-center shafts of capacitor and counterdial are ganged by two gears held in positionby triangular plate bolted to chassis. Enclosurebelow holds power supply components.Silicon rectifiers are mounted in fuse clipsattached to enclosure wall. All power leadsleaving enclosure pass through small, bulkhead-typeceramic capacitors. Primary leadspass through11Hypass~' capacitors on rearwall of chassis. Potentiometers R, and Rz arein corner ol chassis.no the chassis on both sides. This strap is boltedto the two screen terminals of the socket,and to the two diagonal assembly screws of thesocket frame. The ends of the strip are fastenedto the chassis, forming a low impedancereturn path for the screen circuit. The .001fLfd., 5 KV ceramic grid bypass capacitor ismounted directly between the ground strap andthe grid terminal of the tube socket.The r.f. input jack J, is mounted to the rearwall of the chassis directly behind the tubesocket and two .001 fLfd. ceramic capacitorsare connected between the center terminal ofJ, and each filament terminal of the tube socket.Short leads connect the filament terminalsto the filament choke coil L, mounted on theside wall of the chassis, directly above the tubesocket. The power supply components are encasedwithin an auxiliary shield measuring5" x 7 Y:z" (figure 34). All leads leaving thisenclosure pass through .001 }Lfd. ceramic feedthroughcapacitors mounted on the wall of theenclosure. The four silicon rectifiers are mountedin a multiple fuse clip holder placed on theside wall, and the three regulator tubes aremounted on the adjacent wall.Amplifier Wiringand TuningThe power circuits of theamplifier may be wiredwith unshielded wire sincethe r.f. field beneath the chassis is extremelylow. All d.c. leads are cabled and laced as maybe seen in the under-chassis photograph. Leadsto the meter deck pass through the area abovethe chassis in a short section of Y:z-inch alum-inurn conduit, and each meter is bypassed asshown in the schematic. Small insulated tipjacks (Jr.- ]w) are mounted on the rear of thechassis to facilitate voLtage measurements. Inaddition, two insulated, closed-circuit jacks(],, ], ) allow the voltage regulator currents tobe read.Choke coils RFC-2 are wound on a %-inchwooden dowel rod which is removed, leavingthe coils self-supporting. Plate circuit wiringabove the deck is done with silver plated copperstrap.When the amplifier wiring is completed thepower supply should be checked. The regulatortubes are inserted in the correct sockets andthe current flowing through the regulator circuitis measured. The 1500 ohm series resistorin the power supply is adjusted to provide 40milliamperes of screen regulator current. If thecorrect current value is obtained with less •han500 ohms of series resistance, the 1 1-'fd. inputfilter capacitor should be increased in value to8 }Lfd. to boost the supply voltage. The regulatorcurrent divides between the VR-75 andthe bias adjust potentiometer R,, and 25 milliamperesof regulator current is measured at], when the supply is adjusted for 40 rna.at jack J•. After final adjustment has beencompleted, the variable series resistor in thesupply may be replaced with a fixed one ifdesired. Finally, bias control potentiometer R,is set for -60 volts as measured between Jo(minus) and ]a (plus).Switch s, should be placed in the "tune"
<strong>HANDBOOK</strong>Kilowatt Amplifier635Figure 35GENERAL PURPOSE AMPLIFIEROPERATES IN CLASS A, B, OR CMODEThis kilowatt amplifier employs a pair of4-250A's in a pi-network circuit. Mode of operationmay be set by selection of properscreen and bias voltages. Grid, plate, andscreen current meters are mounted on plasticplate behind panel cut-out, and tubes arevisible through shielded panel opening. Acrossbottom of panel (left to right) are bandswitch,grid tuning, plate tuning, loading, andprimary power control circuits. Plate tuningknob is attached to small counter dial.posmon before voltages are applied to thetube. A suitable antenna load should be placedon the amplifier; bias, screen, and plate voltageare applied. The bias control should be variedslightly to produce the correct resting platecurrer,; as indicated in figure 30. Excitationshould never be applied without plate voltageon the amplifier, or without an external antennaload, since the resultant high grid andscreen current would damage the tube. In addition,screen voltage should never be appliedwithout plate voltage. The combination screenand bias supply of the amplifier is designedto be turned on with the plate supply throughthe "relay control" terminal. All voltages maybe left on the amplifier during "VOX" operation,or the supplies may be turned on by the"VOX" circuit. The "tune-operate" swioch s,should not be thrown when excitation is appliedto the rube, since the momentary breakingof the screen to ground path will applythe full excitation power to the control grid ofthe tube during the moment the swioch is intransition from one position to the other. Inpractice, this switching action has been donewith full excitation applied to the tube, butit is not recommended.The exciter should now be connected anda small amount of carrier is inserted. The plateresonant point is found with tuning capacitorCt and the amplifier loading adjustments aremade, gradually increasing the excitation andloading until the single tone conditions outlinedin figure 30 are met. Speaking into themicrophone should produce approximately thesame peak conditions as obtained with carrierinsertion. The linearity of the amplifier may bechecked with the use of two envelope detectorsas described in an earlier chapter.29-9 KilowattAmplifier for Linear orClass C OperationA pair of 4-250A or 4-400A tetrode rubesmay be employed in a pi-coupled amplifiercapable of running one kilowatt input, c-w orplate modulated phone, or two kilowattsp.e.p. for sideband operation. Correct choice ofbias, screen, and exciting voltages will permitthe amplifier to function in either class A, B,or class C mode. The amplifier is designedto operate at plate potentials up to 4000 volts,and excitation requirements for class C operationare less than 2 5 watts.A bandswitching type of pi-network is employedin the plate circuit of such an amplifier,shown in figure 35. The pi-network isan effective means of obtaining an impedancematch between a source of r.f. energy and alow value of load impedance. A properly designedpi-network is capable of transformationratios greater than 10 to 1, and will provideapproximately 30 decibels or more attenuationto the second harmonic output of the amplifieras compared to the desired signal output. Sincerhe second harmonic level of the amplifiertube may already be down some 20 db, theactual second harmonic output of the networkwill be down perhaps 50 db from the fundamentalpower level of the transmitter. Attenuationof the third and higher order harmonicswill be even greater.
636 H.F. Power Amplifiers4-250A4-400A4-250A4-400A500100 20 KV2WwT1=NOTE: SCREEN BYPASS CAPACITORSARE CENTRALAB TYPE 858C1-100 Jl.Ji.fd. HammarlundHF-100c,-200 Jl.!lfd., JOKY variablevacuum capacitor.Jennings UCS-200c,-1500 Jl.ji.fd., variablecapacitor. CardwellB013C 4-Neutralizing capacitor,disc. Millen 15011C,-300 Jl.Jl.fd., mica,1250 voltL1-L10-See coil table,figure 39GND 1, 5 v 115 v. 8+2500-'U 'U 3500 vFigure 36SCHEMATIC, GENERAL PURPOSE KILOWATT AMPLIFIERPC-47 ohm, 2 wattcomposition resistorwound with 6 turns#!Be.RFCJ-2.5 mh. choke.National R-100RFC,-Heavy duty, widebandr.f. choke. Barker& Williamson typeBOORFC.,-VHF choke. OhmiteZ-144S1-Two pole, 6 positionswitch. Two Centra/abPA-17 decks, with PA-30 1 index assembly5 2-Two pole, 6 positionhigh voltage switch.Communication ProductsCo. type BB twogang switchS,-Four pole, three positionswitch. CentralabT1-5 volt, 20 ampere.Stancor P-6492M1-0-50 ma. d.c.TriplettM,-0- 150 ma. d.c.TriplettM,-0-750 ma. d.c.TriplettGears-2 required. BostonGear #G-465 and#G-466The peak voltages encountered across theinput capacitor of the pi-network are the sameas would be encountered across the plate tuningcapacitor of a single-ended tank used inthe same circuit configuration. The peak voltageto be expected across the output capacitorof the network will be less than the voltageacross the input capacitor by the squareroot of the ratio of impedance transformationof the network. Thus if the network is transformingfrom 5000 ohms to 50 ohms, the ra~io of impedance transformation is 100 andthe square root of the ratio is 10, so that thevoltage across the output capacitor is I/ I 0that across the input capacitor.A considerably greater value of maximumcapacitance is required of the output capacitorthan of the input capacitor of a pi-networkwhen transformation to a low impedance loadis desired. For 3.5 Me. operation, maximumvalues of output capacitance may run from500 !J.!J.fd. to I500 !J.f-Lfd., depending upon theratio of transformation. Design informationcovering pi-network circuits is given in an earlierchapter of this Handbook.Illustrated in figures 35- 4I is an up-todateversion of an all-band pi-network amplifier,suited for sideband or class-C operation.The unit is designed for TVI-free operationover this range.CircuitDescriptionThe schematic of the generalpurpose amplifier is shown ·infigure 36. The symmetricalpanel arrangement of the amplifier is shownin the front view (figure 3 5) and the rearview (figure 3 7). A 200 fLI-'fd. variable va·cuum capacitor is employed in the input sideof the pi-network, and a I500 ,.,.,.,.fd. variableair capacitor is used in the low impedance outputside. The coils of the network are switchedin and out of the circuit by a two pole, five
Figure 37REAR VIEW OF GENERAL PURPOSEAMPLIFIER WITH SHIELDREMOVEDThe pi-network circuit is built from an inexpensivehigh voltage rotary switch, and liveinductors. The switch is panel driven by agear and shalt system shown in figure 38.Variable vacuum capacitor is mounted ver·tically between the tubes, directly in back ofthe plate r.f. choke. Neutralizing capacitor isat right, connected to plates of tubes with awide, silver plated copper strop. Meters oreenclosed by aluminum shield partition runningthe width of the enclosure, with conduit corrymgmeter loads to under-chassis area atleft, front of chassis. Metal shells of tubebases ore grounded by spring contacts.position high voltage ceramic rotary switch.Each coil is adjusted for op~imum circuit Q,resulting in no tank circuit wmpromise in efficiencyat the higher frequencies. A close-upof the tank circuit is shown in the introductoryphotograph at the chapter head, and completecoil data is given in figure 39. The plateblocking capacitor is made of two .001 11fd.,5 KV ceramic capacitors connected in series.Special precautions are taken to insure operatingstability over the complete range ofamplifier operation. The screen terminals ofeach tube socket are jumpered together withVs" copper strap and a parasitic choke (PC)is inserted between the center of the strap andthe screen bypass capacitor. In addition, sup-pressor resistors are placed in the screen leadsafter the bypass capacitor to isolate the sensitivescreen circuit from the external powerleads. A third parasitic choke is placed betweenthe grid terminals of the tubes and the tunedgrid circuit.The five coils of the grid circuit are enclosedin a small aluminum shield placed adjacentto the tube sockets (figure 38 and figure40). The amplifier is neutralized by a capacitivebridge system consisting of neutralizingFigure 38PLACEMENT OFPARTS IN UNDER-CHASSIS AREAGrid tuned circuit is enclosedin separate enclosureat left. Bandswitchprojects out the rear ofcase 1and is gear drivenby same shalt that actuatesthe plate bandswitch.Switches are driventhrough right-anglegear drives and gears.Output capacitor of pinetworkis shielded fromrest of under-chassiscomponents.The screen terminals ofeach tube socket arestrapped together with%" copper ribbon, andlow inductance screenbypass capacitor isgrounded t o socketmounting bolt. Screenparasitic choke mountsbetween strap and capacitorterminal. Allpower leads beneath thechassis ore run in shieldedbraid, grounded tochassis at convenientpoints. B-plus lead ismade of section of RG-8/U coaxial c
638 H.F. Power Amplifiers<strong>THE</strong> <strong>RADIO</strong>FIGURE 39COIL TABLE FOR KILOWATT AMPLIFIERf----------------------------1GRID COILSl 1- (80 METERS); 46 TURNS #24 £ 1 3/4" 0/A ., 1" LONG ONAMPH£NOL POLY'STYRENE FORM.l2-(40 METERS): 30 TURNS #24 £, 3/4" 0/A., 3/4" LONGON AMPHENOL POLYSTYRENE FORM.l3-{20 METERS): 12 TURNS, B~W 3011 MINIOUCTOR,3/4" DIA., 3/4" LONG.l4-(15 METERS): ?TURNS, B&WJOTO MINIDUCTOR,3/4" OrA., 7/8" LONG.lS-(10 METERS): 5 TURNS, AS ABOVE.ALL COl LS HAVE 3 TURN LINI
<strong>HANDBOOK</strong>Kilowatt Amplifier 6394-250A/4-400A OPERATING CHARACTERISTICS(2 TUBES)ITEMMODEP_:!'_~-E V0_0~~-_--~ 3oo0 T -=-~00 __Li_Soi-~ ~?O_?_P~~T:_c~R~~ ~~) !--- 110-42_~3~-440_J~-~? __ j_----=-3?_~R~E~o__:_~~G~- 1 _ 6oo _f-__soo _Lsoo 1 saos-inch silver plated copperstrap as seen in the introductory photographat the beginning of this chapter.AmplifierNeutralizationAfter the amplifier is wiredand checked, it should beneutralized. This operationcan be accomplished with no power leads attachedto the unit. The tubes are placed intheir sockets, and about 10 watts of 30 Me.r.f. energy is fed into the plate circuit of theamplifier, via the coaxial output plug }. Theplate and grid circuits are resonated to thefrequency of the exciting voltage with the aidof a grid-dip meter. Next, a sensitive r.f. voltmeter,such as a 0 - 1 d-e milliammeter inseries with a 1N34 crystal diode is connectedto the grid input receptacle (J,) of the amplifier.The reading of this meter will indicatethe degree of unbalance of the neu.tralizingcircuit. Start with a minimum of applied r.f.excitation to avoid damaging the meter or thediode. Resonate the plate and grid circuits formaximum meter reading, then vary the settingof neutralizing capacitor C until the readingof the meter is a minimum. Each change in c,should be accompanied by re-resonating thegrid and plate tank circuits. When a point ofminimum indication is found, the capacitorshould be locked by means of the auxiliary setscrew.Complete neutralization is a function of theefficiency of the screen bypass system, andsubstitution of other capacitors for those n01tedin the parts list is not recommended. Mica,disc-type, or other form of bypass capacitorshould not be substituted for the units specified,as the latter units have the lowest value ofinternal inductance of the many types testedin this circuit.Bias and The amplifier requires -60 toScreen Supply -110 volts of grid bias, andplus 300 to 600 volts ofscreen potential for optimum characteristicswhen working as a class AB1 linear amplifier.Screen voltage for class C operation(phone) is 400 volts. The voltage may beraised to 500 volts for c.w. operation, if desired,although the higher screen voltage doeslittle to enhance operation. Approximately -120volts cut-off bias is required for either phoneor c-w operation. A suitable bias and screenpower supply for all modes of operation isshown in figure 41, together with an operatingchart for all operating voltages. The supplyfurnishes slightly higher than normal screenvoltage which is dropped to the correct valueby an adjustable series resistor, R, This seriesresistor is adjusted for 30 milliamperes of currentas measured in meter jack ]> when thesupply is disconnected from the amplifier.Series bias resistor R, is adjusted for the samecurrent in jack } under the same conditions.The value of protective bias may now be setby adjusting potentiometer R,.Additional bias is required for class C operationwhich is developed across series resistorR,. Switch S, is open for cla>s C operation andclosed for sideband operation.It is imperative that the screens of the tet-
Figure 424CX-1 OOOA ALL BAND AMPLIFIERPROVIDES TWO KILOWATT P.E.P.FOR SIDEBAND, OR ONE KILOWATTFOR PHONE C.W. OPERATIONGrid, plate, and output meters are across topof panel. Counter dial at center drives variablevacuum tuning capacitor, with loadingcontrol directly beneath it. Meter switch isat left, with bandswitch at right. Entire amplifieris screened to reduce unwanted radiation.circuit whenever the screen current reachesapproximately 100 milliamperes.rode amplifier tubes be protected from excessivecurrent that could occur during tuningadjustments, or during improper operation ofthe amplifier. The safest way to accomplishthis is to include an overload relay that willopen the screen circuit whenever the maximumscreen dissipation point is reached. Two 4-250A tubes or 4-400A tubes have a total screendissipation rating of 70 watts, therefore relayRY-1 should be adjusted to open the screen29-1 0 A 2 KilowattP.E.P. All-band AmplifierDescribed in this section is an efficient allbandlinear amplifier suited for SSB, phone,and c.w. operation up to the maximum legalpower input limit. A 4CX-IOOOA ceramicpower tetrode is employed in the basic circuitshown earlier in this chapter in figure IIC.The Eimac 4CX-lOOOA JS a ceramic and4CX-IOOOA2000 OUTPUT20KV L1 Jz47K50020 KV10 K01cABC-.-:J~ DE -I'JII----
All-band Amp I ifier 641Figure 44END VIEW OF AMPLIFIER4CX-1 OOOA sub-chassis is shown, with platecircuit choke mounted at rear. External blowerprovides forced air to cool tube seals viaentrance hole in reat of chassis. Bias adjustmentpotentiometer is in foreground, betweentube and meter switch.metal, forced air-cooled, radial beam tetrodewith a rated maximum plate dissipation of1000 watts. It is a low voltage, high currenttube specifically designed for class AB 1 r-flinear amplifier service where its high gainand low distortion characteristics may be usedto advantage. At its maximum rated platevoltage of 3000, it is capable of producing1680 watts of single-tone output power in SSBservice. Maximum grid dissipation of the 4CX-1000A is zero watts. The design features whichmake the tube capable of maximum power operationwithout driving the grid into the positiveregion also make it necessary to avoid positivegrid operation.Circuit The circuit of the all-band am-Description plifier is shown in figure 43.A resistance loaded, untunedgrid input circuit is employed in conjunctionwith a pi-network plate output circuit. Griddriving requirements are 60 volts peak acrossresistor R, which has a value of 100 ohms.This corresponds to 36 watts p.e.p. all of whichis dissipated in resistor R,. For normal voiceoperation, a 20 watt non-inductive resistor maybe used. A simple 1N66 diode voltmeter isincorporated in the grid circuit to monitor theexcitation voltage. Grid and screen current arealso measured. The d.c. grid bias voltage isobtained from a small selenium-type half-wavepower supply contained within the amplifier.The plate circuit is designed around the newBarker & Williamson type 852 bandswitchingturret, especially designed for high current,low voltage tubes of this type. The input tuningcapacitor of the network is a 500 fLfLfd.variable vacuum unit, and the output loadingcontrol is a 1500 fLpJd. variable air capacitor.For operation into low impedance loads in the80 meter band, an additional 1500 fLfLfd. fixedmica capacitor may be placed in parallel withthe output loading capacitor, C. A second1N66 diode voltmeter is placed across the outputcircuit as a tuning aid.Screen Grid Tetrode tubes may exhibit re-Operation versed (negative) screen currentto a greater or lesser degree dependingupon individual tube design. Thischaracteristic is prominent in the 4CX-1000Aand, under some operating conditions, indicatednegative screen current of the order of 25milliamperes may be encountered. In general,high values of negative screen current will bemeasured at the lower values of plate potential.It is difficult to measure screen dissipationin the presence of secondary emission of thistype. The usual product of current and voltagemay not provide an accurate picture of actualscreen dissipation. Experience has shown thatthe screen will operate within the limits establishedfor this tube if the indicated screencurrent, plate voltage, and drive voltage approximatethe values given in the operatingchart of figure 48.The screen supply voltage must be maintainedconstant for any values of negative andpositive screen current that may be encountered.Dangerously high plate currents may flowif the screen power supply exhibits a risingvoltage characteristic with negative screen cur-
642 H.F. Power Amplifiersrent. Stabilization may be accomplished in severalways. A combination of voltage regu!at'O'rtubes may be connected across the screen supply,or a heavy bleeder resistor may be placedacross the supply that will draw approximately70 milliamperes at the operating potentia] ofthe screen. Screen voltage is not particularlycritical, especially at plate potentials of 2500volts and higher. However, the screen voltageshould be adjusted for best linearity of thestage when the tube is operated at 2000 voltsplate potential. A 3 50 volt, 150 rna. supplyutilizing choke input and bled to 70 milliamperes,in conjunction with a 100 watt primary"Variac" or variable voltage transformer is asatisfactory source of screen voltage. Three VR-105 regulator tubes passing 40 rna. may beemployed for a fixed-voltage screen supply.Amplifier Layout of the amplifier mayConstruction be seen in figures 42, 44, 45,46, and 47. A 121;4" x 19"'rack pane! is employed to form the front of anenclosure 14" deep. The enclosure is built of\12-inch aluminum angle stock and the top,sides and back are covered with perforatedaluminum sheet. The bottom of the enclosureis formed from a solid sheet of aluminum.Only a very small chassis is required to mountthe 4CX-1000A tube socket, filament transformer,and grid circuit components. A chassismeasuring 6" x 3" x 14" with one end cutoff is employed (figure 47). This chassis ismounted at one end of the enclosure and supportsa small aluminum angle bracket holdingthe two pi-network capacitors. The bandswitchingplate tank inductor is mounted di-Figure 45TOP VIEW OF4CX-1000AAMPLIFIERTetrode tube fits in Eimocair socket on smallchassis at one end olenclosure. Pi-networkcapacitors are mountedon aluminum brocket attachedto chassis, andore panel driven throughflexible shaft couplers.Heavy duty, high-C bandswitchinginductor is atopposite side ol assem ..bly, with output circuitdiode voltmeter mountedabove it on rear of enclosure.Extremely low harmoniccontent of amplifier removesneed of metershields. Amplifier may berun without enclosurewith no sign ol TVI onany band in normal TVsignal area.
Figure 46REAR VIEW OF CHASSIS ASSEMBLYAir inlet, power receptacle and coaxial inputjack are located on reot of amplifier chassis.Pi-network capacitor assembly is at left, withRaypar wideband plate r-t choke in toreground.Plate blocking capacitor is made oftour 500 111dd. TV -type capacitors mountedin 11 Sandwich" between stator plate of vacuumcapacitor and round aluminum plate.rectly to the aluminum bottom plate of theenclosure as seen in figure 45.The low impedance plate tank circuit requiresa higher than normal value of platecoupling capacitor. Four 500 1-'-1-'-fd., 20 KVTV-type capacitors are therefore mounted betweentwo aluminum plates to form a 20001-'-/Lfd. "sandwich" which is bolted to the rearstator terminal of tuning capacitor C. The 10meter inductor section of coil L, bolts directlyto the aluminum plate, as does the plate leadof the 4CX-1000A.Special breechlock terminal surfaces are usedon the tube in place of the conventiond hase.Three tabs protrude through the ceramic envelopefor each tube element, providing a lowi:J.ductance termination. The new Eimac SK-800 socket having a built-in screen bypasscapacitor and an air vent should be employedwith this tube.The rated heater voltage for the 4CX-1000Ais 6.0 volts and the operating voltage, as measuredat the socket, should be maintained withinplus or minus 5% of this value if long lifeoperation is to be obtained. Note that the cathodeand one side of the heater are internallyconnected.Sufficient cooling must be provided for theanode and ceramic-to-metal seals to maintainoperating temperatures below 200 degrees,Centigrade. At sea level, with an inlet air temperatureof 20oC., a minimum of 35 cubicfeet per minute of air flow is required. In thiscase, the chassis is pressurized and a 50 cubicfoot/minute blower is connected to the air inleton the rear of the chassis by means of ashort length of automobile radiator hose. Thecooling air must be maintained on the ceramicseals during standby periods when only heatervoltage is applied to the tube.AmplifierOperationAmplifier operating characteristicsare given in figure 48. Thebias potentiometer R, is adjustedto provide a zero-s,ignal plate current of 250milliamperes, At this value, the bias voltagewill be approximately -55 volts. Resting platecurrent and screen voltage may both be variedover small limits to provide the greatest linearityrange. Maximum single tone pl~te currentis 1.0 ampere. Good linearity may be obtainedat 2000 watts p.e.p. at a plate voltage of2500 and maximum plate current of 800 rna.Care should be taken not to apply excitationto the amplifier in the presence of screen voltagebut with no plate voltage present. A relayinterlock may be applied to the screen circuitto open it in the absence of plate voltage. Gridexciting voltage should be limited to about 60volts, and grid current should be monitored tosee that it does not exceed 0.5 milliamperesduring tune-up.Figure 47UNDER-CHASSIS VIEW OF4CX-l OOOA AMPLIFIERBias supply is placed at one end of chassis.Filament transformer is mounted next to airsocket and short, heavy strap leads are usedto eliminate voltage drop. Special air sockethas built-in screen bypass capacitor to ensurelow inductance ground path.
644 H.F. Power Amplifiers<strong>THE</strong> <strong>RADIO</strong>FIGURE 48)oPERATING CHARACTERISTICS, 4CX-1000A AMPLIFIERPLATE VOLTAGE 2000 2500 3000SCREEN VOLTAGE 325 325 325GRID VOLTAGE_ -55 -55 -551--ZERO SIGNAL:IP(MA.) 250 250 250SINGLE TONE !P(MA) 1000 1000 900ZERO SIGNAL· !G2(MA) -2 -2 -2SINGLE TONE:IG2(MA) 30 30 30SINGLE TONE PEAK 55R-FGRID VOLTSSINGLE TONE DRIVINGPOWER (WAiTS)55 55-~-~---0 0 0SINGLE TONE POWER1080 1460 1680~TPUT (WATTS J --NOTE: ADJUST bRIO VOLTAGE TO OBTAIN SPECIFIEDZERO-SIGNAL PLATE CURRENT.29-11 A High PowerPush-pull Tetrode AmplifierThe push-pull tetrode amplifier still retainsa high degree of popularity in spite of thetrend towards single-ended pi-network amplifiers.Regardless of the power level of thepush-pull amplifier, the physical design mustsatisfy two important requirements. First, theplate and grid circuits of the amplifier must besymmetrical with respect to each tube; andsecond, the input and output circuits must beisolated from each other. Finally, after thephysical layout of the amplifier has been de-termined, steps must be taken to eliminate alltendencies towards parasitic oscillation.Described in this section is a push-pull amplifierintended for operation in the 13 · 15Me. region. Push-pull 4-IOOOA tubes are employed,running at their maximum rated input.Other tubes, such as the 4-400A, 4-250A, or4-125A may be substituted in the design, scalingdown the various components to suitablesizes for the reduced power level. This amplifierdesign is suitable for use in the 3 . 30 Me.frequency region. The lower limit of operationis restricted by the amount of tuning capacityin the grid and plate tank circuits. Paddingcapacitors placed across the variable tuning capacitorsmay be employed to reach the lowerfrequency portions of the HF spectrum. Theupper frequency limit of operation is determinedby the residual screen lead inductanceof the particular tubes used in the circuit. Atfrequencies up to 30 Me., no neutralizing difficultiesare encountered with any of the previouslymentioned tubes.Circuit The complete schematic of theDescription push-pull tetrode amplifier isshown in figure 50. Grid, filam~nt,and auxiliary metering circuits of theamplifier are enclosed beneath the chassis inan "electrically tight" box. All leads leavingthis box are suitably filtered to prevent leakageof energy into (or out of) the enclosure. Theplate circuit of the amplifier is contained in aseparate enclosure above the main chassis.A small degree of feedback exists withinthe tetrode tubes due to the minute value ofgrid-plate capacity. It is therefore necessary tocross-neutralize the tubes by means of twocapacitor tabs mounted near the envelope ofthe tube (figure 53). These capa,citors are adjustedfor minimum r-f feedback at the highestoperating frequency of the amplifier.Any tendency towards parasitic oscillationis eliminated by the use of small parasiticchokes placed in the grid and screen leads ofeach tetrode tube. Excellent parasitic suppressionis achieved by strapping the screen ter-Figure 49HIGH POWER PUSH-PULL AMPLIFIERUSES 4-1 OOOA TUBESThis amplifier design is typical of push-pullstages regardless of power level. Tubes atevisible through screened openings in panel.Controls ate (top to bottom): Output linktuning, variable link coupling, plate circuittuning, and grid circuit tuning. Amplifier ishoused in special 2l" rack. Filament metersfot each tube are on auxiliary panel holdingbias and screen supply.
<strong>HANDBOOK</strong>P-P Tetrode Amplifier645C24-IOOOA/4-400ARr.B1 82 ~RFC1, ·~RFc 2}12HO Kl:X X Y Y flffl tlj ":" ""'tc:;:,Tl xTj r;~;EN BYPASS CAPACITORS4RE CENTRAL A& TYPE 858Co~115V,'\_, -BIAS 115 v. '\., +SCREENC 7 ~ CB '~ C 9 B +Figure 50SCHEMATIC, PUSH-PULL HIGH POWER AMPLIFIERC,-700- 700 !l!lfd. "butter-fly" capacitor. Barker & Williamson type JCXCz, C3-Neutraliring capacitors. 1" x 3" aluminum plate mounted near tubeenvelope (figure 53)C,A-B-60- 60 Jl!lfd. split-stator variable vacuum capacitor. Eimac VVC-60-20. Seetext for substitutec,-soo ll!lfd. Johnson 500E20C,-C.,--0.7 !lfd., 600 volt feedthrough capacitor. Centralab "HY-pass"RFC,-VHF-type choke. National R-60, or Ohmite Z-50RFC,-Heavy duty plate choke. Surplus choke from AN/ART-13 transmitter, orBarker & Williamson type BOOPC-Parasitic suppressor. 3 turns # 74 wire, V:z-inch diam. in parallel with 47 ohm,5 watt carbon 11 Globat 11 resistorminals of the tetrode socket together (figure52) and inserting a parasitic choke betweenthe screen terminal of the socket and the screenbypass capacitor.Eimac air sockets should be employed withthe larger tubes, and a blast of air directed atthe base seal of each tube is required. Twosmall 115-volt blower motors mounted on therear of the chassis direct air streams throughsections of radiator hose to the two tube sock-Figure 51REAR VIEW OF AMPLIFIERPlate tank circuit is placed between tetrodetubes which are mounted in air sockets. Splitstatorvariable vacuum capacitor is panelmounted and rotors are grounded to chassisby heavy copper strap. Tank coil is mountedon 10" ceramic insulators. Individual filamenttransformers for the 4-lOOOA tubes areat rear of chassis. Each tube requires 7.5 voltsat 2 I amperes.Swinging link is panel driven through Millenright-angle drive unit, and flexible, braidedleads connect link to antenna resonating capacitorand coaxial receptacle mounted intop screen of amplifier.Dual blowet motot is mounted to teat ofchassis. All power leads (including filamentleads to meters) are brought out throughbulkhead-type capacitors.
Figure 53CLOSE-UP OF PLATE CHOKE ANDNEUTRALIZING CAPACITORNeutralizing capacitor is made of tab solderedto threaded rod mounted in insulating buttonaffixed to chassis.ets. It is necessary to ground the metal basering of each tube to the chassis, since this ringforms part of the screen shielding circuit. Ifgrounding clips are not furnished with thetube socket, they may be made from a shortlength of %-inch wide brass strap formed tospring against the base ring.Excellent circuit symmetry is achieved bythe use of "butter-fly" -type tuning capacitorsin the grid and plate circuits. A 100-wattplug-in coil may be employed in the grid circuit,while heavy-duty plug-.in coils will sufficein the plate circuit up to the kilowatt level.If extremely high plate voltage is to beused, the Eimac dual vacuum variable ca;pacitor(VVC-60-20) should be employed to preventflash-over. The less expensive Barker & Williamson"butter-fly" capacitor may be employedfor plate voltages of 3 500 or less.AmplifierConstructionIf 4-250A or 4-125A tubesare employed, a 15" x 17" x6" chassis may be used. The4-1000A tubes require a 17" x 19" x 6"chassis. Panel height is determined by the componentsmounted above the chassis deck. Thetop of the amplifier compartment should clearthe plate coil by at least the diameter of thecoil to prevent flash-over and to hold the coilQ to the highest possible value. The amplifierenclosure is made of ;/;!-inch aluminum anglestock covered with perforated aluminum. Theunder-chassis area of the amplifier is sealedwith a piece of solid aluminum sheet.Plate circuit wiring is done with ;/;!-inchsilver plated copper strap, and for the higherpower levels all connections in this circuitshould e.ither be bolted or silver-soldered. Allgrid circuit wiring is done with # 10 copperwire, and all low voltage d.c. leads and a.c.wiring is done in shielded braid, the braid beinggrounded to the chassis at each end ofthe lead. A short length of RG-8/U coaxialline is used for the connection between the coaxialinput plug and the link circuit of thegrid coil.The reactance of the antenna link is tunedout by means of a 500 l"l"fd. variable capacitorplaced in the ground return of the linkline. This capacitor is adjusted for maximumloading with the loosest possible degree oflink coupling to the amplifier tank coil.AmplifierOperationExcitation and grid bias are appliedto the amplifier and aflashlight lamp is connected tothe link output connections. The neutralizingtabs are varied in position with respect to theenvelope of the tube after the grid and platecircuits have been tuned to resonance. Adjustmentsare made to the capacitors until the degreeof feedback is reduced to a minimum.Figure 52UNDER-CHASSIS AREA OFPUSH-PULL AMPLIFIERBlower motors are connected to air socketsby short lengths of radiator hose. Filamentleads are run in braid, grounded at each end.Shield plate is used over bottom of chassis toprovide "electrically tight" enclosure. Screenterminals of each socket are strapped to·gether, and choke PC attaches to center ofstrap.
CHAPTER THIRTYSpeech andAmplitude Modulation EquipmentAmplitude modulation of the output of atransmitter for radiotelephony may be accomplishedeither at the plate circuit of the finalamplifier, commonly called high-level AM orsimply plate modulation of the final stage, orit may be accomplished at a lower level. Lowlevelmodulation is accompanied by a platecircuitefficiency in the final stage of 30 to 45per cent, while the efficiency obtainable withhigh-level AM is about twice as great, runningfrom 60 to 80 per cent. Intermediate values ofefficiency may be obtained by a combinationof low-level and high-level modulation; cathodemodulation of the final stage is a commonway of obtaining combined low-level and highlevelmodulation.High-level AM is characterized by a requirementfor an amount of audio power approximatelyequal to one-half the d-e input to theplate circuit of the final stage. Low-level modulation,as for example grid-bias modulation ofthe final stage, requires only a few watts ofaudio power for a medium power transmitterand 10 to 15 watts for modulation of a stagewith one kilowatt input. Cathode modulationof a stage normally is accomplished with anaudio power capability of about 20 per centof the d-e input to the final stage. A detaileddiscussion of the relative advantages of thedifferent methods for accomplishing amplitudemodulation of the output of a transmitter 1sgiven in an earlier chapter.Two trends may be noted in the design ofsystems for obtaining high-level AM of thefinal stage of amateur transmitters. The firstis toward the use of tetrodes in the outputstage of the high-power audio amplifier whichis used as the modulator for a transmitter. Thesecond trend is toward the use of a high-levelsplatter suppressor in the high-voltage circuitbetween the secondary of the modulation transformerand the plate circuit of the modulated;rage.30-1 ModulationTetrodeModulatorsIn regard to the use of tetrodes,the advantages of these tubeshave long been noted for use inmodulators having from 10 to 100 watts output.The 6V6, 6L6, and 807 tubes have servedwell in providing audio power outputs in thisrange. Recently the higher power tetrodes suchas the 4-65A, 813, 4-125A, and 4-250A havecome into more general use as high-level audioamplifiers. The beam tetrodes offer theadvantages of low driving power (even downto zero driving power for many applications)as compared to the moderate driving power requirementsof the usual triode tubes havingequivalent power-output capabilities.On the other hand, beam tetrode tubes requireboth a screen-voltage power supply anda grid-bias source. So it still is expedient inmany cases to use zero-bias triodes or even647
648 Speech and A. M. Equipment <strong>THE</strong> <strong>RADIO</strong>low-mu triodes such as the 304TL in manymodulators for the medium-power and highpowerrange. A list of suggested modulatorcombinations for a range of power output capabilitiesis given in conjunction with severalof the modulators to be described.Increasing theEffective ModulationPercentageIt has long been knownthat the effective modulationpercentage of atransmitter carrying unalteredspeech waves was necessarily limitedto a rather low value by the frequent highamplitudepeaks which occur in a speechwaveform. Many methods for increasing theeffective modulation percentage in terms ofthe peak modulation percentage have beensuggested in various publications and subsequentlytried in the field by the amateurfraternity. Two of the first methods suggestedwere Automatic Modulation Control and VolumeCompression. Both these methods weregiven extensive trials by operating amateurs;the systems do give a degree of improvementas evidenced by the fact that such arrangementsstill are used in many amateur stations. Butthese systems fall far short of the optimum becausethere is no essential modification of thespeech waveform. Some method of actuallymodifying the speech waveform to improve theratio of peak amplitude to average amplitudemust be used before significant improvementis obtained.It has been proven that the most serious effecton the radiated signal accompanying overmodulationis the strong spurious-sideband radiationwhich accompanies negative-peak clipping.Modulation in excess of 100 per cent inthe positive direction is accompanied by noundesirable effects as far as the radiated signalis concerned, at least so long as the linearmodulation capability of the final amplifier isnot exceeded. So the problem becomes mainlyone of constructing a modulator-final amplifiercombination such that negative-peak clipping(modulation in excess of 100 per cent in anegative direction) cannot normally take placeregardless of any reasonable s pee c h inputlevel.AssymetricalSpeechThe speech waveform of thenormal male voice is characterized,as was stated before,by high-amplitude peaks of short duration. Butit is also a significant characteristic of thiswave that these high-amplitude peaks are poledin one direction with respect to the average amplitudeof the wave. This is the "lopsided" orassymetrical speech which has been discussedand illustrated in an earlier chapter.The simplest method of attaining a highaverage level of modulation without negativepeak clipping may be had merely by insuringthat tbese high-amplitude peaks always arepoled in a positive direction at the secondaryof the modulation transformer. This adjustmentmay be achieved in the following manner:Couple a cathode-ray oscilloscope to theoutput of the transmitter in such a manner thatthe carrier and its modulation envelope maybe viewed on the scope. Speak into the microphoneand note whether the sharp peaks ofmodulation are poled upward or whether thesepeaks tend to cut the baseline with the "brightspot" in the center of the trace which denotesnegative-peak clipping. If it is not obviouswhether or not the existing polarity is correct,reverse the polarity of the modulating signaland again look at the envelope. Since a pushpullmodulator almost invariably is used, theeasiest way of reversing signal polarity is toreverse either the leads which go to the gridsor the leads to the plates of the modulatortubes.When the correct adjustment of signal polarityis obtained through the above procedure,it is necessarily correct only for the specificmicrophone which was used while making thetests. The substitution of another microphonemay make it necessary that the polarity be reversed,since the new microphone may be connectedinternally in the opposite polarity tothat of the original one.Law-LevelSpeech ClippingThe low-level speech clipperis, in the ideal case, avery neat method for obtainingan improved ratio of average-to-peakamplitude. Such systems, used in conjunctionwith a voice-frequency filter, can give a veryworthwhile improvement in the effective modulationpercentage. But in the normal amateurtransmitter their operation is often less thanideal. The excessive phase shift between thelow-level clipper and the plate circuit of thefinal amplifier in the normal transmitter resultsin a severe alteration in the square-waveoutput of the clipper-filter which results froma high degree of clipping. The square-waveoutput of the clipper ends up essentially as adouble saw-tooth wave by the time this wavereaches the plate of the modulated amplifier.The net result of the rather complex action ofthe clipper, filter, and the phase shift in thesucceeding stages is that the low-level speechclipper system does provide an improvement
HA.NDBOOKDesign 649MODULATOR+B +BMOD. R.f. 115 V.fiNALEFigure 1HIGH-LEVEL SPLATTER SUPPRESSORThe high-vacuum diode acts as a series limiterto suppress negative-peak clipping int.~e modulated r-f amplifier as a result oflarge amplitude negative-peak modulatingsignals. In addition, the low-pass filter followingthe diode suppresses the transientswhich result from the peak-clipping action ofthe diode. Further, the filter attenuates allharmonics generated within the modulatorsystem whose frequency lies above the cutofffrequency of the filter. The use of anappropriate value of capacitor, determinedexperimentally as d is c us s e d in ChapterFifteen, across the primary of the modulationtransformer (C.1J introduces further attenua ..tion to high-frequency modulator harmonics.Chokes suitable for use at L are manufacturedby Chicago Transformer Corp.. The correctvalues of capacitance for C1, C:, C,, and C "are specified on the installation sheet for thesplatter suppressor chokes for a wide varietyof operating conditions.in the effective modulation percentage, but itdoes not insure against overmodulation. Anextensive discussion of these factors, alongwith representative waveforms, is given inChapter Fifteen. Circuits for some recommendedclipper-filter systems will also be foundin the same chapter.High-LevelOne practicable methodSplatter Suppressor for the substantial eliminationof negative-peakclipping in a high-level AM transmitter is theso-called high-level splatter suppressor. Asfigure 1 shows it is only necessary to add ahigh-vacuum rectifier tube socket, a filamenttransformer and a simple low-pass filter to anexisting modulator-final amplifier combinationto provide high-level suppression.The tube, V,, serves to act as a switch tocut off the circuit from the high-voltage powersupply to the plate circuit of the final amplifer as soon as the peak a-c voltage acrossthe secondary of the modulation transformerhas be~ome equal and opposite to the d-e voltagebe:ng applied to the plate of the final amplifiersta~e. A single-section low-pass filterserves to filter out the high-frequency componentsresu!t:ng from the cli~oping action.Tube V, may be a receiver rectifier with a5-volt filament for any but the highest powertransmitters. The 5Y3-GT is good for 125 rna.p!ate current to the final stage, the 5R4-GYand the 5U4-G are satisfactory for up to 250rna. For high-power high-voltage transmittersthe best t"Jbe is the high-vacuum transmittingtube type 836. This tube is equivalent inshape, filament requirements, and average-currentcapabilities to the 866A. However, it isa vacuum rectifier and utilizes a large-sizeheater-type dual cathode requiring a warm-uptime of at least 40 s e c o n d s before currents h o u 1 d be passed. The tube is rated at anaverage current of 2SO rna. For greater currentdrain by the final amplifier, two or more 836tubes may be placed in parallel.The filament transformer for the cathode ofthe splatter-suppressor tube must be insulatedFigure 2TOP VIEW OF <strong>THE</strong>6L6 MODULATOR
Figure 3UNDERCHASSISOF <strong>THE</strong>6L6 MODULATORA 4-connector plug isused lor filaments andp I a t e voltage to thespeech amplifier, while a6-wire terminal strip isused lor the high-voltageconnections and the transmitfer"controlswitch.for somewhat more than twice the operatingd-e voltage on the plate modulated stage, toallow for a f a c t o r of safety on modulationpeaks. A filament transformer of the type normallyused with high-voltage rectifier tubeswill be suitable for such an application.30-2 Design of SpeechAmplifiers and ModulatorsA number of representative designs forspeech amplifiers and modulators is given inthis chapter. Still other designs are includedin the descriptions of other items of equipmentin other chapters. However, those persons whowish to design a speech amplifier or modulatorto meet their particular needs are referred toChapter Six, Vacuum Tube Amplifiers, for adetailed discussion of the factors involved inthe design of such amplifiers, and for tabularmaterial on recommended operating conditionsfor voltage and power amplifiers.10 to 120 Watt It is difficult to surpassModulator with the capabilities of theBeam Power Tubes reliable beam powertube when an audiopower output of 10 to 120 watts is requiredof a modulator. A pair of 616 tubes operatingin such a modulator will deliver good platecircuit efficiency, require only a very smallamount of driving power, and they impose noserious grid-bias problems.Circuit Included on the chassis of theDescription modulator shown in figures 2and 3 are the speech amplifier,the driver and modulation transformers for thePOWER CONNECTIONSA-G~OUNDB-6.3V.c- 0+250-300 vD- BIASE-8+250-750 v(SEE FIGURE 5):.... 6 3 v.* 0 MATCHED PAIR OF RESISTORS, 1 o/o HQ_l_~ ..PL 1 ~:==;==;=Pri CWS1A\___ - - - - - - -- - - -- - __,:::,.A B C 8+ TOFINALFigure 4SCHEMATIC OF BEAM POWER TUBE MODULATORM-0- 250 d.c. milliammeterT z-- 11 Poly-pedance 11 Modulation transformer60-waff level = Stancor A-3893, or UTC S-20T1-Driver transformer. Stancor A-4701, or UTC S-10.725-waff level = Stancor A-3894
output tubes, and a plate current milliammeter.The power supply has not been included. The6SJ7 pentode first stage is coupled throughthe volume control to the grid of a 6]5 phaseinverter. The output of the phase inverter iscapacitively coupled to the grids of a 6SN7-GTwhich acts as a push-pull driver for the outputtubes. Transformer coupling is used betweenthe driver stage and the grids of the outputtubes so that the output stage may be operatedeither as a class AB, or class AB, amplifier.The OutputStageEither 616, 616-G or 807tubes may be used in the outputstage of the modulator. Asa matter of fact, either 6V6-GT or 6F6-Gtubes could be used in the output stage ifsomewhat less power output is required. The807 tube is the transmitting-tube counterpartof the 6L6 and carries the same ratings andrecommended operating conditions as the 6L6within the ratings of the 6L6. But the 807does have somewhat greater maximum ratingswhen the tube is to be used for ICAS (IntermittentCommercial and Amateur Service) operation.The 6L6 and 807 retail to the amateurfor essentially the same price, although the 807is available only from transmitting tube distributors.The 6L6-G tube retails for a somewhatlower price; hence it is expedient to purchase6L6-G tubes if 360 to 400 volts is themaximum to be used on the output stage, or807 tubes if up to 750 volts will be applied.Tabulated in figure 5 are a group of recommendedoperating conditions for different tubetypes in the output stage of the modulator. Incertain sets of operating conditions the tubeswill be operated class AB,, that is with in-General Purpose Modulator 651FIGURE 5RECOMMENDED OPERATING CONDITIONS FORMODULATOR OF FIG.4 FOR DIFFERENT TUBE TYPESPLATE SCREEN GRID~~~~~D !u~~~T 6~r~0~ !TUBESCLASSV1.V2 VOLTS VOLTS BIASIE I (CI (DI (OHMS} (MA.) (WATTS) I6V6GT AB1 250 250 -15 10,000 170-80 ; 106V6GT AB1 285 285 -19 8,000 70-95 156L6 AB 1 380 270 -23 6,600 1 es-r3s 276L6 AB2 380 270 -23 3,800 85-205 47807 AB1 600 300 -34 I 10,000 35-140 I 58807 AB1 750 300 -35 12,000 f 30-140 75807 AB2 750 300 -35 7,300 30-240 120creased plate current with signal but with nogrid current. Other operating conditions specifyclass AB, operation, in which the plate currentincreases with signal and grid currentflows on signal peaks. Either type of operationis satisfactory for communication work.30-3 General PurposeTriode Class B ModulatorHigh level class B modulators with poweroutput in the 125 to 500 watt level usuallymake use of triodes such as the 809, 811,8005, 805, or 810 tubes with operating platevoltages between 750 and 2000. Figures 6,7, and 8 illustrate a general purpose modulatorunit designed for operation in this powerrange. Figure 8 gives a group of suggestedoperating conditions for various tubes. Thesize of the modulation transformer will ofcourse be dependent upon the amount of audiopower developed by the modulator. In the caseJFigure 6REAR VIEW OF <strong>THE</strong>GENERAL-PURPOSE MODULATOR••'
l652 Speech and A. M. Equipment6SJ7.005c-6N7684-G TIVIT2l[~o~e684-GV2.J+-,_-10 JJF + 4.7 K 10 ..UF + + 5 K450-.r505Y3-GTCHI25 K""10,--_- ---,IIB+ 1000-1250 V.L_+_ - ~~~'!~%~(./::: i~~%iEo,TERMINALS.*=MATCHED PAIRRESISTORS, 1%Figure 7SCHEMATIC OF GENERAL PURPOSE MODULATORM-0- 500 ma.T 4-Suitable for tubes used.T,-Driver transformer. Stancor A-4761For 811-A's = 6.3 volt, 8 amp. Stancor P-630811T 2 -Modulation transformer.For 810's = 10 volt, 10 amp. Stancor P-6461300 Poly-pedance11 watt rating, Stancor A-3898.CH,-14 henry, 100 ma. UTC S-19. Stancor C-1001500 watt rating, Stancor A-3899.R,-1 K, 10 watts, adjustable. Set for plate currentT 3-360- 0-360 volts, ISO mo. Stancor PC-8410of 80 ma. (no signal) to 684-G tubes(approximately 875 ohms).v,,v,-see figure 8.of tl:-:e 500 watt modulator (figures 9 and 10)the size and weight of the components requireths.t the speech amplifier be mounted on ascp1rate chassis. For power levels of 300 wattsor less it is possible to mount the completespeech system on one chassis.~- S:G:T:-0~~~~~ ~GS CONDITIONS--lIFOR GENERAL PURPOSE MODULATOR~sEsTP"C:.r~ GRID PLATE PLATE ro SINE WAVE IV1, v-~ I vOLTAGE I BIAS CURRENT PLATE LOAD pOWEROUTPUTf-·- ---+--- __ (VOLTS) (MA.l (OHMS) (WATTS)I 809'_j' 700 ! 0 70-250 6,200 -~~811:A ~~5~i o 30-350 5,IOO I75 I~-~ _ _:co-+ o 145·350 7,400 2.45 -~'811-A I 1250 I 0 50-350 9,200 310~A I I_'OO ~4.5 32·3I5 I2,400 I 340 II 8~250 I 0 148·400 6,700 300 !805 1 , 500 --j:---I-6-+---84___4_0_0 +-8-.2-0-0-+--3-7 0----j~~o~~---5_o-+-6o_·_4_2_o+_I'-_P_o_o_+-_4_5_o_---l~ 2500 -75 50-420 I7,500 500~5 I 1500 -67 40-330 9,800 ~Circuit Descriptionof General PurposeModulatorThe modulator unitshown in figure 6 iscomplete except for thehigh voltage supply requiredby the modulator tubes. A speech amplifiersuitable for operation with a crystalmicrophone is included on the chassis alongwith its own power supply. A 6S]7 is usedas a high gain preamplifier stage resistancecoupled to a 6N7 phase inverter. The audiolevel is controlled by a potentiometer in theinput grid circuit of the 6N7 stage. Push-pull6B4-G low fL triodes serve as the class B driverstage. The 6B4's are coupled to the grids ofthe modulator tubes through a conventionalmulti-purpose driver transformer. Cathode biasis employed on the driver stage which is capableof providing 12 watts of audio power forthe grid circuit of the modulator.The modulator illustrated in figures 9 and10 is designed for use with class B 810 triodetubes operating at a plate potential of 2500volts. Maximum audio power available is 500watts, sufficient to 100% modulate a transmitterrunning one kw. input. The modulator maybe driven by the simple speech amplifier of figure12, or by the clipper-amplifier of figure 15.
Figure 9REAR VIEW OF 810SOO-WATT CLASS BMODULATORBias control is directly belowthe 810 tubes with biassupply to right.The modulator chassis includes a small lowimpedance bias supply capable of deliveringapproximately -7 5 volts under the changingload conditions imposed by the varying gridcircuit impedance of the class B modulator.A low pass audio filter network is placedafter the modulator stage to reduce harmonicsDRIVER~OR MER810generated in the audio system which wouldnormally show up as side-band "splatter" onthe phone signal. Frequencies above 3 500cycles are attenuated 15 decibels or more bythe simple pi-network filter shown in figure10.TO R-F AMP-65 TO 75 V.____."'B+2500 V5Y3GTFigure 10SCHEMATIC OF SOO WATT TRIODE MODULATOR WITH BIAS SUPPLYT1-Driver transformer, 15 watts. Stancor A-4761T,-Modulation transformer, 500 watt. Chicago Transformer Co. CMS-3T J-5 volts, 3 amp. Stan cor P-6467T;-Tapped bias transformer, 15-100 volts. UTC S-51r,-10 volt, 10 amp. Stancor P-6461CH ,-500 ma. "splatter choke" Chicago Transformer Co. SR-500CHz-7 henry, 150 ma. Stancor C-1710M-0- 500 d.c. milliammeterS,A-8-High voltage switch (see text)
654 Speech and A. M. Equipment <strong>THE</strong> <strong>RADIO</strong>5001 K 1'Q'KV100 A (--~ I+------' I8- B+ TO2500V OSCILLOSCOPEFigure 11TEST SETUP FOR 500-WATTMODULATORFor c-w operation the secondary of the classB modulation transformer is shorted out andthe filament and bias circuits of the modulatorare disabled. Switch s,A must have 10,000volt insulation rating. A suitable switch maybe found in the war surplus BC-306A antennaloading unit.All low voltage connections to the modulatorare brought out to a six terminal phenolicstrip on the back of the chassis. A 0-500 rna.d-e meter is placed in the filament circuit ofthe 810 tubes. The meter is placed across a50 ohm, 1 watt resistor so that the filamentreturn circuit of the modulator is not brokenif the meter is removed.The modulation transformer, T,, is designedfor plate-to-plate loads of either 12,000 ohmsor 18.000 ohms when a 6250 ohm load isplaced across the secondary terminals. The 810tubes are correctly matched when the 18,000ohm taps are used at a plate potential of2500 volts, or when the 12,000 ohm taps areused at a plate potential of 2000 volts or 2250volts.ModulatorConstructionBecause of the great weight ofthe modulator components it isbest to use a heavy-duty steelchassis. A 13" x 17" x 4" chassis (Bud CB-643), a 14" steel panel (Bud PS-1257G) anda pair of Bud MB-449 mounting brackets makeup the assembly for this particular modulator.As seen in figure 9, the CMS-3 modulationtransformer is mounted in the left-front cornerof the unit. The secondary terminals of T, areto the front of the chassis, clearing the frontpanel by about Y2 ". To the right of T, isplaced the high level audio filter choke, CHr.The two 81 0 tubes are mounted in back of T,.To the right of the 810 tubes is the 10 voltfilament transformer, T .. To the right of T, isthe 5Y3-GT bias rectifier tube. Between T,and CH, are mounted the mica bypass capacitorswhich make up the high level filter network.Two .003 /Lfd., 5000 volt mica capacitorsare paralleled for C and also for (,. C is madefrom a .001 /Lfd. capacitor and a .0015 /Lfd. capacitorwhich are connected in parallel. Allof these capacitors are mounted upon a plywoodbracket which insulates them from themetal chassis. This prevents insulation breakdownwithin the capacitors which might occurif they were fastened direct! y to the metalchassis.The bias supply components are mountedbeneath the four-inch deep chassis. Placementof these pares is not critical. The bias adjustmentcontrol, R,, is mounted on the back lipof the chassis as are the two high voltage terminals.Millen 37001 high voltage connectorsare used for the two high voltage leads. HighvoltageTV wire should be employed for allleads in the 810 plate circuit.ModulatorAdjustmentWhen the modulator has beenwired and checked, it should berested before being used withan r-f unit. A satisfactory rest ser-u p is shownin figure 11. A common ground lead shouldbe run between the speech amplifier and themodulator. Six 1000 ohm 100 watt resistorsshould be connected in series and placed acrossthe high voltage terminals of the modulatorunit to act as an audio load. The first step is toplace the 810 tubes in their sockets and turnswitch S, to the "phone" position. The 810filaments should light, and switch section S"'should remove the short across the secondaryof T,. Rr should be adjusted to show -75volts from each 810 grid terminal to groundas measured with a high resistance voltmeter.If an oscilloscope is available, it should be coupledto point "A" on the load resistor (figure11 ) through a 500 /L/Lfd. ceramic TV capacitorof 10,000 volts rating. The case of the oscilloscopeshould be grounded to the commonground point of the modulator.A plate potencial of 2500 volts is now appliedto the modulator, and Rr adjusted fora resting plate current of 50 milliamperes asread on the 500 milliampere meter in the cathodecircuit of the modulator. Be extremelycareful during these adjustments, since theplate supply of the modulator is a lethal weapon.Never touch the modulator when the platevoltage supply is on! Be sure you employ theTV blocking capacitor between the oscilloscopeand the plate load resistors, as theseload resistors are at high voltage potential!If a high resistance a-c voltmeter is availablethat has a 2000 volt scale, it should be clipped
<strong>HANDBOOK</strong>1 0-Watt Amplifier-Driver6556B4GDRIVERTRANSFORMER- ...,,,,-(')t,ltk'::>-~~~~~5-c.:-111»
656 Speech and A. M. Equipment <strong>THE</strong> <strong>RADIO</strong>6SJ76J5+4~8~~K,lWB+2.50V. BIAS B+ 1.500-3000V.AT15t.AA. (SEETABLEI)Figure 13PUSH-PULL 304TL CLASS AB 1 ,MODULATORA three wire shielded cable should be usedto connect the 6B4-G tubes to the driver transformerlocated at the grids of the class Btubes. This cable may be any reasonablelength up to 2 5 or 30 feet. Any of the modulatorconfigurations shown in figure 8 may bedriven with this simple speech amplifier.30-5 500-Watt304TL ModulatorOrdinarily, few amateurs would be inclinedto design the high-level stages of their transmittersaround type 304TL tubes. Although the304TL unquestionably is an excellent tube,with its extremely high transconductance andpower sensitivity, its price and capabilitiesare in excess of those required even for a kilowattamateur transmitter. But with enormoussupplies of these tubes available at a relativelylow price on the surplus market it becomeseconomical to consider their use in high-powerTABLE I304TL OPERATING CHARACTERISTICS, CLASS ABI(NO DRIVING POWER REQUIRED)D.C. PLATE VOLTS 1500 2000 2.&00 3000!;RID BIAS * -118 -170 -2.30 -290ZERO-SIGNALPLATE CURRENT 270 200 180 130(MA.)MAX. SIGNALPLATE CURRENT 572 548 483 444('-'A.)**PLATE-TO-PLATELOAD (OHMS)2.540 5300 8500 12000PEAK A.F.GRID VOLTS 118 170 230 290(PER TUBE)MAX. POWEROUTPUT (WATTS)258 490 flO 730*ADJUST TO GIVE STATED ZERO-SIGNAL PLATE CURRENT** VALUES GIVEN FOR SINE-WAVE MODULATION. FOR VOICEWAVE FORMS, MAXIMUM CURRENT WILL BE APPROXIMATELYTWO-THIRDS SINE-WAVE VALUES.Figure 14amateur transmitters. In fact, with the use ofthese 304TL's the cost of heating power forfilaments becomes a much more significantfigure than the cost of the tubes.The 304TL is ideally suited for use as amodulator for a high-power amateur transmitter.In fact, due to the relatively low amplificationfactor of the tube (about 12 ) the304TL may even be used as a class A triodeHeising modulator. Such an arrangement ispracticable for modulating a medium-powertransmitter when a 1500 to 2000 volt platesupply is available.The class AB, operating characteristics ofthe 304TL tube are shown in figure 14. An interestingfact to be observed is that underthese conditions no grid driving power is requiredby the 304TL tubes. It is necessaryonly to supply the correct amount of grid drivingvoltage. The amounts shown may easilybe obtained from a single 6]5 tube, and theusual push-pull triode driver stage for the highpowered modulator may be eliminated. A suitablemodulator-driver unit which will deliver500 watts of audio from two 304TL tubes operatingclass AB, is shown in figure 13.The filaments of the 304TL tubes may beconnected either in series or parallel for either5 volt, 50 ampere operation, or 10 volt, 25 ampereoperation. To conserve filament powerduring standby periods, the filaments of the304TL tubes may be either turned off, or maybe dropped to one-half voltage by means of adropping resistor in the primary circuit of thefilament transformer. This resistor may beshorted out during periods of transmission bya small relay actuated by the transmitting controlsystem.The gain of the speech amplifier is sufficientso that an inexpensive crystal microphonemay be used with the modulator. The
<strong>HANDBOOK</strong> 15-Watt Clipper-Amplifier 65712AX7 6AL5 12AU7 6B4-G TzUNLESS O<strong>THE</strong>RWISE INDICATED;ALL RESISTORS 0.5 WATTALL CAPACITORS IN .JJF.RESISTORS MARKED WITH ASTERISK *ARE BALANCED PAIRS.T1- 450-0-450 VOLTS AT 105 MA.CHICAGO TRANSFORMER PSR-105FILS.T2.-CLASS B DRIVER TRANSFORMERCHICAGO TRANSFORMER CDS-1T3 -125 V. AT 15 MA.STANCOR PS-8415LPF-2-LOW PASS FILTER UNITCHICAGO TRANS. LPF-2Ll- 12 HI 150MA.CHICAGO TRANS. RC-12150 2.A. g Tl T31jSR-50MA. REPLACEMENT TYPE'. [f3AG ~ 2 .SELENIUM RECTIFIER~>-.. ---~-------------_J11ov.rv----~------------------------~GND.Figure 15SCHEMATIC, 15 WATT CLIPPER-AMPLIFIER"phone-c.w." switch is connected so as toshort the modulation transformer and to openthe filament circuit of the 304TL tubes duringc-w operation.30-6 A 15-WattClipper - AmplifierThe near-ultimate in "talk power" can beobtained with low level clipping and filteringcombined with high level filtering. Such amodulation system will have real "punch,"yet will sound well rounded and normal. Thespeech amplifier described in this sectionmakes use of low level clipping and filteringand is specifically designed to drive a pair ofpush-pull 810 modulators such as shown inSection Three.CircuitDescriptionThe schematic of the speechamplifier-clipper is shown infigure 15. A total of six tubes,including a rectifier are employed and the unitdelivers 15 watts of heavily clipped audio.A 12AX7 tube is used as a two stage microphonepre-amplifier and delivers approximately20 volts ( r.m.s.) audio signal to the6ALS series clipper tube. The clipping levelis adjustable between 0 db and 15 db by clippingcontrol, R,. Amplifier gain is controlledby R,, in the grid circuit of the second sectionof the 12AX7. A low pass filter having a 3500cycle cut-off follows the 6AL5 clipper stage,with an output of 5 volts peak audio signalunder maximum clipping conditions. A doubletriode12AU7 cathode follower phase-inverterfollows the clipper stage and delivers a 125volt r.m.s. signal to the push-pull grids of the6B4-G audio driver tubes. The 6B4-G tubesoperate at a plate potential of 3 3 0 volts andhave a -68 volt bias voltage developed by asmall selenium rectifier supply applied to theirgrid circuit. An audio output of 15 watts is developedacross the secondary terminals of theclass B driver transformer with less than 5 percent distortion under conditions of no clipping.A 5U4-G and a choke input filter network provideunusually good voltage regulation of thehigh voltage plate supply.AmplifierConstructionThe clipper-amplifier may bebuilt upon the same chassis asthe power supply, providedthe low level stages of the amplifier are spacedaway from the power transformers and filterchokes of the supply. All capacitors and resistorsof the audio section should be mounted asclose to the respective sockets as is practical. Forminimum hum pickup, the filament leads to thelow level stages should be run in shielded braid.
658 Speech and A M. Equipment <strong>THE</strong> <strong>RADIO</strong>Those resistors in the 12AU7 phase inverterplate circuit and the grid circuit of the 6B4-Gtubes should be matched to achieve best phaseinverter balance. The exact value of the pairedresistors is not important, but care should betaken that the values are equal. Random resistorsmay be matched on an ohmmeter to findtwo units that are alike in value. When thesematched resistors are soldered in the circuit,care should be taken that the heat of the solderingiron does not cause the resistors toshift value. The resistors should be held firmlyby the lead to be soldered with a long nosepliers, which will act as a heat-sink betweenthe soldered joint and the body of the resistor.If this precaution is taken the two phase mverter outputs will be in close balance.Adjustment ofthe SpeechAmplifierWhen the wiring of thespeech amplifier has beencompleted and checked, theunit is ready to be tested.Before the tubes are plugged in the amplifier,the bias supply should be energized and thevoltage across the 600 ohm bleeder resistorshould be measured. It should be -68 volts.If it is not, slight changes in the value of theseries resistor, R,, should be made until thecorrect voltage appears across the bleeder resistor.The tubes may now be inserted in theamplifier and the positive and cathode voltageschecked in accordance with the measurementsgiven in figure 15. After the unit has beentested and is connected with the modulator,R, should be set so that it is impossible to overmodulatethe transmitter regardless of the settingof R1. The gain control ( R1) may then beadjusted to provide the desired level of clippingconsistent with the setting of R,.30-7 A 200-Watt811-A De-luxe ModulatorOne of the most popular medium power r-famplifier stages consists of a single tetrodetube, such as the 4-125A, 813, or 7094 operatingat a plate potential of 1200- 1700 voltsand a plate current of 150-275 milliamperes.Such an amplifier requires a minimum of r-fdriving power, allows an input of 300 to 400watts, and yet employs power supply componentsthat are relatively modest in cost. The5-db signal increase between a 300 watt transmitterand a 1000 watt transmitter is veryexpensive when one considers the additionalcost of modulator and power supply equipment.Additional economy may be achieved ifthe modulator and final amplifier are operatedfrom the same power supply. The new series ofChicago-Standard plate transformers providevoltage ranges in the 1000 to 1500 volt regionand are well suited for the combinationof this modulator and the aforementioned r-ftubes. Within this voltage range, the 811-Atriode is an excellent choice for the modulatortubes. Zero bias operation may be had up toFigure 16REAR VIEW OF811-A MODULATORModulator tubes and voltageregulator are at right withhigh level filter at center ofchassis. Plug-in speech amplifieris to left of clipper.Gain and clipping controlsare atop the small chassis.6L6/5881 is used as cathodefollower driver stage lormodulator.
<strong>HANDBOOK</strong> 811-A Modulator 659lOOK R2 CLIPPER~ r--NV----11• ---1W+ +!~~I !~;I= ~1.BK~6.3v. li2-57KS0412L _______ _ •IHC+-±-''-+.----ct--cc-~--:-~-cc--1co ~L g_ __ - --- - -- - _jNOTES502 $04 CONNECTIONS1-}6.3 V. ""V OUT2-I-ALL RESISTORS l/2 WATT UNLESSO<strong>THE</strong>RWISE SPECIFIED.2-SW!TCH S1 (PHONE-CW) SHOWNIN PHONE POSITION.3-RYl SHOWN ENERGIZED.6L6GB/5B81 3- GROUND, B-4-} 1!5 V "\...INs-6- 8+350 v; TO EXCITER7-} CW-PHON£ SWITCHa-10-}1 19- GROUND, 8-_ RY1 CONTROL12-M/C. CONTROL PUSH-TO-TALK CIRCUIT'+---,--i
660 Speech and A. M. Equipment <strong>THE</strong> <strong>RADIO</strong>Two 8 11-A tubes are employed in the classB stage. When operated at 1000 volts, no biassupply is needed. At voltages of 1200 or abov~,approximately 9 volts of bias is required. Th1sis supplied by a voltage divider composed ofa 20K, 10 watt resistor and a 2D21 thyratrontube. When the miniature 2D21 is connectedas a triode, it acts as a voltage regulator tubewith a constant voltage drop of almost 9 voltsfrom plate to cathode. The tube will regulateover 300 milliamperes of current while maintaininga reasonably constant voltage dropacross its terminals. The center tap of the811-A filament transformer {Ts) is thus heldat a positive potential with respect to ground.Since the center tap of the 811-A driver transformer( T,) is grounded, the modulator tubesare biased at a constant negative voltage equalto the voltage drop across the 2D21 regulatortube in the cathode circuit of the class B stage.The plate to plate load impedance of the811-A tubes when operating at 1500 volts isapproximately 12,000 ohms. A multi-matchtype modulation transformer may be employedif desired, but in this case a Stancor A-3829unit was used. This transformer is designed tomatch the plate-to-plate load impedance of9,000 ohms to a secondary load of 5000 ohms.With the 12,000 ohm load of the 811-A tubes,a secondary load of 7,500 ohms must be usedto maintain the same primary to secondary impedanceratio. This secondary load can be obtainedwith a single 7094 tube operating at1500 volts and 200 milliamperes of plate current( 300 watts input). Other tubes and loadimpedances can also be used, providing ther-f input to the modulated stage does not exceed400 watts. For example, a 4-125A tubeoperating at 2000 volts and 165 rna. ( 330watts) may be modulated by this audio unit.The secondary winding of the modulationtransformer can pass a maximum of 300 milliampereswith safety.The audio output from the 811-A stage ispassed through a high level low-pass "splattersuppressor" which attenuates all audio frequenciesabove 3500 cycles. The use of bothlow level and high level audio filters doesmuch to reduce the broad sidebands and cochannelinterference that seems to be so commonon the amateur phone bands.A high voltage relay RY, is employed toshort the secondary of the modulation transformerand remove plate potential from themodulator tubes for c-w operation. The relayis actuated by the "phone-c.w." switch on thefront panel of the modulator. Other segmentsof this switch turn off the modulator filamentsand provide extra contacts to controlauxiliary equipment.A 350 volt supply is incorporated in themodulator unit to power the speech amplifierand driver stage and to provide power forthe r-f exciter stages of the transmitter. Thevarious power and control leads are broughtout to a multi-connector plug mounted on therear of the modulator deck.Figure 18UNDER-CHASSIS VIEW OF811-A MODULATORHigh voltage relay is between 81!-A tubesockets and low voltage components are at' opposite end of chassis.
<strong>HANDBOOK</strong>811-A Modulator 661ModulatorConstructionThe modulator is constructedupon a steel chassis measuring8" X 1 r X 2". A lOY;(aluminum panel is bolted to the chassis withtwo mounting brackets to form a rugged assembly.Placement of the major parts may beseen in figures 16 and 18. The modulationtransformer T, and the 811-A tubes occupythe right end of the chassis, balanced in weightby the power transformer T, and modulatorfilament transformer T, at the opposite end ofthe chassis. The center space is taken by theplug-in speech amplifier, the high level splatterfilter assembly and the 5881 driver stage.The speech amplifier is constructed as a separateunit on a small aluminum utility boxmeasuring 5" x 3" x 2". The bottom of thebox holds two male plugs which match tworeceptacles mounted on the amplifier chassis.The speech amplifier, therefore, may be wiredand tested as a separate unit. Clipping andaudio level controls are mounted atop theamplifier box as long usage of clipper circuitshas proven that these controls need not be readjustedonce they are properly set.The phone-c.w. switch, relay RY-, and varioussmall components are mounted beneaththe chassis (figure 18) . The input receptaclefor the speech amplifier box is located adjacentto the microphone receptacle on the frontpanel of the modulator making the interconnectinglead less than two inches long. Alsoplaced beneath the chassis are the filter chokefor the low voltage supply and the various bypassand filter capacitors.Wiring and Testingthe ModulatorThe speech amplifiershould be wired first.The small resistors andcapacitors are mounted either between the tubesocket pins, or between terminals of smallphenolic tie-point strips. Transformer T, isfastened within the amplifier box and is wiredin the circuit after all other wiring is completed.Plugs PL, and PL, are mounted on thebottom portion of the box; the plug pins arewired to the proper points of the speech amplifierwith short lengths of wire that allowthe bottom plate to be removed for inspectionand testing without the necessity of unsolderingany connections to the plugs.The modulator chassis should be wired next.All leads to T,, R Y -1, and the low pass filtershould be carefully insulated from the chassis.High voltage ''5000 volt test" cable should beemployed for these connections. The capacitorsthat make up the high level audio filterare mounted directly to the terminals of thefilter choke which is mounted above the chassison Y2-inch ceramic insulators. High voltageconnections to the modulator are made throughMillen 37001 safety terminals.When the wiring has been completed andchecked, the 12AX7, 6AL5, 12AU7, 5881,and 5 V 4-GB tubes should be inserted in theirsockets and the speech amplifier is pluggedinto the modulator receptacles. The verticalamplifier of an oscilloscope should be connectedto one grid terminal of the 811-A stage.Plate voltage of the 5881 should be approximately3 70 volts. A low level 1000 cycle tone(approximately 0.0 5 volts, r.m.s.) is appliedto the amplifier input. The output level of thespeech amplifier is controlled by the settingof the clipping control R, and the audio gainis controlled by potentiometer R, in the gridcircuit of the 12AX7. The clipping controlshould be set so that not more than 60 voltsr.m.s. is developed from one 811-A grid toground. The modulator tubes may now beplugged in their sockets. A 7K, 200 watt resistorshould be placed between the "H.V.Out" and "H.V. In" terminals, serving as adummy load, and 1500 volts applied to thelatter terminal. With no audio signal theresting plate current of the modulator stageshould be approximately 15 milliamperes,kicking up to about 160 milliamperes underfull output conditions. Final adjustment of theclipping control may be made when the modulatoris placed in use with the r-f section ofthe transmitter. Potentiometer R, is then adjustedto limit the peak modulation level undersme wave modulation to approximately 90%.T1DRi~ERSTAGEru11---~""""'""-±V2Figure 19ZERO BIAS TETRODE MODULATORELIMINATES SCREEN AND BIASSUPPLIESTDMODULATORLOADLow driving power and simplicity are keyfeatures of this novel modulator. Tubes rangingin size from 6AQS's to 813's may beemployed in this circuit.T ,-Class B driver transformerT,-Modulation transformerVI, v,-6AQ5, 6L6, 807, 803, 813, etc.R,, R,-Not used with 803 and 813
662 Speech and A. M. Equipment8-WATT ISPEECHAMP.(FIG.72)803'5hMOPERATING CHARACTERISTICS 8+ 2500 v.X XEG-G {RMS):: 170-VOLTS-7DRIVING POWER::: 1-8 WATTSRESTING PLATE CUR.= SOMA.MAX. PLATE CUR.-340 MA. ~ T 3POWER OUTPUT= 510 WATTS (0?501SUPPRESSOR VOLT -:::.280-340 V. ~ ~115 v. '\...Figure 20INEXPENSIVE 500 WATTMODULATOR USING 803 TUBESZs=6 2SKTOMODULATORLOADTJ- 11 Poly-pedanceu Class B driver transformer2: I ratio. Stancor A-476 IT,-500 watt output transformer. IBK primary,6.25K secondary. Chicago CMS-3T,-JO volts, 10 amperes. Stancot C-6461M-0-500 ma.30-8 Zero BiasTetrode ModulatorsClass B zero bias operation of tetrode tubesis made possible by the application of thedriving signal to the two grids of the tubes asshown in figure 19. Tubes such as the 6AQ5,6L6, 807, 803, and 813 work well in thiscircuit and neither a screen supply nor a biassupply is required. The drive requirementsare low and the tubes operate with excellentplate circuit efficiency. The series grid resistorsfor the small tubes are required to balancethe current drawn by the two grids, but are notneeded in the case of the 80 3 and 813 tubes.Of great interest to the amateur is the circuitof figure 20, wherein 803 tubes are usedas high level modulators. These tubes will deliver500 watts of audio in this configuration,yet they require no screen or bias supply, andcan be driven by an 8 watt amplifier stage.The use of 80 3 tubes (in contrast to 813 's)requires a higher level of driving power whichis offset by the fact that these tubes can oftenbe purchased "surplus" for less than four dollars.A pair of 6B4 tubes operating with cathodebias (figure 12) will suffice as a drivingstage for the 803's. The power supply ofthe speech amplifier provides high voltage forthe suppressors of the modulator stage.Shown in figure 21 is a high level modulatorusing 813 tubes. A full 500 watts ofaudi,o may be obtained at 2500 volts plate potential.Grid driving power is 5.5 watts. Asingle 807 operating as a cathode followerat 400 volts will provide sufficient drive forthe modulator stage. Plate to plate load im·pedance for the 813's is not critical. The ChicagoCMS-3 500 watt modulation transformerhaving a primary impedance of 18,000 ohmshas been used with success, although the optimumplate load impedance of the modulatoris closer to 20,000 ohms.TOMODULATORLOADDRIVER STAGE, OPEFI.ATI NG VOLTS,MEASURED TO GROUND.NOTES1-EXACT VALUE OF 807 CATHODE RESISTORDEPENDS UPON RESISTANCE OF PRIMARYWINDING OF T2. ADJUST RESISTOR FORC4THOD£ BIAS OF 26.5 VOLTS 1 PLATE CUR ~RENT OF 53-55 MA.2-RMS OPERATING VOLTAGES AT MAXIMUMOUTPUT SHOWN ON SCHEMATIC.X X + 2500 V.i1 T411.5V. 1'\..Figure 21500 WATT MODULATOR USING 813 TUBEST,-1:3 interstage transformer. Stancor A-53T,-"Poly-pedance" Class B driver transformer. 2:1 ratio. Stancor A-4761T,-500 watt output transformer. IBK primary, 6.25K secondary. Chicago CMS-3T,,-JO volts, 10 amperes, Stancor C-6461M-0-500 ma.350 watts of audio are obtainable from this circuit at plate potential of 2000 volts.
CHAPTER THIRTY-ONEDespite the fact that complete transmittersmay be purchased at moderate cost many amateursstill prefer to build their station equipment.The experience gained in constructingsuch equipment cannot be duplicated by anyother means.Shown in this chapter are two completephone and c.w. transmitters that are well suitedas ··building projects" for the advancedamateur who has had experience building otherequipment. The transmitter designs are uniquein that equipment of this type and powerrating is not readily available on the commercialmarket. The first transmitter covers the 50Me. and 144 Me. VHF bands and is capableof 300 watts input (phone) and 450 wattsinput ( c.w.). The second transmitter coversthe 3.5 Mc.-29.7 Me. amateur bands and runs375 watts input (phone) and 500 watts input( c.w.). Both transmitters are complete, includingmodulators and power supplies and aresystem-designed for ease of operation and reliability.The cost of either unit is quite low,considering the power level involved.31-1 A 300 WattPhone/C-W Transmitterfor 50/144 Me.Transmitters operating in the VHF region presentsome specialized design problems that mustbe solved before reliable, efficient operationcan be obtained. The first problem concernsthe relative inefficiency of tuned and coupledcircuits in this region. The amount of r-f powerthat can be lost in simple circuits is extremelyhigh unless special pains are taken in the designand layout of the VHF stages. Radiation loss,skin effect, dielectric losses, lead inductance,distributed capacity, and parasitic ground returnsall exist at the lower frequencies, but onlyin the higher portion of the spectrum do theireffects assume such vital importance, and onlyat these frequencies does their cure and eliminationbecome so essential. As a result, thenecessity for close correlation between the electricaldesign and the mechanical properties ofthe VHF circuit cannot be overemphasized.663
Figure 1COMPACT300 WATTTRANSMITTER FORSIX AND TWOMETERS IS IDEALFOR VHFENTHUSIASTThe transmitter is builtin twa sections. Upperdeck contains VFO, r.f.stages and push-pull 826amplifier. VFO tuningdial is at right, withbandswitch and meterselector switch below it.At center ore the multipliertuning controls,with buffer and finalamplifier tuning controlsat the left.Lower deck holds modulatorand power supply.Transmitter co n t r o Iswitches ore along loweredge of the panel.The second problem concerns the vacuumtube. A combination of factors is at work inthe common tube which tend to limit the efficiencyof the tube as the frequency of operationis raised. Inter-electrode capacitance, leadinductance, transit time, and input circuit loadingare some of the factors that cripple vacuumtube performance in the VHF region.The drop in tube operation efficiency showsup in the form of plate dissipation. Reducingthe size of the vacuum tube to combat thevarious operational problems reduces its powerhandling capability, since only small elementareas are then present for heat dissipation.Tetrode vs.TriodeThe combination of inter-electradecapacitance and lead inductancein the simple tetrodetube produce a phenomenon called self-neutralization.At some one particular frequencyin or near the VHF region the tetrode tube isinherently self-neutralized due to the circuitelements within the tube structure and the externalinductance of the screen-to-ground path.Below this frequency normal neutralizing proceduresapply; but above this frequency specialneutralizing circuits are required. These circuitsare usually frequency sensitive. The self-neutralizingfrequency of the less expensive screengrid tubes usually falls between 20 - 50 Me.,requiring that the tubes be re-neutralized whenthe frequency of operation is changed from50 Me. to 144 Me., and often when the operatingfrequency is moved about within oneof these bands. The newer external anodetetrodes are free of this problem as their selfneutralizationfrequency falls well above the144 Me. amateur band.The triode tube, on the other hand, has nofrequency of self-neutralization. The maximumfrequency at which neutralization is effectiveis dependent upon the inductance of the tubeleads, the electron transit time, and the interelectrodecapacitance. Rather than a point inthe spectrum at which the neutralization problemsabruptly change, the neutralization pointof the triode tube merely grows more obscureas the operating frequency is raised, untilabove a certain frequency neutralization is nolonger possible. Fortunately, small, compact,inexpensive triode tubes exist that are capableof efficient operation well into the VHF region,and two tubes of this general type areemployed in this transmitter. Special circuitsare also used to ensure that the tubes remain inneutralization over the operating range.TransmitterCircuitryA block diagram of the r-fcircuitry is shown in figure 2.The transmitter is divided intotwo sections. The exciter section ( V1-V,) deliversabout ten watts of power over the frequencyranges of 50- 54 Me. and 144- 148Me. A 5 763 is employed either as a crystalcontrolled oscillator or as a variable frequencyoscillator, operating in the 8 - 9 Me. region.Two separate oscillator tuned circuits are employedas the tuning rates for the two bandsare different. The ''2 meter" VFO covers therange of 8.0 - 8.222 Me., and the "6 meter"VFO tunes 8.333- 9.0 Me. The tuning capacitorsof the two circuits are ganged for tuningease. Tubes V, and V, multiply the overallVFO range to the 48 - 54 Me. spectrum. Atthis point a simple switching system permitsv, to drive either the intermediate amplifier
<strong>HANDBOOK</strong> 50/144 Me. Transmitter 665(8--9MC.)0V46360 (144-148MC.)V1 V2. V3 ~-l5763 (8-0MC.) 5763 (10-18MC.) 6360ve( 144-148MC.) V5(48-54MC;~~ I/ I{j cso-s4Mc.J rr, ..,_,,8.,c.>'-1SOMC. 826 (50-54 MC. ) 9903/5894---,II)/_......::..../..:::.----- ---_JIV6826826V1I1-------- -2- 1 44 M C. _____,®L10(SOMC.)f+---- i 144 MC. ----+!Figure 2BLOCK DIAGRAM OF VHF TRANSMITTERA-General transmitter circuit, showing operating frequencies of the various stages. Y 4 is a plug-infrequency multiplier.B-Quarter-wave tank circuit is employed in amplifier plate configuration at 144 Me. Shorting baris removed for 50 Me., and C,-L, resonates at this frequency.C-Linear half-wave tank section is employed as interstage coupler at 144 Me. Inductor L1o resonatescircuit to 50 Me.©tube y, directly for 50 - 54 Me. operation, orvia a tripler stage (V,) for 144- 148 Me.operation. A 9903/5894 (Vs) is employedas the driver stage £or the push-pull triodefinal amplifier. This driver tube operates as astraight amplifier on both bands.A unique coupling circuit, shown in figure2C, is employed between the 9903/5894 andthe push-pull 826 stage. The plates of thebuffer and grids of the final stage are coupledro the ends of a loaded, half-wave 144 Me.transmission line. The 50 Me. coil is placedat the electrical center of the 144 Me. line.Tuning capacitor c, is tapped on the line at apoint which permits each circuit to be resonated.Thus the combination L,A-B, L10, C, iscapable of tuning the 2 meter and 6 meterranges simultaneously, eliminating the needfor plug-in coils or coil switching. Neutralizationof the 9903/5894 is not required if adequateinter-stage shielding is used.The power amplifier stage employs two type826 tubes in push-pull. These compact triodetubes are well suited to VHF work as theirlead inductance is small and their inter-electrodecapacitance is reasonably low. The tubeshave adequate plate dissipation and a high reserveof filament emission. The amplifier hasbeen run at one kilowatt input ( 2000 voltsat 500 milliamperes) for c.w. operation forextended periods of time without apparentdamage to the tubes, but this practice is notrecommended for maximum tube life.A modified form of two band plate .circuitis employed in the amplifier stage, as shownin figure 2B. A quarter-wave 144 Me. transmissionline is employed, tuned to frequencyby "butter-fly" capacitor Cs. The "cold" endof the line is shorted by a heavy strap whichis removed for 50 Me. operation. At the latterfrequency, resonance is established by capacitorCs and coil L,, which is mounted at the
TO La,(BELOW)(48-54 MC.)-I ...,0:::::lVl3~(8.0-8.222MC.)~( 8 . .3.33- 9. 0 MC.)(SOMC.)144 MC.TRIPLER IBOX-------,..IIIL-.f-B+ 270 V.()0:::::lVl"""' ...,cn6' """':::::lBIAS#2 (-3SV.)(SEE FIG. 8)B+60QV,CzA-B-25- 25 p.p.fd. Bud LC-1661C:, Cs-15- 15 p.p.fd. Bud LC-1660C,, c,-8- 8 p.p.fd. Bud LC-1659C.-10- 10 p.p.fd. "butterfly," Johnson JJMBJJC7-35- 35 p.p.fd. Bud LC-1667Cs--25- 25 p.p.fd. "butterfly," B&W JCX-25E,te stator removed in each sectionBIAS#1 (-60V.)(SEE FIG. 8)V7csi144 MC.SHORTIN
<strong>HANDBOOK</strong>50/144 Me. Transmitter 667Figure 4REAR VIEW OF VHF TRANSMITTERFinal amplifier compartment is at right withantenna receptacles and loading capacitors onrear of enclosure. Ventilation holes are drilledabove 826 tubes to aid air circulation. Entiretop should be replaced with perforated materialif extended periods of operation areencountered. Buffer compartment is to leftof large enclosure, with 9903/5894 tubemounted in inverted position. Grid coil ( Ls,see figure 7) plugs in holder at top of compartment.Grid tuning capacitor is panel drivenby phenolic shalt extension. The 144 Me.multtplier box is in position behind buffercompartmentVariable frequency oscillator is at left ofchassis, with plug-in crystals mounted on theside. Oscillator tube V 1 can be seen in frontof enclosure. Low voltage power supply componentsare in left corner of chassis.B-plus end of the transmissiOn line. The 144Me. linear lines (LnA-B) serve merely as connectionsbetween the tubes and the coil at 50Me. An attempt was made to do away withthe shorting bar, permitting the amplifier tankcircuit to operate in the same manner as thebuffer circuit which requires no componentchanges when the operating frequency ischanged from band to band. It was found thatthe resulting drop in plate circuit efficiencywas so high as to be unacceptable. Wing nutsare therefore provided to permit rapid removalof the shorting bar when 50 Me. operationis desired.AmplifierNeutralizationCircuitBecause of electron transittime, and the combination oflead inductance and interelectrodecapacitance the us-ual cross-neutralization system is not effectiveover the range of 50- 148 Me. The amplifiermay easily be neutralized at either end of thespectrum, but the correct capacitor settings at50 Me. are not the same as the settings for144 Me. In either case, however, one settingof the capacitors is sufficient for the stage toremain in complete neutralization across oneamateur band. For a time the amplifier wasre-neutralized each time the frequency of operationwas shifted from band to band, butthis chore was finally eliminated by the useof the neutralization system shown in the schematicof figure 3. Conventional cross-neutralization(capacitors Cn, Cu) is employed at 144Figure 5CLOSE-UP OF INTERIOR OFVHF FINAL AMPLIFIERTwo 826 triode tubes are used in unique pushpullcircuit for 50 Me. and 144 Me. operation.Linear tank circuit is employed for 144 Me.with removeable shorting bar in foreground."Butterfly 11tuning capacitor is mounted onceramic insulators above linear tank and connectedto it by means of copper straps on rearstator terminals. 50 Me. inductor is placed at"cold" end of 2-meter tank.Air is drown into enclosure through ventilationholes drilled below 826 sockets. Note thatthe sockets are split to reduce possibility offlash-over.At left of enclosure are 6360 (V-3) andsockets SO- 1 and S0-2 for 144 Me. plug-infrequency multiplier. The 50 Me. adapterPL-3 is in Dloce. Oscillator tuning slugs areseen on right side of oscillator compartmentat left. 50 Me. coil is in place atop buffercompartment.
Figure 6UNDER-CHASSISVIEW OF 300WATT VHFTRANSMITTERSmall 115-vo/t blowermotors for 826 tubes arein upper /eft corner ofchassis. Buffer plate lineL,A-8 occupies left-handportion of chassis. Allleads are kept clear ofthis area. Under-chassispower leads are run inshielded braid to reducer-f pick-up. An11L"shaped shield partitionisolates frequency doublerstages from bufferplate line. Capacitors c,and Cs are ganged, asare capacitors c, and C,.Power supply componentsoccupy lower right-handcorner of chassis.The top of the 9903/-5894 buffer tube projectsthrough hole atcenter of chassis.Me., and a simple link neutralization circuit(L,,'. L,,) is employed to complete neutralizationat 50 Me. This double neutralizationciKuit results in absolute stability of operationon both bands. Since the 50 Me. platecoil L,, is decoupled from the 144 Me. tankcircuit by the shorting bar, very little spuriouscoupling exists between the two neutralizingcircuits and adjustments are simple and maybe made once and then fol'gotten.The BandchangingSystemA frequency shift between50 Me. and 144 Me. maybe made in a matter ofminutes. Switch s, selects the correct crystal orVFO range. The plate coil La is broadly resonantover the 8 - 9 Me. range and requiresno adjustment after initial setting. The 5 763and 6360 plate circuits may be resonated nothe correct frequency for either band withoutthe necessity of coil changing. For 144 Me.operation, however, it is necessary to removethe 50 Me. plug, PL-3 (figure 7) and insertin its place the 144 Me. tripler box (figure 3).In addition, grid coil L, of the 9903/5894stage must be changed. The amplifier platecircuit shorting strap is removed for 144 Me.Finally, the buffer and amplifier plate tankcircuits must be resonated to frequency.Separate antenna coupling systems are employedfor each band. Two coupling loops areused, the 144 Me. loop (La) is coupled tothe quarter-wave tank circuit, and the 50 Me.loop ( L1s) is coupled to plate coil L12.Power Supplyand ModulatorThe low voltage supplies arelocated on the rear of the r-fdeck, and are diagrammed infigure 8. A voltage doubler selenium supplyprovides 600 volts for the 9903/5894 stageand 270 volts for the exciter. A second seleniumrectifier supply provides negative biasfor the buffer and final amplifier stages. Isolatingdiodes Do and D, reduce interaction betweenthe two bias voltages.The high voltage plate supply and modulatorare located on a separate deck. The supplyprovides 1500 volts at 500 rna. for the modulatorand final amplifier stage. The modulatorconsists of a pair of 811-A tubes, operatingFigure 7CLOSE UP OF PLUG-INCOMPONENTS144 Me. plug-in tripler box containing 6360(V-4) is at center. Tube is mounted at angleon small plate with ventilation holes drilledin box above top of tube. 9-pin plug PL, ismade from portion of a Vector socket. 50 Me.plug PL, is at upper left, with 144 Me. coilLs directly below it. 50 Me. coil Ls is at right.See figure 12 for coil data.
Figure 8SCHEMATIC, LOW VOLTAGE SUPPLIES AND METER SWITCHING CIRCUITr,-600/300 v., c.t., 360 ma. Use 300 volt taps for voltage doubler service. Triad P-3.T:-125 volts, 50 ma. Stancor PA-8421T,-7.5 volts, 8 a. Stancor P-6138T,-6.3 volts, 6 a. Stancor P-30360,-,-,-,_Se/enium rectifier, 100 ma., RMS input voltage: 175. Maximum peak inverse voltage: 495.Sarkes-Tarzian Model 108o,-,-,-Se/enium rectifier, 75 ma., RMS input voltage: 130. Sarkes-Tarzian Model 758, 8:-Smal/ 115-vo/t blower motor.Sz-2 pole, 6 position rotary switch.S0,-8 prong octal socket (see figures 3 and 10)class B. Any one of the speech amplifiersshown in chapter 30 is suitable for use withthe modulator. The complete schematic of thisdeck is shown in figure 10.High voltage primary circuits are energizedby relay RY, after the plate circuit controlswitch S, is closed. External relay control terminalsare provided for auxiliary circuits.MeteringCircuitsA 0-100 d.c. mHliammeter (M,,f.igure 10) is employed to monitorthe low voltage circuitry. A sixposition switch ( S,, figure 8) places the meteracross 150 ohm shunns in the plate and gridcircuits as shown in the table of figure 8.Meter M, reads plate current of the modulatorand final r-f amplifier. This meter is placedin a high potential circuit, and should have abakelite case and an insulated zero-set control.TransmitterConstructionThe overall layout of the r-fsection can be seen in figures4, 5, and 6. The foundationconsists of a 13" x 1 7" x 3" aluminum chassisfirmly fastened to a 121,4" x 19" relay rackpanel. The oscillator compartment (figure 9)measures 2" x 4" x 6". The grid circuit compartmentfor the 9903/5894 is 3" x 4" x 5"in size, and the plug-,in 144 Me. tripler boxmeasures 2" x 3" x 3Yz". The push-pull 826final amplifier is enclosed in an aluminumbox measuring 6" x 7" x 12". This compartmentis mounted clear of the chassis on Yzinchmetal posts. Care must be taken to seethat each oorner of the box makes a good electricalground with the chassis. The plug-intripler box fits into two tube sockets (SO-land S0-2) placed in line with the extensionshaft that drives buffer grid capacitor Ce. Thespecial sockets with 50 Me. plug PL-3 insertedin SO-l can be seen in figure 5.The 9903/5894 buffer socket is mounted tothe "top" of the grid circuit compartment withthe plate leads projecting out of matching holescut in the "bottom" of the box and the chassis.
670 Transmitter Construction<strong>THE</strong> <strong>RADIO</strong>Figure 9INTERIOR,HIGH STABILITYVFO FOR VHFTRANSMITTERTwo gang oscillator tuningcapacitor C1A-Bispanel mounted ot top olenclosure. Variable trimmingcapacitors are atthe right, with slug-tunedcoils directly below.Bandswitch S1A-B isplaced at bottom ol lrontpanel, with crystal socketsat lelt. The two leadsto oscillator tube socketpass through large rubbergrommet mounted onchassis deck.Alter alignment is completed,a drop ol nailpolish is placed on endol coil slug to eliminatemechanical vibration olslug. Note that rear olcase is bolted in placewith eight sheet metal~~re~~:~~s~~~idos~n;~;~~tube socket is in loreground,partially hiddenby selenium rectifier.Ventilation holes are drilled in the top of thisbox, and grid coil Ls mounts in banana plugsmounted on a polystyrene plate atop the box.Two 826 sockets are mounted at the frontend of the amplifier box on metal spacers.Trouble occurred with the sockets flashingover on occasional modulation peaks. Accordingly,that portion of the socket holding thetwo plate pins was cut away and the pin connectorswere bolted directly to the ends of theplate circuit straps, L11A and L11B. The backof a hacksaw blade and cutting compoundare used to cut grooves across the top andbottom surfaces of the socket. A sharp blowwith a hammer will fracture the socket alongthe groove lines.Ventilation holes are drilled in the box andchassis directly below the tubes and are coveredwith window screen to reduce spurious energyradiation through the openings.The two I Ys" wide aluminum straps thatserve as the 144 Me. tuned circuit are supportedat the "B-plus" end on two solid ceramicinsulators. Two more insulators of the sameheight serve to mount the "butter-fly" capacitorto the box. Small copper strips connect thetank straps to the stator terminals of the variablecapacitor. The 50 Me. coil and the re-Figure 11REAR VIEW,POWER SUPPLY CHASSISPower supply is built upon steel chassis. Powertransformer and modulation transformer areat rear ol chassis. Three 4 Jlfd., 2KV littercapacitors are connected in parallel to provideadequate dynamic: reserve. One capacitoris placed under the chassis. Insulated, highvoltage terminal is employed lor 8-plus lead.Rectifiers are at right, lront ol chassis, withmodulator tubes at lelt-lront.
<strong>HANDBOOK</strong>50/144 Me. Transmitter671866B+TOFINALTOSPEECHAMPLIFIER1010W504TO R.F. DECK, 50301cFigure 10SCHEMATIC, POWER SUPPLY AND MODULATOR DECKr,-1775/1500 each side c.t., 625 ma., ICAS. 115/230 volt primary. Stancor P-8029Tz-2.5 v., 10 a., IOKV insulation. Stancor P-3060T,-6,3 v., 10 a. Stancor P-63081 4 -300 watt "polypedance" modulation transformer. Stancor A-3898. Primary impedance J2K,secondary impedance 6KTs- 11 Polypedance 11 driver transformer. Stancor A-4762RY,-DPDT relay, 10 ampere contacts, J 15 v. a.c. coilM,-0 - 500 ma.M,-o- 100 ma.S 4 A-B-2 pole, 3 position high voltage switch. (Found in "war surplus" BC-306A antenna tuning unit),or equivalentCH,-500 ma. swinging choke, 2-76 h. Stancor C-1405B,-,-,-1 15 volt pilot lampsso,-8 prong octal socketmovable shorting bar are mounted at the "Bplus"end of the assembly. The rotor of the"butter-fly" capacitor is left floating and ispanel driven through a high voltage shaftcoupling. The neutralizing capacitors (Cu, C12)are mounted on a triangular piece of polystyreneor other good insulating material andplaced between the tube sockets. Constructionof these capacitors is shown in figure 13.The 144 Me. antenna link "coil" is a Ushaped pickup loop mounted beneath the lineartank circuit, while the 50 Me. pickup coil isinserted between the turns of plate coil L12.The single turn neutralizing coil 114 is placednear one end of L,, as shown in the photograph.Placement of the major components beneaththe chassis may be seen in figure 6. The lineartank circuit L,A-B runs in zig-zag fashionfrom the plate terminals of the buffer tube V,to two ceramic standoff insulators mounted onthe bottom plate of the power amplifier enclosure.Spacing berween the lines is held constant,and they are supported at the bufferend on two 1" ceramic insulators. The buffertuning capacitor C, is mounted to the chassisberween the lines and panel driven with aphenolic shaft. The rotor of this capacitor isnot grounded. The stators of C, are connectedto the lines with short lengths of flexible strap.Any grid current unbalance in the final amplifierstage can be corrected by varying the
672 Transmitter Construction<strong>THE</strong>RA D I 0FIGURE 12COIL CHART FOR 50-144 MC. TRANSMITTER6-32 BOLTy 'l 1- ( 144 MC. COIL) 36 T.# 24 E., 1/2" OIA. MILLEN 69046 FORM.l2-(50MC. COIL) 28T.#24E., t/2"01A. M/LtEN 69046 FORM.l3- 52 TURNS# 28 E. MILLEN 69046 FORM.L4- 22 T. CENTER TAP, 3/4" Dl A., 32 T. P.l. 8& W 307 2LS-15 T. CENTER TAP, 1/2" DIA., t6T.P.I. 8&W 3003LINK COIL: ONE TURN HOOKUP WIRE.L6-21/2T.CENTERTAP,l/2"01A. 16T.P.I. B&W3003LINK COIL: ONE TURN HOOKUP WIRE.L7-7T.CENTERTAP, 5/B"DIA., t6T.P.r. B&WJ007LINK COIL: ONE TURN HOOKUP WIRE.l8- (50MC.) 7 T, CENTER TAP, 3/4" OIA., 8 T.P.I, B&W 3010LINK COIL: ONE TURN HOOKUP WIRE.( 744 MC.) 3 1/4' X .02." X , 12.5" COP. STRAP BENT "U".(S££ FIG. 7)LINK: 1 TURN HOOKUP WIRE BENT TO FIT LB.l9A-B- II" #-8 COPPER WIRE, OR .125" DIA. ROO, PLUS 3"' OF1/2-"'WIOE FLEXIBLE COPPER STRAP (GRID LEAD) PLUS3/4" FLEXIBLE COPPER STRAP (PLATE LEAD). ADJUSTSPACING FOR RESONANCE. (SEE FIG. 6 FOR LA 'I' OUT)l10-5T.,3/4"DIA.,#14E. WIRE. ADJUST SPACING OF TURNSFOR RESONANCE AT 50 MC. WITH C7 AT FULL CAPACITY.L11A-B-10" OF 1/8" X 1 118" ALUMINUM STRAP, SPACED 3 3/8"CENTER TO CENTER. SHORTING BAR MADE OF SAME MATERIAL, ATTACHED 6 718" FROM PLATE END TO CENTEROF BAR. PLATE PIN RECEPTACLES REMOVED FROM SOCKETAND ATTACHED TO ENDS OF Ll NES (SEE FIGURE 5)l 12- (SO MC. COIL) 6 T., 1 1/4 DIA. # 6 COPPER WIRE, OR .125 11TUBING. SPACE TURNS TO RESONATE TO 50 MC. WITHC8 NEAR FULL CAPACITY.l13-19"' 0F*12.E. WIRE BENT TO "U" SHAPE TO FORM 5• X 2"LOOP WITH 3 1/2." LEADS. PLACE UNDER l11A-B.l14-l TURN, 2" DIA., 10 KV. HOOKUP WIRELIS-1 TURN, 1/2"' DIA. 1tl6 E., WITH INSULATED SLEEVING.PLACE AT CENTER OF l10.l16- 2 TURNS, 1 1/4" 01 A., 10 KV HOOKUP WIRE.position of the taps on the lines until balancedcurrent is obtained. The tuning range of thecapacitor on 144 Me. is reduced by movingthe taps closer to the electrical center of rheline.An "L" -shaped shield plate passes aroundthe edge of tripler capacitor C.-Cs to shield thelow power stages from the buffer plate lines.The under-chassis shielding is completed whena bottom plate is bolted to the chassis.The power supply section occupies the rearportion of the chassis. High voltage rectifiersD,.,.,., are mounted above the deck (figure4) and the diodes of the bias supply aremounted below the chassis.Figure 9 shows the interior of the VFOcompartment. The two oscillator coils aremounted on the side of the box, across fromthe four crystal sockets. The trimmer capacitorsare mounted above their respective coils. Thegrid and cathode leads to oscillator tube V,pass through a lar,ge rubber grommet mountedin the bottom of the enclosure.The Power andModulator DeckThe power s u p p 1 y ismounted upon a steelchassis measuring 13" xA- 3/8"' X 2" CERAMIC INSULATOR, TAPPED AT EACHEND FOR 6-32 MACHINE SCREWS.B,C-1 118"' DIA. BRASS DISC WITH 3/8" CLEARANCE HOLEFOR CERAMIC INSULATOR. LOWER DISC SUPPORTED ON3/8" CERAMIC INSULATORS. USE FLAT HEAD SCREWS.0- 3/8" CERAMIC INSULATORS.E- 1 1/8' 1 DISC MADE OF TWO LAYERS OF .015" THICKTEFLI)N HELD ATOP C BY CELLOPHANE TAPE.F- TWO 6/32 NUTS PLACED UNDER STRAP. STRAP ANDNUTS SOLDERED TO BRASS PLATE, HOLE IN PLATEpASSES 6/32 SOL T.Figure 13CONSTRUCTION OFNEUTRALIZING CAPACITORFOR 826 STAGETwo capacitors are required (C11 and Ca).Capacitors have top brace plate made ofpolystyrene sheet or other good VHF insulatingmaterial. Plate is supported on two ceramicinsulators which form center support forcapacitors. Plates of neutralizing capacitorsare made of brass discs. The grid disc is supportedon short ceramic insulators, and theplate disc is supported from the polystyreneplate by 6-32 machine screw. Plate encirclesceramic insulator with a close fit. Plate leadand two 6-32 nuts are soldered to top plate.Mounting screw passes through this assembly,and also through hole drilled in fop plate.After capacitors have been adjusted, top plateis eemented to center ceram1c insulator withclear nail polish to retain adjustment.17" x 3", as shown in figure 11. The modulatorand rectifier tubes are placed near thefront panel, with the modulation transformerand power transformer to the rear ofthe chassis. The swinging choke is placed betweenthe tubes at the front-center of thechassis. Smaller components are mounted beneaththe chassis. High voltages are applied tothe transmitter by primary relay RY,, controlledby switch S, and an optional externalrelay control circuit.TransmitterAdjustmentThe tuned circuits of the transmittercan be set to their properoperaJting range with theaid of a grid-dip oscillator. All r-f tubes mustbe in their sockets. Coil L, is set to 8.5 Me.,and circuit C-L, is adjusted to 17 Me. withc, set at half-capacity. The 7 p.p.fd. trimmer
<strong>HANDBOOK</strong> De Luxe All-Band Transmitter 673capacitor across L, is adjusted to resonate theplate circuit of the first tripler stage to 50 Me.with Ca at hal£-capaoity, and Cs-L, is tuned tothe same frequency in the same manner with50 Me. plug PL-1 placed in socket SO-l. Thegrid coils of the buffer stage are next adjustedto frequency.The next step is to adjust the plate circuit ofthe buffer stage. Coil Lo is attached to thetransmission line about 2" from the feedthroughterminals, and capacitor C. is tappedon the line about 1" closer to the buffer tubethan is the coil. Resonance of the lines can beadjusted by changing the length of the flexibleplate leads to the buffer tube, by varyingthe spacing of the lines, or by changing thespacing between the chassis and the lines. Thetuning range of C. may be varied by shiftingthe position of the taps on the lines. Coil L,ois placed at the "cold" point of the line, whichmay be found by touching the line with a pencillead when the transmitter is operating.Place the coil at the spot on the line that hasthe least "fire" when the transmitter is operatingon 144 Me.As soon as grid current can be measuredon the power amplifier, the stage may be neutralized.The transmitter is first neutralizedon 144 Me. by adjusting capacitors Cu and C,.The B-plus lead should be removed for thisoperation. Operating frequency is then changedto 50 Me. and link coils L" and Ls are adjustedfor complete neutralization on the six meterband. The 144 Me. neutralization should thenbe rechecked. The links have very little effectat 144 Me., since the shorting bar effectivelyremoves L, and L, from the circuit.However, the adjustment of the capacitors affectsthe setting of the 50 Me. links, so thelatter must be readjusted after each change inthe settings of Cu and C,,.Reduced voltage is now applied to the amplifierstage and it is loaded into a suitable antennasystem. The links of coils L,, Lc. L,, andL should be adjusted for maximum grid drive.Loose coupling should be used as overcouplingreduces excitation to the power amplifier andresults in "double tuning" of the stages. Bufferplate current runs about 100 milliamperes,and each amplifier tube should develop approximately3 5 milliamperes of grid current.The 144 Me. antenna link (L3) is adjustedby varying the length of the loop and its positionto the lines. It is mounted beneath thelines and spaced about %" away from them.Fine antenna loading may be controlled byloading capacitors Co and Co. The amplifiermay be loaded to 200 rna. for phone operation,or 250 rna. for c.w. operation. The overall amplifierefficiency is better than 50%, delivering160 watts to a dummy load with a power inputof 300 watts at 144 Me. Efficiency is betterthan 60% at 50 Me.Modulator plate current rises to about 150rna. on peaks for 100% modulation of the finalamplifier.31-2 A De-LuxeTransmitter for the3.5-29.7 Me. RangeShown in this section is a de-luxe phone/c.w. transmitter for operation in the 80, 40,20, 15, and 10 meter amateur bands. Thetransmitter runs 500 watts input on c.w. and3 7 5 watts input on phone, and is completelyself-contained in two small relay racks as illustratedin figure 14. The r.f. section of thetransmitter is designed to be virtually TVIproof,and modern techniques such as speechFigure 14DE-LUXE STATIONFOR <strong>THE</strong> DX MANEmployes VFO-controlled,bandswitching, Phone-CWtransmitter running up to500 watts.This compact, all-bandtransmitter employs a7094 tetrode in a pi-networkamplifier, modulatedby 811-A's. Clickless 1break-in keying is useafor c.w., and speechclipping and limiting areemployed for AM phone.Modulator and powersupply are in the rackat left, and entire r-fsection is in ri~tht·handrack. High stab11ity VFOensures drift-free frequencycontrol. Transmitteris designed for maximumharmonic suppressionand is virtually TVIproof
674 Transmitter Construction <strong>THE</strong> <strong>RADIO</strong>Vi V3 V4 Vs ANT.Vs12AX7(3.5-30MC.)tetL--------~--- -- __ _JV712AU7V12811-AV96AL5V1012AU7V11(3.5-30 ;aIelV66AL5MIC.L.V.V14082V~>082V1s2021V19666-A115/230V."vFigure 15BLOCK DIAGRAM OF ALL-BAND 500 WATT TRANSMITTERR-f section employs bandswitching exciter for coverage of 10, 15, 20, 40, and 80 meter bands. Variablefrequency oscillator operates in 1.7 Me. region to achieve maximum isolation, Tuning and antennaloading are aided by 6ALS vacuum tube voltmeter monitoring output of amplifier stage.Low level speech clipping and filtering, combined with high level audio filter imports "punch" to phonesignal, yet holds sidebands to a minimum. Single high voltage supply powers modulator and final amplifier.dipping and dickless, break-in keying are mcorporated in the transmitter.A high stability variable frequency oscillatoris employed for frequency control, the dialof which is directly calibrated for each band.Band changing is a simple operation and maybe accomplished in a few seconds.Transmitter Circuitryand LayoutA block diagram of thetransmitter is given infigure 15. Nineteentubes are used in the equipment: seven in ther-f section, six in the audio section, and s1x mthe power supply.The RCA 7094 beam power tube is employedin the final amplifier stage. This compacttube has high perveance and high powergain. It can be operated at full input to 60Me., and has a maximum plate dissipation of125 watts. In addition, it has triple base-pinconnections for the screen grid to permit goodr-f grounding and large plate radiating finsfor effective cooling. The compact size makesit especially effective in the high frequencyportions of the communication spectrum. Drivingrequirements are modest and permit theuse of a simplified bandswitching exciter.Figure 16VFO-EXCITER DECK CONTAINSALL IMPORTANT OPERATINGCONTROLSLaw level r-f stages are built upon 8"xl7"x2"aluminum chassis. Keyer timing potentiometerR, is at left, with excitation potentiometerR,, contra/ switch s,, and bandswitch s, inline, left ta right. VFO dial is centered on thepanel.Output connector P 1 is mounted on polystyrener,late atop shielded enclosure at right. Shieldngis made from Reynold's Do-it-yourself 1111perforated aluminum sheet.
I - - - - - - - - -v,- - - -IV2.6AC7V42E26L-:fcII- -- --1--.".-~!'.-----'1',,-1§_0_1_-- J4--t---t--' PL 1+300V.2KiOW50K 7K4W tOWMIN.--- MA.X.GAl D DRIVE+R·F OUTPUTTOP-A:X>z0m00;ill;6.3V.V712AU715KSTANDBY(B)R 2 ~-o---vl VIMI-"IOJ0:::::1o._-I .....0:::::1Vl3
Figure 18INTERIOR OF VFOSHOWINGPLACEMENT OFMAJOR.COMPONENTSThe VFO is built upontwo !Is-inch dural plates,one forming the baseand the other formingthe front of the oscillatorenclosure. Inductor L,and padding capacitorsare mounted to frontwall, and the variabletuning capacitor is boltedto base. Oscillatortube (V-1) projects fromfar wall of enclosure toremove as much heatfrom tuned circuit aspossible. Entire VFO ismounted upon miniatureshock mounts and dial Isattached to VFO unit,rather than to rackpanel. 6AG7 and 2E26tubes are at /eft, with2E26 plate choke and5KV coupling capacitorin foreground.The variable frequency oscillator and exciterstages are built upon one chassis deck, asshown in figures 16, 18, 19, and 20. Thecomplete schematic of this unit is given infigure 17. The variable oscillator covers therange of 1.75 Me.- 1.85 Me., permitting fullcoverage of all amateur bands except the extremetop of the 28 Me. band, and the topportion of the 80 meter band. Readjustmentof the oscillator padding capacitor providescomplete coverage of these segments, if desired.The VFO consists of a 6U8 pentodetriode( V1) operating as a "hot cathode" oscillatorand triode cathode follower. Extremelyhigh-C is employed in the tuned circuit of theVFO to obtain a good order of stability. A2000 /L/Lfd. precision ceramic capacitor Ca,forms the large portion of the tuning capacitance,and the coil L, is wound upon a ceramicform having a very low temperature coefficient.The cathode follower stage providesexcellent circui't isolation for the oscillator,combined with a minimum of plate circuitloading. Figure 18 shows the layout of the majorosciHator components. The circuit is builtupon an Ys" thick aluminum plate, bolted toa small panel of the same material. The sides,back, and top of the oscillator enclosure aremade of aluminum sheet. The complete unitis mounted upon miniature rubber shockmounts. The VFO dial is firmly affixed to theenclosure and actually does not make physicalcontact with the panel at all. The frequencydetermining unit is thereby protected from joltsand jars which might cause a "warble" in thetransmitter frequency. The r-f output from thecathode follower is 5 volts, r.m.s.Three tubes are employed in the multiplierand d1:1iver stages of the transmitter. The first6AC7 doubler stage ( V,) operates into abroadly resonant slug-tuned coil L,. Whentuning the exciter unit, this coil is peaked at3.6 Me. The stage will then deliver substantiallyconstant ou~put over the mnge from 3.2to 4.0 Me. The 2E26 stage (V.) operates asan amplifier on the 3.5 Me. band and as adoubler on the 7 Me. band, driven directlyfrom the 6AC7 buffer-doubler.Figure 19REAR VIEW OF EXCITER DECKMultiplier tubes are in the foreground. Adjustmentholes lor the multiplier trimmer capacitorsare on side of chassis. !12 -inch anglestock is run around chassis and panel to formfoundation to which the perforated enclosureis attached. Keyer tube V, is placed in rearcorner. Potentiometer R, and auxiliary keyjack are on chassis wall below keyer tube.Short lead extends from 2E26 blocking capacitorup to connecting plug P, mounted onenclosure.
De Luxe All-Band Transmitter 677For operation on 20, 15, and 10 meters,switch S,D disconnecrs the 2E26 from the6AC7 stage, and connects it to an auxiliary6AG7 frequency multiplier (Va). The multiplieroperates as either a doubler, a tripler, oras a quadvupler. Its output circuit, therefore, iseither tuned to 7 Me., 10.5 Me., or 14 Me.Coil La in the plate circuit of the 6AG7 resonateswith residual circuit capacities to 14Me. Switch section S,C of the bandswitch addsadditional capacity to lower the resonant frequencyof this circuit to 10.5 Me. or 7 Me.The 2E26 stage operates as a doubler to 14Me., receiving 7 Me. excitation from the 6AG7stage. When the 6AG7 is tuned to 10.5 Me.,the 2E26 serves as a doubler to the 21 Me.band. For 28 Me. output from the 2E26, 14Me. excitation is received from the 6AG 7stage, and the 2E26 again functions as adoubler.V F 0 >------.... 4 "+-Figure 20UNDER-CHASSIS VIEW OF EXCITERAll power wiring is run in shielded braid, grounded tothe chassis at convenient points. Voltage regulatortube V" is mounted beneath the chassis on smallangle brackets. RY, and RY, are mounted to the innerchassis wall, directly behind keyer panel control R,.R-t circuits are at the opposite end of chassis, groupedaround bandswitch s,. The slugs of coils L, and L, areadjustable from atop the chassis. Metal plate coversbottom of chassis to reduce spuriou$ radiation fromr.t. circuits.Keying andControl CircuitA simplified schematic of thekeying and control circuitfor the exciter is shown infigure 21. A break-in type keyer is employedfor c.w. operation. The unit is completely actuatedby the transmitter key. When the key ispressed the transmitter is energized and maybe keyed in a normal manner. The keyer"holds" the transmitter power supplies onwhile the key is manipulated. When the operatorstops, the keyer turns off the transmitterafter a short pause. Thus, the transmitter remainson between keyed characters and words,47K~~~+300V.V26AC7B+ TO VFO~ lT-- + 1.50 V. REc;.l-c7094 SCREEN CIRCUIT (SEE FIG. 23)SWITCH 52100L528 A= TRANSMIT RY5B =STANDBYC=TUNE}TO METER-105 v.NOTES'RELAY CONTACTS RY3ANORMALLY OPEN.RELAY CONTACTS RY3BNORMALLY CLOSED.Figure 21KEYER AND CONTROL CIRCUITSClickless, break-in keying is provided by this simple control circuit. See text tor full details of operation
678 Transmitter Construction<strong>THE</strong> <strong>RADIO</strong>Figure 22500 WATT POWERAMPLIFIER HASSYMMETRICALPANEL LAYOUTThe 7094 amplifier stageIs enclosed in a separatedeck. Plate milliammeteris at lett, with 0·1 d.c.milliammeter at rightwhich may be inserted invarious power leads inexciter or amplifier.Large center knobs are(lett) pi-network switchand (tight) main amplifiertuning control.Controls across the bottomare (lett to right):Pi-network loading capacitor,auxiliary pi-networkcapacitor switch,grid circuit tuning ca·pacitor, and meterswitch. Pi-network switch5 4 is ganged with gridturret switch s, eliminatingone panel control.but will return to a stand-by position after thekeying action has been completed. The "turnoff"time may be varied to suit the operator'staste. For phone operation, the time delay circuitis eliminated, and the push-to-talk circuitactuates the keyer and control relays directly.The whole sequence may be over-ridden andmanual operation restored by means of a "manual-automatic"switch on the power supplychassis. Finally, a "transmit- tune- standby"switch S,A removes screen voltage from thefinal amplifier stage for tune-up purposes.Referring to figure 21, the sequence of operationis as follows:1-Key up, switch S,B on "transmit."The right-hand section of the keyer tubeV, is cut off by the negative bias on thecathode. Potentiometer R, is adjusted for1 rna. current through 15K resistor toground ( -15 volts to chass,is at point A).6AC7 ( y,) is cut off.2-Key is depressed.Right- hand section of V, conducts heavilyand cathode voltage rises quickly, cuttingoff left section of V, and removing blockingbias of 6AC7 ( v,). Plate current ofright-hand section of v, operates RY, andcharges 40 fLfd. capacitor through 1N92diode. The relay closes, placing B-plusvoltage on oscillator stage and operatingthe high voltage primary circuit relay circuit.This circuit may be rendered inoperativeon "standby" position by switch S,B,enabling VFO to be used for frequencycheck or "spotting" purposes.3-Key is released .Right-hand section of V, cuts off immediately.Left-hand section draws 1 rna. andcuts off 6AC7 buffer tube. The 1N92 isback-biased by rise in 12AU7 plate voltage,keeping charge on 40 fLfd. capacitor.Relay R y, remains closed until capacitoris discharged through shunting resistances.Relay then opens, turning off high voltagesupplies of transmitter. "Release" delay iscontrolled by potentiometer R,_4-Phone OperationSwitch s, on modulator deck is closed forphone operation, placing the coil of relayR y, in the circuit. When the key is closed,or the press-to-talk circuit activated, R YsAshorts out the time delay circuit of RY,,and at the same time R YsB removes ashort across the screen modulation chokein the final amplifier.Exciter plate voltage is obtained from themodulator power supply (See Chapter 30,figure 17), and bias and amplifier voltage arederived from the main power supply, locatedon a separate chassis. The plate relay RY, maybe actuated by switch s, or by the keyer circuit.
<strong>HANDBOOK</strong> De Luxe All-Band Transmitter 679V50017094 SKY L5 J2.J1~? G~x.. ~1 i7ER (o}--...,---_,.--t---
680 Transmitter Construction<strong>THE</strong> <strong>RADIO</strong>Figure 24REAR OBLIQUE VIEW OFPI-COUPLED AMPLIFIERDural angle strips are mounted around edgeof 8"xJ7"x2" aluminum chassis and panel towhich TV/ enclosure is attached with selltappingsheet metal screws. Meters are encasedin aluminum cups, with r.f. filtersmounted on the back of each case. Meterleads are 10 KV television cable, enclosed inshielded braid. The 7094 tetrode is mountedat rear of chassis, with neutralizing capacitorc, to /eft, and plate choke at right. Connectionsto plate of tube are made with flexiblecopper strap.Filament transformer is at /eft of chassis, withblower motor hidden behind it. Output capacitorof pi-network is below deck.plate circuit of the amplifier, employing oneof the new Barker & Williamson 500 wattband-switching assemblies. A two-gang broadcastcapacitor is employed for the low impedanceoutput loading control. Switch S, allowsthe insertion of an extra 1000 fLI..Jd. capacitorfor operation into low impedance loadsat 3.5 Me. The output circuit of the networkis monitored by a 6AL5 vacuum-tube voltmeter(Vo) which may be employed duringtune-up and loading adjustments.All power leads beneath the chassis areshielded in flexible braid, grounded at convenientpoints to the chassis. The meters areenclosed in spun aluminum cups that are boltedto the panel of the sta.ge. Each meter leadis bypassed to the chassis with a .001, 5KVceramic capacitor and a VHF c,hoke is placedin series with the meter terminal. A small 115-volt blower motor is mounted at one end ofthe chassis (figure 25) to direct a coolingmovement of air across the envelope of thetube.Placement of the major components beneaththe chassis may be seen in figure 26. Analuminum partition separates the pi-networkcomponents from the tube socket and the gridcircuit. The grid coil switch ( s,) is driven bymeans of a dial cord and drum from the platebandswitch inductor shaft ( L) atop the chassis.The cord passes through two holes in thechassis deck. The main bands1witch control onthe amplifier panel thus actuates both thegrid and plate switches. Safety relay RY, ismounted on the rear apron of the chassis, andan aluminum plate covers the complete underchassisarea. A one-inch hole is cut in thisplate directly below the stator terminal of gridtuning capacitor C, and a banana jack ismO'Ilnted to a poly/ityrene plate bolted inplace over the hole. This forms the low capacityinput terminal, ],. A mating hole is cut inthe top of the exciter screen and covered witha second polystyrene plate having a bananaplug mounted in the center. When the amplifieris slipped in the rack the plug and jackmate, fovming the connection between the2E26 plate and the tuned tank in the gridcircuit of the final amplifier.Power Supplyond ModulatorThe power supply is builtupon a separate chassis, asshown in figure 27. Two866A rectifier tubes are employed with achoke input filter to supply 1500 volts at500 milliamperes for the r-f amplifier andmodulator. "Hash" suppression chokes areplaced in the rectifier plate leads to reduce thehigh frequency "buzz" sometimes super-imposedupon the carrier by merccury vapor tubes.Sufficient capacity must be used in the filtersystem to afford good dynamic stability to thesupply under keyed loads, or under conditionsFigure 25LEFT OBLIQUE VIEW OFPI-COUPLED AMPLIFIERBlower motor to cool 7094 tube is locatedbetween front panel and filament transformer.High voltage and coaxial antenna terminalsare on right of rear wall of chassis. Additionof perforated screen makes enclosure radiation-proof.
<strong>HANDBOOK</strong>of heavy modulation. Eight 60 f.tfd., 450 volttubular electrolytic capacitors are connected inseries parallel to form a 30 f.tfd., 1800 voltunit. A small bias supply delivering -105volts for the 7094 amplifier and keyer is alsoincluded on the chassis. Layout and wiring ofthe supplies are shown in figures 27 and 29.The modulator has been previously shownin Chapter 30, figure 17. The speech amplifiersupply provides 3 70 volts for exciter operation.One section of the "phone-c.w." relay (RYsA)shorts out the time delay circuit on the exciterdeck for phone operation, and a high voltagerelay R y, is employed to remove the modulatorfor c.w. operation. Six volts is also suppliedto the exciter for filament service. Thisfilament circuit is balanced to ground to reduceresidual hum in the speech amplifier, sothe ground circuit cannot be employed as afilament return in the exciter.Exciter Tune-upand OperationEach deck of the transmittershould be tested as an individualunit before the transmitteris tested in entirety. Screen voltage ofthe 2E26 stage should always be set at zeroby means of potentiometer R, (figure 1 7)when the exciter is operated without the finalamplifier, since the 2E26 plate circuit is incomplete,and excessive screen current will bedrawn by the 2E26 unless the, screen potentialis removed.Oscillator calibration may be adjus,ted byvarying the spacing of the turns of coil L, orFigure 27REAR VIEW OF POWER SUPPLYPrimary power leads pass through bulkheadtypefilter capacitors to prevent radiation ofTVI-producing harmonics from power line.Voltage r~'!.'i~i!ofn ct:,~~r ~~r ch':fs~~s. supply isFigure 26UNDER-CHASSIS VIEW OFPOWER AMPLIFIERAll power wiring beneath the chassis is placedwithin braid to reduce pickup of r.f. energy.Note that shield divides area into two sections.Smaller area contains output capacitor of pinetwork,6ALS diode voltmeter and high voltageleads. Larger area contains grid circuitcomponents and meter switch. Sensitive relayRYs is mounted to wall of the chassis behindthe grid tuning capacitor. Three .001, 5 Kv.ceramic screen bypass capacitors are mountedclose to socket terminals. Grid turret switch isdriven from plate turret atop chassis.by changing the value of the auxiliary paddingcapacitor c,. Alignment of the 6AC7 and6AG 7 stages is done by observation of thegrid current of the 2E26 stage, as measuredacross shunt resistor R, in the grid return circuit.This circuit is not connected to the meterswitch, since it requires only one preliminaryreading during initial tune-up. Coil L, ispeaked at about 3.6 Me. with padding capacitorabout half meshed. If grid current to the2E26 is excessive, the 15K resistor shuntedacross coil L, may be reduced in value. Gridcurrent should remain between 2 and 3 milliamperesacross any band.After 80 meter alignment is completed,bandswitch S, is set to the 10 meter position.The slug of coil L, is adjusted for maximumgrid current across the 28 Me. band. Coil L,resonates to 14 Me. for this operation. Thebandswitch is next set to the 15 meter posi,tionand the VFO tuned to 21.2 Me. Padding ca-
682 Transmitter Construction<strong>THE</strong> <strong>RADIO</strong>866-A CHt3S Ksow866-A~sow+-------(] B-2701 N92 2 w 082-105 v.SOe20 K..,-w} 6.3 II.GNO24} 11511./"VRY4AUTOMArtC} TO RELAY CONTROLFigure 28SCHEMATIC, TRANSMITTER POWER SUPPLYc,, C.-Bulkhead type, 0.1 Jlfd. 600 volt capacitor. Sprague "Hypass," or equivalentCH,-10 henry at 300 ma. Chicago R-103RFC-"Hash" suppression choke. Millen 77866RY 4-DPDT relay, 20 amp. contacts, 115 v. a-c coilSO,....-Eight pin receptacle, Cinch-JonesT,-1710/1430 each side c.t., 300 ma., CCS. Chicago P-15121 2-2.5 v., 10 a., 10 KV insulation. Stancor P-3060Ts-125 v., 50 ma., 6.3 v., 2a. Stancor PA-8421pacitor C is adjusted for correct grid currentacross the 21 Me. band. As a last step, thebandswitch is set to 20 meters, the VFO tunedto 14.2 Me. and capacitor Co adjusted for uniformgrid drive across the 20 meter band.The consistency of grid drive across thevarious amateur bands is controlled by theshunt resistors placed across coils L, and L,.The lower the value of these resistors, the moreuniform is the grid drive across the variousbands. At the same time, the grid current ofthe 2E26 stage drops as these resistors arelowered in value. The values given in the schematicgive good results on all bands, sincethe grid current of the 2E26 may vary overa 2: 1 ratio across an amateur band with littlechange in output from the stage.Keying of the exciter is smooth and clickless.Buffer keying is controlled by the adjustmentof potentiometer R,. The buffershould turn on before the 2E26 stage ,and remainon after the 2E26 has been cut-off. Timedelay before the transmitter is turned off afterkeying is controlled by potentiometer R,. Currentflowing through relay coil RY, may bevaried by changing the two plate load resistorsof the right-hand section of keyer tubeV,. The plate load resistor is made up of a15K and a 10K resistor. The 10K resistor maybe removed and a 10K relay coil may be insertedin its place if an extra relay is desiredfor operation of auxiliary circuits. Spring tensionof relay RY, may be varied for optimumrelay operation after the circuit has been adjusted.Exciter keying should be relatively soft asthe addition of the amplifier stage will sharpenit to some extent. The 0.1 JLfd. waveformingcapacitor in the bias return lead of the2E26 may be increased in value to soften theexciter keying to compensate for the sharpeningaction of the class-C amplifier followingthe keyed stage. In actual practice, the keyingof the exciter should be adjusted so that dotsrun together when sent on a bug at a rate ofabout 25 w.p.m.The keyer may be disabled if it is desiredto make adjustments to the r.f. section of theexciter. Keyer tube V, is removed from thesocket and pins 1 and 7 are grounded. RelayRY, is then held closed by placing a piece ofcardboard under the back contacts.
<strong>HANDBOOK</strong> De Luxe All-Band Transmitter 683PL7PL4R.F. AMPLIFIERMODULATOR12345678 1 2 3 4 5 6 7 8 9 10 11 12I'.,.'-Yl I~ ~ ? ? ? 'Il't.!~~li>--t-',---;::f-o I1~~k~ f- __, 43r __,2P~E 1 RSUPPL yPL 8I·Ib ·I I A A ,I1 2. 3 4 s e 1 8 9 10 11 1213 14 15EXCITERPleEXCITERPls® NOTE: ;a~o~~Af-t~ir~:~~~:':i;:::r:REGROUND LEAD.@S££FIG.17{ ~Y2AND 21. S2B RY3SEE FIG. 28~SOe~·--I1SOs12 7 10PLs I Y Y~'I~Figure 29UNDER-CHASSIS LAYOUT OFPARTS IN POWER SUPPLYThe 866-A sockets are sub-mounted on !lrinch ceramic insulators. Eight electrolytic capacitorsare mounted on phenolic plate andbanded with insulated strap to form highvoltage filter capacitor. Bias transformer isnext to rectifier tube sockets. All high voltageconnections are made with S KV televisiontypewire.PreliminaryAmplifierAdjustmentThe amplifier stage may beneutralized by supplying excitationto the input circuit untilrectified grid current flows. Ajumper is placed across the antenna outputjack ], and plate tuning capacitor Co is resonatedto frequency. Neutralizing capacitor C,is now adjusted until the tuning of the platecapacitor has no effect upon the reading ofthe rectified grid current.Preliminary tune-up into an antenna maybe done with the control swi~ch ( s,, figure 17)in the "stand-by" position. Screen voltage isremoved from the final amplifier screen circuit.Also, screen voltage cannot be applied to thetube unless relay RY, (fig. 23) is closed byapplying plate voltage to the amplifier stage.PHONE-CW SWITCH S 1s ££F/G.17c HAP.30SECTION A- CONTROLS RY3(PA. SCREEN RElAY& rtMEDEl.AY)SECiiON B- CONTROLS RY1(MOD. B+ SHORTING RELAY)SECTION C- CONTROLSMODULATOR FIL. CIRCUITSECTION 0 ~CONTROLS 8+TO SPEECH AMPLI Fl E RCantral Cablesand Pawer LeadsPL41Figure 30INTERCONNECTINGCABLES FORTRANSMITTERSEEFIG.\ 'l yb rl[=0PL7After testing is completed,the units should be placedin the racks or cabinets andcontrol cables made up as shown in figure 30.Wire sizes should be observed, as several of theleads carry large values of filament current. Inaddition, a separate grounding lead should berun between the individual chassis as a safetymeasure. Do not depend upon a metal relayrack for a ground return. All external cablesshould be laced for neat appearance. A typicallayout of this equipment is shown in figure 14.
CHAPTER THIRTY-TWOIn view of the high cost of iron-core componentssuch as go to make up the bulk of apower supply, it is well to consider carefullythe design of a new or rebuilt transmitter interms of the minimum power supply requirementswhich will permit the desired performanceto be obtained from the transmitter. Carefulevaluation of the power supply requirementsof alternative transmitter arrangementswill permit the selection of that tran~mitterarrangement which requires the minimum ofpower supply components, and which makesmost efficient use of such power supplies asare required.32-1 Power SupplyRequirementsA power supply for a transmitter or for aunit of station equipment should be designedin such a manner that it is capable of deliveringthe required current at a specified voltage,that it has a degree of regulation consistentwith the requirements of the application, thatits ripple level at full current is sufficientlylow for the load which will be fed, that its internalimpedance is sufficiently low for the684job, and that none of the components shall beoverloaded with the type of operation contemplated.The meeting of all the requirements of theprevious paragraph is not always a straightforwardand simple problem. In many casescompromises will be involved, particularlywhen the power supply is for an amateur stationand a number of components already onhand must be fitted into the plan. As muchthought and planning should be devoted to thepower-supply complement of an amateur stationas usually is allocated to the r-f and a-fcomponents of the station.The arrival at the desig;n for the power supplyfor use in a particular application may bestbe accomplished through the use of a seriesof steps, with reference to the data in thischapter by determining the values of componentsto be used. The first step is to establishrhe operating requirements of the powersupply. In general these are:1. Output voltage required under full load.2. Minimum, normal, and peak output current.3. Voltage regulation required over the currentrange.
Requirements 685Figure 1POWER SUPPLYCONTROL PANELA well designed supply controlpanel has separate primaryswitches and indicator lampsfor the filament and plate circuits,overload circuit breaker,plate voltage control switchand primary circuit fuses.4. Ripple voltage limit.5. Rectifier circuit to be used.The output voltage required of the powersupply is more or less established by the operatingconditions of the tubes which it will supply.The current rating of the supply, however,is not necessarily tied down by a particulartube combination. It is always best to designa power supply in such a manner that it willhave the greatest degree of flexibility; thisprocedure will in many cases allow an existingpower supply to be used without changeas a portion of a new transmitter or other itemof station equipment. So the current rating ofa new power supply should be established bytaking into consideration not only the requirementsof the tubes which it immediately willfeed, but also with full consideration of thebest matching of power supply components inthe most economical current range which stillwill meet the requirements. It is often longruneconomy, however, to allow for any likelyadditional equipment to be added in the nearfuture.Current-RatingConsiderationsThe mtmmum current drainwhich will be taken from apower supply will be, in mostcases, merely the bleeder current. There aremany cases where a particular power supplywill always be used with a moderate or heavyload upon it, but when the supply is a portionof a transmitter it is best to consider the mini-mum drain as that of the bleeder. The minimumcurrent drain from a power supply is ofimportance since it, in conjunction with thenominal voltage of the supply, determines theminimum value of inductance which the inputchoke must have to keep the voltage from soaringwhen the external load is removed.The normal current rating of a power supplyusually is a round-number value chosen on thebasis of the transformers and chokes on handor available from the catalog of a reliable manufacturer.The current rating of a supply tofeed a steady load such as a receiver, a speechamplifier, or a continuously-operating r-f stageshould be at least equal to the steady drain ofthe load. However, other considerations comeinto play in choosing the current rating for akeyed amplifier, an amplifier of SSB signals,or a clas,s B modulator. In the case of a supplywhich will feed an intermittent load suchas these, the current racings of the transformersand chokes may be less than the maximumcurrent which will be taken; but the currentratings of the rectifier tubes to be used shouldbe at least equal to the maximum current whichwill be taken. That is to say that 300-ma.transformers and chokes may be used in thesupply for a modulator whose resting currentis 100 ma. but whose maximum current atpeak signal will rise to 500 ma. However, therectifier tubes should be capable of handlingthe full 500 rna.
686 Power SuppliesThe iron-core components of a power supplywhich feeds an- intermittent load may be chosenon the basis of the current as averagedover a period of several minutes, since it isheating effect of the current which is of greatestimportance in establishing the ratings ofsuch components. Since iron-core componentshave a relatively large amount of thermal inertia,the effect of an intermittent heavy currentis offset to an extent by a key-up periodor a period of low modulation in the case of amodulator. However, the current rating of arectifier tube is established by the magnitudeof the emission available from the filament ofthe tube; the maximum emission must not beexceeded even for a short period or the rectifiertube will be damaged. The above considerationsare predicated, however, on the assumptionthat none of the iron-core components willbecome saturated due to the high intermittentcurrent drain. If good-quality components ofgenerous weight are chosen, saturation willnot be encountered.Voltage The general subject of voltageRegulation regulation can really be dividedinto two sub-problems, which differgreatly in degree. The first, and morecommon, problem is the case of the normalpower supply for a transmitter modulator,where the current drain from the supply mayvary over a ratio of four or five to one. Inthis case we desire to keep the voltage changeunder this varying load to a matter of 10 or15 per cent of the operating voltage under fullload. This is a quite different problem fromthe design of a power supply to deliver somevoltage in the vicinity of 250 volts to an oscillatorwhich requires two or three milliamperesof plate current; but in this latter casethe voltage delivered to the oscillator must beconstant within a few volts with small variationsin oscillator current and with large variationsin the a-c line voltage which feeds theoscillator power supply. An additional voltageregulation problem, intermediate in degree betweenthe other two, is the case where a loadmust be fed with 10 to 100 watts of power ata voltage below 500 volts, and still the voltagevariation with changes in load and changesin a-c line voltage must be held to a fewvolts at the output terminals.These three problems are solved in the normaltype of installation in quite different manners.The high-power case where output voltagemust be held to within 10 to 15 per cent isnormally solved by using the proper value ofinductance for the input choke and proper<strong>THE</strong> <strong>RADIO</strong>value of bleeder at the output of the powersupply. The calculations are simple: the inductanceof the power-supply input choke atminimum current drain from the supply shouldbe equal in henries to the load resistance onthe supply (at minimum load current) dividedby 1000. This value of inductance is calledthe critical inductance and it is the minimumvalue of inductance which will keep the outputvoltage from soaring in a choke-inputpower supply with minimum load upon theoutput. The minimum load current may bethat due to the bleeder resistor alone, or itmay be due to the bleeder plus the minimumdrain of the modulator or amplifier to whichthe supply is connected.The low-voltage low-current supply, such aswould be used for a v.f.o. or the high-frequencyoscillator in a receiver, usually is regulatedwith the aid of glow-discharge gaseousregulatortubes. These regulators are usuallycalled "VR tubes." Their use in various typesof power supplies is discussed in Section 32-7.The electronically-regulated power supply,such as is used in the 10 to 100 watts poweroutput range, also is discussed later on in thischapter and examples are given.Ripple The ripple-voltage limitationConsiderations imposed upon a power supplyis determined by the loadwhich will be fed by the supply. The tolerableripple voltage from a supply may vary fromperhaps 5 per cent for a class B or class Camplifier which is to be used for a c-w stageor amplifier of an FM signal down to a fewhundredths of one per cent for the plate-voltagesupply to a low-level voltage amplifier ina speech amplifier. The usual value of ripplevoltage which may be tolerated in the supplyfor the majority of stages of a phone transmitteris between 0.1 and 2.0 per cent.In general it may be stated that, with 60-cycle line voltage and a single-phase rectifiercircuit, a power supply for the usual stages inthe amateur transmitter will be of the chokeinputtype with a single pi-section filter followingthe input choke. A c-w amplifier orother stage which will tolerate up to 5 per centripple may be fed from a power supply whosefilter consists merely of an adequate-size inputchoke and a single filter capacitor.A power supply with input choke and asingle capacitor also will serve in most casesto feed a class B modulator, provided the outputcapacitor in the supply is sufficiently large.The output capacitor in this case must becapable of storing enough energy to supply the
<strong>HANDBOOK</strong>peak-current requirement of the class B tubeson modulation peaks. The output capacitor forsuch a supply normally should be between 4JLfd. and 20 JLfd.Capacitances larger than 20 JLfd. involve ahigh initial charging current when the supplyis first turned on, so that an unusually largeinput choke should be used ahead of the capacitorto limit the peak-current surge throughthe rectifier tubes. A capacitance of less than4 JLfd. may reduce the power output capabilityof a class B modulator when it is passing thelower audio frequencies, and in addition maysuperimpose a low-frequency "growl" on theoutput signal. This growl will be apparent onlywhen the supply is delivering a relatively highpower output; it will not be present when modulationis at a low level.When a stage such as a low-level audio amplifierrequires an extremely low value of ripplevoltage, but when regulation is not of importanceto the operation of the stage, the highdegree of filtering usually is obtained throughthe use of a resistance-capacitance filter.This filter usually is employed in addition tothe choke-capacitor filter in the power supplyfor the higher-level stages, but in some caseswhen the supply is to be used only to feedlow-current stages the entire filter of the powersupply will be of the resistance-capacitancetype. Design data for resistance-capacitan~efilters is given in a following paragraph.When a low-current stage requires very lowripple in addition to excellent voltage regulation,the power supply filter often will endwith one or more gaseous-type voltage-regulatortubes. These VR tubes give a high degreeof filtering in addition to their voltage-regulatingaction, as is obvious from the fact thatthe tubes tend to hold the voltage drop acrosstheir elements to a very constant value regardlessof the current passing through the tube.The VR tube is quite satisfactory for improvingboth the regulation and ripple characteristicsof a supply when the current drain willnot exceed 25 to 35 ma. depending upon thetype of VR tube. Some types are rated at amaximum current drain of 30 ma. while othersare capable of passing up to 40 ma. withoutdamage. In any event the minimum currentthrough the VR tube will occur when the associatedcircuit is taking maximum current.This minimum current requirement is 5 ma.for all types of gaseous-type voltage-regulatortubes.Other types of voltage-regulation systems,in addition to VR tubes, exhibit the addedTO FULL- WAVERECTIF'IER~= 2>000 CAPAC~~NCE, CMT 1--f6lJfTO FULL-WAVERECTIFIERTO FULL-WAVERECTIFIERRequirements 687RIPPLE IN TERMS OF C AT FULL LOADFIGURE 2PERCENT Rl PPLE13.18.56.24.0RIPPLE IN TERMS OF LOAD RESISTANCELOAD, OHMS2~000 (BLEEDER ONLY) 0.0215000 0.0425000 10000 0.065000 0. I3000 0.172000 0.25FIGURE 3PERCENT RIPPLERIPPLE IN TERMS OF Ct AND C2 AT FULL LOADC 1C2 PERCENT Rl PPLE2 2 1.23 2 0. 725000 4 4 0.258 0.06FIGURE4characteristic of offering a low value of rippleacross their output terminals. The electronic-typeof voltage-regulated power supplyis capable of delivering an extremely smallvalue of ripple across its output terminals,even though the rectifier-filter system aheadof the regulator delivers a relatively highvalue of ripple, such as in the vicinity of 5 to10 per cent. In fact, it is more or less selfevident that the better the regulation of sucha supply, the better will be its ripple characteristic.It must be remembered that the rippleoutput of a voltage-regulated power supply ofany type will rise rapidly when the load uponthe supply is so high that the regulator beginsto lose control. This will occur in a supply ofthe electronic type when the voltage ahead ofthe series regulator tube falls below a valueequal to the sum of the minimum drop acroosthe tube at that value of current, plus the outputvoltage. In the case of a shunt regulatorof the VR-tube type, the regulating effect willfail when the current through the VR tubefalls below the usual minimum value of about5 ma.Calculation Although figures 2, 3 and 4of Ripplegive the value of ripple voltagefor several more or lessstandard types of filter systems, it is often ofvalue to be able to calculate the value of ripplevoltage to be expected with a particularset of filter components. Fortunately, the approximateripple percentage for normal values
688 Power Supplies<strong>THE</strong> <strong>RADIO</strong>TO F"ULL-WAVERECTIFIER8 HY12 HYFigure 5SAMPLE FILTER FORCALCULATION OF RIPPLEof filter components may be calculated withthe aid of rather simple formulas. In the twoformulas to follow it is assumed that the linefrequency is 60 cycles and that a full wave ora full-wave bridge rectifier is being used. Forthe case of a single-section choke-input filteras illustrated in figure 2, or for the ripple atthe output of the first section of a two-sectionchoke input filter the equation is as folJows,118Per cent ripple=--LC-1where LC is the product of the input chokeinductance in henrys (at the operating currentto be used) and the capacitance which followsthis choke expressed in microfarads.In the case of a two-section filter, the percent ripple at the output of the first section isdetermined by the above formula. Then thispercentage is multiplied by the filter reductionfactor of the following section of filter. Thisreduction factor is determined through the useof the following formula:LC-1Filter reduction factor = ---1.76Where LC again is the product of the inductanceand capacitance of the filter section.The reduction factor will turn out to be a decimalvalue, which is then multiplied by the percentageripple obtained from the use of thepreceding formula.As an example, take the case of the filterdiagrammed in figure 5. The LC product of thefirst section is 16. So the ripple to be expectedat the output of the first section will be: 118/(16-1) or 118/15, which gives 7.87 per cent.Then the second section, with an LC productof 48, will give a reduction factor of: 1.76/(48-1) or 1.76/47 or 0.037. Then the ripplepercentage at the output of the total filter willbe: 7.87 times 0.03 7 or slightly greater than0.29 per cent ripple.Resistance·CapacitanceFiltersIn many applications where thecurrent drain is relatively small,so that the voltage drop acrossthe series resistors would not beexcessive, a filter system made up of resistorsand capacitors only may be used to advantage.In the normal case, where the reactance of theshunting capacitor is very much smaller thanthe resistance of the load fed by the filter system,the ripple reduction per section is equalto 11 (21rRC). In terms of the 120-cycle ripplefrom a full-wave rectifier the ripple-reductionfactor becomes: 1.33/RC where R is expressedin thousands of ohms and C in microfarads.For 60-cycle ripple the expression is: 2.66/RCwith R and C in the same quantities as above.Filter System Many persons have noticed,Resonance particularly when using an inputchoke followed by a 2-pJd.first filter capacitor, that at some value ofload current the power supply will begin tohum excessively and the rectifier tubes willtend to flicker or one tube will seem to takeall the load while the other tube dims out. Ifthe power supply is shut off and then againstarted, it may be the other tube which takesthe load; or first one tube and then the otherwill take the load as the current drain isvaried. This condition, as well as other lessobvious phenomena such as a tendency for thefirst filter capacitor to break down regardlessof its voltage rating or for rectifier tubes tohave short life, results from resonance in thefilter system following the high-voltage rectifier.The condition of resonance is seldom encounteredin low-voltage power supplies sincethe capacitors used are usually high enoughso that resonance does not occur. But in highvoltagepower supplies, where both choke inductanceand filter capacitance are more expensive,the condition of resonance happensfrequently. The product of inductance and capacitancewhich resonates at 120 cycles is1. 77. Thus a 1-JLfd. capacitor and a 1. 77 henrychoke will resonate at 120 cycles. In almostany normal case the LC product of any sectionin the filter system will be somewhat greaterthan 1.77, so that resonance at 120 cycles willseldom take place. But the LC product forresonance at 60 cycles is about 7 .1. This is avalue frequently encountered in the input sectionof a high-voltage power supply. It occurswith a 2-JLfd. capacitor and a choke which has3.55 henrys of inductance at some currentva~ue. With a 2-JLfd. filter capacitor followingthis choke, resonance will occur at the currentvalue which causes the inductance of the choketo be 3.55 henrys. When this resonance doesoccur, one rectifier tube (assuming mercury-
<strong>HANDBOOK</strong>Rectification Circuits 689vapor types) will dim and the other will becomemuch brighter.Thus we see that we must avoid the LCproducts of 1.77 and 7.1. With a swinging-typeinput choke, whose inductance varies over a5-to-1 range, we see that it is possible forresonance to occur at 60 cycles at a low valueof current drain, and then for resonance to occurat 120 cycles at approximately full loadon the power supply. Since the LC productmust certainly be greater than 1. 77 for satisfactoryfiltering along with peak-current limitationon the rectifier tubes, we see that witha swinging-type input choke the LC productmust still be greater than 7.1 at maximum currentdrain from the power supply. To allow areasonable factor of safety, it will be well tokeep the LC product at maximum current drainabove the value 10.From the above we see that we can just getby with a '2-f.tfd. first capacitor following theinput choke when this choke is of the moreor-lessstandard 5-25 henry swinging variety.Some of the less expensive types of chokesswing from about 3 to 12 henrys, so we seethat the first capacitor must be greater than 2f.tfd. to avoid resonance with these chokes. A3-f.tfd. capacitor may be used if available, buta standard 4-f.tfd. unit will give a greater safetyfactor.32-2 RectificationCircuitsThere are a large number of rectifier circuitsthat may be used in the power supplies for stationequipment. But the simpler circuits aremore satisfactory for the power levels up tothe maximum permitted the radio amateur. Figure6 shows the three most common circuitsused in power supplies for amateur equipment.Half-WaveRectifiersA half-wave rectifier, as shown infigure 6A, passes one half of thewave of each cycle of the alternatingcurrent and blocks the other half. Theoutput current is of a pulsating nature, whichcan be smoothed into pure, direct current bymeans of filter circuits. Half-wave rectifiersproduce a pulsating current which has zerooutput during one-half of each a-c cycle; thismakes it difficult to filter the output properlyinto d.c. and also to secure good voltage regulationfor varying loads.Full-Wave A full-wave rectifier consists ofRe~:tifiers a pair of half-wave rectifiersworking on opposite halves of thecycle, connected in such a manner that eachE o.c.= 0.4.5 EA. c.I A. c.= I D.c.TE D.C.lE D.C.= 0.9 E A.C.I A.c.=o.7o7lo.c.Eo.c.= o.9 EA.c.[ A.C =I D.C.Figure 6MOST COMMON RECTIFIER CIRCUITS(A) shows a half-wave reetifier circuit, (B) isthe standard full-wave rectifier circuit usedwith a dual rectilier or two rectifier tubes,and (C) is the bridge rectifier circuit.half of the rectified a-c wave is combined inthe output as shown in figure 7. This pulsatingunidirectional current can be filtered toany desired degree, depending upon the particularapplication for which the power supplyis designed.A full-wave rectifier may consist of twoplates and a filament, either in a single glassor metal envelope for low-voltage rectificationor in the form of two separate tubes, each havinga single plate and filament for high-voltagerectification. The plates are connected acrossthe high-voltage a-c power transformer winding,as shown in figure 6B. The power transformeris for the purpose of transforming the110-volt a-c line supply to the desired second-
690 Power Supplies<strong>THE</strong> <strong>RADIO</strong>f\1\PvTRANSFORMER:V\ [\ [\] v vSECONDARYVOLTAGERECTIFIED VOL TAC;EPLATE N°1~IFI£JJ VOLTAGEPLATE N• 2tVV\/VVV'\COMBINEDRECTIFI£0VOLTAGEPLATES N°1 &. 2.0~--~--~--~--~L---~--~+~SMOO<strong>THE</strong>D\IOLTAGEMTER FIRST SECTIONOF FILTER0+~-------------------------D.C. VOLTAGEA VAl LABLE FOR<strong>RADIO</strong> USEFigure 7FULL-WAVE RECTIFICATIONShowing transformer secondary voltage, therectified output of each tube, the combinedoutput of the rectifier, the smoothed voltageafter one section of filter, and the substantiallypure d.c. output of the rectifier-filterafter additional sections of filter.ary a-c voltages for filament and plate supplies.The transformer delivers alternating currentto the two plates of the rectifier tube; oneof these plates is positive at any instant duringwhich the other is negative. The centerpoint of the high-voltage transformer windingis usually grounded and is, therefore, at zerovoltage, thereby constituting the negative Bconnection.While one plate of the rectifier tube is conducting,the other is inoperative, and vice versa.The output voltages from the rectifier tubesare connected together through the commonrectifier filament circuit. Thus the plates alternatelysupply pulsating current to the output(load) circuit. The rectifier tube filamentsor cathodes are always positive in polaritywith respect to the plate transformer in thistype of circuit.The output current pulsates 120 times persecond for a full-wave rectifier connected toa 60-cycle a-c line supply; hence the output ofthe rectifier must pass through a filter tosmooth the pulsations into direct current. Filtersare designed to select or reject alternatingcurrents; those most commonly used in a-cpower supplies are of the low-pass type. Thismeans that pulsating currents which have afrequency below the cutoff frequency of thefilter will pass through the filter to the load.Direct current can be considered as alternatingcurrent of zero frequency; this passesthrough the low-pass filter. The 120-cycle pulsationsare similar to alternating current incharacteristic, so that the filter must be designedto have a cutoff at a frequency lowerthan 120 cycles (for a 60 cycle a-c supply).Bridge The bridge rectifier (figure 6C)Rectification is a type of full-wave circuitin which four rectifier elementsor tubes are operated from a single high-voltagewinding on the power transformer.While twice as much output voltage can beobtained from a bridge rectifier as from a center-tappedcircuit, the permissible output currentis only one-half as great for a given powertransformer. In the bridge circuit, four rectifiersand three filament heating transformerwindings are needed, as against two rcx:tifiersand one filament winding in the center-tappedfull-wave circuit. In a bridge rectifier circuit,the inverse peak voltage impressed on any onerectifier tube is halved, which means that tubesof lower peak inverse voltage rating may beused for a given voltage output.32-3 Standard PowerSupply CircuitsChoke input is shown for all three of thestandard circuits of figure 6, since choke inputgives the best utilization of rectifier-tube andpower transformer capability, and in additiongives much better regulation. Where greateroutput voltage is a requirement, where the loadis relatively constant so that regulation is notof great significance, and where the rectifiertubes will be operated well within their peakcurrentratings, the capacitor-input type of filtermay be used.The capacitor-input filter gives a no-loadoutput voltage equal approximately to the peakvoltage being applied to the rectifier tubes.At full-load, the d-e output voltage is usuallyslightly above one-half the secondary a-c voltageof the transformer, wi,th the normal valuesof capacitance at the input to the filter. Withlarge values of input capacitance, the outputvoltage will run somewhat higher than ther-m-s secondary voltage applied to the tubes,but the peak current flowing through the rectifiertubes will be many times as great as thed-e output current of the power supply. Thehalf-wave rectifier of figure 6A is commonlyused with capacitor input and resistance-capacitancefilter as a high-voltage supp.ly for acathode-ray tube. In this case the current drain
<strong>HANDBOOK</strong>Standard Circuits 691] llf3~:::~=@ HALF AND FULL VOLTAGE BRIDGE POWER SUPPLY@ TWO VOLTAGE BRIDGE POWER SUPPLYto•o.e~00•o.e~z© TWO TRANSFORMER POWER SUPPLY@ CENTER TAPPED METHOD FOR UNTAPPED TRANSFORMERS®TWO VOLTAGE POWER SUPPLY® SPECIAL FILTER CIRCUIT FOR BRIDGE RECTIFIERFigure 8SPECIAL SINGLE-PHASE RECTIFICATION CIRCUITSA description of the application and operation of each of these special circuits isnccompanying text.given in theis very small so that the peak-current ratingof the rectifier tube seldom will be exceeded.The circuit of figure 6B is most commonlyused in medium-voltage power supplies sincethis circuit is the most economical of filamenttransformers, rectifier tubes and sockets, andspace. But the circuit of figure 6C, commonlycalled the bridge rectifier, gives better transformerutilization so that the circuit is mostcommonly used in higher powered supplies.The circuit has the advantage that the entiresecondary of the transformer is in use at alltimes, instead of each side being used alternatelyas in the case of the full-wave rectifier.As a point of interest, the current flow throughthe secondary of the plate transformer is a substantiallypure a-c wave as a result of bettertransformer utilization, instead of the pulsatingd-e wave through each half of the powertransformer secondary in the case of the fullwaverectifier.The circuit of figure 6C will give the greatestvalue of output power for a given transformerweight and cost in a single-phase powersupply as illustrated. But in attempting tobridge-rectify the whole secondary of a transformerdesigned for a full-wave rectifier, inorder to obtain doubled output voltage, makesure that the insulation rating of the transformerto be used is adequate. In the bridgerectifier circuit the center of the high-voltagewinding is at a d-e potential of one-half thetotal voltage output from the rectifier. In anormal full-wave rectifier the center of thehigh-voltage winding is grounded. So in thebridge rectifier the entire high-voltage secondaryof the transformer is subjected to twicethe peak-voltage stress that would exist if thesame transformer were used in a full-wave rectifier.High-quality full-wave transformers willwithstand bridge operation quite satisfactorilyso long as the total output voltage from thesupply is less than perhaps 4500 volts. Butinexpensive transformers, whose insulationis just sufficient for full-wave operation, willbreak down when bridge rectification of theentire secondary is attempted.Special SinglePhose RectificationCircuitsFigure 8 shows six circuitswhich may provevaluable when it is desiredto obtain more than
692 Power Supplies <strong>THE</strong> <strong>RADIO</strong>PRIMARY=Eo= 1.11 EsIs = 0.517 I D.C.RIPPLE FREQUENCY= 3FRIPPLE PERCENT= 18.3~~~~ ~"c,vL~~~~) = ~:: ~~@ 6-PHASE STAREo= 1.35 EsIs = 0.408 I o.c.RIPPLE FREQUENCY = IFRIPPLE PERCENT= 4.2~~:: ~~~~::: ) = ::~: ~~Figure 9COMMONPOLYPHASERECTIFICATIONCIRCUITSThese circuits are usedwhen polyphase power isavailable for the platesupply of a high-powertransmitter. The circuitat (B) is also called athree-phase full-waverectification system. Thecircuits are described inthe accompanying text.Eo= 2.34 EsIs = o.a1e I o.c.Rl PPLE FREQUENCY = IFRIPPLE PERCENT= 4.2~~~~ ~~~~~!: ) = ~:~! ~~© 6-PHASE BRIDGEone output voltage from one plate transformeror where some special combination of voltagesis required. Figure SA shows a more or lesscommon method for obtaining full voltage andhalf voltage from a bridge reCtification circuit.With this type of circuit separate input chokesand filter systems are used on both outputvoltages. If a transformer designed for usewith a full-wave rectifier is used in this circuit,the current drain from the full-voltagetap is doubled and added to the drain from thehalf-voltage tap to determine whether the ratingof the transformer is being exceeded. Thusif the transformer is rated at 1250 volts at 500rna. it will be permissible to pull 250 rna. at2500 volts with no drain from the 1250-volttap, or the drain from the 1250-volt tap maybe 200 rna. if the drain from the 2500-volttap is 150 rna., and so forth.Figure SB shows a system which may beconvenient for obtaining two voltages whichare not in a ratio of 2 to 1 from a bridge-typerectifier; a transformer with taps along thewinding is required for the circuit however.With the circuit arrangement shown the voltagefrom the tap will be greater than one-halfthe voltage at the top. If the circuit is changedso that the plates of the two rectifier tubesare connected to the outside of the windinginstead of to the taps, and the cathodes of theother pair are connected to the taps insteadof to the outside, the total voltage ouput ofthe rectifier will be the same, but the voltageat the tap position will be less than half thetop voltage.An interesting variable-voltage circuit isshown in figure SC. The arrangement may beused to increase or decrease the output voltageof a conventional power supply, as representedby transformer T1, by adding anotherfilament transformer to isolate the filamentcircuits of the two rectifier tubes and addinganother plate transformer between the filamentsof the two tubes. The voltage contribution ofthe added transformer T, may be subtractedfrom or added to the voltage produced by T1
<strong>HANDBOOK</strong>Standard Circuits 693simply by reversing the double-pole doublethrowswitch S. A serious disadvantage of thiscircuit is the fact that the entire secondarywinding of transformer T, must be insulatedfor the total output voltage of the power supply.An arrangement for operating a full-waverectifier from a plate transformer not equippedwith a center tap is shown in figure 8D. Thetwo chokes L, must have high inductance ratingsat the operating current of the plate supplyto hold down the a-c current load on thesecondary of the transformer since the totalpeak voltage output of the plate transformeris impressed across the chokes alternately.However, the chokes need only have half thecurrent rating of the filter choke L, for a certaincurrent drain from the power supply sinceonly half the current passes through eachchoke. Also, the two chokes L, act as inputchokes so that an additional swinging chokeis not required for such a power supply.A conventional two-voltage power supplywith grounded transformer center tap is shownin figure 8E. The output voltages from thiscircuit are separate and not additive as inthe circuit of figure 8B. Figure SF is of advantagewhen it is desired to operate Class Bmodulators from the half-voltage output of abridge power supply and the final amplifierfrom the full voltage output. Both L, and L,should be swinging chokes but the total drainfrom the power supply passes through L whileonly the drain of the final amplifier passesthrough L,. Capacitors C and c, need be ratedonly half the maximum output voltage of thepower supply, plus the usual safety factor.This arrangement is also of advantage in holdingdown the "key-up" voltage of a c-w transmittersince both L and L, are in series, andtheir inductances are additive, insofar as the"critical inductance" of a choke-input filteris concerned. If 4 fLfd. capacitors are used atboth C and C, adequate filter will be obtainedon both plate supplies for hum-free radiophoneoperation.PolyphaseRectificationCircuitsIt is usual practice in commercialequipment installationswhen the power drain from aplate supply is to be greaterthan about one kilowatt ~o use a polyphase rectificationsystem. Such power supplies offerbetter transformer utilization, less ripple outputand better power factor in the load placedupon the a-c line. However, such systems requirea source of three-phase (or two-phasewith Scott connection) energy. Several of themore common polyphase rectification circuitswith their significant characteristics areshown in figure 9. The increase in ripple frequencyand decrease in percentage of rippleis apparent from the figures given in figure 9.The circuit of figure 9C gives the best transformerutilization as does the bridge circuitin the single-phase connection. The circuithas the further advantage that there ,is no averaged-e flow in the transformer, so that threesingle-phase transformers may be used. A tapat half-voltage may be taken at the junctionof the star transformers, but there will be d-eflow in the transformer secondaries with thepower supply center tap in use. The circuit offigure 9A has the disadvantage that there is anaverage d-e flow in each of the windings.Rectifiers Rectifying elements in high-voltageplate supplies are almost invariablyelectron tubes of either the high-vacuumor mercury-vapor type, although seleniumor silicon rectifier stacks containing a largenumber of elements are often used. Low-voltagehigh-current supplies may use argon gasrectifiers ( Tungar tubes), selenium rectifiers,or other types of dry-disc rectification elements.The xenon rectifier tubes offer someadvantage over mercury-vapor rectifiers forhigh-voltage applications where extreme temperatureranges are likely to be encountered.However, such rectifiers (3B25 for example)are considerably more expensive than theirmercury-vapor counterparts.Peak Inverse Plate In an a-c circuit, the maxi-Voltage and PeakPlate Currentmum peak voltage or currentis V 2 or 1.41 timesthat indicated by the a-cmeters in the circuit. The meters read the rootmean-square( r.m.s.) values, which are thepeak values divided by 1.41 for a sine wave.If a potential of 1,000 r.m.s. volts is obtainedfrom a high-voltage secondary windingof a transformer, there will be 1 ,410-volts peakpotential from the rectifier plate to ground. Ina single-phase supply the rectifier tube hasthis voltage impressed on it, either positivelywhen the current flows or "inverse" when thecurrent is blocked on the other half-cycle. Theinverse peak voltage which the tube will standsafely is used as a rating for rectifier tubes.At higher voltages the tube is liable to arcback, thereby destroying or damaging it. Therelations between peak inverse voltage, totaltransformer voltage and filter output voltagedepend upon the characteristics of the filterand rectifier circuits (whether full- or halfwave,bridge, single-phase or polyphase, etc.).
694 Power Supplies1 [j~ ~+ 0._______.._]=L_.~R=LINEVOLTS-HEATERVOLTSHEATER AMPERES...._:__. -LINE RECTIFIER®SELENIUMLINE· RECTIFIER=0-, --ic__+~--~-c_,....._~::OLTAG~DOUBLER- + FULL-WAVEI C2 -'--------'- +Cl®SELENIUMRECTIFIERVOLTAGEQUADRUPLERFigure 10TRANSFORMER LESS POWER-SUPPLYCIRCUITSCircuits such as shown above are also frequentlycalled line-rectifier circuits. Seleniumrectifiers, vacuum diodes, or gas diodesmay be used as the rectifying elements inthese circuits.Rectifier tubes are also rated in terms ofpeak plate current. The actual direct load currentwhich can be drawn from a given rectifiertube or tubes depends upon the type of filtercircuit. A full-wave rectifier with capacitor inputpasses a peak current several times thedirect load current.In a filter with choke input, the peak currentis not much greater than the load currentif the inductance of the choke is fairly high(assuming full-wave rectification).A full-wave rectifier with two rectifier elementsrequires a transformer which deliverstwice as much a-c voltage as would be thecase with a hlf-wave rectifier or bridge rectifier.Mercury-VaporRectifier Tubes<strong>THE</strong> <strong>RADIO</strong>The inexpensive mercury-vaportype of rectifier tube isalmost universally used inthe high-voltage plate supplies of amateur andcommercial transmitters. Most amateurs arequite familiar with the use of these tubes butit should be pointed out that when new orlong-unused mercury-vapor tubes are firstplaced in service, the filaments should be operatedat normal temperature for approximatelytwenty minutes before plate voltageis applied, in order to remove all traces ofmercury from the cathode and to clear anymercury deposits from the top of the envelope.After this preliminary warm-up with a newtube, plate voltage may be applied within 20to 30 seconds after the time the filamentsare turned on, each time the power supplyis used. If plate voltage should be appliedbefore the filament is brought to full temperature,active material may be knocked fromthe oxide-coated filament and the life of thetube will be greatly shortened.Small r-f chokes must sometimes be connectedin series with the plate leads of mercury-vaporrectifier tubes in order to preventthe generation of radio-frequency hash. Theser-f chokes must be wound with sufficientlyheavy wire to carry the load current and musthave enough inductance to attenuate the r-fparasitic noise current to prevent it from flowingin the filter supply leads and then beingradiated into nearby receivers. Manufacturedmercury-vapor rectifier hash chokes are availablein various current ratings from the ]amesMillen Company in Malden, Mass., and fromthe ]. W. Miller Company in Los Angeles.When mercury-vapor rectifier tubes are operatedin parallel in a power supply, smallresistors or small iron-core choke coils shouldbe connected in series with the plate lead ofeach tube. These resistors or inductors tendto create an equal division of plate current betweenparallel tubes and prevent one tube fromcarrying the major portion of the current.When high vacuum rectifiers are operated inparallel, these chokes or resistors are not required.TransformerlessPower SuppliesFigure 10 shows a group offive different types of transformerlesspower supplieswhich are operated directly from the a-c line.Circuits of the general type are normally foundin a.c.-d.c. receivers but may be used in lowpoweredexciters and in test instruments. Whencircuits such as shown in (A) and (B) areoperated directly from the a-c line, the rec-
<strong>HANDBOOK</strong>tifier element simply rectifies the line voltageand delivers the alternate half cycles of energyto the filter network. With the normal typeof rectifier tube, load currents up to approximately75 ma. may be employed. The d-evoltage output of the filter will be slightlyless than the r-m-s line voltage, dependingupon the particular type of rectifier tube employed.With the introduction of the miniatureselenium rectifier, the transformerless powersupply has become a very convenient sourceof moderate voltage at currents up to perhaps500 ma. A number of advantages are offeredby the selenium rectifier as compared to thevacuum tube rectifier. Outstanding amongthese are the factors that the selenium rectifieroperates instantly, and that it requires noheater power in order to obtain emission. Theamount of heat developed by the selenium rectifieris very much less than that produced byan equivalent vacuum-tube type of rectifier.In the circuits of figure 10 (A) , (B) and(C), capacitors C and C, should be ratedat approximately 150 volts and for a normaldegree of filtering and capacitance, should bebetween 15 to 60 JLfd. In the circuit of figure10D, capacitor C should be rated at 150volts and capacitor c, should be rated at 300volts. In the circuit of figure 10E, capacitorsC and C, should be rated at 150 volts andc, and C should be rated at 300 volts.The d-e output voltage of the line rectifiermay be stabilized by means of a VR tube.However, due to the unusually low internalresistance of the selenium rectifier, transformerlesspower supplies using this type of rectifyingelement can normally be expected togive very good regulation.Voltage-Doubler Figures 10C and 10D illusCircuits trate two simple voltagedoublercircuits which willdeliver a d-e output voltage equal approximatelyto twice the r-m-s value of the power linevoltage. The no-load d-e output voltage isequal to 2.82 times the r-m-s line voltagevalue. At high current levels, the output voltagewill be slightly under twice the line voltage.The circuit of figure 1 OC is of advantagewhen the lowest level of ripple is requiredfrom the power supply, since its ripple frequencyis equal to twice the line frequency.The circuit of figure 10D is of advantage whenit is desired to use the grounded side of thea-c line in a permanent installation as the returncircuit for the power supply. However,with the circuit of figure 10D the ripple frequencyis the same as the a-c line frequency.Standard Circuits 695OUTS I DE COLLECTORINSIDE COLLECTORPHENOLIC WASHERBASE PLATE® SELENIUM RECTIFIER CELL100,.---9080~ 70)-u 50zw 40u 300::LL 20w , 000®/IY/VIf-fJ PHASE ~r' PHASE50 100 150 200 2.50RELATIVE LOAD CURRENT,PERCENT OF FULL LOAD.Figure 11<strong>THE</strong> SELENIUM RECTIFIERA-The selenium rectifier is a semi-conductorstack built up of nickel plated aluminumdiscs coated on one side with seleniumalloy.8-Rectifier efficiency is high, reaching 70%for single phase service, dropping slightlyat high current densities.VoltageQuadrupler300The circuit of figure 10E illustratesa voltage quadrupler circuitfor miniature selenium rectifiers.In effect this circuit is equivalent totwo voltage doublers of the type shown in figure10D with their outputs connected in series.The circuit delivers a d-e output voltage underlight load approximately equal to four timesthe r-m-s value of the line voltage. The noloadd-e output voltage delivered by the quadrupleris equal to 5.66 times the r-m-s linevoltage value and the output voltage decreasesrather rapidly as the load current is increased.In each of the circuits in figure 10 whereselenium rectifiers have been shown, conventionalhigh-vacuum rectifiers may be substitutedwith their filaments connected in seriesand an appropriate value of the line resistoradded in series with the filament string.32-4 Selenium andSilicon RectifiersSelenium rectifiers are characterized by longlife, dependability, and maintenance-free operationunder severe operating conditions. The
696Power Supplies<strong>THE</strong> <strong>RADIO</strong>f-~- 6~~ 40(/) 2Zf-- ...J 0wo~> -2
<strong>HANDBOOK</strong>Mobile Power Supply 697from 600 to 900 amperes per square inch ofeffective barrier layer. The usable current densitydepends upon the general construction ofthe unit and the ability of the heat sink toconduct heat from the crystal. The small sizeof the crystal is illustrated by the fact that arectifier rated at 15 amperes d.c., and 150amperes peak surge current has a total cellvolume of only .00023 inches. Peak currentsare extremely critical because the small massof the cell will heat instantaneously and couldreach failure temperatures within a time lapseof microseconds. The assembly of a typicalsilicon cell is shown in figure 13.OperatingCharacteristicsThe reverse direction of a siliconrectifier is characterizedby extremely high resistance,up to 10" ohms below a critical voltage point.This point of avalanche voltage is the regionof a sharp break in the resistance curve, followedby rapidly decreasing resistance (figure15A). In practice, the peak inverse workingvoltage is usually set at least 20% below theavalanche point to provide a safety factor.The forward direction, or direction of lowresistance determines the majority of powerloss within the semi-conductor device. Figure15 B shows the static forward current characteristicsversus applied voltage. The thresholdvoltage is about 0.6 volts.Since the forward resistance of a semi-conductoris very low, any unbalance betweenthreshold voltages or internal voltage dropwould cause serious unbalance of load distributionand ultimate failure of the overloadedsection. A small resistance should thereforebe placed in series with each half wave sectionFigure 14MINIATURE SEMI-CONDUCTORTYPE RECTIFIERRaytheon CK-777 power rectifier bolts tochassis to gain large "heat sink" area. Lowinternal voltage drop and high efficiencypermit small size of unit.operating in parallel to balance the load currents.Some interesting and practical semi-conductorpower supplies are shown in figure 16.Remember that the circuits of figure 16A andB, and those of figure 10 are "hot" with respectto one side of the power line.32-5 100 Watt MobilePower Supplywwa::u ~ 1ci~Wf>z~w1.
698 Power Supplies,.4---IfL-...--*L-----o + 20~ v.1$0MA.DUAL VOLTAGE DOUBLERWITH COMMON B-MINUS+32~V.450MA.@ HIGH CURRENT SUPPLY© 900 WATT HIGH VOLTAGE SUPPLYFigure 16SEMI-CONDUCTOR POWER SUPPLIESA-Voltage quadruplet circuit. II point "A" is taken as ground instead of point "8," supply will deliver530 volts at 150 mo. from 115 volt a-c line. Supply is "hot" to line.B-Voltage triplet delivering 325 volts at 450 mo. Supply is "hot" to line.C-900 watt supply lor sideband service may be made from two voltage quadruplers working in seriesfrom inexpensive "distribution-type" transformer. Supply features good dynamic voltage regulation.PARTS LIST:o,_sarkes Tarzian Model 150 selenium cell or Model M-500 silicon cell.Dz-Sarkes Tarzian Model 500 selenium cell or Model M-500 silicon cell.T1-Power distribution transformer, used backwards. 230/460 primary, 115/230 secondary, 0.75 KVA.Chicago PCB-24750.ments, delivering 100 watts at various voltagesto run both the mobile transmitter and receiver.The input power required by the- supplyis almost directly proportional to the outputpower drain, and only a small amount ofpower is wasted to actuate the vibrator reeds.The large demand for efficient and powerfulmobile radio equipment has led to thedevelopment of new heavy-duty, vibrator-typepower supply components; these are used toadvantage in this unique design.The Split-ReedVibratorThe new split-reed dual-interruptervibrator overcomes thepower capacity limitations ofthe older type vibrators. In addition, this vibratorpermits the design of power suppliesrequiring no component changes for operationfrom either the 6- or 12-volt d.c. power systemswith which most automobiles areequipped.Until recently, the majority of vibrator supplieshave been designed around the synchonoustype vibrator. A simplified diagram ofsuch a vibrator is shown in figure 18A. Oneset of vibrator contacts switches the batterycurrent alternately through two opposed primarywindings of a transformer, thus inducinga square-wave a.c. voltage in the secondary.This a.c. secondary voltage is then rectified by asecond set of contacts on the vibrator armature.The limitations of this circuit are lowpower handling capacity and the need to usea different number of turns on the transformerprimary when the battery voltage is changedfrom six to twelve volts.In the split-reed vibrator (figure 18B), twosets of double-throw contacts are electricallyisolated from each other. Each set has muchgreater current carrying capacity than thesynchronous vibrator. However, a power transformerhaving two center-tapped primarywindings is required. One set of contactsswitches the d.c. power alternately betweenhalves of one primary winding, and the otherset simultaneously switches the other winding.Therefore, the primaries can be connected inparallel for 6-volt operation, or in series for12-volt operation. No wiring changes areneeded in the supply if the battery is properlyconnected to the power input terminals.The Selenium A selenium rectifier systemRectifier System is employed in this supply.Since the vibrator contactsopen and close abruptly, the periodically in-
Figure 17100 WATT MOBILEPOWER SUPPLYThis high efficiency mobilesupply features 6 orJ 2 volt input, and willcompletely power boththe mobile transmitterand receiver. 450, 300,and 240 volts are available.At right is controlbox for complete mobilesystem. Overall size ofsupply is only 6"x6"x6".terrupted voltage impressed on the transformerprimary has practically a square waveform.The secondary voltage also has nearly a squarewaveform as a result and thus the peak voltageon the rectifiers is only slightly higherthan the average voltage of the waveform.This means that the rectifiers in a vibratortypepower supply can be operated on a squarewave voltage close to their maximum peakinverse voltage rating, instead of considerablybelow this rating, as when a sine wave a.c.vooltage is applied to them.Power SupplyCircuitThis power supply has threehigh voltage sections as shownin figure 19. The 250 vo1treceiver supply is shown in figure 19A, the300 volt low voltage transmitter supply infigure 19B, and the 450 volt high voltagetransmitter supply JS shown in figure 19C.Note that each rectifier has two sections. Thepart numbers correspond to the markingsshown in the complete schematic of figure20. In the 2 50 volt circuit, the full transformersecondary voltage is applied to a fullwave rectifier consisting of half of rectifiersSR, and SR,. These two rectifiers form a commonportion of all three rectifier circuits. Thejunction of the rwo rectifiers is grounded andthe positive 250 volt output is taken fromthe center-tap transformer lead. This is theopposite of the usual full wave rectifier circuit.The 300 volt d.c. output is obtained froma bridge rectifier circuit (figure 19B) consistingof one-half of SR, and SR, in theground legs, plus SRa in the two bridge legsfrom which the positive voltage is obtained.Another bridge rectifier circuit is employedfor the 450 volt d.c. output, again with theSYNCHRONOUSVIBRATOR.----------VIBRATORTRANSFORMERSPLIT REEDVI BRA TORt-----LIJIJ' tI ,- ~I I I I® Lllllllllll + 12 VOLT CONNECTIONSt 6 VOLT CONNECTIONSFigure 18SUPPLY FEATURES <strong>THE</strong> NEW SPLIT-REED VIBRATOR CAPABLE OF HANDLING100 WATTS OF POWER AT EI<strong>THE</strong>R 6 OR 12 VOLTSA-simplified diagram of typical synchronous vibrator and rectifier circuit.B-Simplified diagram of typical split-reed vibrator and 6- 12 volt d-e. changeover system.
700 Power Supplies <strong>THE</strong> <strong>RADIO</strong>T! T! T!~oSR1A+ 11~ + SR2A+ •I++SR1ASR2A250 VOLT SUPPLY 300 VOLT SUPPLY 450 VOLT SUPPLY® @ ©Figure 19SIMPLIFIED CIRCUITS OF RECTIFIER SYSTEMS USED IN HIGH EFFICIENCYMOBILE SUPPLYA-Full wave, 250 volt rectifier.B-300 volt bridge circuit tapped across portion of the high voltage winding.C-450 volt bridge rectifier circuit.two grounded legs of the circuit passingthrough SR, and SR,. The other two sectionsof these rectifiers form the legs of the bridgefrom which the positive d.c. voltage is taken.Since there is little ripple voltage in thed.c. output from the rectifiers, a single 40fLfd. filter capacitor on the 300 volt sectionof the supply is adequate. Two 100 f'fd. capacitorsare placed in series across the 450volt section and a capacitor input filter isemployed for the low voltage section.The ControlCircuitA "push-to-talk" power switchingcircuit is included in thepower supply so that the 250volt output may be applied to a mobile receiveror converter. This is accomplished withone pair of contacts on a d.p.d.t. relay, RY,,which also turns on the 300 volt output to atransmitter exciter when its relay coil is energized.A second d.p.d.t. relay, RY,, turnson the 450 volt output to a transmitter finalamplifier and modulator when its coil is energized.It also provides a power-reducing featurewhen the coil is not energized by applying300 volts to the 450 volt output terminaiand 250 volts to the 300 volt output terminal.Since all rectifiers have high voltage onthem continuously when the supply is operating,the idling current flow through the unusedrectifiers when receiving is reduced toalmost zero by disconnecting them from thefilter capacitors and bleeder resistors by theaction of the relays.A special 20 volt secondary winding on thetransformer provides transmitter negative biasthrough rectifier SR, and a 50 f'fd. filter capacitor.-, Circuit Details- In this power supply,Low Voltage Section changing from 6- to 12-volt operation is accomplishedwith a 12-pin Cinch-Jones type 300power plug and socket, P, and J,, respectively.A separate low voltage input cable is usedfor each voltage as shown in figure 20. Leadsfrom p, to the main power relay RYa shouldbe made as short as possible to reduce thevoltage drop. This is particularly importantwith a six volt power source, where a cableresistance of only 0.04 ohm will cause a onevolt primary drop when the power supplyis opera.ting at full load.The six volt input plug connects the alternatepairs of vibrator contacts and the halvesof the transformer primary windings in parallel.These units are connected in series fortwelve volt operation as shown in figure 18B.Six volts for the relay coils is supplied fromthe power cable through pin 6 on ],. With thetwelve volt plug, a voltage dropping resistor,R,, is connected in series with the relay coils.If the power supply is to be operated exclusivelyfrom the higher primary voltage, relayshaving twelve volt coils should be used,thus eliminating R,.There usually is sparking at the vibratorcontacts so var>ious capacitors and resistorsare incorporated in the primary circuit to suppressany radio noise generated by vibratorRction. These components are placed doseto the pins of the vibrator socket.ComponentPartsThe heart of this power supplyis the vibrator transformer designedspecially for two-way mobileradio equipment. It is readily availablefrom many of the 3000 General Electric Co.mobile radio service stations, or it may beordered from the address given in the partslist of figure 20.Power Supply All components except theConstruction power transformer are enclosedinside a 6"x6"x6" aluminumutility box for maximum protection againstdirt and dust. A sub-chassis made from 1116-
<strong>HANDBOOK</strong>Mobile Power Supply 701TSt_;_---...--...--+o +450/300 v.6 TRANSMIT.56K2WGROUND+ 300V56 K 5 TRANSMIT.2W2. 3 4 5 6 7 8 9 10 11 12.l lfl yy ll jl H I " • "'""'GNDTOAUTO6V. TO P2RY3o---LF1 T06VOLTS..._------+-0 + 2.50 v'3 TRANSMIT.TO S3,P22 ?. 4 56 7 6 9 101112 (SEEFIG.238)y jj ~ I ,, "''" "'''GND TO AUTO12.v.ro PzRv3..,..-LFz ro 12. voLrsTO 53, Pz(S££ FIG. 238)1- RELAYS SHOWN !NUNENERGIZED POSITION.2.-ALL RESISTORS 1 WATT UNLESS O<strong>THE</strong>RWISE NOTED.*-MOUNT AT iRANSFORMER TERMINAL.Figure 20SCHEMATIC, 6-12 VOLT MOBILE SUPPLYEz-Vibrator, dual interrupter, split-reed; 6 volt -~oils, 716.c.p.s. reed frequency, 7 pin base. Mallory type1701, Oak V-6853, Radiart 5722, or G.E. A-7141584-P3.F z-30 ampere cartridge fuse.F:-15 ampere cartridge fuse.Jz-12 pin male chassis plug. Cinch-Jones P-312-ABP z-12 pin female cable socket. Cinch-Jones S-312-CCTLz-7.5 henry, 100 ma. Stancor C-1421L,--7 Jlh., I amp. Ohmite 2-50 r-f chokeRYz, RY,-D.P.D.T. relay, 6 volt coil. Advance MG/2c/6VD, Potter & Brumfield MR-11D-6V, Guardian200-6D Coil and 200-2 assembly, or Ohmite DOSX-158T, or equiv.SR1, SRz--Two section selenium rectifier, 150 ma., 380 volts peak inverse per section, connected fordoubler service. G.E. A-7144141-P2, or two Federal 1005-A rectifiers in series for each leg (8 in all).SRs-Two section selenium rectifier, 150 mo., 380 volts peak inverse per section, connected for center-tapservice. G.E. A-7144141-Pl, or two Federal 1005-A rectifiers with red terminals connected together.SR,-150 ma., 64 volt peak inverse selenium rectifier. G. E. A-7140806-PI or federal !015.Tz-Vibrator-type power transformer. Dual center-tapped 6 volt primaries. Secondaries: 420 volts c.t.with 150 volt taps, 300 ma., and 20 volt, !50 ma. bias voltage winding. G.E. B-7486449-Pl.TS,-8 terminal barrier strip. Cinch-Jones 8-141-Y.Note: General Electric parts may be ordered from: D. S. Clark, G.E. Co., Product Service Renewal PartsSection of Communication Products, 509 Kent St., Utica, N.Y. Current prices plus shipping chargesare: Tz, $14.70; Ez, $5.75; SR,, SR:, SR,, $7.75 each; SR 4, $1.40.inch thick aluminum is cut as shown in figures21 and 22. A cut-out in one corner isnecessary to clear the lower portion of thepower transformer. All chassis drilling andthe cutouts for the power transformer, powerinput plug, and output terminal strip shouldbe completed before the chassis is permanentlyfastened in place with small aluminumangle brackets, two inches from the bottom ofthe box.One eleotrolytic capacitor, C, is mountedon an insulated plate furnished with the capacitor.When twisting the locking lugs, makesure that they do not come near the metal
Figure 21UNDER-CHASSISVIEW OFPOWER SUPPLYThe vibrator socket ishidden by the resistorsand disc ceramic capacitorsof the hash filter.R-F choke L: and the25 p.fd. capacitors areabove these parts on thewall of the cose. ResistorR, is mounted to thewall, with SR,, SR:, andSR, in line below it.chassis. The can of this capacitor is more than200 volts "above ground" and should be insulatedwith a fibre sleeve.The large seven pin socket into which thevibrator plugs should have two soldering lugsplaced under each mounting bolt for the bypasscapacitors which are later wired to thesocket pins. If the power supply is to bemounted with the chassis in a vertical plane,pins 1 and 4 of the vibrator socket should bein a vertical plane. The smaller componentsbelow the chassis should not be installed untilparts above the chassis have been assembledand wired.The transformer primary leads are wiredto the input plug and vibrator socket withshortest possible leads before other parts arewired. Placement of the under-chassis partsmay be seen in figure 21. Layout of the partsis not critical.Testing the After the wmng has beenSupply checked, the power supply shouldbe checked at half input voltage,but with full voltage applied to the vibratorcoil. This is done by temporarily removingthe jumper between pins 7 and 8 on thetwelve volt power plug, P1. Pin 8 is thenFigure 22TOP VIEW OFPOWER SUPPLYINTERIORRelays RY, and RY: areat center above choke L,.Power transformer T 1ismounted on end of cabinet.Across top of~7:,~;~r(le~~Jo ffi~Z:) ~~~pacitors
Transistorized Power Supply703FROMBATTERY2. WIRE CABLE#6 FOR 6V.#10FOR12V.®4 WIRE HEATER& RELAY CABLELOW VOLTAGE INPUT PLUG J2J3 FROM H.V. SUPPLYFigure 24CABLE HARNESS, MAIN POWERRELAY, RY 3 , AND REAR OFPOWER CONTROL BOXHif1h voltage for the receiver, plus the controlsw1tch circuits for RY, and RY:, are broughtinto the control box from the power supplythrough J,.High voltage leads for the transmitter arerun directly from the supply,, but the transmitterheater power and 'transmit-receive"control circuit is run to the transmitterthrough control box circuit h (figure 23A).6 VOLT POWER CABLE11Pli£TO RY3~OV~~:;eRs,COl L FROM P1(FIG. 20) (FIG 20)®P2.©12 VOLT POWER CABLEIii j lifDITO RY 3COl L(FIG.20)TO HEATERS,~~;~L~~(FIG.20)Figure 23CONTROL WIRINGA-Block diagram of suggested power andcontrol cables and switching system formobile power supply.8-Schematic diagram of suggested controlbox including power plugs for changin!Jheaters for 6 or 12 volt operation. s, JS11transmit-receive switch," S: is ''high-low"switch, and s, is main power switch forRY,.C-6 and 12 volt power cables.jumped to pin 9. The cable is attached to ;, onthe supply and to a six volt power source. Approximatelyhalf the rated voltage should bemeasured at the output terminal strip if thesupply has been properly wired.Replace the original connections on thepower plug and test the supply with fullinput voltage. A 2500 ohm, 100 watt resistor,or four 2 5 watt 115 volt lamp bulbs in seriesmake a good load resistance. The output voltagesshould measure close to 450, 300, and240 volts under load. Additional .02 JLfd. bypasscapacitors at RY,, the output terminalstrip, and the control box should eliminateany hash still present during reception. Everyexperienced "mobileer" will agree that noisein each mobile installation usually must beeliminated on a "search and filter" basis.This power supply may be operated in conjunction with thesuggested configuration shown inInstallationin the Carfigure 23A. Note that a separate cable isrecommended for the heater power circuit toreduce vibrator hash pickup, and to m1mmizeheater voltage variations when the supply isswitched from receive to transmit.Provision has been made for changing themobile receiver and transmitter power circuitsfor either six or twelve volt operation as shownin the schematic of the control box in figure23B. An eight contact plug and socket automaticallymake the proper connections whenthe six or twelve volt plug is attached.32-6 TransistorizedPower SuppliesThe vibrator type of mobile supply achievesan overall efficiency in the neighborhood of70%. The vibrator may be thought of as amechanical switch reversing the polarity ofthe primary source at a repetition rate of 120transfers per second. The switch is actuated bya magnetic coil and breaker circuit requiringappreciable power which must be supplied bythe primary source.One of the principal applications of thetransistor is in switching circuits. The transistormay be switched from an "off" conditionto an "on" condition with but the ap-
704 Power Supplies <strong>THE</strong> <strong>RADIO</strong>1fPIIE- DIIE-- + +® ® © @Figure 25TRANSISTORS CAN REPLACE VIBRATOR IN MOBILE POWER SUPPLY SYSTEMA-Typical vibrator circuit.8-Vibrator can be represented by two single-pole single throw switches, or transistors.C-Push-pu/1 square wave ''oscil/ator 11 is driven by special feedback windings on power transformer.D-Addition of bias in base-emitter circuit results in oscillator capable of starting under full load.plication of a minute exciting signal. \\"henthe transistor is nonconductive it may be consideredto be an open circuit. When it is ina conductive state, the internal resistance isvery low. Two transistors properly connected,therefore, can replace the single pole, doublethrow mechanical switch representing the vibrator.The transistor switching action is manytimes faster than that of the mechanical vibratorand the transistor can switch an appreciableamount of power. Efficiencies in theneighborhood of 95% can be obtained with 28volt primary-type transistor power supplies,permitting great savings in primary powerover conventional vibrators and dynamotors.TransistorOperationThe transistor operation resemblesa magnetically coupled multi-vibrator,or an audio frequencypush-pull square wave oscillator (figure 2 5C).A special feedback winding on the power transwl9~ .SVOLTS ,rRISETIME=70..US26 40 -,--- _,_ ~200-200o:.,~, _____~ 3s~-- r--1'-----' 11s I0 30--- If-~ 25--j 20-8 15-~ 10w 5 -~~ o~--'-cnIO.SVOLTTIME---Figure 26BASE-COLLECTOR WAVEFORM OFSWITCHING CIRCUIT,FOR 12 VOLT CIRCUITSquare waveshape produces almost ideal11switching action. Small Spike" on leadingedge of pulses may be reduced by propertransformer design.former provides 180 degree phase shift voltagenecessary to maintain oscillation. In thisapplication the transistors are operated as onoffswitches; i.e., they are either completingthe circuit or opening it. The oscillator outputvoltage is a square wave having a frequencythat is dependent upon the driving voltage,the primary inductance of the power transformer,and upon the peak collector currentdrawn by the conducting transistor. Changesin transformer turns, core area, core material,and feedback turns ratio have an effect on thefrequency of oscillation. Frequencies in commonuse are in the range of 120 c.p.s. to3,500 c.p.s.The power consumed by the transistors 1srelatively independent of load. Loading theoscillator causes an increase in input currentthat is sufficient to supply the required powerto the load and the additional losses in thetransformer windings. Thus, the overall efficiencyactually increases with load and is greatestat the heaviest load the oscillator will supply.A result of this is that an increase in loadproduces very little extra heating of the transistors.This feature means that it is impossibleto burn out the transistors in the event ofa shorted load since the switching action merelystops.Transistor The power capability of thePower Rating transistor is limited by theamount of heat created by thecurrent flow through the internal resistanceof the transistor. When the transistor is conductingthe internal resistance is extremelylow and little heat is generated by currentflow. Conversely, when the transistor is in acut-off condition the internal resistance is veryhigh and the current flow is extremely small.Thus, in both the '"on" and "off" conditions
<strong>HANDBOOK</strong> Transistorized Power Supplies 7052N278r-----------..-----.-o+250V. AT25 K5W50 MA :;;:12WATTSTOTAL POWER= 87 WATTS2N278 OR 2N301At----?----~--0+275 V. AT 125 MA = 35 WATTS25K"'iOW®Figure 27PRACTICAL TRANSISTOR POWER SUPPLIEST ,--Chicago DCT -2 transistor transformerT ,-Chicago DCT -I transistor transformerD1-Ds-Sarkes-Tarzian M-500 silicon rectifier or equivalent.the transistor dissipates a mm1mum of power.The important portion of the operating cycleis that portion when the actual switching fromone transistor to the other occurs, as this isthe time during which the transistor may bepassing through the region of high dissipation.The greater the rate of switching, in general,the faster will be the rise time of the squarewave (figure 2 6) and the lower will be theinternal losses of the transistor. The averagetransistor can switch about eight times thepower rating of class A operation of theunit. Two switching transistors having 5 wattclass A power output rating can thereforeswitch 80 watts of power when working atoptimum switching frequency.Self-StartingOscillatorsThe transistor supply shown infigure 25C is impractical becauseoscillations will not startunder load. Base current of the proper polarityhas to be momentarily introduced intothe base-emitter circuit before oscillation willstart and sustain itself. The addition of a biasresistor (figure 25D) to the circuit resultsin an oscillator that is capable of startingunder full load. R, is usually of the order of10-50 ohms while R, is adjusted so that approximately100 milliamperes flows throughthe circuit.The current drawn from the battery by thisnetwork flows through R, and then dividesbetween R, and the input resistances of thetwo transistors. The current flowing in theemitter-base circuit depends upon the valueof input resistance. The induced voltage acrossthe feedback winding of the transformer is asquare wave of such polarity that it forwardbiasesthe emitter-base diode of the transistorthat is starting to conduct collector current,and reverse-biases the other transistor. Theforward-biased transistor will have a very lowinput impedance, while the input impedanceof the reversed-biased transistor will be quitehigh. Thus, most of the starting current drainedfrom the primary power source will flowin R, and the base-emitter circuit of the forward-biasedtransistor and very little in theother transistor. It can be seen that R, must notbe too low in comparison to the input re-
706 Power Supplies<strong>THE</strong> <strong>RADIO</strong>Figure 2835 WATT TRANSISTORPOWER SUPPLYTwo 2N30 I A power transistors are used inthis midget supply. Transistors are mountedon sanded portion of chassis deck which actsas ''heat sink." See text tor details.sistance of the conducting transistor, or elseit will shunt too much current from the transistor.When switching takes place, the transformerpolarities reverse and the additionalcurrent now flows in the base-emitter circuitof the other transistor.The PowerTransformerThe power transformer in atransistor-type supply is designedto reach a state of maximum fluxdensity (saturation) at the point of maximumtransistor conductance. When this state isreached the flux density drops to zero andreduces the feedback voltage developed in thebase winding to zero. The flux then reversesbecause there is no conducting transistor tosustain the magnetizing current. This changeof flux induces a voltage of the opposite polarityin the transformer. This voltage turnsthe first transistor off and holds the secondtransistor on. The transistor instantly reachesa state of maximum conduction, producing astate of saturation in the transformer. Thisaction repeats itself at a very fast rate. Switchingtime is of the order of 5 to 10 microseconds,and saturation time is perhaps 200to 2,000 microseconds. The collector waveformof a typical transistor supply is shownin figure 26. The rise time of the wave isabout 5 microseconds, and the saturation timeis 500 microseconds. The small ''spike" atthe leading edge of the pulse has an amplitudeof about 2 1;2 volts and is a product ofswitching transients caused by the primaryleakage reactance of the transformer. Propertransformer design can reduce this "spike" toa minimum value. An excessively large "spike"can puncture the transistor junction and ruinthe unit.32-7 TwoTransistorizedMobile SuppliesThe new Chicago-Standard Transformer Co.series of power transformers designed to workin transistor-type power supplies permits theamateur and experimenter to construct efficientmobile power supplies at a fractionof their former price. Described in this sectionare two power supplies designed around theseefficient transformers. The smaller supply delivers35 watts (275 volts at 125 milliamperes)and the larger supply delivers 8 5 watts(500 volts at 125 milliamperes and 250 voltsFigure 29SCHEMATIC, TRANSISTORPOWER SUPPLY FOR12 VOLT AUTOMOTIVESYSTEMT ,-Transistor power transformer.12 volt primary, to provide 275volts at 125 ma. Chicago StandardDCT-1.D,-D,--Sarkes-Tarzian silicon rectifier,type M-500, or equivalent.25KTIY27!S v.AT12.5MA,- 12 V. BATTERY +
<strong>HANDBOOK</strong>Components 707at 50 milliamperes, simultaneously). Bothpower units operate from a 12 volt primarysource.The 35 WattSupplyThe 35 watt power unit usestwo inexpensive RCA 2N301AP-N-P-type power transistorsfor the switching elements and four silicondiodes for the high voltage rectifiers. Thecomplete schematic is shown in figure 29.Because of the relatively high switching frequencyonly a single 20 1-'fd. filter capacitoris required to provide pure d.c.Regulation of the supply is remarkably good.No load voltage is 310 volts, dropping to275 volts at maximum current drain of 125milliamperes.The complete power package is built uponan aluminum chassis-box measuring 5V
708Power Supplies<strong>THE</strong> <strong>RADIO</strong>2N278470~- 12 V. BATTERY+2.5 K..,-w-SOK10WFigure 31SCHEMATIC,85 WATTTRANSISTORPOWER SUPPLYFOR 12 VOLTAUTOMOTIVESYSTEMT1-Transistor powertransformer. 12 voltprimary to provide 275volts at 125 ma.~&ir~r standardD,-D,-Sarkes-Tarziansilicon rectifier, typeM-500paper. Some types of paper capacitors arewax-impregnated, but the better ones, especiallythe high-voltage types, are oil-impregnatedand oil-filled. Some capacitors are rated bothfor flash test and normal operating voltages;the latter is the important rating and is themaximum voltage which the capacitor shouldbe required to withstand in service.The capacitor across the rectifier ciw 1it ina capacitor-input filter should have a tva ~kingvoltage rating equal at least to 1.41 times ther-m-s voltage output of the rectifier. The remainingcapacitors may be rated more nearlyin accordance with the d-e voltage.The electrolytic capacitor consists of twoaluminum electrodes in contact with a conductingpaste or liquid which acts as an electrolyte.A very thin film of oxide is formed on thesurface of one electvode, called the anode.This film of oxide acts as the dielectric. Theelectrolytic capacitor must be correctly connectedin the circuit so that the anode alwaysis at a positive potential with respect to theelectrolyte, the latter actually serving as theother electrode (plate) of the capacitor. A reversalof the polarity for any length of timewill ruin the capacitor.The dry type of electrolytic capacitor usesan electrolyte in the form of paste. The dielectricin electrolytic capacitors is not perfect;these capacitors have a much higher directcurrent leakage than the paper type.The high capacitance of electrolytic capacitorsresults from the thinness of the filmwhich is formed on the plates. The maximumvoltage that can be safely impressed acrossthe average electrolytic filter capacitor is between450 and 600 volts; the working voltageis usually rated at 450. When electrolytic ca-pacitors are used in filter circuits of highV'Oltage supplies, the capacitors should be connectedin series. The positive terminal of onecapacitor must connect to the negative terminalof the other, in the same manner asdry batteries are connected in series.It is not necessary to connect shunt resistorsacross each electrulytic capacitor sectionas it is with paper capacitors connected inseries, because electrolytic capacitors havefairly low internal d-e resistance as comparedto paper capacitors. Also, if there is any variationin resistance, it is that electrolytic unitin the poorest condition which will have thehighest leakage current, and therefore the voltageacross this capacitor will be lower thanthat across one of the series connected unitsin better condition and having higher internalresistance. Thus we see that equalizing resistorsare not only unnecessary across seriesconnectedelectrolytic capacitors but are actuallyundesirable. This assumes, of course,similar capacitors by the same manufacturerand of the same capacitance and voltage rating.It is not advisable to connect in serieselectrolytic capacitors of different make orratings.There is very little economy in using electrolyticcapacitors in series in circuits wheremore than two of these capacitors would berequired to prevent voltage breakdown.Electrolytic capacitors can be greatly reducedin size by the use of etched aluminumfoil for the anode. This greatly increases thesurface area, and the dielectric film coveringit, but raises the power factor slightly. Forthis reason, ultra-midget electrolytic capacitorsordinarily should not be used at full rated d-evoltage when a high a-c component is present,
<strong>HANDBOOK</strong>such as would be the case for the input capacitorin a capacitor-input filter.BleederResistorsA heavy duty resistor should beconnected across the output of afilter in order to draw some loadcurrent at all time. This resistor avoids soaringof the voltage at no load when swingingchoke input is used, and also provides a meansfor discharging the filter capacitors when noexternal vacuum-tube circuit load is connectedto the filter. This bleeder resistor should normallydraw approximately 10 per cent of thefull load current.The power dissipated in the bleeder resistorcan be calculated by dividing the square ofthe d-e voltage by the resistance. This poweris dissipated in the form of heat, and, if theresistor is not in a well-ventilated position,the wattage rating should be higher than theactual wattage being dissipated. High-voltage,high-capacitance filter capacitors can hold adangerous charge if not bled off, and wirewoundresistors occasionally open up withoutwarning. Hence it is wise to place carbon resistorsin series across the regular wire-woundbleeder.When purchasing a bleeder resistor, be surethat the resistor will stand not only the requiredwattage, but also the voltage. Some resistorshave a voltage limitation which makesit impossible to force sufficient current throughthem to result in rated wattage dissipation.This type of resistor usually is provided withslider taps, and is designed for voltage dividerservice. An untapped, non-adjustable resistoris preferable as a high voltage bleeder, and isless expensive. Several small resistors may beconnected in series, if desired, to obtain therequired wattage and voltage rating.Transformers Power transformers and filamenttransformers normallywill give no trouble over a period of manyyears if purchased from a reputable manufacturer,and if given a reasonable amount ofcare. Transformers must be kept dry; even asmall amount of moisture in a high-voltageunit will cause quick failure. A transformerwhich is operated continuously, within itsratings, seldom will give trouble from moisture,since an economically designed transformeroperates at a moderate temperaturerise above the temperature of the surroundingair. But an unsealed transformer which isinactive for an appreciable period of time ina highly humid location can absorb enoughmoisture to cause early failure.Filter ChokeCoilsSpecial Supplies 709Filter inductors consist of acoil of wire wound on a laminatediron core. The size ofwire is determined by the amount of directcurrent which is to flow through the chokecoil. This direct current magnetizes the coreand reduces the inductance of the choke coil;therefore, filter choke coils of the smoothingtvpe are built with an air gap of a sma[,l fractionof an inch in the iron core, for the purposeof preventing saturation when maximumd.c. flows through the coil winding. The "airgap" is usually in the form of a piece of fiberinserted between the ends of the laminations.The air gap reduces the initial inductance ofthe choke coil, but keeps it at a higher valueunder maximum load conditions. The coil musthave a great many more turns for the sameinitial inductance when an air gap is used.The d-e resistance of any filter choke shouldbe as low as practicable for a specified valueof inductance. Smaller filter chokes, such asthose used in radio receivers, usually havean inductance of from 6 to 15 henrys, and ad-e resistance of from 200 to 400 ohms. Ahigh d-e resistance will reduce the output voltage,due ro the voltage drop across each chokecoil. Large filter choke coils for radio transmittersand Class B amplifiers usually haveless than 100 ohms d-e resistance.32-9 Special PowerSuppliesA complete transmitter usually includes oneor more power supplies such as grid-biaspacks, voltage-regulated supplies, or transformerlesssupplies having some special characteristic.RegulatedSuppliesV-R TubesWhere it is desired in a circuitto stabilize the voltage supplyto a load requiring not more thanperhaps 20 to 25 rna., the glowdischargetype of voltage-regulator tube canbe used to great advantage. Examples of suchcircuits are the local oscillator circuit in areceiver, the tuned oscillator in a v-f-o-, theoscillator in a frequency meter, or the bridgecircuit in a vacuum-tube voltmeter. A numberof tubes are available for this applicationincluding the OA3/VR75, OB3/VR90,OC3/VR105, OD3/VR150, and the OA2and OB2 miniature types. These tubes stabilizethe voltage across their terminals to 7 5,90, 105, or 150 volts. The minature typesOA2 stabilize to 150 volts and OB2 to 108volts. The types OA2, OB2 and OB3/VR90
710 Power Supplieshave a maximum current rating of 30 rna.and the other three types have a maximumcurrent rating of 40 rna. The minimum currentrequired by all six types to sustain aconstant discharge is 5 ma.A V R tube (common term applied to allglow-discharge voltage regulator tubes) maybe used to stabilize the voltage across a variaNeload or the voltage acros·s a constant loadfed from a varying voltage. Two or more VRtubes may be connected in series to provideexactly 180, 210, 255 volts or other combinationsof the voltage ratings of the tubes.It is not recommended, however, that VRtubes be connected in parallel since both thesnriking and the regulMed voltage of theparalleled tubes normally will be sufficientlydifferent so that only one of the tubes willlight. The remarks following apply generallyto all the VR types although some examplesapply specifically to the OD3/VR150 type.A device requiring say, only 50 volts canbe stabilized against supply voltage variationsby means of a VR-105 simply by putting asuitable resistor in series with the regulatedvoltage and the load, dropping the voltagefrom 105 to 50 volts. However, it should beborne in mind that under these conditionsthe device will not be regulated for vcvryingload; in other words, if the load resistancevaries, the voltage across the load will vary,even though the regulated voltage remains at105 volts.To maintain constant voltage across a varyingload resistance there must be no seriesresistance between the regulator tube and theload. This means that the device must beoperated exactly at one of the voltages obtainableby seriesing two or more s·imi!ar ordifferent VR tubes.A VR-150 may be considered as a stubbornvariable resistor having a range of from 30,000to 5000 ohms and so intent upon maintaininga fixed voltage of 150 volts across its terminalsthat when connected across a voltagesource having very poor regulation it will instantlyvary tts own resistance within the limitsof 5000 and 30,000 ohms in an attemptto maintain the same 150 volt drop across itsterminals when the supply voltage is varied.The theory upon which a VR tube operates iscovered in an earlier chapter.It is paradoxical that in order to do a goodjob of regulating, the regulator tube must befed from a voltage source having poor regulation(high series resistance) . The reason for~his presently will become apparent.<strong>THE</strong> <strong>RADIO</strong>If a high resistance is connected across theVR tube, it will not impair its ability to maintaina fixed voltage drop. However, if the loadis made too low, a variable 5000 to 30,000ohm shunt resistance (the VR-150) will notexert sufficient effect upon the resulting resistanceto provide constant voltage exceptover a very limited change in supply voltageor load resistance. The tube will supply maximumregulation, or regulate the largest load,when the source of supply voltage has highinternal cr high series resistance, because avariation in the effective internal resistanceof the VR tube will then have more controllingeffect upon the load shunted across it.In order to provide greatest range of regulation,a VR tube (or two in series) should beused with a series resistor (to effect a poorlyregulated voltage source) of such a value thatit will permit the VR tube to draw from 8 to20 rna. under normal or average conditionsof supply voltage and load impedance. Formaximum control range, the series resistanceshould be not less than approximately 20,000ohms, which will necessitate a source of voltageconsiderably in excess of 150 volts. However,where the supply voltage is limited, goodconnrol over a limited range can be obtainedwith as little as 3000 ohms series resistance.If it takes les·s than 3000 ohms series resistanceto make the VR tube draw 15 to 20rna. when the VR tube is connected to theload, then the supply voltage is not highenough for proper operation.Should the current through a VR -15 0, VR-10 5, or VR-7 5 be allowed to exceed 40 rna.,the life of the tube will be shortened. If thecurrent falls below 5 rna., operation will becomeunstable. Therefore, the tube must operatewithin this range, and within the twoextremes will maintain the voltage within 1.5per cent. It takes a voltage excess of at least10 to 15 per cent to "start" a VR type regulator;and to insure positive starting each timethe voltage supply should preferably exceedthe regulated output voltage raning about 20per cent or more. This usually is automaticallytaken care of by the fact that if sufficientseries resistance for good regulation is employed,the voltage impressed across the VRtube before the VR tube ionizes and startspassing current is quite a bit higher than thestarting voltage of the tube.When a VR tube is to be used to regulatethe voltage applied to a circuit drawing lessthan 15 ma. normal or average current, thesimplest method of adjusting the series resist-
<strong>HANDBOOK</strong>Special Supplies 711EsD.C. SUPPLYVOLTAGERs, SERIES RESISTORVR TUBEFigure 32STANDARD VR-TUBE REGULATORCIRCUITThe VR-tube regulator will maintain the voltageacross its terminals constant within afew volts for moderate variations in RL or E • 8See text for discussion of the use of VRtubes in various circuit applications.RLl-OADance is to remove the load and vary the seriesres,istor until the VR tube draws about 30 rna.Then connect the load, and that is all there isto it. This method is particularly recommendedwhen the load is a heater type vacuum tube,which may not draw current for several secondsafter the power supply is turned on.Under these conditions, the current throughthe VR tube will never exceed 40 rna. evenwhen it is running unloaded (while the heatertube is warming up and the power supplyrectifier has already reached operating temperature).Figure 32 illustrates the standard glow dischargeregulator tube circuit. The tube willmaintain the voltage across RL constant towithin 1 or 2 volts for moderate variations inRL or Es.Voltage RegulatedPower SuppliesWhen it is desired to stabilizethe potential acrossa circuit drawing morethan a few milliamperes it is advisable touse a voltage-regulated power supply of thetype illustrated in f,igure 33 rather than glowdischarge tubes.A 6AS7-G is employed as the series controlelement, and type 816 mercury vapor rectifiersare used in the power supply section. The6AS 7 -G acts as a variable series resistancewhich is controlled by a separate regulatortube much in the manner of a-v-e circuits orinverse feedback as used in receivers and a-famplifiers. A 6SH7 controls the operating biason the 6AS7 -G, and therefore controls the internalresistance of the 6AS7 -G. This, in turn,controls the output voltage of the supply, whichcontrols the plate current of the 6SH7, thuscompleting the cycle of regulation. It is apparentthat under these conditions any changein the output vol.tage will tend to "resist itself,"much as the a-v-e system of a receiverresists any change in S'ignal strength deliveredto the detector.Because it is necessary that there always bea moderate voltage drop through the 6AS7-Gin order for it to have proper control, the restof the power supply is designed to deliver asmuch output voltage as possible. This callsfor a low resistance full-wave rectifier, a highcapacitance output capacitor in the filter systemand a low resistance choke.Reference voltage in the power supply isobtained from a VR-150 gaseous regulator.Note that the 6.3-volt heater winding for the6SH7 and the 6AS7-G tubes is operated ata potential of plus 150 volts by connectingthe winding to the plate of the VR-150. Thisprocedure causes the heater-cathode voltageof the 6SH7 to be zero, and permits an outputvoltage of up to 450 since the 300-voltheater-to-cathode rating of the 6AS7 -G is notexceeded with an output voltage of 450 fromthe power supply.The 6SH7 tube was used in place of themore standard 6SJ7 after it was found thatthe regulation of the power supply could beimproved by a factor of two with the 6SH7in place of the 6SJ7. The original version ofthe power supply used a 5R4-GY rectifiertube in place of the 816's which now areused. The excessive drop of the 5R4-GY resultedin loss of control by the regulator por-Figure 33SCHEMATIC OF VOLTAGEREGULATED POVfER SUPPLYT 1-6 J 5 or 520 volts each side of c.t., 300ma. Stancor P-804 1.T r-5 volts at 3 amp., 6.3 volts at 6 amp.Stancor P-5009.CH-4 henry at 250 ma. Stancor C-1412.
712 Power Supplies<strong>THE</strong> <strong>RADIO</strong>T1V1115V.50-60'\..8,3V.COMPONENTST1 y, CHI CH2 c. C2 R1APPROXIMATEOUTPUT VOLTAGEMAX. 6.3V.NO FULL CURRENT FILAMENLOAD LOAD10 H. IOH,350-0-35010JJF,4~ V. 20JJF,450V.STANCOR STANCOR CORNELL- CORNELL-STANCOR PC~B409 SYJ·ca C-1001 C-1001DUBILIER OUBILIERMERIT P-3152 MERIT MERIT BR-1045 BR-2045C-2993 C-299335K,10W 310 240 80MA. 3A.3-13 H 7 H. IOJJF,450V, 201JF375-0-3751 45QV,STANCOR STANCORSTANCOR PC-8411 5Y3·c;T C-!718 C-1421CORNELL- CORNELL-DUBILIER DUB/L/ERMERIT P-2954 ME/UT MERITC-3187BR-1045C-3180BR-204535K,10W 330 230 140MA. 4.5A.2-12 H. 4 H. IOJJF 450 V. 10JJF,450 V.400-0-400 STANCOR STANCOR CORNELL- CORNELL-5U4-G C-1402 C-1412STANCOR PG-8413 DUBILIER DUB ILlERMERIT MERIT BR-1045 BR-104535K,IOW 300 270 250MA. 5A.c-31a9 C-3 182525-0-525 5-25 H. 20 H.5U4-t;BUTC S-40UTC S-32 urc s-3110.UF 1 6QOV. IO.UF,600 V,MALLORY MALLORY 35K,10W 460 37> 240MA. 4A.TC-92 TC-92600-0-600 5-25 H.8JJF. 600 v. 81JF, 600 v.20 H.5R4-GY SPRAGUE SPRAGUE 35K,25W. .40 410 200r.4A. 4A.UTC S-41 UTC S-32 UTC S-31CR-86 CR-86900-0-900 S-25H. 20H.5R4·CO.Yurc s-45UTC S-32 UTC S-3141JF, 1 KV. 81JF, 1 KV.SPRAGUE SPRAGUE 50K,2.5W. 630 ••o175r.4 ....CR-41 CR-81Figure 34DESIGN CHART FOR CHOKE-INPUT POWER SUPPLIEStion with an output voltage of about 390with a 225-ma. drain. Satisfactory regulationcan be obtained, however, at up to 450 voltsif the maximum current drain is limited to150 rna. when using a 5R4-GY rectifier. Ifthe power transformer is used with the tapsgiving 520 volts each side of center, and ifthe maximum drain is limited to 225 rna.,a type 83 rectifier may be used as the powersupply rectifier. The 615-volt taps on thepower transformer deliver a voltage in excessof the maximum ratings of the 83 tube. Withthe 83 in the power supply, excellent regulationmay be obtained with up to about 420volts output if the output current is limitedto 225 rna. But with the 816's as rectifiersthe full capabilities of all the componentsin the power supply may be utilized.If the power supply is to be used with anoutput voltage of 400 to 450 volts, the full615 volts each side of center should be appliedto the 816's. However, the maximumplate dissipation rating of the 6AS7 -G willbe exceeded, due to the voltage drop acrossthe tube, if the full current rating of 250 rna.is used with an output voltage below 400volts. If the power supply is to be used withfull output current at voltages below 400 voltsthe 520-volt taps on the plate transformershould be connected to the 816's. Some variationin the output range of the power supplymay be obtained by varying the values of theresistors and the potentiometer across the output.However, be sure that the total platedissipation rating of 26 watts on the 6AS7-Gseries regulator is not exceeded at maximumcurrent output from the supply. The total dissipationin the 6AS7-G is equal to the currentthrough it (output current plus the currentpassing through the two bleeder strings)multiplied by the drop through the tube (voltageacross the filter capacitor minus the outputvoltage of the supply).32-10 Power SupplyDesignPower supplies may either be of the chokeinput type illustrated in figure 34, or the capacitorinput type, illustrated in figure 35. Capacitorinput filter systems are characterizedby a d-e supply output voltage that runs from0.9 to about 1.3 times the r.m.s. voltage ofone-half of the high voltage secondary windingof the transformer. The approximate regulationof a capacitor input filter system isshown in figure 36. Capacitor input filtersystems are not recommended for use with
<strong>HANDBOOK</strong> Special Supplies 713T1V111!> v,so-eo"'Tl2.60-0-2.60STANCOR PC~8404MERIT P-3148VI5Y3-GT375-0-375STANCOR PC-8411 SY3-CTMERIT P-2.954435-0-43.5MERIT' P-315155U4-G600-0-8005R4-GYSTANCOR PC-8414900-0-900UTC S-45~R4-GYCOMPONENTSCHICl10 H. 2.0.UF,4SOV.STANCORCORNELL-C-1001DUSIL!ERMERITC-2993BR-20457 H.STANCOR 10.UF,600 V.C-1.12.1 MALLORYMERIT TC-92C-31804H.STANCOR 8.UF,800V.C-1412 SPRAGUEMERIT CR-86C-31824 H.STANCOR 4.UF, 1 KY.C-1412 SPRAfiUEMERIT CR-41c-Jtsz2.0 H.UTC S-314lJF, t.!IKV.SPRAGUECR-415C220.UF, 4SO v.CORNELL-DUB ILlERBR-2.045R1APPROXIMATEOUTPUT VOLTAGE MAX. 6.3V.NO FULL CURRENT FILAMENTLOAD LOAD35K,10W 340 240 80MA. 3A.10JJF,800 v.MALLORY' 35K,t0W 480 3>0 12..5MA. 4.SA.TC-9281JF 1 &00 V.SPRAGUE 35K,Z5W 800 400 22.5 MA. 8 A.CR-888.UF, 1 KV.SPRAfiUE 50K,25W 800 800 ZOOMA. 8A.CR-81llJF. 1.bKV.SPRA,UE 7!1K 2!1W 1200 910 1!10 t.4A. -CR-815Figure 35DESIGN CHART FOR CAPACITOR-INPUT POWER SUPPLIESmercury vapor rectifier tubes, as the peak rectifiercurrent may run as high as five or sixtimes the d-e load current of the power supply.It is possible, however, to employ type872-A mercury vapor rectifier tubes in capacitorinput circuits wherein the load currentis less than 600 milliamperes or so, andwhere a low resistance bleeder is used to holdthe minimum current drain of the supply toa value greater than 50 milliamperes or so.Under these conditions the peak plate currentof the 872-A mercury vapor tubes will notbe exceeded if the input filter capacitor is4 /Lfd. or less.Choke input filter systems are characterizedby lower peak load currents ( 1.1 to 1.3 timesthe average load current) than the capacitorinput filter, and by better voltage regulation.Design Charts for capacitor and choke inputfilter supplies for various voltages and loadcurrents are shown in figures 34, 35, and 37.The construction of power supplies for transmitters,receivers and accessory equipment isa relatively simple matter electrically sincelead lengths and placement of parts are ofminor importance and since the circuits themselvesare quite simple. Under-chassis wiringof a heavy-duty supply is shown in figure 38.Bridge SuppliesSome practical variations ofthe common bridge rectifiercircuit of figure 6 are illustrated in figures 39and 40. In many instances a transmitter ormodulator requires two different supply voltages,differing by a ratio of about 2:1. Asimple bridge supply such as shown in figure39 will provide both of these voltages froma simple broadcast "replacement-type" powertransformer. The first supply of figure 39 isample to power a transmitter of the 6CL6-807type to an input of 60 watts. The second supplywill run a transmitter running up to 120watts, such as one employing a pair of 6146MAX -0.9 1.0 1.1 1.2 1.3 t.4RATIO OF D.C. OUTPUT VOLTAGE TO R.M.S VOLTAGEOF i SECONDARY WINDING OF PLATE TRANSFORMER.Figure 36APPROXIMATE REGULATION OFCAPACITOR-INPUT FILTER SYSTEM
714 Power Supplies <strong>THE</strong> <strong>RADIO</strong>TzCOMPONENTST• Tz V1-V2 CH1 CH2 c. C2 R1APPROXIMATEOUTPUT VOL TAG£ MAX.CURRENTNO FULLLOAD LOAD (I CAS)11.50-0-11~0 2.5V.,10A.6 H. IOH. 4JJF, 1.5KV. 8JJF, 1.5 KV.888-ACHICAGO TRANS. CHI. TRAN.CHI.TRAN. CHI. TRAN. SANGAMO SANGAMO868-A40K,75W 1150 1000 350t.4A.P-!07 F-210 R-G3 R-103 llts-4 7115-81710-0-1710 2.5V., lOA.6 H. IOH. 4lJF, 2 KY. 8.UF, 2 KY.886-ACHICAGO TRANS. CHI. TRAN.CHI. TRAN. CHI.TRAN. SANGAMO SANGAMO MK,75W 1700see-Auoo 4Z5MA.P-1512 F-210H R-tJ5 R-105 7120-4 7120-B2900-0-2900 5 V., lOA.6 H. 6 H. 4.UF, JKV, 4JJF, 3KV.872-ACHICAGO TRANS. CHI. TRAN. CHI. TRAN. CHI. TRAN. SANGAMO SANGAMO70K 2750 2500 700 MA.P-2126872-AF-510HR-tJl R-tJT 7130-
<strong>HANDBOOK</strong> Special Supplies 7156.3 v.COMPONENTSFULL LOAD VOLl'. ~AX. CURRENTT1 V1-V2. V3 C1-C2. C3-C4 CH1 CH2 R1-R2 R3 HV#1 HV#2 #1 #2360-0-360STANCORPC-84106X5-GT5V4·G16JJF450 v.16JJF. IOH. 8 H.100 K20K,10W.450V. 12.0MA. SOMA. IW600 240 100 MA. 40MA.T2.,T3=6.3V., 1 A.STANCOR P-6134400-0-400STANCDRPC-84126AX5-GT 5U4-GB16JJF.450 v.15JJF, 10 H. 8 H. 100 K4SOV. 225MA. 75MA. 20K lOW. , v625 2.60 115 MA 50 MA.115V.S0-60'\..oFigure 39DUAL VOLTAGE BRIDGE POWER SUPPLIESfigure 40, a d-e supply voltage of about 1900volts at a current of 500 milliamperes may bedrawn from such a transformer. Do not attemptto use a smaller transformer than the2-KV A rating, as the voltage regulation ofthe unit will be too poor for practical purposes.For higher voltages, a pole transformer witha 4400 volt primary and a 110/220 volt secondarymay be reversed to provide a d-e platesupply of about 3800 volts.Commercial plate transformers intended forfull wave rectifier service may also be usedin bridge service provided that the insulationat the center-tap point of the high voltagewinding is sufficient to withstand one-half ofthe r.m.s. voltage of the secondary winding.Many high voltage transformers are specificallydesigned for operation with the center-tapof the secondary winding at ground potential;consequently the insulation of the winding atthis point is not designed to withstand highvoltage. It is best to check with the manufacturerof the transformer and find out if theinsulation will withstand the increased volt-age before a full wave-type transformer tsutilized in bridge rectifier service.32-11 300 Volt,SO Ma. Power SupplyThere are many applications in the laboratoryand amateur station for a simple lowdrain power supply. The most common appli-Figure 40HIGH-VOLTAGEPOWER SUPPLYT1COMPONENTSFULLFULL LOADLOA~VOLTAGE CURRENTT2 V1-Y4 CHI C1 R1 (I CAS)22.00-VOLT20H.UTC10.JJF. BKPOLE TRANSFOR~ER 866-A ~OOMA. 1900 ~00 MA.S-71 2.~00urc s-31 v. 2.00W.2. KVA10H.1/TC8.UF 200 KUTC CG-308 LS-121-Y 6800 v. 300W.3~00·0·3~00 872-Aj~gi:-~016000 ~OOMA.
Figure 41TYPICAL LIGHT DUTYPOWER SUPPLYThis 300 volt, 50 milliampere power supplymay be used to run signal generators Ire~quency meters, small receivers, etc. Swifch SzI see schematic) is placed on the rear of thechassis near the line cord.cation in the amateur station for such a supplyis for items of test equipment such as thetype LM and BC-221 frequency meters, forfrequency converters to be used in conjunctionwith the station receiver, and for auxiliaryequipment such as high selectivity i-f stripsor variable frequency oscillators. Equipmen~ssuch as these may be operated from a supplydelivering 250 to 300 volts at up to 50 milliamperesof plate current. A filament source of6.3 or 12 volts may also be required, as willa source of regulated voltage.The simple power supply illustrated in figures41 and 42 is capable of meeting theserequirements. A two section capacitor inputfilter system is employed to provide minimumripple content and a switch ( S,) isprovided to insert a VR-150 regulator tubein the circuit to provide 150 volts at approximately3 5 milliamperes.A separate 6.3 volt filament transformer isconnected in series with the 6.3 volt windingof power transformer T, to provide 12.6volts at 1.2 amperes for operating "12 volt"tubes or a string of two "6 volt" tubes connectedin series. The secondary winding ofT, must be polarized correctly to provide 12.6volts across the two windings.Resistor R, must be adjusted to "fire" thevoltage regulator tube. It should be adjustedso that approximately 15 milliamperes passthrough the tube. For maximum permissiblecurrent drain from the high voltage tap ofthe supply the VR tube should be switchedout of the circuit.32-12 500 Volt,200 MilliamperePower SupplyThe availability of inexpensive "TV-type"components makes possible the constructionof a sturdy 100 watt power supply at a relativelymodest price. This power unit is suitablefor operating a 50 watt phone/c-w transmitter,a small modulator, or a 100 watt sidebandlinear amplifier. Maximum intermittentoutput rating is 500 volts at 200 milliamperesand 6.3 volts at 7 amperes.The circuit of the supply is shown in figure44 and is designed around a voltagedoublertype power transformer ( StancorP-8336) having a 117 volt, 280 milliamperesecondary and three 6.3 volt filamentwindings. The transformer has a 140 wattcore and is conservatively rated. In quadruplerservice, 100 watts of power may be takenfrom the high voltage secondary winding providedthe filament current drain is held to40 watts. Under these limitations, the transformercan run for an hour or so beforethe core becomes warm to the touch.Four replacement-type selenium rectifiersare used in a simple quadrupler circuit. Although500 rna. rectifiers are shown, 200 rna.units will work as well. Five ohm surge resistorsare employed to limit the starting currentof the filter system. Regulation of theT15Y3-GTFigure 42SCHEMATIC, LIGHT DUTY SUPPLYT ,-350 - 0 - 350 volts at 50 milliamperes, 5volts at 2 amperes, 6.3 volts at 3 amperes11tep/acement 11 transformer. 'Tz-6.3 volts at 1.2 amp. Stancor P-6134CH-4.5 henry at 50 mo. Stancor C-1706
717T1Figure 43100 WATT "ECONOMY MODEL"POWER SUPPLYEmploying a TV-type voltage doubler transformerand inexpensive selenium rectifiers,this supply delivers 500 volts at 200 milliampereswith good regulation.The selenium rectifiers are mounted on along 6-32 bolt fastened to two upright postsmounted at each front corner of the chassis.supply is good, the voltage dropping from600 volts at no load to 500 volts at maximumcurrent drain.The supply is built upon a steel chassismeasuring 9" x 6" x 2". The filter capacitorsare mounted in the under-chassis area onphenolic tie-point strips. All transformer leadsare left their full length so that the transformerwill be in usable shape in the eventit is eventually used in a different piece ofequipment.32-13 1500 Volt,425 MilliamperePower SupplyOne of the most popular and also one ofthe most convenient power ranges for amateurequipment is that which can be suppliedfrom a 1500 volt power unit with a currentcapability of about 400 ma. The r-f amplifierof an A-M phone transmitter ( 1500 voltsat 250 ma.) capable of 375 watts input andits companion modulator ( 1500 volts at 20-200 ma.) can both be run from a supply ofthis rating. The use of this supply for SSBwork will permit a p.e.p. of about 600 watts( 1500 volts at 400 ma.) with the new lowvoltage, high current RCA 7094 tetrodes.This voltage will be found to be very economicalwhen the cost of power supply componentsis computed. A jump in supply voltageto 2000 will almost double the cost ofthe various components. Unless full kilowattoperation is intended, 1500 volts is a veryconvenient and relatively economical compromisevoltage.11sv. rv"'-----;:::::;=.;:::::::::::=::;-----1:~6.3 v.REGULATIONLOAD(MA) VOLTS0 600100 550200 saoFigure 44SCHEMATIC, ECONOMY SUPPLYT1-l 17 volt secondary rated at 280 ma fordoubler service. Three 6.3 volt fi/a;,entwindings. Stancor P-8336.CH,-8.5 henry at 200 ma. Stancor C-1721.SR,-SR,-;-500 ma. selenium rectifier. SarkesTarz•an or equivalent.The schematic of a typical supply is showni~ figure 46. Primary power source may bee1~her 115 or 230 volts, the latter providingslightly better power supply regulation. Thetwo trans£ormer primaries are connected inseries for 230 volt operation, or in parallelfor .115 volt operation. In addition the primanesmay be connected in series for halfvoltageoperation on 115 volts as shown. Thesupply provides 750 volts at 400 ma. underthis operating condition.For optimum dynamic voltage regulationunder varying loads such as imposed by sidebandor class B modulator equipment theoutput filter capacitor of the supply shouldFigure 45UNDER-CHASSIS VIEWOF ECONOMY SUPPLYTwo 20 yfd., 450 volt capacitors are connected1n parallel for each 80 !lfd. unit.
718 Power Supplies<strong>THE</strong> <strong>RADIO</strong>RYI 866.._ __ ,_ __ ,_o ,~oo/7~ov.AT 425 MA.llli.sow35";owFigure 46SCHEMATIC, 1500 VOLT, 425 MILLIAMPERE SUPPLY1,_1110- 0- 1710 volts, 425 ma. 115/230 volt primary. Chicago P-1512T.-2.5 volt, 10 ampere, 10 KV insulation. Stancor P-3060CHJ-6 henry at 300 ma., CCS. Chicago R-63RFC-"Hash" suppression choke. Millen 77866be as large as is practical. Occasionally 601-'fd., 2000 volt capacitors can be picked upon the "surplus" market for a few dollars,although their new price would give mostamateurs pause for thought. An inexpensiveand reliable substitute may be made up of agroup of replacement-type tubular electrolyticcapacitors connected in series-parallel as shownin the schematic. Eight 30 1-'fd., 450 voltcapacitors connected in series parallel willprovide an effective value of 15 p;fd., at aworking voltage of 1800. This is the minimumvalue suitable for sideband operation.Sixteen capacitors will provide 30 f'fd., at1800 volts.A power supply of this type should bebuilt upon a heavy steel chassis, and all wiringmust be done with 10,000 volt TV-typeplastic insulated wire. R-F chokes should beplaced in the plate leads of the mercury vaporrectifiers as shown, to reduce the tendencythese tubes have of breaking into oscillationover a portion of the operating cycle. Oscillationot this type will produce a 120 cycle"buzz" on the sidebands of the signal. Theparasitic is eliminated by the use of thechokes.32-14 A Dual VoltageTransmitter SupplyThe majority of high voltage transformershave tapped secondary windings, similar tothe transformer shown in the schematic offigure 47. Separate rectifier and filter systemsmay be used with the transformer to providetwo different output voltages provided the~otal wattage drain from the supply does notexceed the wattage rating of the transformer.The drain may be divided between the twosupply systems in any manner desired. Theintermittent rating of Tz (figure 4 7) is 750watts and the continuous duty rating is 600watts. Under CCS rating, the supply can provide(for example) 2000 volts at 160 rna.for the operation of an 813 r-f amplifier at320 watts input, and 1750 volts at 20-150rna. for the operation of 811-A class B modulators.Under intermittent duty rating, the813 amplifier can run at 400 watts input(phone) and 500 watts ( c-w) without overloadingthe supply.A remote switch is used to energize theplate circuit relay of the supply. An auxiliaryantenna relay is also operated by the "transmit"switch.32-15 A KilowattPower SupplyShown in figure 48 is the schematic of apower supply capable of delivering 2500 voltsat a continuous current drain of 500 milliamperes,or 700 milliamperes with an intermittentload. The supply is designed to powera kilowatt amplifier operating at 2500 voltsand 400 rna., in conjunction with a 500 wattmodulator operating at 2500 volts at a vary-
<strong>HANDBOOK</strong>Special Supplies 719CH1(MOD,)'-1---1--0+ 1750 v.AT 150 MA.R1(R.F.)"--1.----..---o + 2000 v.R~AT 160MA.TRANSMITSWITCHANTENNARELAYFigure 47DUAL VOLTAGE POWER SUPPLYr,, T,--2.5 volts at 5 amperes. Stancor P-6733T,-2400-2700 volts each side center tap at 375 ma. ICAS. Stancor P-8032CH,-3 to 17 henry, 300 ma. Stancor C-1403CHz----8 henry, 300 ma., Stancor C-1 4 73R" R,-70K., 700 wattRFC,, RFC,-"Hash" suppression choke. J. W. Miller 7865 twin chokes (2 req.)RY,-SPST relay, 715 volt coil.ing current drain of 50-300 rna. Specifically,the supply is employed with a transmitterhaving a pair of 4-250A tetrode tubes in theclass C stage, and a pair of 810 modulatortubes. For sideband work, the supply may beused to power a 1750 watt p.e.p. linear amplifier,such as the 4CX-1000A amplifier shownin an earlier chapter.Because the total weight of the componentsis over 150 pounds, the supply should bebuilt directly on the bottom of a relay rackinstead of upon a steel chassis.The r-f hash suppression chokes RFC andRFC, are fastened directly to the high voltageterminals of the plate transformer. The two872-A rectifier tubes are so located that theleads from the r-f chokes to the plate capsare only about three inches long.A 0.15 ,_,fd., 5000 volt paper capacitor isused to resonate the filter choke to approximately120 cycles at a bleeder current of 25milliamperes. When full load current isdrawn, the inductance of the filter chokedrops, detuning the parallel resonant circuit.Improved voltage regulation is gained by thisaction; the no load voltage increases only200 volts over the full load voltage.872A·sT 1-2900-0-2900 VOLTS AT 700 MA., !CAS. CHICAGO P-2126T2.-5 VOLTS, 10 AMP., CHICAGO F-5JOHCH 1-6 HENRIES, 700 MA. CHICAGO R-67C!-0.15.UF, 5000-VOLT.Cz.- THREE 4-uF 3000-VOLTR 1- 100,000 OHMS, 2.00-WATT.Rz.-ELEVEN 0.5 MEG. 2.-WATT RESISTORS IN SERIESRY I- DPST RELAY, 110 V. COl L, 2.0 A. CONTACTS. POTTER 4 BRUMFIELDPRlARFCl, RFC2- HASH FILTER, J.W. MILLER co. N° 7868Figure 48HIGH VOLTAGE POWER SUPPLY
CHAPTER THIRTY-THREEWith a few possible exceptions, such asfixed air capacitors, neutralizing capacitorsand transmitting coils, it hardly pays one toattempt to build the components required forthe construction of an amateur transmitter.This is especially true when the parts are ofthe type used in construction and replacementwork on broadcast receivers, as mass productionhas made these parts very inexpensive.Transmitters Those who have and wish tospend the necessary rime caneffect considerable monetary saving in theirtransmitters by building them from the componentparts. The necessary data is givenin the construction chapters of this handbook.To many builders, the construction is asfascinating as the operation of the finishedtransmitter; in fact, many amateurs get somuch satisfaction out of building a well-performingpiece of equipment that they spendmore time constructing and rebuilding equipmentthan they do operating the equipmenton the air.72033-1 ToolsBeautiful work can be done with metalchassis and panels with the help of only afew inexpensive tools. The time required forconstruction, however, will be greatly reducedif a fairly complete assortment of metal-workingtools is available. Thus, while an array oftools will speed up the work, excellent resultsmay be accomplished with few tools, if onehas the time and patience.The investment one is justified in makingin tools is dependent upon several factors. Ifyou like to tinker, there are many tools usefulin radio construction that you would probablybuy anyway, or perhaps already have,such as screwdrivers, hammer, saws, square,vise, files, etc. This means that the moneytaken for tools from your radio budget can beused to buy the more specialized tools, suchas socket punches or hole saws, taps and dies,etc.The amount of construction work one doe5determines whether buying a large assortment
Tools 721-Aof tools is an economical move. It also determinesif one should buy the less expensivetype offered at surprisingly low prices by thefamiliar mail order houses, "five and ten"stores and chain auto-supply stores, or whetherone should spend more money and get firstgradetools. The latter cost considerably moreand work but little better when new, but willoutlast several sets of the cheaper tools.Therefore they are a wise investment for theexperimenter who does lots of constructionwork (if he can afford the initial cash outlay).The amateur who constructs only an occasionalpiece of apparatus need not be so concernedwith tool life, as even the cheaper grade toolswill last him several years, if they are givenproper care.The hand tools and materials in the accompanyinglists will be found very useful aroundthe home workshop. Materials not listed butordinarily used, such as paint, can best bepurchased as required for each individual job.ESSENTIAL HAND TOOLS ANDMATERIALS1 Good electric soldering iron, about 100watts,Spool rosin-core wire solderEach large, medium, small, and midgetscrewdriversGood hand drill (eggbeater type), preferablytwo-speed1 Pair regular pliers, 6 inchPair long nose pliers, 6 inchPair cutting pliers (diagonals), 5 inch or6 inch1 Ys-inch tube-socket punch1 "Boy Scout" knife1 Combination square and steel rule, 1 foot1 Yardstick or steel pushrule1 Scratch awl or ice pick scribe1 Center punch1 Dozen or more assorted round shank drills(as many as you can afford between no.50 and 14 or Ys inch, depending uponsize of hand drill chuck)Combination oil stoneLight machine oil (in squirt can)Friction tape1 Hacksaw and blades1 Medium file and handle1 Cold chisel ( V2 inch tip)1 Wrench for socket punch1 HammerHIGHLY DESIRABLE HAND TOOLSAND MATERIALS1 Bench vise (jaws at least 3 inch)1 Spool plain wire solder1 Carpenter's brace, ratchet type1 Square-shank counters,ink bit1 Square-shank taper reamer, small1 Square-shank taper reamer, large (the tworeamers should overlap; Y2 inch and Ysinch size will usually be suitable)Ys inch tube-socket punch (for electrolyticcapacitors)1 1-3/16 inch tube-socket punch1 Ys-inch tube-socket punch1 Adjustable circle cutter for holes to 3 inch1 Set small, inexpensive, open-end wrenches1 Pair tin shears, 10 or 12 inch1 Wood chisel ( Y2 inch tip)1 Pair wing dividers1 Coarse mill file, flat 12 inch1 Coarse bastard file, round, Y2 or % inch1 Set allen and spline-head wrenches6 or 8 Assorted small files; round, half-roundor triangular, flat, square, rat-tail4 Small "C" clampsSteel wool, coarse and fineSandpaper and emery cloth, coarse, medium,and fineDuco cementFile brushUSEFUL BUT NOT ESSENTIAL TOOLSAND MATERIALS1 Jig or scroll saw (small) with metal-cuttingbladesSmall wood saw (crosscut teeth)Each square-shank drills: Ys, 7/16, and Y2inchTap and die outfit for 6-32, 8-32, 10-32and 10-24 machine screw threads4 Medium size "C" clampsLard oil (in squirt can)KeroseneEmpire clothClear lacquer ("industrial" grade)Lacquer thinnerDusting brushPaint brushesSheet celluloid, Lucite, or polystyrene1 Carpenter's plane1 Each "Spintite" wrenches, ;4, 5/16, 11/32to fit the standard 6-32 and 8-32 nutsused in radio workScrewdriver for recessed head type screwsThe foregoing assortment assumes that theconstructor does not want to invest in themore expensive power tools, such as drill press,
<strong>THE</strong> <strong>RADIO</strong>Figure 1SOFT ALUMINUMSHEET MAY BE CUTWITH HEAVYKITCHEN SHEARSgrinding head, etc. If power equipment is purchased,obviously some of the hand tools andaccessories listed will be superfluous. A drillpress greatly facilitates construction work,arid it is unfortunate that a good one costs asmuch as a small transmitter.Not listed in the table are several specialpurposeradio tools which are somewhat of aluxury, but are nevertheless quite handy, suchas various around-the-corner screwdrivers andwrenches, special soldering iron tips, etc.These can be found in the larger radio partsstores and are usually listed in their mail ordercatalogs.If it is contemplated to use the newer andvery popular miniature series of tubes ( 6AK5,6C4, 6BA6, etc.) in the construction of equipmentcertain additional tools will be requiredto mount the smaller components. Miniaturetube sockets mount in a %-inch hole, while9-pill sockets mount in a %-inch hole. Greenleesocket punches can be obtained in thesesizes, or a smaller hole may be reamed to theproper size. Needless to say, the punch ismuch the more satisfactory solution. Mountingscrews for miniature sockets are usually ofthe 4-40 size.Metal Chassis Though quite a few more toolsand considerably more timewill be required for metal chassis construction,much neater and more satisfactory equipmentcan be built by mounting the parts on sheetmetal chassis instead of breadboards. This typeof construction is necessary when shielding ofthe appartus is required. A front panel and aback shield minimize the danger of shock andcomplete the shielding of the enclosure.Figure 2CONVENTIONALWOOD EXPANSIONBIT IS EFFECTIVE INDRILLING SOCKETHOLES IN REYNOLDSDO-IT-YOURSELFALUMINUM
<strong>HANDBOOK</strong>Material 723-AFigure 3SOFT ALUMINUMTUBING MAY BEBENT AROUNDWOODEN FORMBLOCKS. TO PREVENT<strong>THE</strong> TUBE FROMCOLLAPSING ONSHARP BENDS, IT ISPACKED WITHWET SAND.33-2The MaterialElectronic equipment may be built uponfoundation of wood, steel or aluminum. Thechoice of foundation material is governed by therequirements of the electrical circuit, the weightof the components of the assembly, and the financialcost of the project when balanced againstthe pocketbook contents of the constructor.Breadboard The simp!est method of constructingequipment is to lay itout in breadboard fashion, which consists offastening the various components to a boardof suitable size with wood screws or machinebolts, arranging the parts so that importantleads will be as short as possible.Breadboard construction is suitable for testingan experimental layout, or sometimes forassembling an experimental unit of test equipment.But no permanent item of station equipmentshould be left in the breadboard form.Breadboard construction is dangerous, sincecomponents carrying dangerous voltages areleft exposed. Also, breadboard construction isnever suitable for any r-f portion of a transmitter,since it would be substantially impossibleto shield such an item of equipment forthe elimination of TVI resulting from harmonicradiation.Dish type construction is practically thesame as metal chassis construction, the maindifference lying in the manner in which thechassis is fastened to the panel.Special For high-powered r-f stages,Frameworks many amateur constructors preferto discard the more conventionaltypes of construction and employ insteadspecial metal frameworks and bracketswhich they design specially for the parts whichthey intend to use. These are usually arrangedto give the shortest possible r-f leads and tofasten directly behind a relay rack panel bymeans of a few bolts, with the control shaftsFigure 4A WOODWORKINGPLANE MAY BE USEDTO SMOOTH ORTRIM <strong>THE</strong> EDGES OFREYNOLDSDO-IT-YOURSELFALUMINUM STOCK.
724-A Workshop Practice<strong>THE</strong> <strong>RADIO</strong>DESK~----- 6'0" --I. legs and stringersaluminumangle W'xW'2. Top-flush door3. Shelves- ¥.1" plywoodFigure 6TVI ENCLOSURE MADE FROMSINGLE SHEET OFPERFORATED ALUMINUMReynolds Metal Co. "Do-it-yourself" aluminumsheet may be cut and folded to form TVIproofenclosure. One-hal# inch lip on edges isbolted to center section with 6-32 machinescrews.Figure SINEXPENSIVE OPERATING DESK MADEFROM ALUMINUM ANGLE STOCK, PLYWOOD AND A FLUSH-TYPE DOORprojecting through corresponding holes in thepanel.Working withAluminumThe necessity of employing"electrically tight enclosures"for the containment of TVIproducingharmonics has led to the generaluse of aluminum for chassis, panel, and enclosureconstruction. If the proper type ofaluminum material is used, it may be cut andworked with the usual woodworking toolsfound in the home shop. Hard, brittle aluminumalloys such as 24ST and 61ST should beavoided, and the softer materials such as 2Sor Y:zH should be employed.A new market product is Reynold's Do-ityourselfaluminum, which is being distributedon a nationwide basis through hardware stores,lumber yards and building material outlets.This material is an alloy which is temper selectedfor easy working with ordinary tools.Aluminum sheet, bar and angle stock may beobtained, as well as perforated sheets for ventilatedenclosures.Figures 1 through 4 illustrate how this softmaterial may be cut and worked with ordinaryshop tools, and fig. 5 shows a simple operatingdesk that may be made from aluminum anglestock, plywood and a flush-type six foot door.33-3TV I-ProofEnclosuresArmed with a right-angle square, a tin-snipsand a straight edge, the home constructor willfind the assembly of aluminum enclosures aneasy task. This section will show simple constructionmethods, and short cut·s in producingenclosures.The simplest type of aluminum enclosureis that formed from a s•ingle sheet of perforatedmaterial as shown in figure 6. Thetop, sides, and back of the enclosure are ofone piece, complete with folds that permitthe formed enclosure to be bolted togetheralong the edges. The top area of the enclosureshould match the area of the chassis to ensurea close fit. The front edge of the enclosureis attached to aluminum angle stripsthat are bolted to the front panel of theunit; the sides and back can either be boltedto matching angle S•trips affixed to the chassis,or may s'imply be attached to the edge of thechassis with s.elf-tapping sheet metal screws.Enclosures of this type are used on the allbandtransmitter described in chapter 31.A more sophisticated enclosure is shownin figure 7. In this assembly aluminum anglestock is cut to length to form a frameworkupon which the individual sides, back, andtop of the enclosure are bolted. For greateststrength, small aluminum gusset plates shouldbe affixed in each corner of the enclosure.The complete assembly may be held togetherby no. 6 sheet metal screws.
<strong>HANDBOOK</strong>Openings 725-ARegardless of the type of enclosure to bemade, care should be taken to ensure that alljoints are square. Do not assume that all prefabricatedchassis and panel are absolutely trueand square. Check them before you start toform your shield as any dimensional errors inthe foundation will cause endless patching andcutting after your enclosure is bolted together.Finally, be sure that paint is removed fromthe panel and chassis at the point the enclosureattaches to the foundation. A clean,metallic contact along the seam is requiredfor maximum harmonic suppression.33-4 EnclosureOpeningsOpenings into shielded enclosures may bemade simply by covering them by a piece ofshielding held in pla:ce by sheet metal screws.Openings through vertical panels, however,usually require a bit more attention to preventleakage of harmonic energy through thecrack of the door which is supposed to sealthe opening. A simple way to provide a panelopening is to employ the Bud ventilated doorrack panel model PS-814 or 815. The grilleopening in this panel has holes small enoughin area to prevent sel'ious harmonic leakage.The actual door opening, however, does notseal tightly enough to be called TVI-proof.In areas of high TV signal strength wherea minimum of operation on 28 Me. is contemplated,the door is satisfactory as is. ToaccompHsh more complete harmonic suppression,the edges of the opening should be linedwith preformed contact finger stock manufacturedby Eitel-McCullough, Inc., of San Bruno,Calif. Eimac finger stock is an excellent meansof providing good .contact continuity whenusing components with adjustable or movingcontact surfaces, or in acting as electrical"weatherstrip" around access doors in enclosures.Harmonic leakage through such a sealedopening is reduced to a negligible level. Themating surface to the finger stock should befree of paint, and should provide a good electl'icalconnection to the stock.A second method of re-establishing electvicalcontinuity ac!'oss an access port is roemploy Metex shielding around the matingedges of the opening. Metex is a flexible knittedwire mesh which may be obtained invarious sizes and shapes. This r-f gasket materialis produced by Metal Textile Corp.,Roselle, N.J. Metex is both flexible and resilientand conforms to irregularities in matingsurfaces with a minimum of closing pressure.Figure 7TVI-PROOF ENCLOSURE" BUILT OFPERFORATED ALUMINUM SHEETAND ANGLE STOCK33-5 Summationof the ProblemThe creation of r-f energy is accompaniedby harmonic generation and leakage of fundamentaland harmonic energy from the generatorsource. For practical purposes, radio frequencypower may be considered as a form of bothelectrical and r-f energy. As electrical energy,it will travel along any convenient conductor.As r-f energy, it will radiate directly from thesource or from any conductor connected to thesource. In view of this "dual personality" ofr-f emanations, there is no panacea for all£orms of r-f energy leakage. The cure involvesboth filtering and shielding: one to block thepaths of conducted energy, the other to preventthe leakage of radiated energy. The propercombination of filtering and shielding can reducethe radiation of harmonic energy from asignal source some 80 decibels. In most cases,this is sufficient to eliminate interference causedby the generation of undesirable harmonics.
726-A Workshop Practice33-6 ConstructionPracticeChassisLayoutThe chassis first should be coveredwith a layer of wrapping paper,which is drawn tightly down ona11 sides and fastened with scotch tape. Thisallows any number of measurement lines andhole centers to be spotted in the correct positionswithout making any marks on tihechassis itself. Place on it the parts to bemounted and play a game of chess with them,trying different arrangements until all the gridand plate leads are made as short as possible,tubes are clear of coil fields, r-f chokes are insafe positions, etc. Remember, especially ifyou are going to use a panel, that a good mechanicallayout often can accompany soundelectrical design, but that the electrical designshould be given first consideration.All too often parts are grouped to give asymmetrical panel, irrespective of the arrangementbehind. When a satisfactory arrangementhas been reached, the mounting holes may bemarked. The same procedure now must befollowed for the underside, always being carefulto see that there are no clashes betweenthe two (that no top mounting screws comedown into the middle of a paper capacitor onthe underside, that the variable capacitor rotorsdo not hit anything when turned, etc.).When all the holes have been spotted, theyshould be center-punched through the paperinto the chassis. Don't forget to spot holes forleads which must also come through thechassis.For transformers which have lugs on thebottoms, the clearance holes may be spottedby pres•sing the transformer on a piece of paperto obtain impressions, which may thenbe transferred to the chassis.Punching In cutting socket holes, one canuse either a fly-cutter or socketpunches. These punches are easy to operateand only a few precautions are necessary. Theguide pin should fit snugly in the guide hole.This increases the accuracy of location of thesocket. If this is not of great importance, onemay well use a drill of 1/32 inch larger diameterthan the guide pin. Some of the puncheswill operate without guide holes, but the latteralways make the punching operntions simplerand easier. The only other precaution is to besure the work is properly lined up before applyingthe hammer. If this is not done, thepunch may slide sideways when you strike andthus not only shear the chassis but also taker!JOCOOO~B 8§ ~g o,troQQ_Qdj0DRILL HOLES SLIGHTLYINSIDE DASHED OUTLINEOF DESIRED HOLE.<strong>THE</strong> <strong>RADIO</strong>IIY.>."""'"-'~""_j®BREAK OUTPIECE INSIDEDRILL HOLES.MAKING RECTANGULAR CUTOUTFigure 8D0FILESMOOTHoff part of the die. This is easily avoided byalways making sure that the piece is parallelto the faces of the punch, the die, and thebase. The latter should be an anvil or otheroolid base of heavy material.A punch by Greenlee forces socket holesthrough the chassis by means of a screw turnedwith a wrench. It is noiseless, and works muchmore easily and accurately than most others.The male part of the punch should be placedin the vise, cutting edge up and the femaleportion forced against the metal with a wrench.These punches can be obtained in sizes toaccommodate all tube sockets and even largeenough to be used for meter holes. In theoctal socket sizes they require the use of a Ysinch center hole to accommodate the bolt.TransformerCutoutsCutouts for transformers andchokes are not so simply handled.After marking off the partto be cut, drill about a Y4-inch hole on eachof the inside corners and tangential to theedges. After burring the holes, clamp the pieceand a block of cast iron or steel in the vise.Then, take your burring chisel and insert it inone of the corner holes. Cut out the metal byhitting the chisel with a hammer. The blowsshould be light and numerous. The chisel actsagainst the block in the same way that the twoblades of a pair of scissors work against eachother. This same process is repeated for theother sides. A file is used to trim up the completedcutout.Another method is to drill the four cornerholes large enough to take a hack saw blade,then saw instead of chisel. The four holes permitnice looking corners.Still another method is shown in figure 8.When heavy panel steel is used and a drillpress or electric drill is available, this is themost satisfactory method.RemovingBurrsIn both drilling and punching, aburr is usually left on the work.There are three simple ways of
<strong>HANDBOOK</strong>Construction Practice 727-Aremoving these. Perhaps the best is to take achisel (be sure it is one for use on metal) andset it so that its bottom face is parallel to thepiece. Then gently tap it with a hammer. Thisusually will make a clean job with a littlepractice. If one has access to a counterbore,this will also do a nice job. A countersink willwork, although it bevels the edges. A drill ofseveral sizes larger is a much used arrangement.The third method is by filing off theburr, which does a good job but scratches theadjacent metal surfaces badly.MountingComponentsThere are two methods in generaluse for the fastening oftransformers, chokes, and similarNUMBERED DRILL SIZESCorrect forDl-TappingDRILL a meter Clears Steel orNUMBER (in.) Screw Brasst1 .. .2282 ..... ·············--- .221 12-243 ··-- ................ .213 14-244 ····· .209 12-205 ··-· ................. .2056 -- ----- ---------- .2047 ....8-- ......... .201............. .1999 ·-- --- .19610* __ -- .193 10-3211 ····· .191 10-2412* --------------------- .18913 --· .18514 ... ---------------- .18215 .... --------------- .18016 ··-·- -- -------- .177 12-2417 ---- .17318* __ ····- .169 8-3219 -- .166 12-2020 ······················ .16121•--- ............. .159 10-3222 ···-- ................ .1572324--- ............ .154.15225* ····- -- ............ .149 10-2426 ...................... .14727 ·-- ................ .14428• --- ................. .140 6-3229*. --- ................ .136 8-3230 ····· ·····----------- .12831 .. ----------- .12032 .. --- ............ .11633* __ -- ····· .113 4-36 4-4034 --· .11135• --------------····-·· .110 6-3236 --------·-············ .1063738-----· .104-------------------- .10239* -·----------------·-· .100 3-4840 ·-------------- .09841 ---·---··--- .09642• .093 4-36 4-4043 ---------------······- .089 2-5644 ..... ·····--·----·- .08645*-- .082 3-48tUse next size larger drill for tappingbakelite and similar composition materials(plastics, etc.).*Sizes most commonly used in radio construction.Figure 9pieces of apparatus to chassis or breadboards.The first, using nuts and machine screws, isslow, and the commercial manufacturing practiceof using self-tapping screws is gainingfavor. For the mounting of small parts suchas resistors and capacitors, "tie points" arevery useful to gain rigidity. They also contributematerially to the appearance of finishedapparatus.Rubber grommets of the proper size, placedin all chassis holes through which wires areto be passed, will give a neater appearing joband also will reduce the possibility of shortcircuits.Soldering Making a strong, low-resistancesolder joint does not mean justdropping a blob of solder on the two parts tobe joined and then hoping that they'll stick.There are several definite rules that must beobserved.All parts to be soldered must be absolutelyclean. To clean a wire, lug, or whatever it maybe, take your pocket knife and scrape it thoroughly,until fresh metal is laid bare. It is notenough to make a few streaks; scrape until thepart to be soldered is bright .Make a good mechanical joint before applyingany solder. Solder is intended primarilyto make a good electrical connection; mechanicalrigidity should be obtained by bending tht'wire into a small hook at the end and nippingit firmly around the other part, so that it willhold well even before the solder is applied.Keep your iron properly tinned. It is impossibleto get the work hot enough to takethe solder properly if the iron is dirty. To tinyour iron, file it, while hot, on one side untila full surface of clean metal is exposed. Immediatelyapply rosin core solder until a thinlayer flows completely over the exposed surface.Repeat for the other faces. Then take aclean rag and wipe off all excess solder androsin. The iron should also be wiped frequentlywhile the actual construction is going on; ithelps prevent pitting the tip.Apply the solder to the work, not to theiron. The iron should be held against the parts~o be joined until they are thoroughly heated.The solder should then be applied against theparts, and the iron should be held in placeuntil the solder flows smoothly and envelopesthe work. If it acts like water on a greasyplate, and forms a ball, the work is not sufficientlyclean.The •ompleted joint must be held perfectlystill until the solder has had time to solidify.If the work is moved before the solder has be-
728-A Workshop Practice-COILWINDING COIL ON INSULATING FORMWINDING "AIR-SUPPORTED" COILFigure 10Figure 11come completely solid, a "cold" joint will result.This can be identified immediately, becausethe solder will have a dull "white" appearancerather than one of shiny "silver."Such joints tend to be of high resistance andwill very likely have a bad effect upon a circuit.The cure is simple, merely reheat thejoint and do the job correctly.Wipe away all surplus flux when the jointhas cooled if you are using a paste type flux.Be sure it is non-corrosive, and use it withplain (not rosin core) solder.Finishes If the apparatus is constructed ona painted chassis (commonly availablein black wrinkle and gray wrinkle), thereis no need for application of a protective coatingwhen the equipment is finished, assumingthat you are careful not to scratch or mar thefinish while drilling holes and mounting parts.However, many amateurs prefer to use unpainted(zinc or cadmium plated) chassis, becauseit is much simpler to make a chassisground connection with this type of chassis.A thin coat of clear "linoleum" lacquer maybe applied to the whole chassis after the wiringis completed to retard rusting. In localitiesnear the sea coast it is a good idea to lacquerthe various chassis cutouts even on a paintedchassis, as rust will get a good start at thesepoints unless the metal is protected where rhedrill or saw has exposed it. If too thick a coatis applied, the lacquer will tend to peel. Itmay be thinned with lacquer thinner to permitapplication of a light coat. A thin coat willadhere to any clean metal surface that is nottoo shiny.An attractive dull gloss finish, almost velvetycan be put on aluminum by sand-blastingit with a very weak blast and fine particlesand then lacquering it. Soaking the aluminumin a solution of lye produces somewhat thesame effect as a fine grain sand blast.There are also several brands of dull glossblack enamels on the market which adherewell to metals and make a nice appearance.Airdrying wrinkle finishes are sometimes successful,but a bake job is usually far better.Wrinkle finishes, properly applied, are verydumble and are pleasing to the eye. If youlive in a large community, there is probablyan enameling concern which can wrinkle yourwork for you at a reasonable cost. A very attractivefinish, for panels especially, is tospray a wrinkle finish with aluminum paint.In any painting operation (or plating, either,for that matter) , the work should be verythoroughly cleaned of all greases and oils.To protect brass from tarnish, thoroughlycleanse and remove the last trace of grease bythe use of potash and water. The brass mustbe carefully rinsed with water and dried; butin doing it, care must be taken not to handleany portion with the bare hands or anythingelse that is greasy. Then lacquer.Winding Coils Coils are of two general types,those using a form and "airwound"types. Neither type offers any particularconstructional difficulties. Figure 10 illustratesthe procedure used in form windinga coil. If the winding is to be spaced, thespacing can be done either by eye or a stringor another piece of wire may be wound simultaneouslywith the coil wire and removed afterthe winding is in place. The usual procedureis to clamp one end of the wire in a vise, attachingthe other end to the coil form and withthe coil form in hand, walk slowly towards thevise winding the wire but at the same timekeeping a strong tension on the wire as theform is rotated. After the coil is wound, ifthere is any possibility of the turns slipping,the completed coil is either entirely coatedwith a coil or Duco cement or cemented 1nthose spots where slippage might occur.
Figure 12GOOD SHOPLAYOUT AIDSCAREFULWORKMANSHIPBuilt in a corner of agarage, this shop has allfeatures necessary forelectronic work. Test instrumentsare arrangedon shelves above bench.Numerous outlets reduce11haywire" produced bytangled line cords. Notshown in/icture are drillpress an sander at endof left benchV-h-f and u-h-f coils are commonly woundof heavy enameled wire on a form and thenremoved from the form as in figure 11. Ifthe coil is long or has a tendency to buckle,strips of polystyrene or a similar material maybe cemented longitudinally inside the coil. Dueallowance must be made for the coil springingout when removed from the form, when selectingthe diameter of the form.On air wound coils of this type, spacing betweenturns is accomplished after removalfrom the form, by running a pencii, the shankof a screwdriver or other round object spirallybetween the turns from one end of the coil tothe other, again making due allowance forspring.Air-wound coils, approaching the appearanceof commercially manufactured ones, can beconstructed by using a round wooden formwhich has been sawed diagonally from endto end. Strips of insulating material are temporarilyattached to this mandrel, the wirethen being wound over these strips with thedesired separation between turns and cementedto the strips. When dry, the split mandrel maybe removed by unwedging it.33-7 Shop LayoutThe size of your workshop is relatively unimportantsince the shop layout will determineits efficiency and the ease with whichyou may complete your work.Shown in figure 12 is a workshop builtinto a lO'xlO' area in the corner of a garage.The workbench is 32" wide, made up of fourstrips of 2"x8" lumber supported on a solid729-Aframework made of 2"x4" lumber. The topof the workbench is covered with hard-surfaceMasonite. The edge of the surface is protectedwith aluminum "counter edging" strip,obtainable at large hardware stores. Two woodenshelves 12" wide are placed above thebench to hold the various items of test equipment.The shelves are bolted to the wall studswith large angle brackets and have woodenend pieces. Along the edge of the lower shelfa metal "outlet strip" is placed that has an115-volt outlet every six inches along itslength. A similar strip is run along the backof the lower shelf. The front strip is usedfor equipment that is being bench-tested, andthe rear strip powers the various items of testequipment placed on the shelves.At the left of the bench is a storage binfor small components. A file cabinet can beseen at the right of the photograph. This necessaryitem holds schematics, transformer datasheets, and other papers that normally arelost in the usual clutter and confusion.The area below the workbench has twostorage shelves which are concealed by slidingdoors made of Y.,-inch Masonite. Heavier tooJs,and large components are stored in this area.On the floor and not shown in the photographis a very necessary item of shop equipment:a large trash receptacle.A compact and efficient shop built in onehalfof a wardrobe closet is shown in figure13. The workbench length is four feet. Thewp is made of 4"x6" lumber sheathed withhard surface Masonite and trimmed with"counter edging" strip. The supporting struc-
730-A Workshop PracticeFigure 13COMPLETE WORKSHOP IN A CLOSET!Careful layout permits complete electronicworkshop to be placed in one-half of a wardrobecloset. Work bench is built atop an unpaintedthree-shelf bookcase.ture is made from an unpainted three-shelfbookcase. A 2"x2" leg is placed under thefrom corners of the bench to provide maximumstability.Atop the bench, a small wooden frameworksupports needed items of test equipment anda single shelf contains a 115-volt '"outlet strip."The instruments at the top of the photo areplaced on the wardrobe shelf.When not in use, the doors of the wardrobeare closed, concealing the workshop completelyfrom view.
CHAPTER THIRTY-FOURElectronicTest Equipment721-BAll amateur stations are required by law tohave certain items of test equipment avuihLblewithin the station. A c-w station is requiredto have a frequency meter or other means inaddition to the transmitter frequency controlfor insuring that the transmitted signal is ona frequency within one of the frequency bandsassigned for such use. A radiophone station isrequired in addition to have a means of determiningthat the transmitter is not being modulatedin excess of its modulation capability,and in any event not more than 100 per cent.Further, any station operating with a powerinput greater than 900 watts is required tohave a means of determining the exact inputto the final stage of the transmitter, so as toinsure that the power input to the plate circuitof the output stage does not exceed 1000watts.The additional test and measurement equipmentrequired by a station will be determinedby the type of operation contemplated. It isdesirable that all stations have an accuratelycalibrated volt-ohmmeter for routine transmitterand receiver checking and as an assistancein getting new pieces of equipment into operation.An oscilloscope and an audio oscillatormake a very desirable adjunct to a phone stationusing AM or FM transmission, and are anecessity if single-sideband operation is contemplated.A calibrated signal generator is almosta necessity if much receiver work is contemplated,although a frequency meter of LMor BC-221 type, particularly if it includes internalmodulation, will serve in place of thesignal generator. Extensive antenna work 1nvariablyrequires the use of some type of fieldstrengthmeter, and a standing-wave meter ofsome type is very helpful. Lastly, if much v-h-fwork is to be done, a simple grid-dip meterwill be found to be one of the most used itemsof test equipment in the station.34-1 Voltage,Current and PowerThe measurement of voltage and current inradio circuits is very important in proper maintenanceof equipment. Vacuum tubes of thetypes used in communications work must beoperated within rather narrow limits in regardto filament or heater voltage, and they mustbe operated within certain maximum limits inregard to the voltage a~d current on otherelectrodes.Both direct current and voltage are mostcommonly measured with the aid of an instrumentconsisting of a coil that is free to rotatein a constant magnetic field (d'Arsonval typeinstrument). If the instrument is to be used forthe measurement of current it is called an ammeteror milliammeter. The current flowingthrough the circuit is caused to flow throughthe moving coil of this type of instrument. Ifthe current to be measured is greater than 10milliamperes or so it is the usual practice tocause the majority of the current to flowthrough a by-pass resistor called a shunt, onlya specified portion of the current flowingthrough the moving coil of the instrument.The calculation of shunts for extending therange of d-e milliammeters and ammeters isdiscussed in Chapter Two.
722-B Test Equipment<strong>THE</strong> <strong>RADIO</strong>+fO +100 +2.50 +1000fOK lOOK 250K I MEG.@+fO +100 +250 +1000tOK 90K 150 K 750KfOK@©Figure 1MULTI-VOLTMETER CIRCUITS(A) shows o circuit whereby individual multiplierresistors are used lor each range. (8) isthe more economical 11 Series multipliet 11 cit ..cuit. The same number ol resistors is required,but those for the higher ranges haveJess resistance, and hence are Jess expensivewhen precision wirewound resistors areto be used. (C) shows a circuit essentiallythe same as at (A), except that a rangeswitch is used. With a 0-500 d-e microammetersubstituted for the 0-1 milliammetershown above, all resistor values would bemultiplied by two and the voltmeter wouldhave a "2000-ohm-per-volt" sensitivity. Simi·larly, if a 0-50 d-e microammeter were to beused, all resistance values would be multi ..plied by twenty, and the voltmeter wouldhave a sensitivity of 20,000 ohms per volt.A direct current voltmeter is merely a d-emilliammeter with a multiplier resistor in serieswith it. If it is desired to use a low-rangemilliammeter as a voltmeter the value of themultiplier resistor for any voltage range maybe determined from the following formula:1000 ER=-Iwhere: R =multiplier resistor in ohmsE = desired full scale voltageI = full scale current of meter In rna.The sensitivity of a voltmeter is commonlyexpressed in ohms per volt. The higher theohms per volt of a voltmeter the greater itssensitivity. When the full-scale current drainof a voltmeter is known, its sensitiVIty ratingin ohms per volt may be determined by:1000Ohms per volt=--I+Figure 2VOLT-OHMMETER CIRCUITWith the switch In position 1, the 0-1 milliammeterwould be connected directly to theterminals. In position 2 the meter would readfrom 0-100,000 ohms, approximately, with aresistance value of 4500 ohms at half scale.(Note: The half-scale resistance value of anohmmeter using this circuit is equal to theresistance in series with the battery insidethe instrument.) The other four taps are voltageranges with 10, 50, 250, and 500 voltsfull scale.Where I is the full-scale current drain of theindicating instrument in milliamperes.Multi-RangeMetersIt is common practice to connecta group of multiplier resistorsin the circuit with a single indicatinginstrument to obtain a multi-rangevoltmeter. There are several ways of wiringsuch a meter, the most common ones of whichare indicated in figure 1. With all these methodsof connection, the sensitivity of the meterin ohms per volt is the same on all scales.With a 0-1 milliammeter as shown the sensitivityis 1000 ohms per volt.Volt-Ohmmeters An extremely useful piece oftest equipment which shouldbe found in every laboratory or radio stationis the volt.ohmmeter. It consists of a multirangevoltmeter with an additional fixed resistor,a variable resistor, and a battery. A typicalexample of such an instrument is diagrammedin figure 2. Tap 1 is used to permit use ofthe instrument as an 0-1 d-e milliammeter.Tap 2 permits accurate reading of resistors upto 100,000 ohms; taps 3, 4, 5, and 6 arefor making voltage measurements, the fullscale voltages being 10, 50, 250, and 500volts r~pectively.The 1000-ohm potentiometer is used tobring the needle to zero ohms when the terminalsare shorted; this adjustment shouldalways be made before a resistance measurementis taken. Higher voltages than 500 canbe read if a higher value of multiplier resistoris added to an additional tap on the switch.The proper value for a given full scale readingcan be determined from Ohm's law.Resistances higher than 100,000 ohms can-
<strong>HANDBOOK</strong>Ohmmeters723-Bnot be measured accurately with the circuitconstants shown; however, by increasing theohmmeter battery to 45 volts and multiplyingthe 4000-ohm resistor and 1000-ohm potentiometerby 10, the ohms scale also will bemultiplied by 10_ This would permit accuratemeasurements up to 1 megohm_0-1 d-e milliammeters are available wi~hspecial volt-ohmmeter scales which make individualcalibration unnecessary. Or, specialscales can be purchased separately and substitutedfor the original scale on the milliammeter.Obviously, the accuracy of the instrumenteither as a voltmeter or as an ammeter can beno better than the accuracy of the milliammeterand the resistors.Because volt-ohmmeters are so widely usedand because the circuit is standardized to aconsiderable extent, it is possible to purchasea factory-built volt-ohmmeter for no more thanthe component parts would cost if purchasedindividually. For this reason no constructiondetails are given. However, anyone alreadypossessing a suitable milliammeter and desirousof incorporating it in a simple voltohmmetershould be able to build one from theschematic diagram and design data given here.Special, precision (accurately calibrated) multiplierresistors are available if a high degreeof accuracy is desired. Alternatively, goodquality carbon resistors whose actual resistancehas been checked may be used as multiplierswhere less accuracy is required.Medium- andLow-RangeOhmmeterMoM ohmmeters, including theone just described, are notadapted for accurate measurementof low-resistances-in theneighborhood of 100 ohms, for instance.The ohmmeter diagrammed in figure 3 wasespecially designed for the reasonably accuratereading of resistances down to 1 ohm. Twoscales are provided, one going in one directionand the other scale going in the otherdirection because of the different manner inwhich the milliammeter is used in each case.The low scale covers from 1 to 100 ohms andthe high scale from 100 to 10,000 ohms. Thehigh scale is in reality a medium-range scale.For accurate reading of resistances over 10,000ohms, an ohmmeter of the type previouslydescribed should be used.The 1-100 ohm scale is useful for checkingtransformers, chokes, r-f coils, etc., which oftenhave a resistance of only a few ohms.The calibration scale will depend upon theinternal resistance of the particular make of1.5V.D. P.O. T.SWITCH0y.-----+4~V?7~0 n.RxLEADS0-1.!) MA .Figure 3SCHEMATIC OF A LOW-RANGEOHMMETERA description of the operation of this circuitis given in the text. With the switch in theleft position the half-scale reading of themeter will occur with an external resistanceof 1000 ohms. With the switch in the rightposition, half-scale deflection will be obtainedwith an external resistance equal tothe d-e resistance of the milliammeter (20 to50 ohms depending upon the make of instrument).1.5-ma. meter used. The instrument can becalibrated by means of a Wheatstone bridge ora few res·istors of known accuracy. The lattercan be series-connected and parallel-connectedto give sufficient calibration points. A handdrawnhand-calibrated scale can be cementedover the regular meter scale to give a directreading in ohms.Before calibrating the instrument or usingit, the test prods should always be touched togetherand the zero adjuster set accurately.Measurement ofAlternating Currentand VoltageThe measurement of alternatingcurrent andvoltage is complicatedby two factors; first, thefrequency range covered in ordinary communicationchannels is so great that calibration ofan instrument becomes extremely difficult; second,there is no single type of instrumentwhich is suitable for all a-c measurements-asthe d'Arsonval type of movement is suitablefor d-e. The d'Arsonval movement will notoperate on a-c since it indicates the averagevalue of current flow, and the average valueof an a-c wave is zero.As a result of the inability of the reliabled'Arsonval type of movement to record an alternatingcurrent, either this current must berectified and then fed to the movement, or aspecial type of movement which will opemtefrom the effective value of the cuuent can beused.For the usual measurements of power frequencya.c. ( 25-60 cycles) the iron-vane instrumentis commonly used. For audio frequencya.c. ( 50-20,000 cycles) a d'Arsonval+
724-B Test Equipment<strong>THE</strong> <strong>RADIO</strong>This means that a v.t.v.m. may be used forthe measurement of a-v-e, a-f-c, and discriminatoroutput voltages where no loading of thecircuit can be tolerated.:to---+--{Figure 4SLIDE-BACK V-T VOLTMETERBy connecting a variable source of voltagein series with the input to a conventionalv-t voltmeter, or in series with the simpletriode voltmeter shown above, a slide-backa-c voltmeter for peak voltage measurementcon be constructed. Resistor R should beabout 1000 ohms per volt used at battery B.This type of v-t voltmeter has the advantagethat it can give a reading of the actual peakvoltage of the wave being measured, withoutany current drain from the source of voltage.instrument having an integral copper oxideor selenium rectifier is usually used. Radio frequencyvoltage measurements are usually madewith some type of vacuum-tube voltmeter,while r-f current measurements are almost invariablymade with an instrument contammga thermo-couple to convert the r.f. into d.c.for the movement.Since an alternating current wave can havean almost infinite variety of shapes, it caneasily be seen that the ratios between thethree fundamental quantities of the wave(peak, r.m.s. effective, and average after rectification)can also vary widely. So it becomesnecessary to know beforehand just which qualityof the wave under measurement our instrumentis going to indicate. For the purposeof simplicity we can list the usual types ofa-c meters in a table along with the characteristicof an a-c wave which they will indicate:Iron-vane, thermocouple-r.m.s.Rectifier type (copper oxide or selenium)-average after rectification.V.t.v.m.-r.m.s., average or peak, dependingupon design and calibration.Vacuum-TubeVoltmetersA vacuum-tube voltmeter is essentiallya detector in whicha change in the signal placedupon the input will produce a change in theindicating instrument (usually a d'Arsonvalmeter) placed in the output circuit. A vacuumcubevoltmeter may use a diode, a triode, or amulti-element tube, and it may be used eitherfor the measurement of a.c. or d.c.When a v.t.v.m. is used in d-e measurementit is used for this purpose primarily becauseof the very great input resistance of the device.A-C V-T There are many different typesVoltmeters of a-c vacuum-tube voltmeters,all of which operate as sometype of rectifier to give an indication on a d-einstrument. There are two general types: thosewhich give an indication of the r-m-s value ofthe wave (or approximately this value of acomplex wave), and those which give an indicationof the peak or crest value of thewave.Since the setting up and calibration of awide-range vacuum-tube voltmeter is rathertedious, in most cases it will be best to purchasea commercially manufactured unit. Severalexcellent commercial units are on themarket at the present time; also kits for homeconstruction of a quite satisfactory v.t.v.m.are available from several manufacturers.These feature a wide range of a-c and d-e voltagescales at high sensitivity, and in additionseveral feature a built-in vacuum-tube ohmmeterwhich will give indications up to 500or 1000 megohms.Peak A-C V-T There are two common typesVoltmeters of peak-indicating vacuumtubevoltmeters. The first isthe so-called slide-back type in which a simplev.t.v.m. is used along with a conventionald-e voltmeter and a source of bucking bias inseries with the input. With this type of arrangement(figure 4) leads are connected tothe voltage to be measured and the slider resistorR across the bucking voltage is backeddown until an indication on the meter (calleda false zero) equal to that value given withthe prods shorted and the bucking voltagereduced to zero, is obtained. Then the valueof the bucking voltage (read on V) is equalto the peak value of the voltage under measurement.The slide-back voltmeter has thedisadvantage that it is not instantaneous in itsindication-adjustments must be made forevery voltage measurement. For this reason theslide-back v.t.v.m. is not commonly used, beingsupplanted by the diode-rectifier type ofpeak v.t.v.m. for most applications.High-VoltageDiode PeakVoltmeterthe constantsA diode vacuum-tube voltmetersuitable for the measurementof high values of a-c voltage isdiagrammed in figure 5. Withshown, the voltmeter has two
<strong>HANDBOOK</strong>Power Measurement 725-B2X2/8792.~ v.A.C.VOLTAGEFROMSOURCEWITHD.C.RETURNPATHC1R1:~~fUV.Figure 5SCHEMATIC OF A HIGH-VOLTAGE PEAKVOLTMETERA peak voltmeter such as diagrammedabove is convenient lor the measurementof peak voltages at fairly highpower levels from a source of moderatelylow impedance.c,-.001-JJ.fd. high-voltage micaC2-1.0JJ.fd. high-voltage paperR1-500,000 ohms (two 0.25-megohm liz-wattin series)R 2-l.O megohm (four 0.25-megohm liz-wattin series)T-2.5 v., 1.75 a. filament transformerM-0-1 d-e milliammeter5 lll-w-5-p-d-t toggle switch5-5-p-s-t toggle switch(Note: c, is a by-pass around c,, the inductivereactance of which may be appreciableat high frequencies.)ranges: 500 and 1500 volts peak full scale.Capacitors C and c, should be able to withstanda voltage in excess of the highest peakvoltage to be measured. Likewise, R, and R,should be able to withstand the same amountof voltage. The easiest and least expensiveway of obtaining such resistors is to use severallow-voltage resistors in series, as shownin figure 5. Other voltage ranges can be obtainedby changing the value of these resistors,but for voltages less than several hundredvolts a more linear calibration can be obtainedby using a receiving-type diode. A calibrationcurve should be run to eliminate theappreciable error due to the high internal resistanceof the diode, preventing the capacitorfrom charging to the full peak value of thevoltage being measured.A direct reading diode peak voltmeter ofthe type shown in figure 5 will load the sourceof voltage by approximately one-half the valueof the load resistance in the circuit (R,, or R,plus R,, in this case). Also, the peak voltagereading on the meter will be slightly less thanthe actual peak voltage being measured. Theamount of lowering of the reading is determinedby the ratio of the reactance of thestorage capacitance to the load resistance. Ifa cathode-ray oscilloscope is placed across theterminals of the v.t.v.m. when a voltage is beingmeasured, the actual amount of the loweringin voltage may be determined by inspectionC+CONVENTIONALHIGH-SENSITIVITYVOLTMETERFigure 6PEAK-VOLTAGE MEASUREMENTCIRCUITThrough use of the arrangement shown aboveit is possible to make accurate measurementsof peak a-c voltages, such as across the secondaryof a modulation transformer, with aconventional d-e multi-voltmeter. CapacitorC and transformer T should, of course, be insulatedfor the highest peak voltage likely tobe encountered. A capacitance of 0.25-JJ.fd.at C has been found to be adequate. Thehigher the sensitivity of the indicating d-evoltmeter, the smaller will be the error betweenthe indication on the meter and the actualpeak voltage being measured.of the trace on the c-r tube screen. The peakpositive excursion of the wave will be slightlyflattened by the action of the v.t.v.m. Usuallythis flattening will be so small as to be negligible.An alternative arrangement, shown in figure6, is quite convenient for the measurement ofhigh a-c voltages such as are encountered inthe adjustment and testing of high-poweraudio amplifiers and modulators. The arrangementconsists simply of a 2X2 rectifier tubeand a filter capacitor of perhaps 0.25-pJd. capacitance,but with a voltage rating highenough that it is not likely to be puncturedas a result of any tests made. Cathode-rayoscilloscope capacitors, and those for electrostatic-deflectionTV tubes often have ratingsas high as 0.25 J.Lfd. at 7500 to 10,000 volts.The indicating instrument is a conventionalmulti-scale d-e voltmeter of the high-sensitivitytype, preferably with a sensitivity of 20,000 or50,000 ohms per volt. The higher the sensitivityof the d-e voltmeter used with therectifier, the smaller will be the amount offlattening of the a-c wave as a result of therectifier action.Measurementof PowerAudio frequency or radio frequencypower in a resistivecircuit is most commonly andmost easily determined by the indirect method,i.e., through the use of one of the followingformulas:P=EI P=E'/R P=I'RThese three formulas mean that if any two ofthe three factors determining power are known
726-B Test Equipment<strong>THE</strong> <strong>RADIO</strong>Figure 7100-WATT DUMMY LOAD SUITABLE FORRANGER, OR SIMILAR TYPETRANSMITTER(resistance, current, voltage) the power beingdissipated may be determined. In an ordinary120-volt a-c line circuit the above formulasare not strictly true since the power factor ofthe load must be multiplied into the resultora direct method of determining power suchas a wattmeter may be us,ed. But in a resistivea-f circuit and in a resonant r-f circuit thepower factor of the load is taken as beingunity.For accurate measurement of a-f and r-fpower, a thermogalvanometer or thermocoupleammeter in series with a non-inductive resistorof known resistance can be used. The metershould have good accuracy, and the exactvalue of resistance should be known with accuracy.Suitable dummy load resistors areavailable in various resistances in both 100and 250-watt ratings. These are virtually noninductive,and may be considered as a pureresistance up to 30 Me. The resistance ofthese units is substantially constant for allvalues of current up to the maximum dissipationrating, but where extreme accuracy is required,a correction chart of the dissipationcoefficient of resistance (supplied by the manufacturer)may be employed. This chart showsthe exact resistance for different values ofcurrent through the resistor.Sine-wave power measurements (r-f or single-frequencyaudio) may also be madethrough the use of a v.t.v.m. and a resistor ofknown value. In fact a v.t.v.m. of the typeshown in figure 6 is particularly suited to thiswork. The formula, P = E'/R is used in thiscase. However, it must be remembered thata v.t.v.m. of the type shown in figure 6 indicatesthe peak value of the a-c wave. Thisreading must be converted to the r-m-s orheating value of the wave by multiplying itby 0. 707 before substi,tuting the voltage valuein the formula. The same result can be obtainedby using the formula P = E'/2R.Thus all three methods of determining power,ammeter-resistor, voltmeter-resistor, andvoltmeter-ammeter, give an excellent crosscheckupon the accuracy of the determinationand upon the accuracy of the standards.Power may also be measured through theuse of a calorimeter, by actually measuring theamount of heat being dissipated. Through theuse of a watt-r-cooled dummy load resistor thismethod of power output determination is beingused by some of the most modern broadcaststations. But the method is too cumbersomefor ordinary power determinations.Power may also be determined photometricallythrough the use of a voltmeter, ammeter,incandescent lamp us,ed as a load resistor, anda phowgraphic exposure meter. With thismethod the exposure meter is used to determinethe relative visual output of the lamprunning as a dummy load resistor and of thelamp running from the 120-volt a-c line. Arheostat in series with the lead from the a-cline to the lamp is used to vary its light intensityto the same value (as indicated by the exposuremeter) as it was putting out as a dummyload. The a-c voltmeter in parallel with thelamp and ammeter in series with it is thenused to determine lamp power input by:P = EI. This method of power determinationis satisfactory for audio and low frequencyr.f. but is not satisfactory for v-h-f work becauseof variations in lamp efficiency due touneven heating of the filament.Dummy Loads Lamp bulbs make poor dummyloads for r-f work, in general,as they have considerable reactance above 2Me., and the resistance of the lamp varies withthe amount of current passing through it.A suitable r-f load for powers up to a fewwatts may be made by paralleling 2-watt compositionresistors of suitable value to make
<strong>HANDBOOK</strong> Measurement of Constants 727-Ba 50-ohm resistor of 5 or 10 watts dissipation.The resistors should be mounted in a compactbundle, with short interconnecting leads.For powers of 100 and 250 watts, theOhmite models D-101 and D-251 low inductanceresi5tors may be used as dummy antennas.A D-101 resistor mounted on a smallmetal chassis with a r-f ammeter and coaxialplug is shown in figure 7. The Ohmite D-250resistor may be employed in such a configurationfor higher power.A dummy load capable of taking the outputof a plate-modulated !-kilowatt transmitter isshown in figure 8. The load is made up of ten160-watt, 500-ohm Ohmite type 2409 noninductiveresistors. These resi5tors are mountedbetween two aluminum end plates measuring4 inches in diameter. The assembly is thenend mounted by means of 1" ceramic insulatorsto an aluminum chassis conuining ther-f ammeter and an S0-239 coaxial receptacle.This dummy load presents a good approximationof a 50-ohm, 1600-watt resistor upto approximately 8 Me. Above this frequency,it becomes increasingly reactive. It may beused up to 30 Me., if it is remembered that~he reading of the r-f ammeter no longer isan absolute indication of the resistive powerdissipation of the load. For use on the 20,15 and 10 meter bands, it is recommendedthat the coaxial line coupling the dummy loadto the rransmi tter either be rdtrioted to alength of two feet or less, or else be madean electrical half-wavelength long. This willinsure that a minimum of reactance will becoupled back into the transmitter.34-2 Measurementof Circuit ConstantsThe measurement of the resistance, capacitance,inductance, and Q (figure of merit) ofthe components used in communications work·can be divided into three general methods:the impedance method, the substitution or resonancemethod, and the bridge method.The Impedance The impedance method ofMethod measuring inductance andcapacitance can be likened tothe ohmmeter method for measuring resistance.An a-c voltmeter, or milliammeter in serieswith a resistor, is connected in series withthe inductance or capacitance to be measuredand the a-c line. The reading of the meter willbe inversely proportional to the impedance ofthe component being measured. After the meterhas been calibrated it will be pos,sible toR- TEN 500-0HM, 160-WATT NON-INDUCTIVE RESISTORSIN PARALLEL (OHM/TE 2.409)Figure 81.6-KILOWATT DUMMY LOAD SCHEMATICobtain the approximate value of the impedancedirectly Jirom the scale of the meter. If thecomponent is a capacitor, the value of impedancemay be taken as its reactance at themeasurement frequency and the capacitancedetermined accordingly. But the d-e resistanceof an inductor must also be taken into considerationin determining its inductance. Afterthe d-e resistance and the impedance havebeen determined, the reactance may be determinedfrom the formula: XL= VZ'-R 2 •Then the inductance may be determined from:L = XL/21Tf.The Substitution The substitution method isMethod a satisfactory system forobtaining the inductance orcapacitance of high-frequency components. Alarge variable capacitor with a good dial havingan accurate calibration curve is a necessityfor making determinations by this method.If an unknown inductor is to be measured, it isconnected in parallel with the standard capacitorand the combination tuned accurately tosome known frequency. This tuning may beaccomplished either by using the tuned circuitas a wavemeter and coupling it to the tunedcircuit of a reference oscillator, or by usingthe tuned circuit in the controlling position ofa two terminal oscillator such as a dynatronor transitron. The capacitance required to tunethi,s first frequency is then noted as C. Thecircuit or the oscillator is then tuned to thesecond harmonic of this first frequency andthe amount of capacitance again noted, thisr:me as C2. Then the distributed capacitanceacross the coil (including all stray capacitances)is equal to: Co= (C,-4C,)/3.R
728-B Test EquipmentFigure 9"'~--~·:=RAj~- "52Rx; ~: Rs®TWO WHEATSTONE BRIDGE CIRCUITSThese circuits are used lor the measurementof d-e resistance. In (A) the 11 tatio arms" R 8and R A are fixed and balancing of the bridgeis accomplished by variation of the standardR • 8The standard in this case usually consistsof a decade box giving resistance in1-ohm steps from 0 to 1110 or to 11,110 ohms.In (B) a fixed standard is used for each rangeand the ratio arm is varied to obtain balance,A calibrated slide-wire or potentiometer calibratedby resistance in terms of degrees isusually employed as RA and RB. It will benoticed that the formula for determining theunknown resistance from the known is thesame in either case.This value of distributed capacitance isthen substituted in the following formula alongwith the value of the standard capacitance foreither of the two frequencies of measurement:1L=-----47T'f,' ( C + Co)The determination of an unknown capacitanceis somewhat less complicated than theabove. A tuned circuit including a coil, theunknown capacitor and the standard capacitor,all in parallel, is resonated to some convenientfrequency. The capacitance of the standardcapacitor is noted. Then the unknown capacitoris removed and the circuit re-resonatedby means of the standard capacitor. The differencebetween the two readings of the standardcapacitor is then equal to the capacitanceof the unknown capacitor.34-3 Measurementswith a BridgeExperience has shown th~t one of the mostsatisfactory methods for measuring circuit constants(resistance, capacitance, and inductance)at audio frequencies is by means of the a-cbridge. The Wheatstone (d-e) bridge is alsoone of the most accurate methods for themeasurement of d-e resistance. With a simplebridge of the type shown at figure 9A it isentirely practical to obtain d-e resistance de-0<strong>THE</strong> <strong>RADIO</strong>terminations accurate to four significant figures.With an a-c bridge operating within itsnormal rating as to frequency and range ofmeasurement it is possible to obtain resultsaccurate to three significant figures.Both the a-c and the d-e bridges consist ofa source of energy, a standard or reference ofmeasurement, a means of balancing this standardagainst the unknown, and a means of indicatingwhen this balance has been reached.The source of energy in the d-e bridge is abattery; the indicator is a sensitive galvanometer.In the a-c bridge the source of energyis an audio oscillator (usually in the vicinityof 1000 cycles), and the indicator is usuallya pair of headphones. The standard for the d-ebridge is a resistance, usually in the form ofa decade box. Standards for the a-c bridge canbe resistance, capacitance, and inductance inva.rying forms.Figure 9 shows two general types of theWheatstone or d-e bridge. In (A) ,the so-called"ratio arms" RA and Rn are fixed (usually ina ratio of 1-to-1, 1-to-10, 1-to-100, or 1-to-1,000) and the ~tandard resistor Rs is varieduntil the bridge is in balance. In commerciallymanufactured bridges there are usually two ormore buttons on the galvanometer for progressivelyincreasing its sensitivity as balance isapproached. Figure 9B is the slide wire typeof bridge in which fixed standards are usedand the ratio arm is continuously variable.The slide wire may actually consist of a movingcontact along a length of wire of uniformcross section in which case the ratio ofRA to Rn may be read off directly in centimetersor inches, or in degrees of rotation ifthe slide wire is bent around a circular former.Alternatively, the slide wire may consist oflinear-wound potentiometer with its dial calibratedin degrees or in resistance from eachend.Figure lOA shows a simple type of a-cbridge for the measurement of capacitance andinductance. It can also, if desired, be usedfor the measurement of resistance. The fourarms of the bridge may be made up in a varietyof ways. As before, Rn and R-' make upthe ratio arms of the device and may be eitherof the slide wire type, as indicated, or theymay be fixed and a variable standard used toobtain balance. In any case it is always necessarywith this type of bridge to use a standardwhich presents the same type of impedance asthe unknown being measured: resistance standardfor a resistance measurement, capacitancestandard for capacitance, and inductance stand-
<strong>HANDBOOK</strong>Frequency Measurement 729-Bard for inductance determination. Also, i.r is agreat help in obtaining an accurate balance ofthe bridge if a standard of approximately thesame value as the assumed value of the unknownis employed. Also, the standard shouldbe of the same general type and should haveapproximately the same power factor as the unknownimpedance. If all these precautions areobserved, little trouble will be experienced inthe measurement of resistance and in themeasurement of impedances of the valuesusually used in audio and low radio frequencywork.However, the bridge shown at lOA willnot be satisfactory for the measurement of capacitancessmaller than about 1000 fLfLfd. Forthe measurement of capacitances from a fewmicro-microfarads to about 0.001 1-'fd. a Wagnerg~ounded substitution capacitance bridge ofthe type shown in figure lOB will be foundsatisfactory. The ratio arms RA and Rn shouldbe of the same value within 1 per cent; anyvalue between 2500 and 10,000 ohms forboth will be satisfactory. The two resistorsRc and Rn should be 1000-ohm wire-woundpotentiometers. Cs should be a straight-linecapacitance capacitor with an accurate vernierdial; 500 to 1000 fLfLfd. will be satisfactory.Cc can be a two or three gang broadcast capacitorfrom 700 to 1000 fLfLfd. maximum capacitance.The procedure for making a measurementis as follows: The unknown capacitor Cx isplaced in parallel with the standard capacitorCs. The Wagner ground Rn is varied back andforth a small amount from the center of itsrange until no signal is heard in the phoneswith the switch S in the center position. Thenthe switch S is placed in either of the two outsidepositions, Co is adjusted to a capacitancesomewhat greater than the assumed valueof the unknown Cx, and the bridge is broughtinto balance by variation of the standard capacitorC,. It may be necessary to cut someresistance in at Rc and to switch to the otheroutside position of S before an exact balancecan be obtained. The setting of Cs is thennoted, Cx is removed from the circuit (but theleads which went to it are not changed in anyway which would alter their mutual capacitance),and Cs is readjusted until balance isagain obtained. The difference in the two settingsof Cs is equal to the capacitance of theunknown capacitor Cx.There are many other types of a-c bridgecircuits in common use for measuring inductancewith a capacitance standard, frequencyRAZx= ""J!5:B Zs Xx= =~ XsZx= IMPEDANCE BEING MEASURED, Rs= RESISTANCE COMPONENT OF ZsZs::: IMPEDANCE OF STANDARD, Xx ='REACTANCE COMPONENT OF ZxRx= RESISTANCE COMPONENT Of Zx, Xs:REACTANCE COMPONENT OF Zs@®Figure 10WAGNER/ GROUND.Ro FTWO A-C BRIDGE CIRCUITSThe operation of these bridges is essentiallythe same as those of figure 9 except thata.c. is fed into the bridge instead of d.c. anda pair of phones is used as the indicator in·stead of the galvanometer. The bridge shownat (A) can be used for the measurement ofresistance, but it is usually used for themeasurement of the impedance and reactanceof coils and capacitors at frequencies from200 to 1000 cycles. The bridge shown at (B)is used for the measurement of small valuesof capacitance by the substitution method.Full description of the operation of bothbridges is given in the accompanying text.in terms of resistance and capacitance, andso forth. Terman's Radio Engineers' Handbookgives an excellent discussion of common typesof a-c bridge circuits.34-4 FrequencyMeasurementsAll frequency measurement within theUnited States is based on the transmissions ofStation WWV of the National Bureau ofStandards. This station operates continuouslyon frequencies of 2.5, 5, 10, 15, 20, 25, 30,and 3 5 Me. The carriers of those frequenciesbelow 30 Me. are modulated alternately bya 440-cycle tone or a 600-cycle tone for periodsof four minutes each. This tone is interruptedat the beginning of the 59th minute
730-B Test Equipment <strong>THE</strong> <strong>RADIO</strong>c,-100-tLtLfd. air trimmerc,,C,-0.0003-tLfd. midget micac,-50-tLtLfd. midget micac,-0.002-tLfd. midget micaR,, Rz--100,000 ohms V2 wattL,-10-mh. shielded r-f chokeL:-2.1-mh. r-f chokeX-100-kc. crystalFigure 11SCHEMATIC OF A 100-KC. FREQUENCY SPOTTERof each hour and each five minutes thereafterfor a period of precisely one minute. GreenwichCivil Time is given in code during theseone-minute intervals, followed by a voice announcementgiving Eastern Standard Time.The accuracy of all radio and audio frequenciesis better than one part in 50,000,000.A 5000 microsecond pulse ( 5 cycles of a 1000-cyde wave) may be heard as a tick tor everysecond except the 59th second of each minute.These standard-frequency transmissions ofstation WWV may be used for accurately determiningthe limits of the various amateurbands with the aid of the station communicationsreceiver and a 50-kc., 100-kc., or 200-kc. band-edge spotter. The low frequency oscillatormay be self-excited if desired, but lowfrequency standard crystals have become sorelatively inexpensive that a reference crystalmay be purchased for very little more thanthe cost of the components for a self-excitecloscillator. The crystal has the additional advantagethat it may be once set so that its harmonicsare at zero beat with WWV and thenleft with only an occasional check to see thatthe frequency has not drifted more than afew cycles. The self-excited oscillator, on theother hand, must be monitored very frequentlyto insure that it is on frequency.Using aFrequencySpotterTo use a frequency spotter it isonly necessary to couple the outputof the oscillator unit to theantenna terminal of the receiverthrough a very small capacitance such as mightbe made by twisting two pieces of insulatedhookup wire together. Station WWV is thentuned in on one of its harmonics, 15 Me.will usually be best in the daytime and 5 or10 Me. at night, and the trimmer adjustmenton the oscillator is varied until zero beat isobtained between the harmonic of the oscillatorand WWV. With a crystal referenceoscillator no difficulty will be had with usingthe wrong harmonic of the oscillator to obtainthe beat, but with a self-excited oscillatorit will be wise to insure that the referenceoscillator is operating exactly on 50, 100, or200 kc. (whichever frequency has beenchosen) by making sure that zero beat is obtainedsimultaneously on all the frequenciesof WWV that can be heard, and by notingwhether or not the harmonics of the oscillatorin the amateur bands fall on the approximatecalibration marks of the receiver.A simple frequency spotter is diagrammedin figure 11.34-5 Antenna andTransmissionLine MeasurementsThe degree of adjustment of any amateurantenna can be judged by the study of thestanding-wave ratio on the transmissioll linefeeding the antenna. Various types Oir ; nstrumentshave been designed to measure thes.r.w. present on the transmission line, or tomeasure the actual radiation resistance of theantenna in question. The most important ofthese instruments are the slotted line, thebridge-type s-w-r meter, and the antennascope.The Slotted Line It is obviously impracticalto measure the voltagestanding-waveratio in a length of coaxial linesince the voltages and currents inside the lineare completely shielded by the outer conductorof the cable. Hence it is necessary to insertsome type of instrument into a section of theline in order to be able to asce!1tain the conditionswhich are taking place inside theshielded line. Where measurements of a highdegree of accuracy are required, the slottedline is the instrument most frequently used.Such an instrument, diagrammed in figure 12,is an item of test equipment which could beconstructed in a home workshop which includeda lathe and other metal working tools.
<strong>HANDBOOK</strong>Antenna Measurements 731-OUTPUTTOANTENNAFEED LINEFigure 12DIAGRAMMATIC REPRESENT AT IONOF A SLOTTED LINEThe conductor ratios in the slotted line, includingthe tapered end sections should besuch that the characteristic impedance of theequipment is the same as that of the transmissionline with which the equipment is tobe used. The indicating instrument may beoperated by the d-e output of the rectifiercoupled to the probe, or it may be operatedby the '!·c components of the rectified signal•f the s•gnal generator or transmitter is amplitudemodulated by a constant percentage.Commercially built slotted lines are very expensivesince they are constructed with a highdegree of accuracy for precise laboratory work.The slotted line consists essentially of asection of air-dielectric line having the samecharacteristic impedance as the transmissionline into which it is inserted. Tapered fittingsfor the transmission line connectors at eachend of the slotted line usually are required dueto differences in the diameters of the slottedline and the line into which it is inserted. Anarrow slot from Ys-inch to 0,-inch in widthis cut into the outer conductor of the line. Aprobe then is inserted into the slot so that itis coupled to the field inside the line. Somesort of accurately machined track or lead screwmust be provided to insure that the probemaintains a constant spacing from the innerconductor as it is moved from one end of the~lotted line to the other. The probe usuallyIncludes some type of rectifying elementwhose output is fed to an indicating instrumentalongside the slotted line.The unfortunate part of the slooted-line systemof measurement is that the line must besomewhat over one-half wavelength long atthe test frequency, and for boot results shouldbe a full wavelength long. This requirement iseasily met at frequencies of 420 Me. and abovewhere a full wavelength is 28 inches or less.But for the lower frequencies such an instrumentis mechanically impracticable.Bridge-TypeStanding-WaveIndicatorsThe bridge type of standingwaveindicator is used quitegenerally for making measurementson commercial co-Figure 13RESISTOR-BRIDGESTANDING-WAVE INDICATORThis type of test equipment is suitablefor use with coaxial feed lines.C,-0.001-p,fd. midget ceramic capacitorc, C,-.001 disc ceramicR,, Rr-22-ohm 2-watt carbon resistorsR,-Resistor equal in resistance to the characteristicimpedance of the coaxial transmissionline to be used (I watt)R,,-5000-ohm wire-wound potentiometerRs-!0,000-ohm !-watt resistorRFC-R-f choke suitable for operation at themeasurement frequencyaxial transmissiOn lines. A simplified versionis available from M. C. ]ones Electronics Co.,Bristol, Conn. ("Micro-Match").One type of bridge standing-wave indicatoris diagrammed in figure 13. This type of instrumentcompares the electrical impedance ofthe transmission line with that of the resistorR, which is induded within the unit. Experiencewith such units has shown that the resistorR, should be a good grade of non-inductivecarbon type. The Ohmite "Little Devil"type !'esistor in the 2-watt rating has givengood performance. The resistance at R, shouldbe equal to the characteristic impedance ofthe antenna transmission line. In other words,this resistor should have a value of 52 ohmsfor lines having this characteristic impedancesuch as RG-8/U and RG-58/U. For use withlines having a nominal characteristic impedanceof 70 ohms, a selected "68 ohm" resistorhaving an actual resis,tance of 70 ohmsmay be used.Balance within the equipment is checked bymounting a resistor, equal in value to thenominal characteristic impedance of the lineto be used, on a coaxial plug of the type usedon the end of the antenna feed line. Thenthis plug is inserted into the input receptacleof the instrument and a power of 2 to 4 wattsapplied to the output receptacle on the desired~requency of operation. Note that the signalIS passed through the bridge in the direction
732Test Equipment<strong>THE</strong> <strong>RADIO</strong>o'052 32 08,209f
<strong>HANDBOOK</strong> Reflectometer 733Figure 16INTERIOR VIEW OF COAXIAL REFLECTOMETERThe Reflectometer is a short section of transmission line containing two r-1 voltmeters. Center conductorof line is a section of brass rod soldered to center pins of input and output receptacles. At either endof unit are the crystal diodes, bypass capacitors and terminals. Diode load resistors are at center ofinstrument, grounded to brass alignment rod.reads the reflected component. The magnitudeof standing wave ratio on the transmissionline is the ratio of the incident componentto the reflected component, as shown in figure14. In actual use, calibration of the reflectometeris not required since the relativereading of reflected power indicates the degreeof match or mis-match and all antennaand transmission line adjustments should beconducted so as to make this reading as lowas possible, regardless of its absolute value.The actual meter readings obtained fromthe device are a function of the operating frequency,the sensitivity of the instrument beinga function of transmitter power, increasingrapidly as the frequency of operation is increased.However, the reflectometer is invaluablein that it may be left permanently inthe transmission line, regardless of the poweroutput level of the transmitter. It will indicatethe degree of reflected power in the antennasystem, and at the same time providea visual indication of the power output ofthe transmitter.ReflectometerCircuitThe circuit and assembly informationfor the reflectometerare given in figure 15. Twodiode voltmeters are coupled back-to-back toa short length of transmission line. The combinedinductive and capacative pickup betweeneaoch voltmeter and the line is such that theincident component of the line voltage is balancedout in one case and the inductive componentis balanced out in the other case. Eachvoltmeter, therefore, reads only one wave-component.Careful attention to physical symmetryof the assembly insures accurate and completeseparation of the voltage components by thetwo voltmeters. The outputs of the two voltmetersmay be selected and read on an externalmeter connected to the terminal postsof the reflectometer.Each r-f voltmeter is composed of a loadresistor and a pickup loop. The pickup loopis positioned parallel tv a section of transmissionline permitting both inductive and capacativecoupling to exist between the centerconductor of the line and the loop. The dimensionsof the center conductor and theouter shield of the reflectometer are chosenso that the instrument impedance closelymatches that of the transmission line.ReflectometerConstructionA view of the interior of thereflectometer is shown in figure16. The coaxial input andoutput connectors of the instrument aremounted on machined brass discs that areheld in place by brass alignment rods, tappedat each end. The center conductor is machinedfrom a short section of brass rod, tapered anddrilled at each end to fit over the center pinof each coaxial receptacle. The end discs, therods, and the center conductor should be silverplated before assembly. When the centerconductor is placed in position, it is solderedat each end to the center pin of the coaxialreceptacles.One of the alignment rods is drilled andtapped for a 6-32 bolt at the mid-point, andthe end discs are drilled to hold 1;2-inch
734 Test Equipmentceramic insulators and binding posts, as &hownin the photograph. The load resistors, crystaldiodes, and bypass capaci~ors are finallymounted in the assembly as the last step.The two load resistors should be measuredon an ohmmeter to ensure that the resistancevalues are equal. The exaat value of resistanceis unimportant as long as the two resistorsare equal. The diodes should also be checkedon an ohmmeter to make sure that the frontresistances and back resistances are balancedbetween the units. Care should be taken duringsoldering to ensure the diodes and resisnorsare not overheated. Observe that the resistorleads are of equal length and that eachhalf of the assembly is a mirror-image of theother half. The body of the resistor is spacedabout Ys-inch away from the center conductor.Testing theReflectometerThe instrument can be adjustedon the 28 Me. band.An r-f source of a few wattsand nonreactive load are required. The constructionof the reflectometer is such that itwill work well with either 52- or 72-ohmcoaxial transmission lines. A suitable dummyload for the 52-ohm line can be made of four220 ohm, 2 watt composition resistors (Ohmite"Little Devil") connected in parallel. Clipthe leads of the resistors short and mountthem on a coaxial plug. This assembly providesan eight watt, 55 ohm load, suitablefor use at 30 Me. If an accurate ohmmeteris at hand, the resistors may be hand pickedto obtain four 208 ohm units, thus makingthe dummy load resistors exactly 52 ohms.For all practical purposes, the 55 ohm loadis satisfactory. A 75 ohm, eight watt loadresistor may be made of four 300 ohm, 2watt compos.ition resistors connected in parallel.R-f power is coupled to the reflectometerand the dummy load is placed in the "output"receptacle. The indicator meter is switched tothe ''reflected power" position. The meter readingshould be almost zero. It may be broughtro zero by removing the case of the instrumentand adjusting the position of the loadresistor. The actual length of wire in the resistorlead and its positioning determine the meternull. Replace the case before power is appliedto the reflectometer. The reflectometer is nowreversed and power is applied to the "output"receptacle, with a dummy load attached tothe "input" receptacle. The second voltmeter(forward power) is adjusted for a null readingof the meter in the same manner.<strong>THE</strong> <strong>RADIO</strong>If a reflected reading of zero is not obtainable,the harmonic content of the r-f sourcemight be causing a slight residual meter reading.Coupling the reflectometer to the r-fsource through a tuned circuit ("antennatuner") will remove the offending harmonicand permit an accurate null indication. Besure to hold the r-f input power to a lowvalue to prevent overheating the dummy loadresistors.Using theReflectometerThe bridge may be used up to150 Me. It is placed in thetransmission line at a convenientpoint, preferably be/ore any tuner, balun,or TVI filter. The indicator should be set toread forward power, with a maximum of resistancein the circuit. Power is applied andthe indicator resistor is adjusted for a fullscale reading. The switch is then thrown toread reflected power (indicated as A, figure14). Assume that the forward power meterreading is 1.0 and the reflected power readingis 0.5. Substituting these values in theSWR formula of figure 14 shows the SWR~o be 3. If forward power is always set to1.0 on the meter, the reflected power (A)can be read directly from the curve of figure14 with little error.If the meter is adjusted so as to provide ahalf-,scale reading of the forward power, thereflectometer may be used as a transmitterpower output meter. Tuning adjustments maythen be undertaken to provide greatest meterreading.34-7 Measurementson BalancedTransmission LinesMeasurements made on balanced transmissionlines may be conducted in the same manneras those made on coaxial Jines. In thecase of the coaxial Jines, care must be mkento prevent flow of r-f current on the outersurface of the line as this unwanted componentwill introduce errors in measurementsmade on the line. In like fashion, the currentsin a balanced transmission line mustbe 180 degrees out of phase and balancedwith respect to ground in order to obtain arealistic relationship between incident and reflectedpower. This situation is not alwayseasy to obtain in practice because of the proximityeffects of metallic object's or the earthto the transmission line. All transmission linemeasurements, therefore, should be conductedwith the realization of the physical limitations
<strong>HANDBOOK</strong>S.W.R. Indicator 735Figure 17SKETCH OF <strong>THE</strong> "TWIN-LAMP"TYPE OF S-W-R INDICATORThe short section of line with lamps at eachend usually is taped to the main transmissionline with plastic electrical tape.of the equipment and the measuring techniquethat is being used.Measurements onMolded Parallel-Wire LinesOne of the most satisfactoryand least expensive devicesfor obtaining a rough ideaof the standing-wave ratioon a transmission line of the molded parallelwiretype is the twin-lamp. This ingenious instrumentmay be constructed of new componentsfor a total cost of about 2 5 cents; thisfact alone places the twin-lamp in a class byitself as far as test instruments are concerned.Figure 17 shows a sketch of a twin-lamp indicator.The indicating portion of the systemconsists merely of a length of 300-ohm TwinLead about 10 inches long with a dial lamp ateach end. In the unit illustrated the dial lampsare standard 6.3-volt 150-ma. bayonet-baselamps. The lamps are soldered to the two leadsat each end of the short section of Twin-Lead.To make a measurement the short sectionof line with the lamps at each end is merelytaped to the section of Twin-Lead (or othersimilar transmission line) running from thetransmitter or from the antenna changeover relayto the antenna system. When there are nostanding waves on the antenna transmissionline the lamp toward the transmitter will lightwhile the one toward the antenna will notlight. With 300-ohm Twin-Lead running fromthe antenna changeover relay to the antenna,and with about 200 watts input on the 28-Mc.band, the dial lamp toward the transmitter willlight nearly to full brilliancy. With a standingwaveratio of about 1.5 to 1 on the transmissionline to the antenna the lamp toward theantenna will just begin to light. With a highstanding-wave ratio on the antenna feed lineboth lamps will light nearly to full brilliancy.Hence the instrument gives an indication ofrelatively low standing waves, but when theFigure 18OPERATION OF <strong>THE</strong> "TWIN LAMP"INDICATORShowing current flow resulting from inductiveand capacitive fields in a 11 twin lamp" attachedto a line with a low standing-wave ratio.standing-wave ratio is high the twin-lampmerely indicates that they are high withoutgiving any idea of the actual magnitude.Operation ofthe Twin-LampThe twin lamp operates byvirtue of the fact that the capacitiveand inductive couplingof the wire making up one side of thetwin-lamp is much greater to the transmissionlinelead immediately adjacent to it than tothe transmission-line lead on the other side.The same is of course true of the wire on theother side of the twin-lamp and the transmission-linelead adjacent to it. A further conditionwhich must be met for the twin-lamp tooperate is that the section of line making upthe twin-lamp must be short with respect to aquarter wavelength. Then the current due tocapacitive coupling passes through both lampsin the same direction, while the current due toinductive coupling between the leads of thetwin-lamp and the leads of the antenna transmissionline passes through the two lamps inopposite directions. Hence, in a line withoutreflections, the two currents will cancel inone lamp while the other lamp is lighted dueto the sum of the currents (figure 18).The basic fact which makes the twin-lamp adirectional coupler is a result of the conditionwhereby the capacitive coupling is a scalaraction not dependent upon the direction of thewaves passing down the line, yet the inductivecoupling is a vector action which is dependentupon the direction of wave propagation downthe line. Thus the capacitive current is thesame and is in the same phase for energytravelling in either direction down the line.But the inductive current travels in one directionfor energy travelling in one direction andin the other direction for energy going theother direction. Hence the two currents add atone end of the line for a wave passing towardthe antenna, while the currents add at theother end of the twin-lamp for the waves reflectedfrom the antenna. When the waves are
736 Test EquipmentFigure 19SWR BRIDGE FOR BALANCEDTRANSMISSION LINEA double bridge can be used lor two wiretransmission lines. Bridge is inserted in lineand may be driven with grid-dip oscillator orother low power r-1 source.strongly reflected upon reaching the anrtenna,the reflected wave is nearly the same as thedirect wave, and both lamps will light. Thiscondition of strong reflection from the antennasystem is that which results in a high standing-waveratio on the antenna feed line.Use of theTwin Lamp withVarious Feed LinesThe rwin-lamp is bestsuited for use with antennatransmission linesof the flat ribbon type.Lines with a high power rating are availablein impedances of 75 ohms, 205 ohms, and 300ohms in the flat ribbon type. In addition, onemanufacturer makes a 300-ohm line in twopower-level ratings with a tubular cross sect:on.A twin-lamp made from flat 300-ohmline may be used with this tubular line bytaping the twin-lamp tightly to the tubularline so that the conductors of the twin-lampare as close as possible on each side of the conductorsof the antenna line.34-8 A 11 Balanced 11SWR BridgeTwo resistor-type standing wave indicato.t5may be placed "back-to-back" to ~orm a SWRbridge capable of being used on two wirebalanced transmission lines. Such a bridge isshown in figures 19 and 21. The schematic ofsuch an instrument (figure 20) may be comparedto two of the simple bridges shown infigure 13. When the dual bridge circuit isbalanced the meter reading is zero. This stateis reached when the line currents are equaland exactly 180 degrees out of phase and theSWR is unity.As the condition of the line departs fromthe optimum, the meter of the bridge willshow the degree of departure. When the linecurreJ;J.ts are balanced and 180 degrees out ofphase, the meter will read the true value ofstanding wave ratio on the line. If these conditionsare not met, the reading is not absolute,merely giving an indication of the degreeof mis-match in the line. This handicap is notimportant, since the relative, not the absolute,degree of mis-match is sufficient for transmissionline adjustments to be made.Bridge A suggested method of con-Construction struction of the balanced bridgeis illu~ated in figures 19 and21. The unit is constructed within a box measuring4" x 6" x 2" in size. The 0- 200 d.c.microammeter is placed in the center of the4" x 6" side of the case. The input and outputconnectors of the instrument are placed oneach end of the box and the internal wiringis arranged so that the transmission line, ineffect, passes in one side of the box and outthe other with as little discontinuity as possible.The input and output terminals are mountedon phenolic plates placed over large cutoutsin the ends of the box, thus reducingcircuit capacity to ground to a minimum value.The "SWR-CAL" switch S, is located on oneside of the meter and the "Calibrate" potentiometerR, is placed on the opposite side.The transmission line within the unit is~OUTPUT: 51~RFCT.001MRFCSWR .001......--R1250250R1250R1250R1250R1250R1Figure 20SCHEMATIC OF BRIDGE FORBALANCED LINESM-0-200 d-e microammeterR1-Note: Six 250 ohm resistors are composition,non-inductive units. IRC type BT, orOhmite 11 Little Devil" l-watt resistors maybe used. (see text)$ 1 -DPDT rotary switch. Centralab type 1464INPUT
figure 21INTERIOR VIEW OFBALANCED BRIDGESHOWING PARTSPLACEMENTDiode rectifiers are placedat right angles to theshort section of transmissionline. Both sidesof bridge are balancedto ground by virtue ofsymmetrical constructionof unit.broken by two 250 olun composition resistorsand switch S,. The line segments are made ofshort pieces of # 10 copper wire, running betweenthe various components. Spacing betweenthe wires is held close to three inchesto approximate a 500-600 olun line.The small components of the bridge areplaced symmetrically about the 250 ohm seriesline resistors and the calibrating potentiometer,as can be seen in figure 21. Exact parts placementis not critical, except that the crystaldiodes should be placed at right angles tothe wires of the transmission line to reducecapacative pickup. The two r-f chokes shouldthen be placed at right angles to the diodes. ·The six 250 ohm resistors are checked on anohmmeter and should be hand-picked to obtainunits that are reasonably close in value.If it is desired to use the bridge with a 600ohm line, the value of these resistors shouldbe increased to 300 oluns each. Excess,ive heatshould not be used in soldering either the resistorsor the diodes to ensure that their characteristicswill not be altered by application ofhigh temperatures over an extended period.Testing theBalanced BridgeWhen the instrument iscompleted, a grid-dip metermay be coupled to the inputterminals via a two turn link. Be carefulnot to pin the bridge meter. Place switch S, inthe "Calibrate" (open) position. Set the griddip meter in the 10-Mc. to 20-Mc. range andadjust the link and calibration control R, forfull scale meter reading. A 500 ohm carbonresistor placed across the output terminals ofthe unit should produce a zero meter readingwhen S, is set to the "SWR" position. Variousvalues of resistance may now be placed acrossthe meter terminals to obtain calibration points737for the meter scale. The ratio of the externalresistor to the design value of the bridge willprovide the SWR value for any given meterreading. For example, a 1000 ohm resistorhas a ratio of 1000/500, and will give an indicatedSWR reading of 2. A 1500 ohm resistorwill give an indicated reading of 3, a 2000olun resistor will provide a reading of 4, andso on. Before each measurement is recorded,the calibrate control should be set to a fullscale reading with S, open.This simple system of calibration will leadto slight errors in calibration if the regulationof the r-f source is poor. That is, a change inthe external calibrating resistance will producea varying load on the r-f generator which couldeasily cause a change in the power applied tothe bridge. A separate diode r-f voltmeterplaced across the pickup loop will enable theinput voltage to be held to a constant valueand will provide somewhat more accurllltebridge calibration.Using the The bridge is placed in the transmissionline and driven from aBridgelow powered source having a minimumof harmonic content. Several measurementsshould be made at various frequencieswithin the range of antenna operation. Theselector switch is set to the "Cal." position andthe "Calibrate" potentiometer is adjusted forfull scale meter reading. The switch is thenset to the "SWR" posinion and a reading istaken. This reading and others taken at variousfrequencies may be plotted on a graph to providea "SWR curve" for the particular antennaand transmission line. Antenna adjustmentsand line balancing operations may now beconducted to provide a smooth SWR curve,
738 Test Equipment<strong>THE</strong> <strong>RADIO</strong>1000R11000GUTIIN34L -50R2-- _ __jFigure 23SCHEMATIC, ANTENNASCOPER,-1000 ohm composition potentiometer ohmitetype AB or Allen Bradley type J,linear taperR:-50 ohm, 1-watt composition resistor, IRCtype BT, or ohmite "Little Devil" (seetext)M-0-200 d-e microammeterINFigure 22<strong>THE</strong> ANTENNASCOPEThe radiation resistance of r-f loads connectedacross the output receptacle may be quicklydetermined by a direct dial reading. The Antennascopemay be driven with a grid-diposcillator, covering r-f impedance range of 5to 1000 ohms.with the point of minimum SWR occuring atthe chosen design frequency of the antenna installation.The complete adjustment and checkoutprocedure for an antenna and transmissionline sy,stem is covered in the Beam AntennaHandbook, published by Radio Publications,Inc., Wilton, Conn.34-9 The AntennascopeThe Antennascope is a modified SWRbridge in which one leg of the bridge is composedof a non-inductive variable resistor. Thisresis,tor is calibrated in ohms, and when itssetting is equal to the radiation resistance ofthe antenna under test the bridge is in a balancedstate. If a sensitive voltmeter is con-nected across the bridge, it will indicate avoltage null at bridge balance. The radiationresistance of the antenna may then be read directlyfrom the calibrated resistor of the instrument.When the antenna under test is in a nonresonantor reacnive state, the null indicationon the meter of the Antennascope will be incomplete.The frequency of the exciting signalmust then be moved to the resonant frequencyof the antenna to obtain accurate readingsof radiation resistance from the dial ofthe instrument.A typical Antennascope is shown in figures22 and 24, and the schematic is shown in figure23. A 1000 ohm non-inductive carbon potentiometerserves as the variable leg of thebridge. The other legs are composed of the 50ohm composition resistor and the radiation resistanceof the antenna. If the radiation resistanceof the external load or antenna is 50ohms and the potentiometer is set at mid-scalethe bridge is in balance and the diode voltmeterwill read zero. If the radiation resistanceof the antenna is any value other than 50ohms, the bridge may be balanced to this newvalue by varying the position of the potentiometer.Bridge balance may be obtained withnon-reactive loads in the range of 5 ohms to1000 ohms with this simple circuit. Whenmeasurements are conducted at the resonantfrequency of the antenna system the radiationresistance of the installation may be read directlyfrom the calibrated dial of the Antennascope.Conversely, a null reading of the instrumentwill occur at the resonant frequency,which may easily be found with the aid of acalibrated receiver or frequency meter.
<strong>HANDBOOK</strong>Antennascope739Constructing theAntennascopeThe Antennascope is builtwithin a sheet metal casemeasuring 3" x 6" x 2".The indicating meter is placed at rthe top ofthe case, and the r-f bridge occupies the lowerportion of the box. The input and output coaxialfittings are mounted on each side of thebox and the non-inductive 50 ohm resistor issoldered between the center terminals of thereceptacles.The calibrating potentiometer (R,) ismounted upon a phenolic plate placed over a%-inch hole drilled in the front of the box.This reduces the capacity to ground of the potentiometerto a minimum. Placement of thesmall components within the box may be seenin figure 24. Care should be taken to mountthe crystal diode at right angles to the 50 ohmresistor to reduce capacity coupling betweenthe components.The upper frequency limit of accuracy ofthe Antennascope is determined by the assemblytechnique. The unit shown will workwith good accuracy to approximately 100 Me.Above this frequency, the self-inductance of theleads prevents a perfect null from being obtained.For operation in the VHF region, itwould be wise to rearrange the components toreduce lead length ·to an absolute minimum,and to use Vi-inch copper strap for the r-.fleads instead of wire.Testing theAntennascopeWhen the instrument is completed,a grid-dip meter maybe coupled to the input receptacleof the Antennascope by means of a twoturn link. The frequency of excitation shouldbe in the 10 Mc.-20 Me. region. Couplingshould be adfusted to obtain a half-scale readingof the meter. Various values of 1-wattoomposition resistors up to 1000 ohms arethen plugged into the '"output" coaxial receptacleand the potentiometer is adjusted for anull on the meter. The settings of the potentiometermay now be calibrated in terms ofthe load resistor, the null position indicatingthe value of the test resistor. A calibrated scalefor the potentiometer should be made, asshown in figure 22.Using the The antennascope may beAntennascope driven by a grid-dip oscillator·coupled to it by a two turnlink. Enough coupling should be used to obtainat least a % scale reading on the meter ofthe Antennas.cope with no load connected nothe measuring terminals. The Antennascopemay be considered to be a low range r-f ohm-Figure 24PLACEMENT OF PARTS WITHIN<strong>THE</strong> ANTENNASCOPEWith the length of leads shown this modelis useful up to about 100 Me. Crystal diodeshould be placed at right angles to 50 ohmcomposition resistor.meter and may be employed to determine theelectrical length of quarter-wave lines, surgeimpedance of transmission lines, and antennaresonance and radiation resistance.In general, the measuring terminals of theAntennascope are connected in series with theload at a point of maximum cunrent. Thismeans the center of a dipole, or the base of avertical Vi-wave ground plane antenna. Excitationis supplied to the Antennascope, and thefrequency of excitation and the resistance controlof the Antennascope are both varied untila complete null is obtained on the indicatingmeter of the Antennascope. The frequency of
740 Test Equipment<strong>THE</strong> <strong>RADIO</strong>SHIELDED !NCLOSUR!A SILICON CRYSTAL NOISE GENERATORFigure 26Figure 25SIMPLE SILICON CRYSTAL NOISEGENERATORthe source of excitation is now the resonantfrequency of the load, and the radiation resistanceof the load may be read upon the dial ofthe Antennascope.On measurements on 80 and 40 meters, itmight be found that it is impossible to obtaina complete null on the Antennascope. This isusually caused by pickup of a nearby broadcaststation, the rectified signal of the broadcaststation obscu11ing the null indication onthe Antennascope. This action is only noticedwhen antennas of large size are being checked.34-10 A Silicon CrystalNoise GeneratorThe limiting factor in signal reception above25 Me. is usually the thermal noise generatedin the receiver. At any frequency, however,the wned circuits of the receiver must be accuratelyaligned fo,r best signal-to-noise ratio.Circuit changes (and even alignment changes)in 1he r-f stages of a receiver may do much toeither enhance or degrade the noise figure ofthe receiver. It is exceedingly hard to determinewhether changes of either aLignment orcirc'itry are really providing a boost in signalto-noiseratio of the receiver, or are merely increasingthe gain (and noise) of the unit.A simple means of determining the degreeof actual sensitivity of a receiver is to injecta minute signal in the input circuit and thenmeasure the amount of this signal that isneeded to oveJJCome the inherent receivernoise. The less injected signal needed to overridethe receiver noise by a certain, fixedamount, the more sensitive is the receiver.A simple source of minute signal may be obtainedfrom a silicon crystal diode. If a smalld-e current is passed through a silicon crystalin the direction of highest resistance, a smallbut constant r-f noise (or hiss) is generated.The voltage necessary to generate this noisemay be obtained from a few flashlight celkThe generator is a broad band device and requiresno tuning. If built with short leads, itmay be employed for receiver measurementswell above 150 Me. The noise generator shouldbe used for comparative measurements only,since calibration against a high quality commercialnoise generator is necessary for absolutemeasurements.A Practical Shown in figure 25 is a simNaise Generator pie silicon c r y s t a I noisegenerator. The schematic ofthis uni't is illustrated in figure 26. The 1N21crystal and .001 fLfd. ceramic capacitor areconnected in series directly across the outputterminals of the instrument. Three small flashlightbatteries are wired in series and mountedinside the case, along with the 0-2 d-e milliammeterand the noise level potentiometer.To prevent heat damage to the 1N21 crystalduring the soldering process, the crystal shouldbe held with a damp rag, and the connectionssoldered to it quickly with a very hot iron.Across the terminals (and in parallel with theequipment to be attached to the generator) isa 1-watt carbon resistor whose resistance isequal to the impedance level at which measurementsare to be made. This will usually beeither 50 or 300 ohms. If the noise generatoris to be used at one impedance level only, this
<strong>HANDBOOK</strong>Noise Generator 741TEST SET-UP FOR NOISE GENERATORFigure 27resistor may be mounted permanently insideof the case.Using theNoise GeneratorThe test setup for use ofthe noise generator is shownin figure 27. The noise gen-erator is connected to the antenna terminalsof the receiver under test. The receiver isturned on, the a.v.c. turned off, and the r-fgain control placed full on. The audio volumecontrol is adjusted until the output meter advancesto one-quarter scale. This reading isthe basic receiver noise. The noise generatoris turned on, and the noise level potentiometeradjusted until the noise output voltage of thereceiver is doubled. The more resistance inthe diode circuit, the better is the signal-tonoiseratio of the receiver under test. The r-fcircuits of the receiver may be aligned formaximum signal-to-noise ratio with the noisegenerator by aligning for a 2/1 noise ratio atminimum diode current.
CHAPTER THIRTY-FIVERadio Mathematicsand CalculationsRadiomen often have occasion to calculatesizes and values of required parts. Thisrequires some knowledge of mathematics. Thefollowing pages contain a review of those partsof mathematics necessary to understand andapply the information contained in this book.It is assumed that the reader has had somemathematical training; this chapter is not intendedto teach those who have never learnedanything of the subject.Fortunately only a knowledge of fundamentalsis necessary, although this knowledge mustinclude several branches of the subject. Fortunately,too, the majority of practical applicationsin radio work reduce to the solution ofequations or formulas or the interpretation ofgraphs.Notation ofNumbersArithmeticIn writing numbers in the Arabicsystem we employ ten dif·ferent symbols, digits, or figures:1, 2, 3, 4, 'i, 6, 7, 8, 9, and 0, and placethem in a definite sequence. If there is morethan one figure in the number the position ofeach figure or digit is as important in determiningits value as is the digit itself. When wedeal with whole numbers the righthandmostdigit represents units, the next to the left representstens, the next hundreds, the next thousands,from which we derive the rule that everytime a digit is placed one space further tothe left its value is multiplied by ten.8 1 4 3thousands hundreds tens uni.tsIt will be seen that any number is actually asum. In the example given above it is the sumof eight thousands, plus one hundred, plus742four tens, plus three units, which could bewritten as follows:8I438143thousands I 1 0 x 1 0 x 1 0)hundreds I 10 x 1 0)tensunitsThe number in the units pos1t10n is sometimesreferred to as a first order number, thatin the tens position is of the second order, thatin the hundreds position the third order, etc.The idea of letting the position of the symboldenote its value is an outcome of the abacus.The abacus had only a limited numberof wires with beads, but it soon becameapparent that the quantity of symbols mightbe continued indefinitely towards the left,each further space multiplying the digit'svalue by ten. Thus any quantity, howeverlarge, may readily be indicated.It has become customary for ease of readingto divide large numbers into groups of threedigits, separating them by commas.6,000,000 rather than 6000000Our system of notation then is characterizedby two things: the use of positions to indicatethe value of each symbol, and the use of tensymbols, from which we derive the name decimalsystem.Retaining the same use of positions, wemight have used a different number of symbols,and displacing a symbol one place to theleft might multiply its value by any other factorsuch as 2, 6 or 12. Such other systems havebeen in use in history, but will not be discussedhere. There are also systems in which displacinga symbol to the left multiplies its value by
Decimal Fractions 7430 0.5 44Figure I.AN ILLUSTRATION OF LINEAR FRACTIONS.varying factors in accordance with complicatedrules. The English system of measurementsis such an inconsistent and inferior system.Decimal Fractions Since we can extend anumber indefinitely tothe left to make it bigger, it is a logical step toextend it towards the right to make it smaller.Numbers smaller than unity are fractions andif a displacement one position to the right dividesits value by ten, then the number is referredto as a decimal fraction. Thus a digitto the right of the units column indicates thenumber of tenths, the second digit to the right. represents the number of hundredths, the third,the number of thousandths, etc. Some distinguishingmark must be used to divide unit fromtenths so that one may properly evaluate eachsymbol. This mark is the decimal point.A decimal fraction like four-tenths may bewritten .4 or 0.4 as desired, the latter probablybeing the clearer. Every time a digit is placedone space further to the right it represents aten times smaller part. This is illustrated inFigure 1, where each large division representsa unit; each unit may be divided into ten partsalthough in the drawing we have only so dividedthe first part. The length ab is equal toseven of these tenth parts and is written as 0.7.The next smaller divisions, which should bewritten in the second column to the right ofthe decimal point, are each one-tenth of thesmall division, or one one-hundredth each.They are so small that we can only show themby imagining a magnifying glass to look atthem, as in Figure 1. Six of these divisions isto be written as 0.06 (six hundredths). Weneed a microscope to see the next smaller division,that is those in the third place, which willbe a tenth of one one-hundredth, or a thousandth;four such divisions would be writtenas 0.004 (four thousandths).Ar------------,-------------,------------,-------------r-----------~80.1 0.1 0.1 0.1 0.1G0 0 0 0 0ci ci ci ci ciK 0.1 0.1 0.1 0.10 0 0ci ci ci~0~~~~~~----------~----------L_ ________ _L ________ ~cHFigure 2.IN THIS ILLUSTRATION FRACTIONAL PORTIONS ARE REPRESENTED IN <strong>THE</strong>FORM OF RECTANGLES RA<strong>THE</strong>R THAN LINEARLY.ABCD = I.O;GFED = 0.1; KJEH = 0.01; each small sedlan within KJEH equal• 0.001
744 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>It should not be thought that such numbersare merely of academic interest for very smallquantities are common in radio work.Possibly the conception of fractions may beclearer to some students by representing it inthe form of rectangles rather than linearly(see Figure 2).Addition When two or more numbers areto be added we sometimes writethem horizontally with the plus sign betweenthem. + is the sign or operator indicating addition.Thus if 7 and 12 are to be added to·gether we may write 7+12=19.But if la1ger or more numbers are to be addedtogether they are almost invariably written oneunder another in such a position that the deci·mal points fall in a vertical line. If a numberhas no decimal point, it is still considered asbeing just to the right of the units figure; sucha number is a whole number or integer. Examples:6543253041537270.6543.253.04156.895654325304.15990.1The result obtained by adding numbers iscalled the mm.Subtraction Subtraction is the reverse ofaddition. Its operator is - (themin11.r sign). The number to be subtracted iscalled the minttend, the number from which itis subtracted is the subtrahend, and the resultis called the remainder.subtrahend-minuendremainderExamples:65.4 65.4-32 -32.2133.433.19Multiplication When numbers are to be multipliedtogether we use the X,which is known as the multiplication or thetimes sign. The number to be multiplied isknown as the multiplicand and that by whichit is to be multiplied is the multiplier, whichmay be written in words as follows:multiplicandX multiplierpartial productpartial productproductThe result of the operation is called theproduct.From the examples to follow it will be obviousthat there are as many partial productsas there are digits in the multiplier. In the followingexamples note that the righthandmostdigit of each partial product is placed onespace farther to the left than the previous one.834X 265004166821684834X 20650040001668171804In the second example above it will be seenthat the inclusion of the second partial productwas unnecessary; whenever the multipliercontains a cipher (zero) the next partial productshould be moved an additional space tothe left.Numbers containing decimal fractions mayfirst be multiplied exactly as if the decimalpoint did not occur in the numbers at all; theposition of the decimal point in the product isdetermined after all operations have been completed.It must be so positioned in the productthat the number of digits to its right is equalto the number of decimal place$ in the multiplicandplus the number of decimal places inthe multiplier.This rule should be well understood sincemany radio calculations contain quantitieswhich involve very small decimal fractions. Inthe examples which follow the explanatorynotations "2 places," etc., are not actuallywritten down since it is comparatively easy todetermine the decimal point's proper iocationmentally.5.43xo.n10863 8012 places2 places3.9096 2+2=4 places0.04xo.oo30.000122 places3 places2+3=5 placesDivision Division is the reverse of multiplication.Its operator is the +,which is called the division sign. It is also commonto indicate division by the use of the fractionbar (/) or by writing one number overthe other. The number which is to be dividedis called the dividend and is written beforethe division sign or fraction bar or over thehorizontal line indicating a fraction. The num-
<strong>HANDBOOK</strong>ber by which the dividend is to be divided iscalled the divisor and follows the divisionsign or fraction bar or comes under the horizontalline of the fraction. The answer orresult is called the quotient.quotientdivisor ) dividendordividend 7 divisor= quotientordividend--- = quotientdivisorExamples:126 49834) 10508483449) 2436196216816685004500447644135 remainderNote that one number often fails to divideinto another evenly. Hence there is often aquantity left over called the remainder.The rules for placing the decimal pointare the reverse of those fot multiplication.The number of decimal places in the quotientis equal to the difference between the numberof decimal places in the dividend and that inthe divisor. It is often simpler and clearerto remove the decimal point entirely from thedivisor by multiplying both dividend and di·visor by the necessary factor; that is we movethe decimal point in the divisor as many placesto the right as is necessary to make it a wholenumber and then we move the decimal pointin the dividend exactly the same number ofplaces to the right regardless of whether thismakes the dividend a whole number or not.When this has been done the decimal pointin the quotient will automatically come directlyabove that in the dividend as shown inthe following example.Example: Divide 10.5084 by 8.34. Move thedecimal point of both dividend and divisortwo places to the right.1.26834 ) I 050.848342168166850045004Division 745Another example: Divide 0.000325 by 0.017.Here we must move the decimal point threeplaces to the right in both dividend and divisor.0.01917) 0.32517155153In a case where the dividend has fewer decimalsthan the divisor the same rules still mayLe applied by adding ciphers. For example todivide 0.49 by 0.006 we must move thedecimal point three places to the right. The0.49 now becomes 490 and we write:2816 ')4'9'048106When the division shows a remainder it issometimes necessary to continue the work soas to obtain more figures. In that case ciphersmay be annexed to the dividend, brought downto the remainder, and the division continuedas long as may be necessary; be sure to placea decimal point in the dividend before theciphers are annexed if the dividend does notalready contain a decimal point. For example:80.336) 482.0048201842018This operation is not very often requiredin radio work since the accuracy of the measurementsfrom which our problems startseldom justifies the use of more than threesignificant figures. This point will be coveredfurther later in this chapter.Fractions Quantities of less than one(unity) are called fractions. Theymay be expressed by decimal notation as wehave seen, or they may be expressed as vulgarfractions. Examples of vulgar fractions:2
746 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>numeratordenominatorThe upper pos1t10n of a vulgar fraction iscalled the numerator and the lower positionthe denominator. When the numerator is thesmaller of the two, the fraction is called aproper fraction; the examples of vulgar fractionsgiven above are proper vulgar fractions.When the numerator is the larger, the expressionis an improper fraction, which canbe reduced to an integer or whole numberwith a proper fraction, the whole being calleda mixed number. In the following examplesimproper fractions have been reduced totheir corresponding mixed numbers.7 34= 1 43467- 515 2T= 1 3Adding or SubtractingFractionsExceptfractionswhenaretheverysimple it will usuallybe found much easier to add and subtractfractions in the form of decimals. This rulelikewise applies for practically all other operationswith fractions. However, it is occasionallynecessary to perform various operationswith vulgar fractions and the rulesshould be understood.When adding or subtracting such fractionsthe denominators must be made equal. Thismay be done by multiplying both numeratorand denominator of the first fraction by thedenominator of the other fraction, afterwhich we multiply the numerator and denominatorof the second fraction by the denominatorof the first fraction. This soundsmore complicated than it usually proves inpractice, as the following examples will show.I 1 [lx3 lx2] 3 2 52+3= 2x3 + 3x2 =6+6=63 2 [3 X 5 2 X 4 J IS 8 74-s= 4xs -sx4 =Tcf-20=20Except in problems involving large numbersthe step shown in brackets above is usuallydone in the head and is not written down.Although in the examples shown above wehave used proper fractions, it is obvious thatthe same procedure applies with improperfractions. In the case of problems involvingmixed numbers it is necessary first to convertthem into improper fractions. Example:_!_22x7 + 3 171 1 =7The numerator of the improper fraction isequal to the whole number multiplied by thedenominator of the original fraction, to whichthe numerator is added. That is in the aboveexample we multiply 2 by 7 and then add 3to obtain 17 for the numerator. The denominatoris the same as is the denominator ofthe original fraction. In the following examplewe have added two mixed numbers.2..!..+ 3 __!__..!!_ +..!!- [17x4 +~]7 4- 7 4- 7x4 4x7MultiplyingFractions68 105 173 5=n+n=n = 621All vulgar fractions are multipliedby multiplying the numeratorstogether and the denominatorstogether, as shown in the followingexample:3 2 [3x2] 6 34 X-s-= 4X5 =20=10As above, the step indicated in brackets isusually not written down since it may easilybe performed mentally. As with addition andsubtraction any mixed numbers should befirst reduced to improper fractions as shownin the following example:3 1 3 13 39 13nX 4 -3-=nXT=69=nDivision ofFractionsExample:Fractions may be most easilydivided by inverting the divisorand then multiplying.2 • 3 2 4 85-:-4=-s-X3=15In the above example it will be seen that todivide by % is exactly the same thing as tomultiply by 4/3. Actual division of fractionsis a rather rare operation and if necessary isusually postponed until the final answer is securedwhen it is often desired to reduce theresulting vulgar fraction to a decimal fractionby division. It is more common andusually results in least overall work to reducevulgar fractions to decimals at the beginningof a problem. Examples:3 5a= 0.375 32 = 0.156250.1562532 ) 5.000003 2lio1 60200192so64i6o160
<strong>HANDBOOK</strong>It will be obvious that many vulgar fractionscannot be reduced to exact decimal equivalents.This fact need not worry us, however,since the degree of equivalence can always beas much as the data warrants. For instance,if we know that one-third of an ampere isflowing in a given circuit, this can be writtenas 0. 333 amperes. This is not the exactequivalent of 1/3 but is close enough since itshows the value to the nearest thousandth ofan ampere and it is probable that the meterfrom which we secured our original data wasnot accurate to the nearest thousandth of anampere.Thus in converting vulgar fractions to adecimal we unhesitatingly stop when we havereached the number of significant figures warrantedby our original data, which is veryseldom more than three places (see sectionSignificant Figures later in this chapter).When the denominator of a vulgar fractioncontains only the factors 2 or 5, division canbe brought to a finish and there will be noremainder, as shown in the examples above.When the denominator has other factorssuch as 3, 7, 11, etc., the division will seldomcome out even no matter how long it is continuedbut, as previously stated, this is ofno consequence in practical work since it maybe carried to whatever degree of accuracy isnecessary. The digits in the quotient willusually repeat either singly or i..n groups, althoughthere may first occur one or moredigits which do not repeat. Such fractionsare known as repeating fractions. They aresometimes indicated by an oblique line (fractionbar) through the digit which repeats, orthrough the first and last digits of a repeatinggroup. Example:1 I3 = 0.3333 .... =0-P+ = 0.142857142857 .... = o-}4285/The foregoing examples contained only repeatingdigits. In the following example anon-repeating digit precedes the repeatingdigit:7 I30 = 0.2333 .... = 0.2JWhile repeating decimal fractions can beconverted into their vulgar fraction equivalents,this is seldom necessary in practicalwork and the rules will be omitted here.Powers and When a number is to be mul-Roots tiplied by itself we say thatit is to be squared or to berai.red to the second power. When it is to bemultipled by itself once again, we say thatit is cubed or raiuJ to the third power.Division of Fractions 747In general terms, when a number is to bemultipled by itself we speak of raising to apower or involution; the number of timeswhich the number is to be multiplied by itselfis called the order of the power. Thestandard notation requires that the order ofthe power be indicated by a small numberwritten after the number and above the line,called the exponent. Examples:2' = 2 X 2, or 2 squared, or the secondpower of 22' = 2 X 2 X 2, or 2 cubed, or the thirdpower of 22' = 2 X 2 X 2 X 2, or the fourth powerof 2Sometimes it is necessary to perform thereverse of this operation, that is, it may benecessary, for instance, to find that numberwhich multiplied by itself will give a productof nine. The answer is of course 3. Thisprocess is known as extracting the root orevolution. The particular example which iscited would be written:The sign for extracting the root is V,which is known as the radical sign; the orderof the root is indicated by a small numberabove the radical as in r, which would meanthe fourth root; this number is called theindex. When the radical bears no index, thesquare or second root is intended.Restricting our attention for the momentto square root, we know that 2 is the squareroot of 4, and 3 is the square root of 9. Ifwe want the square root of a number between3 and 9, such as the square root of 5, it isobvious that it must lie between 2 and 3. Ingeneral the square root of such a number cannotbe exactly expressed either by a vulgarfraction or a decimal fraction. However, thesquare root can be carried out decimally asfar as may be necessary for sufficient accuracy.In general such a decimal fraction willcontain a never-ending series of digits withoutrepeating groups. Such a number is anirrational number, such asV5 = 2.2361 ••..The extraction of roots is usually done bytables or logarithms the use of which willbe described later. There are longhand methodsof extracting various roots, but we shallgive only that for extracting the square rootsince the others become so tedious as to makeother methods almost invariably preferable.Even the longhand method for extracting thesquare root will usually be used only if loga-
748 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>rithm tables, slide rule, or table of roots arenot handy.Extracting theSquare RootFirst divide the number theroot of which is to be extractedinto groups of twodigits starting at the decimal point and goingin both directions. If the lefthandmost groupproves to have only one digit instead of two,no harm will be done. The righthandmostgroup may be made to have two digits byannexing a zero if necessary. For example,let it be requited to .find the square root of5678.91. This is to be divided off as follows:y56' 78.91The mark used to divide the groups maybe anything convenient, although the primesign(') is most commonly used for thepurpose.Next .find the largest square which is containedin the .first group, in this case 56. Thelargest square is obviously 49, the square of 7.Place the 7 above the .first group of the numberwhose root is to be extracted, which issometimes called the dividend from analogyto ordinary division. Place the square of this.figure, that is 49, under the .first group, 56,and subtract leaving a remainder of 7.7..rs6' 78.91497Bring down the next group and annex it tothe remainder so that we have 778. Now tothe left of this quantity write down twice theroot so far found (2 X 7 or 14 in this example),annex a cipher as a trial divisor, andsee how many times the result is containedin 778. In our example 140 will go into 7785 times. Replace the cipher with a 5; andmultiply the resulting 145 by 5 to give 725.Place the 5 directly above the second groupin the dividend and then subtract the 725from 778.7 5vs6' 78.9149140 7 78145 X 5 = 7 25The next step is an exact repetition of theprevious step. Bring down the third groupand annex it to the remainder of 53, giving5391. Write down twice the root already53found and annex the cipher (2 x 75 or 150plus the cipher, which will give 1500). 1500will go into 5391 3 times. Replace the lastcipher with a three and multiply 1503 by 3 togive 4509. Place 3 above the third group.Subtract to .find the remainder of 882. Thequotient 75.3 which has been found so far isnot the exact square root which was 'Clesired;in most cases it will be sufficiently accurate.However, if greater accuracy is desired groupsof two ciphers can be brought down and theprocess carried on as long as necessary.140145 X 5 =15001503 X 3 =7 s. 3v56' 78.91497 787 2553 9145 098 82Each digit of the root should be placed directlyabove the group of the dividend fromwhich it was derived; if this is done thedecimal point of the root will come directlyabove the decimal point of the dividend .Sometimes the remainder after a squarehas been subtracted (such as the 1 in the followingexample) will not be sufficiently largeto contain twice the root already found evenafter the next group of .figures has beenbrought down. In this case we write a cipherabove the group just brought down and bringdown another group.7. 0 8 2v5o.16' oo· oo491400 1 16 001408 X 8 = 1 12 6414160 3 36 0014162 X 2 = 2 83 2452 76In the above example the amount 116 was notsufficient to contain twice the root alreadyfound with a cipher annexed to it; that is,it was not sufficient to contain 140. Thereforewe write a zero above 16 and bring downthe next group, which in this example is apair of ciphers.Order ofOperationsOne frequently encounters problemsin which several of the fundamentaloperations of arithmeticwhich have been described are to be performed.The order in which these operations
<strong>HANDBOOK</strong>Order of Operations 749must be performed is important. First al! powersand roots should be calculated; multipli·cation and division come next; adding andsubtraction come last. In the examplewe must first square the 4 to get 16; then wemultiply 16 by 3, making 48, and to theproduct we add 2, giving a result of 50.If a different order of operations were followed,a different result would be obtained.For instance, if we add 2 to 3 we would obtain5, and then multiplying this by the squareof 4 or 16, we would obtain a result of 80,which is incorrect.In more complicated forms such as frac·tions whose numerators and denominators mayboth be in complicated forms, the numeratorand denominator are first found separatelybtfore the division is made, such as in thefollowing example:3 X 4 + 5 X 2 12 + 10 222 X 3 + 2 + 3 = 6 + 2 + 3 = 11= 2Problems of this type are very common indealing with circuits containing several inductances,capacities, or resistances.The order of operations specified above doesnot always meet all possible conditions; if aseries of operations should be performed in adifferent order, this is always indicated byparenthe.re.r or hracketJ. for example:2 + 3 X 4' = 2 + 3 X 16 = 2 + 48 = SO
750 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>E =I X Rwhere E = e.m.f. in voltsI = current in amperesR = resistance in ohmsLetters therefore represent numbers, and forany letter we can read "any number." Whenthe same letter occurs again in the same expressionwe would mentally read "the samenumber," and for another letter "another numberof any value."These letters are connected by the usual operationalsymbols of arithmetic, +, -, X, +,and so forth. In algebra, the sign for divisionis seldom used, a division being usually writtenas a fraction. The multiplication sign, X, isusually omitted or one may write a periodonly. Examples:2 X a X b = 2ab2.3.4.5a = 2 X 3 X 4 X 5 X aIn practical applications of algebra, an expressionusually states some physical law andeach letter represents a variable quantity whichis therefore called a variable. A fixed numberin front of such a quantity (by which it is tobe multiplied) is known as the coefficient.Sometimes the coefficient may be unknown, yetto be determined; it is then also written as aletter; k is most commonly usee\ for this purpose.The NegativeSignIn ordinary arithmetic weseldom work with negativenumbers, although we maybe "short" in a subtraction. In algebra, however,a number may be either negative or positive.Such a thing may seem academic but anegative quantity can have a real existence.We need only refer to a debt being considereda negative possession. In electrical work, however,a result of a problem might be a negativenumber of amperes or volts, indicating that thedirection of the current is opposite to the directionchosen as positive. We shall have illustrationsof this shortly.Having established the existence of negativequantities, we must now learn how to workwith these negative quantities in addition, subtraction,multiplication and so forth.In addition, a negative number !dded to apositive number is the same as subtracting apositive number from it.7 7-3 . 3-- (add) IS the same as (subtract)4 4or we might write it7+C-3l=7-3=4Similarly, we have:a+ (-b)= a- bWhen a minus sign is in front of an expressionin brackets, this minus sign has the effectof reversing the signs of every term within thebrackets:- (a - b) = - a + b- (2a + 3b - 5cl = - 2a - 3b + 5cMultiplication. When both the multiplican_dand the multiplier are negative, the product ISpositive. When only one (either one) is negativethe product is negative. The four possiblecases are illustrated below:+X+=+-X+=-+X-=X-=+Divi.rion. Since division is but the reverse ofmultiplication, similar rules apply for the signof the quotient. When both the dividend a_ndthe divisor have the same sign (both negativeor both positive) the quotient is positive. Ifthey have unlike signs (one positive and onenegative) the quotient is negative.+ +-=+ -=+-= -=++Powers. Even powers of negative numbersare positive and odd powers are negative. Powersof positive numbers are always positive.Examples:-2•=-2X-2=+42'=-2X-2X-2=+4X- 2=- 8Roots. Since the square of a negative numberis positive and the square of a positivenumber is also positive, it follows that a positivenumber has two square roots. The squareroot of 4 can be either + 2 or -2 for ( + 2)x (+2) = +4 and (-2) X (-2) = +4.Addition and Polynomials are quantitiesSubtraction like 3ab' + 4ab' - 7a'b'which have several terms ofdifferent names. When adding polynomials,only terms of the same name can be taken together.7a' + 8 ab' + 3 a'b + 3a 3 - 5 ab' - b'Sa' + 3 ab' + 3 a'b - b' + 3
<strong>HANDBOOK</strong>Collecting terms. When an expression containsmore than one term of the same name,these can be added together and the expressionmade simpler:5 x' + 2 xy + 3 xy' - 3 x' + 7 xy =5 x' - 3 x' + 2 xy + 7 xy + 3 xy' =Multiplicationthem together.2 x' + 9 xy + 3 xy'Multiplication of single termsis indicated simply by writinga X b is written as aba X b' is written as ab'Bracketed quantities are multiplied by asingle term by multiplying each term:a (b + c + d) = ab + ac + adWhen two bracketed quantities are multiplied,each term of the first bracketed quantityis to be multiplied by each term of the secondbracketed quantity, thereby making every possiblecombination.(a + b) (c + dl = ac + ad + be + bdIn this work particular care must be takento get the signs correct. Examples:(a + b) (a - b) = a' + ab - ab - b' =a' - b'(a + bl (a + b) = a' + ab + ab + b' =a'+ lab + b'(a - bl ( a - b) = a' - ab - ab + b' =a•- 2 ab + b'Division It is possible to do longhand divisionin algebra, although it issomewhat more complicated than in arithmetic.However, the division will seldom comeout even, and is not often done in this form.Th!i method ts as follows: Write the terms ofthe dividend in the order of descending powersof one variable and do likewise with the divisor.Example:Divide 5a'b + 21 b' + 2a' -2a - 3b26ab' byWrite the dividend in the order of descendingpowers of a and divide in the same way as inarithmetic.Division 751a'+ 4 ab -7b'2a- 3b) 2a' + 5a'b- 26 ab' + 21b'2a'- 3o'b+ Sa'b - 26ab'+ Sa'b - I lab'14ab' + 21b 314ab' + 21b'Another example: Divide x' - y' by x - yx - y ) x' + 0 + 0 - y 3 ( x' + xy + Yx'- x'y+ x'yx'y -xy'+ xy'- y'xy'- y'Factoring Very often it is necessary to simplifyexpressions by finding a factor.This is done by collecting two or moreterms having the same factor and bringing thefactor outside the brackets:6ab + 3ac = 3a(2b +c)In a four term expression one can take togethertwo terms at a time; the intention is totry getting the terms within the brackets thesame after the factor has been removed:30ac - ISbc + lOad- 6bd =6c (5a - 3b) + 2d (5o - 3b) =(5o - 3bl (6c + 2dlOf course, this is not always possible andthe expression may not have any factors. Asimilar process can of course be followed whenthe expression has six or eight or any evennumber of terms.A special case is a three-term polynomial,which can sometimes be factored by writingthe middle term as the sum of two terms:x' - 7xy + 12y' may be rewritten asx' - 3xy - 4xy + 12y' =x (x - 3y) - 4y (x - 3y) -(x - 4y) (x - 3y)The middle term should be split into twoin such a way that the sum of the two newterms equals the original middle term and thattheir product equals the product of the twoouter terms. In the above example these conditionsare fulfilled for - 3xy -'-- 4 xy = - 7xyand (-3xy) (--4xy) = 12 x'y'. It is notalwayspossible to do this and there are then nosimple factors.
752 Radio Mathematics and Calculotions <strong>THE</strong> <strong>RADIO</strong>Working withPowenand RootsWhen two powers of thesame number are to be multiplied,the exponents areadded.a' X a' = aa X aaa = aaaaa = a• ora2 X aa = a(2+3> =a~:~b' X b = b'c' X c' = c"Similarly, dividing of powers is done bysubtracting the exponents.~- _l>b_bbb - b' b"._ -b(ll-3) - b'b3 - bbb - or b' - -Now we are logically led into some importantnew ways of notation. We have seen thatwhen dividing, the exponents are subtracted.This can be continued into negative exponents.In the following series, we successively divideby a and since this can now be done in twoways, the two ways of notation must have thesame meaning and be identical.a• a• a-'=__!_aa' a'=• a-• = ~.a• a'= 1 a-• = ~.These examples illustrate two rules: ( 1) anynumber raised to "zero'· power equals one orunity; ( 2) any quantity raised to a negativepower is the inverse or reciprocal of the samequantity raised to the same positive power.n' = 1Roots. The product of the square root oftwo quantities equals the square root of theirproduct.Also, the quotient of two roots is equal to theroot of the quotient.Note, however, that in addition or subtractionthe square root of the sum or difference isnot the same as the sum or difference of thesquare roots.2 = Ibut /9='4 = VS = 2.2361Thus, V9- V 4 = 3 -Likewise Va + v'b is not the same as v"Ci+bRoots may be written as fractional powers.Thus Va may be written as a II becausevax va=aand, a\.11 X a'h = a\.11+1-11 = a' = aAny root may be written in this formThe same notation is also extended in thenegative direction:b -'12 _1_- __!__- b 1h- yb c-'13 = !A> = _,,-c v cFollowing the previous rules that exponentsadd when powers are multiplied,~·x ~ = V"''but also a' 12 X a' 12 = a'~-ttherefore a~> = aPowers of powers. When a power is againraised to a power, the exponents are multiplied;(a')'= a•(a')'= au(b-'l' = b"'(b-') -·= b'This same rule also applies to roots of rootsand also powers of roots and roots of powersbecause a root can always be written as a fractionalpower.~=«;for (a'l2)' 12 =a%Removing radicals. A root or radical in thedenominator of a fraction makes the expressiondifficult to handle. If there must be a radicalit should be located in the numeratorrather than in the denominator. The removalof the radical from the denominator is doneby multiplying both numerator and denominatorby a quantity which will remove the radicalfrom the denominator, thus rationalizing it:_1 ___ .;; - 1 _r-Va-VaXVa-avaSuppose we have to rationalize3ah'Va + Vi) In t IScase we must multiplynumerator and denominator by Va - VI),' thesame terms but with the second having theopposite sign, so that their product will notcontain a root.3ava+Jb
<strong>HANDBOOK</strong>ImaginaryNumbersSince the square of a negativenumber is positive and thesquare of a positive number isalso positive, the square root of a negativenumber can be neither positive nor negative.Such a number is said to be imaginary; themost common such number ( v--=-i) is oftenrepresented by the letter i in mathematicalwork or j in electrical work.Y=T = i or j and i' or j' = - 1Imaginary numbers do not exactly correspondto anything in our experience and it isbest not to try to visualize them. Despite thisfact, their interest is much more than academic,for they are extremely useful in many calculationsinvolving alternating currents.The square root of any other negative numbermay be reduced to a product of two roots,one positive and one negative. For instance:or, in general-.r::a =iVaSince i = Y-1, the powers of i have thefollowing values:i 2 = -1i 3 = -1 Xi= -ii'= +1i'=+1Xi=iImaginary numbers are different from eitherpositive or negative numbers; so in addition orsubtraction they must always be accounted forseparately. Numbers which consist of both realand imaginary parts are called complex numbers.Examples of complex numbers:3 + 4i = 3 + 4 v'=1a+ bi =a+ bv=TSince an imaginary number can never beequal to a real number, it follows that in anequality likea + bi = c + dia must equal c and bi must equal diComplex numbers are handled in algebrajust like any other expression, considering ias a known quantity. Whenever powers of ioccur, they can be replaced by the equivalentsgiven above. This idea of having In one equationtwo separate sets of quantities which mustPowers, Roots, Imaginaries 753be accounted for separately, has found a symbolicapplication in vector notation. These arecovered later in this chapter.Equations of theFirst DegreeAlgebraic expressions usuallycome in the form ofequations, that is, one setof terms equals another set of terms. The simplestexample of this is Ohm's Law:E = IROne of the three quantities may be unknownbut if the other two are known, the third canbe found readily by substituting the knownvalues in the equation. This is very easy if itis E in the above example that is to be found;but suppose we wish to find I while E and Rare given. We must then rearrange the equationso that I comes to stand alone to the leftof the equality sign. This is known as solvingthe equation for I.Solution of the equation in this case is donesimply by transposing. If two things are equalthen they must still be equal if both are multipliedor divided by the same number. Dividingboth sides of the equation by R:E _ IR ER-R= I or I= ItIf it were required to solve the equation forR, we should divide both sides of the equationby I.EE- 1 = R orR= 1A little more complicated example is theequation for the reactance of a condenser:1X= 2nfCTo solve this equation for C, we may multiplyboth sides of the equation by C and divideboth sides by XX ~ --- 1 -- x ~ orXX- 2nfC X'1C=~This equation is one of those which requiresa good knowledge of the placing of the decimalpoint when solving. Therefore we give afew examples: What is the reactance of a 2'il'l'fd. capacitor at 1000 kc.? In filling in thegiven values in the equation we must rememberthat the units usea are farads, cycles, andohms. Hence, we must write 25 l'l'fd. as 25millionths of a millionth of a farad or 25 x10 12 farad; similarly, 1000 kc. must be convertedto 1,000,000 cycles. Substituting thesevalues in the original equation, we have
754 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>X = "2-=x:--.3.-. 1"4'x=-1',o"o"'o"'t,o"o"'o-:x:o5~x;;-;-1n:o-=tn110•X = 6.21xfo• x 25 x 1o-u = 6.28 x 25= 6360 ohmsA bias resistor of 1000 ohms should be bypassed,so that at the lowest frequency the reactanceof the condenser is 1/1 Oth of that ofthe resistor. Assume the lowest frequency to be50 cycles, then the required capacity shouldhave a reactance of 100 ohms, at 50 cycles:1C = 2x3.14x50x100 forads1b 6C = 6_ 28x 5000microfaradsc = 32 pfd.In the third possible case, it may be that thefrequency is the unknown. This happens forinstance in some tone control problems. Supposeit is required to find the frequency whichmakes the reactance of a 0.03 pfd. condenserequal to 100,000 ohms. . .First we must solve the equatwn for f. Th1sis done by transposition.1X=21tfeSubstituting known values1f = -..2""n,.=:c...,x,....1f = 2 x 3.14 x 0.03 x 10 • x 100,000 cycles1f = 0_ 01884cycles = 53 cyclesThese equations are known as first degreeequations with one unknown. First degree, becausethe unknown occurs only as a first power.Such an equation always has one possible solutionor root if all the other values are known.If there are two unknowns, a single equationwill not suffice, for there are then an infinitenumber of possible solutions. In the caseof two unknowns we need two independentsimultaneous equations. An example of this is:3x + 5y = 7 4x- lOy= 3Required, to find x and y.This type of work is done either by the JubJtitution method or by the elimination method.In the substitution method we might write forthe first equation:3x = 7- 5y .". x =7- 5y3(The symbol .·. means. therefore or hence).This value of x can then be substituted for xin the second equation making it a single equationwith but one unknown, y.It is, however, simpler in this case to usethe elimination method. Multiply both sides ofthe first equation by two and add it to thesecond equation:6x + lOy= 144x- lOy= 3------addlOx 17 X= 1.7Substituting this value of x in the first equation,we have5.1 + 5y = 7 .-. 5y = 7- 5.1 - 1.9 :.y = 0.38+ I,+ 1;,:·~2000OHMSlIOOOOHMSA~rBflOOOOHMSc 0Figure 3.Itt this simple network the current dividesthrough the 2000-ohm and 3000-ohm resistors.The current through each may be found byusing two simultaneous linear equations. Notethat the arrows indicate the direction of electronflow as explained on page 18.An application of two simultaneous linearequations will now be given. In Figure 3 asimple network is shown consisting of three resistances;let it be required to find the currentsI, and I, in the two branches.The general way in which all such problemscan be solved is to assign directions tothe currents through the various resistances.When these are chosen wrong it will do noharm for the re~ult of the equations will thenbe negative, showing up the error. In this simpleillustration there is, of course, no such difficulty.Next we write the equations for the meshes,in accordance with Kirchhoff's second law. Allvoltage drops in the direction of the curvedarrow are considered positive, the reverse onesnegative. Since there are two unknowns wewrite two equations.1000 (I,+ 1,) + 2000 I.= 6-2000 11 + 3000 I,= 0Expand the first equation3000 I. + 1000 '· = 6'•
<strong>HANDBOOK</strong>Quadratic Equations 755Multiply the second equation by 2 and addit to the third equationFigure 4.A MORE COMPLICATED PROBLEM REQUIRING <strong>THE</strong> SOLUTIONOF CURRENTS IN A NETWORK.This problem Is similar to that in figure 3 butrequires the use of three simultaneous linearequations.Multiply this equation by 39000 I,+ 3000 I,= 18Subtracting the second equation from the first11000 I,= 18I,= 18/11000 = 0.00164 amp.Filling in this value in the second equation3000 I,= 3.28 I,= 0.00109 amp.A similar problem but requiring three equationsis shown in Figure 4. This consists of anunbalanced bridge and the problem is to findthe current in the bridge-branch, I,. We againassign directions to the different currents,guessing at the one marked L. The voltagesaround closed loops ABC ( eq. ( 1)} and BDC( eq. ( 2) ] equal zero and are assumed to bepositive in a counterclockwise direction; thatfrom D to A equals 10 volts (eq. (3)}.(1)-1000 I,+ 2000 I, - 1000 Ia = 0(2)-1000 ( 1,-1,) + 1000 Ia+ 3000 ( 1.+ 1,) =0(3)1000 I,+ 1000 (1,- 1,) - 10 = 0Expand equations (2) and (3)(2)-1000 I, + 3000 I, + 5000 I, = 0(3)2000 I, - 1000 Ia - 10 = 0Subtract equation (2) from equation (1)(a)- 1000 I, - 6000 Ia = 0(b)6000 1, + 9000 I, - 10 = 0Now we have but two equations with twounknowns.Multiplying equation (a) by 6 and addingto equation (b) we have-27000" - 10 = 01 3 = - 1 0/27000 = -0.00037 amp.Note that now the solution is negativewhich means that we have drawn the arrowfor 1:. in Figure 4 in the wrong direction. Thecurrent is 0.37 rna. in the other direction.Second Degree orQuadratic EquationsA somewhat similarproblem in radio wouldbe, if power in wattsand resistance in ohms of a circuit are given,to find the voltage and the current. Example:When lighted to normal brilliancy, a 100 wattlamp has a resistance of 49 ohms; for whatline voltage was the lamp designed and whatcurrent would it take~Here we have to use the simultaneous equations:P = Eland E = IRFilling in the known values:P = El = 100 and E = IR =I X 49Substitute the second equation into the firstequationP = El =
756 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>where x is the unknown and a, b, and c areconstants.This type of equation can sometimes besolved by the method of factoring a threetermexpression as follows:Also: (XL - R 22x 2 + 7x + 6 = 02r + 4x + 3x + 6 = 02x (x + 2) + 3 (x + 2) = 0Definition(2x + 3) (x + 2) = 0 and Use2x, + 3 = 0 x.+2=02x, = -3 X.= -2x, = -1YzX - -b±vb•-4ac- 2ac-a=b_-7±v49-8X6_-7±VJ -7±1- 4 - 4 --4--7x + 1' = -- 4 -- = - P/zX. - -7-1 4 --- 2ab = c% = V R 2 + IXL - Xol 2z• = R 2 + (XL - Xol 1Vc=aand R' = zt - (XL - Xo) •factoring:There are two possibilities when a productis zero. Either the one or the other factorequals zero. Therefore there are two solutions.Since factoring is not always easy, the followinggeneral solution can usually be employed;in this equation a, b, and c are the coefficientsreferred to above.Applying this method of solution to the previousexample:XA practical example involving quadratics isthe law of impedance in a.c. circuits. However,this is a simple kind of quadratic equationwhich can be solved readily without the useof the special formula given above.This equation can always be solved for R,by squaring both sides of the equation. Itshould now be understood that squaring bothsides of an equation as well as multiplyingboth sides with a term containing the unknownmay add a new root. Since we know here thatZ and R are positive, when we square the expressionthere is no ambiguity.and ± (XL - Xo> = " z• - R 2But here we do not know the sign of the solutionunless there are other facts which indicateit. To find either XL or Xc alone it wouldhave to be known whether the one or the otheris the larger.LogarithmsA logarithm is the power (or exponent)to which we must raiseone number to obtain another.Although the large numbers used in logarithmicwork may make them seem difficult orcomplicated, in reality the principal use oflogarithms is to simplify calculations whichwould otherwise be extremely laborious.We have seen so far that every operationin arithmetic can be reversed. If we have theaddition:we can reverse this operation in two ways. Itmay be that b is the unknown, and then wereverse the equation so that it becomesIt is also possible that we wish to know a, andthat b and c are given. The equation then becomesc-b=aWe call both of these reversed operations subtraction,and we make no distinction betweenthe two possible reverses.Multiplication can also be reversed in twomanners. In the multiplicationwe may wish to know a, when b and c aregiven, or we may wish to know b when a andc are given. In both cases we speak of division,and we make again no distinction between thetwo.In the case of powers we can also reversethe operation in two manners, but now theyare not equivalent. Suppose we have the equationIf a is the unknown, and b and c are given,we may reverse the operation by writingThis operation we call taking the root. Butthere is a third possibility: that a and c aregiven, and that we wish to know b. In other
<strong>HANDBOOK</strong>words, the question is "to which power mustwe raise a so as to obtain c?". This operationis known as taking the logarithm, and b is thelogarithm of c to the base a. We write thisoperation as follows:log. c = bConsider a numerical example. We know 2 8 =8.We can reverse this operation by asking "towhich power must we raise 2 so as to obtain8?" Therefore, the logarithm of 8 to the base2 is 3, orlog,8=3Taking any single b~e. such as 2, we mightwrite a series of all the powers of the base nextto the series of their logarithms:Number: 2 4 8 16 32 64 128 256 512 1024Logarithm: 1 2 3 4 5 6 7 8 9 1 0We can expand this table by finding termsbetween the terms listed above. For instance,if we let the logarithms increase with % eachtime, successive terms in the upper serieswould have to be multiplied by the square rootof 2. Similarly, if we wish to increase the logarithmby 1/10 at each term, the ratio betweentwo consecutive terms in the upper serieswould be the tenth root of 2. Now this shortlist of numbers constitutes a small logarithmtable. It should be clear that one could findthe logarithm of any number to the base 2.This logarithm will usually be a number withmany decimals.LogarithmicBasesThe fact that we chose 2 asa base for the illustration ispurely arbitrary. Any basecould be used, and therefore there are manypossible systems of logarithms. In practice weuse only two bases: The most frequently usedbase is 10, and the system using this base isknown as the system of common logarithms,or Briggs' logarithms. The second system employsas a base an odd number, designated bythe letter e; e = 2.71828 .... This is knownas the natural logarithmic system, also as theNapierian system, and the hyperbolic system.Although different writers may vary on thesubject, the usual notation is simply log a forthe common logarithm of a, and log. a (orsometimes ln a) for the natural logarithm ofa. We shall use the common logarithmic systemin most cases, and th6refore we shall examinethis system more clo!iely.Common In the system wherein 10 is theLogarithms base, the logarithm of 10 equals1; the logarithm of 100 equals 2,etc., as shown in the following table:Logarithms 757log 10 = log 10' =log 100 = log 10' = 2log 1,000 =log 10= 3 3log 10,000 -log 10' =4log 100,000 = log 10" = 5log 1 ,000,000 = log l o• = 6This table can be extended for numbers lessthan 10 when we remember the rules of powersdiscussed under the subject of algebra.Numbers less than unity, too, can be writtenas powers of ten.log 1 = log 1 0' = 0log 0.1 = log 1 o-l = - 1log 0.01 =log 1o-• = -2log 0.001 =log 10-' = -3log 0.0001 =log 1o-• = -4From these examples follow several rules:The logarithm of any number between zeroand + 1 is negative; the logarithm of zero isminus infinity; the logarithm of a numbergreater than + 1 is positive. Negative numbershave no logarithm. These rules are trueof common logarithms and of logarithms toany base.The logarithm of a number between thepowers of ten is an irrational number, that is,it has a never ending series of decimals. For instance,the logarithm of 20 must be between1 and 2 because 20 is between 10 and 100; thevalue of the logarithm of 20 is 1.30103 ....The part of the logarithm to the left of thedecimal point is called the characteriJtic, whilethe decimals are called the mantissa. In thecase of 1.30103 .. , the logarithm of 20, thecharacteristic is 1 and the mantissa is .30103 ..Properties ofLogarithmsIf the base of our system isten, then, by definition of alogarithm:10iog•=aor, if the base is raised to the power having anexponent equal to the logarithm of a number,the result is that number.The logarithm of a product is equal to thesum of the logarithms of the two factors.log ob = log o + log bThis is easily proved to be true because, it
N10111213141516171819202122232425262728293031323334353637383940414243444546474849505152535400000041407921139146117612041230425532788301032223424361738023979415043144472462447714914505151855315544155635682579859116021612862326335643565326628672168126902699070767160724373241004304530828117314921790206823302577281030323243344436363820399741664330448746394786492850655198532854535575569458095922603161386243634564446542663767306821691169987084716872517332200860492086412061523181820952355260128333054326334643655383840144183434645024654480049425079521153405465558757055821593360426149625363556454655166466739683069207007709371777259734030128053108991239155318472122238026252856307532843483367438564031420043624518466948144955509252245353547855995717583259446053616062636365646465616656674968396928701671017185726773484017005690934127115841875214824052648287830963304350236923874404842164378453346834829496951055237536654905611572958435955606461706274637564746571666567586848693770247110719372757356021206070969130316141903217524302672290031183324352237113892406542324393454846984843498351195250537855025623574058555966607561806284638564846580667567676857694670337118720272847364Figure 5. FOUR PLACE LOGARITHM TABLES.025306451004133516441931220124552695292331393345354137293909408242494409456447134857499751325263539155145635575258665977608561916294639564936590668467766866695570427126721072927372029406821038136716731959222724802718294531603365356037473927409942654425457947284871501151455276540355275647576358775988609662016304640565036599669367856875696470507135721873007380033407191072139917031987225325042742296731813385357937663945411642814440459447424886502451595289541655395658577558885999610762126314641565136609670267946884697270597143722673087388037407551106143017322014227925292765298932013404359837843962413342984456460947574900503851725302542855515670578658996010611762226325642565226618671268036893698170677152723573167396~-~-~o-~l~~2-'---3-'-455565758596061626364656667686970717273747576777879808182838485868788899091929394959697989974047482755976347709778278537924799380628129819582618325838884518513857386338692875188088865892189769031908591389191924392949345939594459494954295909638968597319777982398689912995674127490756676427716778978607931800080698136820282678331839584578519857986398698875688148871892789829036909091439196924892999350940094509499954795959643968997369782982798729917996174197497757476497723779678687938800780758142820982748338840184638525858586458704876288208876893289879042909691499201925393049355940594559504955296009647969497419786983298779921996574277505758276577731780378757945801480828149821582808344840784708531859186518710876888258882893889939047910191549206925893099360941094609509955796059652969997459791983698819926996974357513758976647738781078827952802180898156822282878351841484768537859786578716877488318887894389989053910691599212926393159365941594659513956296099657970397509795984198869930997457443752075977672774578187889795980288096816282288293835784208482854386038663872287798837889389499004905891129165921792699320937094209469951895669614966197089754980098459890993499786745175287604767977527825789679668035810281698235829983638426848885498609866987278785884288998954900990639117917092229274932593759425947495239571961996669713975998059850989499399983774597536761276867760783279037973804181098176824183068370843284948555861586758733879188488904896090159069912291759227927993309380943094799528957696249671971797639809985498999943998787466754376197694776778397910798080488116818282488312837684398500856186218681873987978854891089659020907491289180923292849335938594359484953395819628967597229768981498599903994899919747475517527770177747846791779878055812281898254831983&184458506856786278686874588028859891589719025907991339186923892899340939094409489953895869633968097279773981898639908995~9996;:o00....0~0.......::::J(1)30.......(){f)0:J0....()0()c0.......0:J{f)-f:I:"'0
<strong>HANDBOOK</strong>was shown before that when multiplying topowers, the exponents are added; therefore,a X b =l01oga X 10 1ogb= 10 (1oga+ >ogb>Similarly, the logarithm of a quotient is thedifference between the logarithm of the dividendand the logarithm of the divisor.log{-= log a -log bThis is so because by the same rules of exponents:~-~- 1Q(Iogn-logb)b - 101ogb -We have thus established an easier way ofmultiplication and division since these operation.rhat•e been reduced to adding and .rubtracting.The logarithm of a power of a number isequal to the logarithm of that number, multipliedby the exponent of the power.log o' = 2 log a and log a' = 3 log aor, in general:log a" = n log aAlso, the logarithm of a root of a numberts equal to the logarithm of that number dividedby the index of the root:log Va =_!_lag anIt follows from the rules of multiplication,that numbers having the same digits but differentlocations for the decimal point, havelogarithms with the same mantissa:log 829 = 2.918555log 82.9 = L918555log 8.29 = 0.918555log 0.829 = -1.918555log 0.0829 = -2.918555lag 829 = log (8.29 X 100) = log 8.29 +log 100 = 0.918555 + 2Logarithm tables give the mantissas of logarithmsonly. The characteristic has to be determinedby inspection. The characteristic isequal to the number of digits to the left of thedecimal point minus one. In the case of logarithmsof numbers less than unity, the characteristicis negative and is equal to the numberof ciphers to the right of the decimal pointplus one.For reasons of convenience in making upLogarithm Tables 759logarithm tables, it has become the rule thatthe mantissa should always be positive. Suchnotations above as -1.918555 really mean(+0.918555 -1) and -2.981555 means( + 0.918555 - 2). There are also some othernotations in use such as1.918555 and 2.918555also 9.918555- 10 8.918555- 107.918555- 10,etc.When, after some addition and subtractionof logarithms a mantissa should come out negative,one cannot look up its equivalent numberor anti-logarithm in the table. The mantissamust first be made positive by adding andsubtracting an appropriate integral number.Example: Suppose we find that the logarithmof a number is -0.34569, then we can transformit into the proper form by adding andsubtracting 11 -1-0.345690.65431 - 1 or - 1.65431Using Logarithm TablesLogarithms are used for calculations involvingmultiplication, division, powers, and roots.Especially when the numbers are large and forhigher, or fractional powers and roots, this becomesthe most convenient way.Logarithm tables are available giving thelogarithms to three places, some to four places, ·others to five and six places. The table to usedepends on the accuracy required in the resultof our calculations. The four place table,printed in this chapter, permits the finding ofanswers to problems to four significant figureswhich is good enough for most constructionalpurposes. If greater accuracy is required a fiveplace table should be consulted. The five placetable is perhaps the most popular of all.Referring now to the four place table, tofind a common logarithm of a number, proceedas follows. Suppose the number is 5576.First, determine the characteristic. An inspectionwill show that the characteristic shouldbe 3. This figure is placed to the left of thedecimal point. The mantissa is now found byreference to the logarithm table. The first twonumbers are 55; glance down the N columnuntil coming to these figures. Advance to theright until coming in line with the columnheaded 7; the mantissa will be 7459. (Note thatthe column headed 7 corresponds to the thirdfigure in the number 5576.) Place the mantissa7459 to the right of the decimal point, makingthe logarithm of 5576 now read 3.7459. Important:do not consider the last figure 6 in the
760 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>N L. 0 • I 4 5 7 9 P.P.250 39 794 811 829 846 863 881 898 915 933 950lSI m 985 *OOPOIP037 *OS4 '071 *088 *10&*123 18252 40 140 157 175 192 209 226 243 261 278 295 I 1.8253 312 329 346 364 381 398 415 432 449 466 2 3.6254 483 500 518 535 552 569 586 603 620 637 3 5.44 7.2255 654 671 688 705 722 739 756 773 790 807 etc.Figure 6.A SMALL SECTION OF A FIVE PLACELOGARITHM TABLE.Logarithms may be found with greater accuracywith such tables, but they are only ofuse when the accuracy of the original datawa11ants greater precision In the figure work.Slightly greater accuracy may be obtained forIntermediate points by Interpolation, as explainedIn the text.number 5576 when looking for the mantissain the accompanying four place tables; infact, one may usually disregard all digits be·yond the first three when determining the man·tissa. (lntupolation. sometimes used to lind alogarithm more accurately, is unnecessary un·less warranted by unusual accuracy in theavailable data.) However, be doubly sure toinclude all ligures when ascertaining the mag·nitude of the characteristic.To lind the anti-logarithm, the table is usedin reverse. As an example, let us lind the antilogarithmof 1.272 or, in other words, findthe n~mber of which 1.272 is the logarithm.Look m the table for the mantissa closest to272. This is found in the first half of the tableand the nearest value is 2718. Write down the~rst two sig_nilicant ligures of the anti-loganthn;by takm~ the ligures at the beginning ofthe lme on wh1ch 2718 was found. This is 18;add to this, the digit above the column inwhich 2718 was found; this is 7. The anti-logarit.hmis 1~7 but we have not yet placed thedeCimal pomt. The characteristic is 1, whichmeans that there should be two digits to theleft of the decimal point. Hence, 18.7 is theanti-logarithm of 1.272.For the sake of completeness we shall alsodescribe the same operation with a live-placetable where interpolation is done by means oftables of proportional parts (P.P. tables)_Therefore we are reproducing here a smallpart of one page of a live-place table.Finding the logarithm of 0.025013 is done asfollows: We can begin with the characteristicwhich is - 2. Next find the first three digits i~the col~mn, headed by N and immediatelyafter .this we see 39, the first two digits of themantissa. Then look among the headings ofthe other columns for the next digit of thenumber, in this case 1. In the column headedby 1 and on the line headed 250, we 'find thenext three digits of the logarithm, 811. So far,t?e logarithm is -2.39811 but this is the loganthmof 0.025010 and we want the logarithmof 0.025013. Here we can interpolate by observingthat the difference between the log of0.02501 and 0.02502 is 829 - 811 or 18 inthe last two significant figures. Looking in' theP.P. table marked 18 we find after 3 the number'i.4 which is to be added to the logarithm.-2.398115.4-2.39816, the logarithm of 0.025013Since ou~ t~ble is only good to live places,we must ehmmate the last figure given in theP.P. table if it is less than 5, otherwise wemust ~dd one to the next to the last figure,roundmg off to a whole number in the P.P.table.Finding the anti-logarithm is done the same~ay but ":ith the procedure reversed. SupposeIt IS requ~red to lind the anti-logarithm of0.40100. Fmd the first two digits in the columnheaded by L. Then one must look for the nextthree digits or the ones nearest to it, in thecolumns after 40 and on the lines from 40 to41. Now here we find that numbers in the?eighborho?d ~f 100 occur only with an asterIsk on the lme JUSt before 40 and still after 39.The asterisk means that instead of the 39 asthe first two digits, these mantissas shouldhave 40 as the first two digits. The logarithm0.40100 is between the logs 0.40088 and0.40106; the anti-logarithm is between 2517and 2518. The difference between the twologarithms in the table is again 18 in thel~st two .ligures and our logarithm 0.40100differs With the lower one 12 in the lastfigures. Look in the P.P. table of 18 whichnumber comes closest to 12. This is foundto be 12.6 for 7 X 1.8 = 12.6. Thereforewe may add the digit 7 to the anti-logarithmalready found; so we have 25177. Nextplace the decimal point according to the rules;There are as many digits to the left of thedecimal point as indicated in the characteristicplus one. The anti-logarithm of 0.40100 is2.5177.In the following examples of the use of logarithms~e sha~l use .only three places from thetables pnnted .u~ th~s chapter since a greaterdegree of preciSJ.On m our calculations wouldn?t be warranted by the accuracy of the datagiven.In a 375 ohm bias resistor flows a currento! 41.5 milliamperes; how many watts are disSipated by the resistor?We write the equation for power in watts:P = I"R
<strong>HANDBOOK</strong>and filling in the quantities m question, wehave:Taking logarithms,P = 0.0415 2 X 375log P = 2 log 0.0415 + log 375log 0.0415 = -2.618So 2 X log 0.0415 = -3.236log 375 = 2.574log P - -1.810antilog = 0.646. Answer = 0.646 wattsCaution: Do not forget that the negativesign before the characteristic belongs to thecharacteristic only and that mantissas are alwayspositive. Therefore we recommend theother notation, for it is less likely to lead toerrors. The work is then written:log 0.0415 = 8.618-102 X log 0.0415 = 17.236-20 = 7.236-10log 375 = 2.574log P = 9.810-10Another example follows which demonstratesthe ease in handling powers and roots.Assume an all-wave receiver is to be built,covering from 550 kc. to 60 me. Can this bedone in five ranges and what will be the requiredtuning ratio for each range if no overlappingis required? Call the tuning ratio ofone band, x. Then the total tuning ratio forfive such bands is x'. But the total tuning ratiofor all bands is 60/0.55. Therefore:X ,_60.- 0.55 or. x -Taking logarithms:1(60lO:SSlog 60 - log 0.55I ogx =5log 60log 0.551.778-1.740---subtract2.038Remember again that the mantissas are positiveand the characteristic alone can be negative.Subtracting -1 is the same as adding + 1.2.038log x =-,-= 0.408x = antilog 0.408 = 2.56The tuning ratio should be 2.56.db01234s678910203040so607080The Decibel 761PowerRatio1.001.261.582.002.513.163.985.016.317.9410.001001,00010,000100,0001,000,00010,000,000100,000,000Figure 7.A TABLE OF DECIBEL GAINS VERSUSPOWER RATIOS.The DecibelThe decibel is a unit for the comparison ofpower or voltage levels in sound and electricalwork. The sensation of our ears due to soundwaves in the surrounding air is roughly proportionalto the logarithm of the energy of thesound-wave and not proportional to the energyitself. For this reason a logarithmic unit is usedso as to approach the reaction of the ear.The decibel represents a ratio of two powerlevels, usually connected with gains or loss dueto an amplifier or other network. The decibelis definedN.b = 10 log ::where Po stands for the output power, P, forthe input power and N.b for the number ofdecibels. When the answer is positive, there isa gain; when the answer is negative, there isa loss.The gain of amplifiers is usually given indecibels. For this purpose both the input powerand output power should be measured. Example:Suppose that an intermediate amplifieris being driven by an input power of 0.2 wattand after amplification, the output is found tobe 6 watts.Po 6.-,.-;-= 0.2 = 30log 30 = 1.48Therefore the gain is 10 X 1.48 = 14.8decibels. The decibel is a logarithmic unit;when the power was multiplied by 30, theppwer level in decibels was increased- by 14.8decibels, or 14.8 decibels added.
762 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>TUBEC:AIN=ASTEP-UPRATIO:: 3.!!1:1Example: In the circuit of Figure 8, the gainin the stage is equal to the amplification in thetube and the step-up ratio of the transformer.If the amplification in the tube is 10 and thestep-up in the transformer is 3. 5, the voltagegain is 35 and the gain in decibels is:20 X log 35 = 20 X 1.54 = 30.8 dbFigure 8.STAGE GAIN.The voltage gain in decibels in this stage isequal to the amplification in the tube plus thestep-up ratio of the transformer, both expressedin decibels.When one amplifier is to be followed byanother amplifier, power gains are multipliedbut the decibel gains are added. If a main amplifierhaving a gain of 1,000,000 (power ratiois 1,000,000) is preceded by a pre-amplifierwith a gain of 1000, the total gain is 1,000,-000,000. But in decibels, the first amplifier hasa gain of 60 decibels, the second a gain of 30decibels and the two of them will have a gainof 90 decibels when connected in cascade.(This is true only if the two amplifiers areproperly matched at the junction as otherwisethere wili be a reflection loss at this pointwhich must be subtracted from the total.)Conversion of power ratios to decibels orvice versa is easy with the small table shownon these pages. In any case, an ordinary logarithmtable will do. Find the logarithm of thepower ratio and multiply by ten to find decibels.Sometimes it is more convenient to figuredecibels from voltage or current ratios or gainsrather than from power ratios. This appliesespecially to voltage amplifiers. The equationfor this isNctb = 20 log :: or 20 log "*where the subscript, o, denotes the output voltageor current and , the input voltage or current.Remember, this equation is true only ifthe volt:tge or current gain in question representsa power gain which is the square of itand not if the power gain which results fromthis is some other quantity due to impedancechanges. This should be quite clear when weconsider that a matching transformer to connecta speaker to a line or output tube doesnot represent a gain or loss; there is a voltagechange and a current change yet the power remainsthe same for the impedance has changed.On the other hand, when dealing with voltageamplifiers, we can figure the gain in astage by finding the voltage ratio from the gridof the first tube to the grid .of the next tube.+8Decibels osPower LevelThe original use of the decibelwas only as a ratio of powerlevels-not as an absolutemeasure of power. However, one may use thedecibel as such an absolute unit by fixing anarbitrary "zero" level, and to indicate anypower level by its number of decibels above orbelow this arbitrary zero level. This is all verygood so long as we agree on the zero level.Any power level may then be converted todecibels by the equation:Nctb = I 0 log _PPoret.where N .. b is the desired power level in decibels,Po the output of the amplifier, P,.,_ thearbitrary reference level.The zero level most frequently used (but notalways) is 6 milliwatts or 0.006 watts. For thiszero level, the equation reduces toN .. b =I 0 log 0 ~o 06Example: An amplifier using a 6F6 tubeshould be able to deliver an undistorted outputof 3 watts. How much is this in decibels?Po 3p,.,, = .006 = 50010 X log 500 = 10 X 2.10 = 21.0Therefore the power level at the output ofthe 6F6 is 27.0 decibels. When the power levelto be converted is less than 6 milliwatts, thelevel is noted as negative. Here we must rememberall that has been said regarding logarithmsof numbers less than unity and the factthat the characteristic is negative but not themantissa.A preamplifier for a microphone is feeding1. 5 milliwatts into the line going to the regularspeech amplifier. What is this power levelexpressed in decibels?decibels = 1 0 log o.:~ 6 =0.0015I 0 log0 _ 006= 10 log 0.25Log 0.25 = -1.398 (from table).Therefore,10 X -1.398 = (10 x -1 = -10)+ (10 X .398 = 3.98); adding the productsalgebraically, gives -6.02 db.The conversion chart reproduced in thischapter will be of use in converting decibels towatts and vice versa.
<strong>HANDBOOK</strong> Decibel-Power Conversion 763606 0505 0404 0302010.3 02 0I 000co-10-1 ocicoOt:f [-.;;rOQ0zz-20 -2 o..J..J-30w>w..J -40nm ...,,...-50 -5 0-60...
764 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>Converting Decibelsto PowerIt is often convenientto be able to convert adecibel value to a powerequivalent. The formula used for this operationisP = 0.006 X antilog ~:where P is the desired level in watts and N.bthe decibels to be converted.To determine tHe power level P from a decibelequivalent, simply divide the decibel valueby 10; then take the number comprising theantilog and multiply it by 0.006; the productgives the level in watts.Note: In problems dealing with the conversionof minu.r decibels to power, it often happensthat the decibel valu~ -:- N., is notdivisible by 10. When this IS the case,the numerator in the factor - !() N"" must b emade evenly divisible by 10, the negative signsmust be observed, and the quotient labeled accordingly.To make the numerator evenly divisible by10 proceed as follows: Assume, for example,that - N .. b is some such value as - 38; tomake this figure evenly divisible by 10, wemust add - 2 to it, and, since we have addeda negative 2 to it, we must also add a positive2 so as to keep the net result the same.Our decibel value now stands, - 40 + 2.Dividing both of these figures by 10, as in theequation above, we have - 4 and +0.2. _Puttingthe two together we have the loganthm-4.2 with the negative characteristic and thepositive mantissa as required.The following examples will show the techniqueto be followed in practical problems.(a) The output of a certain device is ratedat -74 db. What is the power equivalent?Solution:Routine:~~" = -.. ~ 4 (not evenly divisible by 1 0 l-74- 6 +6-80 +6N•" -80 + 610= 10-8.6antilog -8.6 = 0.000 000 04. 006 X 0.000 000 04 =0.000 000 000 24 watt or240 micro-micrawatt(b) This example differs somewhat fromthat of the foregoing one in that the mantissasare added differently. A low-powered amplifierhas an input signal levt:l of -17.3 db. Howmany milliwatts does this value represent?Solution:-17.3- 2.7-20H•b -20 + 2.710= 10+ 2.7+ 2.7-2.27Antilog -2.27 = 0.01860.006 X 0.0186 = 0.000 1116 watt or0.1116 milliwattInput voltages: To determine the requiredinput voltage, take the peak voltage necessaryto drive the last class A amplifier tube to maximumoutput, and divide this figure by the totaloverall voltage gain of the preceding stages.Computing Specifications: From the precedingexplanations the following data can becomputed with any degree of accuracy warrantedby the circumstances:( 1) Voltage amplification( 2) Over all gain in db(3) Output signal level in db( 4) Input signal level in db( 5) Input signal level in watts(6) Input signal voltageWhen a power level is available which mustbe brought up to a new power level, the gainrequired in the intervening amplifier is equalto the difference between the two levels in decibels.If the required input of an amplifier forfull output is -30 decibels and the outputfrom a device to be used is but -45 decibels,the pre-amplifier required should have a gainof the difference, or 15 decibels. Again this istrue only if the two amplifiers are properlymatched and no losses are introduced due tomismatching.Push-PullAmplifiersTo double the output of any cascadeamplifier, it is only necessaryto connect in push-pull thelast amplifying stage, and replace the interstageand output transformers with push-pulltypes.To determine the voltage gain (voltage ratio)of a push-pull amplifier, take the ratio ofone half of the secondary winding of the pushpulltransformer and multiply it by the p. ofone of the output tubes in the push-pull stage;the product, when doubled, will be the voltageamplification, or step-up .Other Units ondZero LevelsWhen working with decibelsone should not immediatelytake for grantedthat the zero level is 6 milliwatts for th@re areother zero levels in use.In broadcast stations an entirely new systemis now employed. Measurements made in
<strong>HANDBOOK</strong>acoustics are now made with the standard zerolevel of 10-'• watts per square em.Microphones are often rated with referenceto the following zero level: one volt at opencircuit when the sound pressure is one millibar.In any case, the rating of the microphone mustinclude the loudness of the sound. It is obviousthat this zero level does not lend itself readilyfor the calculation of required gain in an amplifier.The VU: So far, the decibel has always referredto a type of signal which can readily bemeasured, that is, a steady signal of a singlefrequency. But what would be the power levelof a signal which is constantly varying in volumeand frequency? The measurement of voltagewould depend on the type of instrumentemployed, whether it is measured with athermal square law meter or one that showsaverage values; also, the inertia of the movementwill change its indications at the peaksand valleys.After considerable consultation, the broadcastchains and the Bell System have agreedon the VU. The level in VU is the level indecibels above 1 milliwatt zero level and measuredwith a carefully defined type of instrumentacross a 600 ohm line. So long as wedeal with an unvarying sound, the level in VUis equal to decibels above 1 milliwatt; butwhen the sound level varies, the unit is theVU and the special meter must be used. Thereis then no equivalent in decibels.The Neper: We might have used the naturallogarithm instead of the common logarithmwhen defining our logarithmic unit of sound.This was done in Europe and the unit obtainedis known as the neper or napier. It is stillfound in some American literature on filters.1 neper = 8.686 decibels1 decibel = 0.1151 neperAC Meters WithDecibel ScalesMany test instrumentsare now equipped withscales calibrated in decibelswhich is very handy when making measurementsof frequency characteristics and gain.These meters are generally calibrated for connectionacross a 500 ohm line and for a zerolevel of 6 milliwatts. When they are connectedacross another impedance, the reading on themeter is no longer correct for the zero level of6 milliwatts. A correction factor should beapplied consisting in the addition or subtractionof a steady figure to all readings on themeter. This figure is given by the equation:500db to be added= 10 log Twhere Z is the impedance of the circuit undermeasurement.THIRDQUADRANTTrigonometry 765FOURTHQUADRANTFigure 10.<strong>THE</strong> CIRCLE IS DIVIDED INTOFOUR QUADRANTS BY TWO PERPENDICULAR LINES AT RIGHTANGLES TO EACH O<strong>THE</strong>R.The "northeast" quadrant thus formed isknown os the first quadrant; the others orenumbered consecutively in o counterclockwisedirection.Definitionand UseTrigonometryTrigonometry is the science ofmensuration of triangles. At firstglance triangles may seem tohave little to do with electrical phenomena;however, in a.c. work most currents ,and voltagesfollow laws equivalent to those of thevarious trigonometric relations which we areabout to examine briefly. Examples of theirapplication to a.c. work will be given in thesection on Vectors.Angles are measured in degrees or in radians.The circle has been divided into 360degrees, each degree into 60 minutes, and eachminute into 60 seconds. A decimal division ofthe degree is also in use because it makes calculationeasier. Degrees, minutes and secondsare indicated by the following signs: 0 ,' and".Example: 6° 5' 23" means six degrees, fiveminutes, twenty-three seconds. In the decimalnotation we simply write 8.4 7 o, eight andforty-seven hundredths of a degree.When a circle is divided into four quadrantsby two perpendicular lines passing throughthe center (Figure 10) the angle made by thetwo lines is 90 degrees, known as a right angle.Two right angles, or 180° equals a straightangle.The radian: If we take the radius of a circleand bend it so it can cover a part of the circumference,the arc it covers subtends an anglecalled a radian (Figure 11). Since the circumferenceof a circle equals 2 times the radius,there are 271' radians in 360°. So we have thefollowing relations:1 radian=57° 17' 45''=57.2958° 71'=3.141591 degree= 0.017 4 5 radians71' radians= 180°. 71' /2 radians= 90°'TI' /3 radians= 60°
766 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>In the angle A, Figure 13A, a line is drawnfrom P, perpendicular to b. Regardless of thepoint selected for P, the ratio ajc will alwaysbe the same for any given angle, A. So will allthe other proportions between a, b, and c remainconstant regardless of the position ofpoint P on c. The six possible ratios each arenamed and defined as follows:Figure 11.<strong>THE</strong> RADIAN.A radian is an angle whose arc is exactly equalto the length of either side. Note that theangle Is constant regardless of the length ofthe side and the arc so long as they are equal.A radian equals 57.2958".In trigonometry we consider an angle generatedby two lines, one stationary and theother rotating as if it were hinged at 0, Figure12. Angles can be greater than 180 degrees andeven greater than 360 degrees as illustrated inthis figure.Two angles are complements of each otherwhen their sum ·is 90°, or a right angle. A isthe complement of B and B is the complementof A whenA=(90°-B)and whenB=(90o-A)Two angles are supplements of each otherwhen their sum is equal to· a straight angle, or180°. A is the supplement of B and B is thesupplement of A whenA=(180°-B)andB=(180° -A)sine A= ~catangent A= bcsecant A= b. A bcosme = cbcotangent A =acosecant A = ~aLet us take a special angle as an example.For instance, let the angle A be 60 degrees asin Figure 13B. Then the relations between thesides are as in the figure and the six functionsbecome:sin. 60° = acb ¥2cos 60° =c- = T = 1;2.tan 60° = a = ¥2 V 3 = V3b ¥2¥2 1cot 60° = ¥2 V3 = V3 = % V3sec 60° = ~b1% =2esc 60°c 1a = ¥2 V3 = 2!J V3Another example: Let the angle be 45°, thenthe relations between the lengths of a, b, andc are as shown in Figure 13C, and the sixfunctions are:A?-A~ A~~~~-----------x0Figure 12.AN ANGLE IS GENERATED BY TWO LINES, ONE STATIONARY AND <strong>THE</strong> O<strong>THE</strong>RROTATING.The line OX is stationary; the line with the small arrow at the far end rotates in a counterclockwisedirection. At the position Illustrated in the lelthandmost section of the drawing It makes an angle,A, which is less than 90 o and is therefore In the first quadrant. In the position shown in the secondportion of the drawing the angle A ltas increased to such a value that It now lies In the thirdquadrant; note that an angle can be greater than 180". In the third illustration the angle A is Inthe fourth quadrant. In the fourth position the rotating vector has made more than one completerevolution and is hence in the fifth quadrant; since the fifth quadrant is an exact repetition of thefirst quadrant, its values will be the same as in the lefthandmost portion of the illustration.
<strong>HANDBOOK</strong> Trigonometric Relations 767a-'hVl90"A A A® ©b-0®Figure 13.<strong>THE</strong> TRIGONOMETRIC FUNCTIONS.In the right triangle shown in (A) the side opposite the ongle A is a, while the adjoining sides areb and c; the trigonometric functions of the angle A are completely defined by the ratios of thesides a, b and c. In (8) are shown the lengths of the sides a and b when angle A is 60' and side cis I. In (C) angle A is 45'; a and b equal I, while c equals yf2. In (D) note that c equals a far aright angle while b equals 0.a= Ipc=asin 45°tan 45°sec 45°1Vz=%Vz1cos 45° = Vz = %Vz1 11 =1 cot 45o T = 1VzV2Vzcosec 4 5o = - 1- = VzThere are some special difficulties when theangle is zero or 90 degrees. In Figure 13D anangle of 90 degrees is shown; drawing a lineperpendicular to b from point P makes it fallon top of c. Therefore in this case a = c andb = 0. The six ratios are now:b 0sin90° = ~= 1ca atan 90° = b= 0 = 0oocos 90°. =c- =c-= 0cot 90° = -= 0asec90° =~=~= oo cosec90° =_5_= 1b 0 , aWhen the angle is zero, a= 0 and b =c. Thevalues are then:bsin0° '= ~=~= 0 cos 0° = -= 1c cca 0b btan 0° = b = b = 0 cot 0° =- = -= ooa 0cc csec 0° =b= 1 cosec0° =-=-= 00a 0In general, for every angle, there will be definitevalues of the six functions. Conversely,when any of the six functions is known, theangle is defined. Tables have been calculatedgiving the value of the functions for angles.From the foregoing we can make up a smalltable of our own (Figure 14), giving values ofthe functions for some common angles.Relations BetweenFunctions0 1smA = ---It follows from the definitionsthat1cos A=--cosec Asec A1and tan A =cot AFrom the definitions also follows the relationcos A= sin (complement of A) =sin (90°-A)because in the right triangle of Figure 15,cos A=b/c=sinB and B=90°-A or thecomplement of A. For the same reason:Relations inRight Trianglescot A = tan (90°-A)esc A = sec (90°-A)In the right triangle ofFigure 15, sin A=a/c andby transpositiona= c sin AFor the same reason we have the followingidentities:tan A= a/bcot A= b/aa= b tan Ab =a cot AIn the same triangle we can do the same forfunctions of the angle BAngle Sin Cos. Tan Cot Sec. Cosec.0 0 1 0 00 130° % %v3%v3' V3 2!Jv3' 245° %Vz%Vz 1 1 v'2 V260° %v3' '12 V3 l!3V3 2 2!Jv3'90° 1 0 00 0 00 1Figure 14.Values of trigonometric functions for commonangles in the first quadrant.
768 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>POSITIVE FUNCTIONSSECOND OUADRANT ~IRST OUADRANTsine, cosecall functionsFigure 15.In this figure the sides a, b, and c are usedto define the trigonometric functions of angle8 as well as angle A.sin B = b/ccos B = a/ctan B = b/acot B = a/bFunctions of AnglesGreater than90 Degreesb = c sin Ba= c cos Bb =a tan Ba= b cot BIn angles greater than90 degrees, the valuesof a and b become negativeon occasion in accordancewith the rules of Cartesian coordinates.When b is measured from 0 towardsthe left it is considered negative and similarly,when a is measured from 0 downwards, it isnegative. Referring to Figure 16, an angle inthe second quadrant (between 90° and 180°)has some of its functions negative:sm. A= -ca= pos.atan A = _ b = neg.csec A = _ b = neg.-bcos A=-= neg.c-bcot A = -----;;- = neg.ccosec A = a = pos.For an angle in the third quadrant (180° to270° ), the functions are-asin A = -c- = neg.-atan A = _ b = pos.-bcos A =-c- = neg.-bcot A=-=-apos.tan, cotTHIRD QUADRANTcosine, secantFOURTH QUADRANTFigure 17.SIGNS OF <strong>THE</strong> TRIGONOMETRICFUNCTIONS.The functions listed in this diagram are positive;all other functions are negative.csec A= ·-b =neg.ccosec A =- = neg.-aAnd in the fourth quadrant (270° to 360°):. -asmA =-=neg.c-atan A =b =neg.csec A= b = pos.bcos A = -c- = pos.bcot A = - = neg.-accosec A =- = neg.-aSummarizing, the sign of the functions ineach quadrant can be seen at a glance fromFigure 17, where in each quadrant are writtenthe names of functions which are positive;those not mentioned are negative.THIRDQUADRANTFOURTHQUADRANTFigure 16.TRIGONOMETRIC FUNCTIONS IN <strong>THE</strong> SECOND, THIRD, AND FOURTH QUADRANTS.The trigonometric functions in these quadrants are similar to first quadrant values, bwt thesigns of the fwnctions vary crs listed In the text crnd in Flgwre 17.
<strong>HANDBOOK</strong>Figure 18.SINE AND COSINE CURVES.In (A) we have a sine curvedrawn in Cartesian coordinates.This is the usual representationof an alternating current wavewithout substantial harmonics. In(8) we have a cosine wove;note that it is exactly similarto a sine wove displaced by90° or tt/2 radians.(AI 0' 90'" r-- ~-..V' -·r-,:"...,~·~·~· _,.o-'-~:,.Trigonometric Curves 769180' 270' 360' 450 540" 630' 720'1- t-12: r--.,...,1:1'~-JLr-- - -r- ,_ 1-!--t- ,_ l"s r-!Z t:-::7T .'!1r 2Tr 57>" _,n- 'C!!.,,.""'t·-r-(B)Graphs af TrigonametricFunctionsThe sine tl'ave. Whenwe have the relationy=sin x, where xis anangle measured in radians or degrees, we candraw a curve of y versus x for all values ofthe independent variable, and thus get a goodconception how the sine varies with the magnitudeof the angle. This has been done inFigure l8A. We can learn from this curve thefollowing facts.1. The sine varies between + 1 and -12. It is a periodic curve, repeating itself afterevery multiple of 2'1T or 360°3. Sin x =sin (180°-x) or sin ('TT -x)4. Sin x = -sin (180° + x), or-sin ('TT + x)The cosine ll'ave. Making a curve for thefunction y = cos x, we obtain a curve similarto that for y = sin x except that it is displacedby 90° or "'/2 radians with respect to theY-axis. This curve (Figure 18B) is also periodicbut it does not start with zero. We readfrom the curve:1. The value of the cosine never goes beyond+1 or -12. The curve repeats, after every multipleof 2'7i radians or 360°3. Cos x = -cos (180° -x) or-cos ('
770 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>Figure 21.ANO<strong>THE</strong>R REPRESENTATION OFTRIGONOMETRIC FUNCTIONS.If the radius of a circle is considered as theunit of measurement, then the lengths of thevarious lines shown in this diagram are numericallyequal to the functions marked adjacentto them.The graphs of the secant and cosecant areof lesser importance and will not be shownhere. They are the inverse, respectively, of thecosine and the sine, and therefore they varyfrom + 1 to infinity and from -1 to -infinity.Perhaps another useful way of visualizingthe values of the functions is by consideringFigure 21. If the radius of the circle is the unitof measurement then the lengths of the linesare equal to the functions marked on them.Trigonometric TablesThere are two kinds oftrigonometric tab 1 e s.The first type gives the functions of the angles,the second the logarithms· of the functions.The first kind is also known as the table ofnatural trigonometric functions.These tables give the functions of all anglesbetween 0 and 45°. This is all that is necessaryfor the function of an angle between 45° and90° can always be written as the co-functionof an angle below 4 5°. Example: If we had tofind the sine of 48°, we might writesin 48° = cos (90° -48°) = cos 42°Tables of the logarithms of trigonometricfunctions give the common logarithms (log10)of these functions. Since many of these logarithmshave negative characteristics, one shouldadd -10 to all logarithms in the table whichhave a characteristic of 6 or higher. For instance,the log sin 24° = 9.60931 -10. Logtan 1 o = 8.24192 -10 but log cot 1° =1.75808. When the characteristic shown is lessthan 6, it is supposed to be positive and oneshould not add -10.VectorsA scalar quantity has magnitude only; a11ector quantity has both magnitude and direction.When we speak of a speed of 50 milesper hour, we are using a scalar quantity, butwhen we say the wind is Northeast and has aFigure 22.Vectors may be added as shown in thesesketches. In each case the long vector representsthe vector sum of the smaller vectors.For many engineering applications sufficientaccuracy can be obtained by this methodwhich avoids long and laborious calculations.velocity of 50 miles per hour, we speak of avector quantity.Vectors, representing forces, speeds, displacements,etc., are represented by arrows.They can be added graphically by well knownmethods illustrated in Figure 22. We can makethe parallelogram of forces or we can simplydraw a triangle. The addition of many vectorscan be accomplished graphically as in the samefigure.In order that we may define vectors algebraicallyand add, subtract, multiply, or dividethem, we must have a logical notation systemthat lends itself to these operations. For thispurpose vectors can be defined by coordinatesystems. Both the Cartesian and the polar coordinatesare in use.Vectors Definedby CartesianCoordinatesSince we have seen how thesum of two vectors is obtained,it follows from Figure23, that the vector Zequals the sum of the two vectors ; and j. Infact, any vector can be resolved into vectorsalong the X- and Y-axis. For convenience inworking with these quantities we need to dis-+iBFigure 23.RESOLUTION OF VECTORS.Any vector such as .i may be resolved Infotwo vectors, x and y, along the X- and Yaxes. If vectors are to be added, their respectivex and y components may be added tolind the x and y components of the resultantvector.+
<strong>HANDBOOK</strong>Vectors 771--J-;..,.. __ -t__II.~1 ---• o Y,"'-----~=::.....;:.__ ___ -i.:------7. -----x.-----i:'--X~Figure 24.ADDITION OR SUBTRACTION OFVECTORS.Vectors may be added or subtracted byadding or subtracting their x or y componentsseparately.tinguish between the X- and ¥-component,and so it has been agreed that the ¥-componentalone shall be marked with the letter j.Example (Figure 23):i: = 3 + 4jNote again that the sign of componentsalong the X-axis is positive when measuredfrom 0 to the right and negative when measuredfrom 0 towards the left. Also, the componentalong the Y-axis is positive when measuredfrom 0 upwards, and negative whenmeasured from 0 downwards. So the vector,R, is described asR. = 5- 3iVector quantities are usually indicated bysome special typography, especially by using apoint over the letter indicating the vector, asR..Absolute Valueaf a VectorThe absolute or scalarvalue of vectors such as Zor R in Figure 23 is easilyfound by the theorem of Pythagoras, whichstates that in any right-angled triangle thesquare of the side opposite the right angle isequal to the sum of the squares of the sidesadjoining the right angle. In Figure 23, OABis a right-angled triangle; therefore, the squareof OB (or Z) is equal to the square of OA(or x) plus the square of AB (or y). Thus theabsolute values of Z and R may be determinedas follows:IZI= R+"7I z I = v 3' + 4' = 5I R I = v 5' + 3' = v'T4 = 5.83The vertical lines indicate that the absoluteor scalar value is meant without regard to signor direction.Addition of Vectorsthe two vectorsAn examination of Figure24 will show thatR = Xl + jy,i; = x, + j y,can be added, if we add the X-componentsand the ¥-components separately.R. + i: = x, + x. + i (y, + y,>For the same reason we can carry out subtractionby subtracting the horizontal componentsand subtracting the vertical componentsR. - i: = x, - x, + i (y, - y,>Let us consider the operator j. If we have avector a along the X-axis and add a j in frontof it (multiplying by j) the result is that thedirection of the vector is rotated forward 90degrees. If we do this twice (multiplying byl) the vector is rotated forward by 180 degreesand now has the value -a. Therefore multiplyingby l is equivalent to multiplying by -1.Theni' = - 1 and j = vf=1This is the imaginary number discussed beforeunder algebra. In electrical engineeringthe letter j is used rather than i, because i isalready known as the symbol for current.Multiplying Vectors When two vectors areto be multiplied we canperform the operation just as in algebra, rememberingthat j' = -1.RZ = (x, + jy,) (x, + jy,)= x, x, + jx, y, + j x, y, + i' y, Y•= x, x, - y, y, + j ( x, y, + x, y,)Division has to be carried out so as to removethe j-term from the denominator. Thiscan be done by multiplying both denominatorand numerator by a quantity which will eliminatej from the denominator. Example:R x, jy, (x, jy,) (x, - jy,)z= x, + jy, = (x, + jy,) (x,- jy,)_ x,x, + y,y, + j ( x,y, -- x," + y,"x,y,)Polar Coordinates A vector can also be definedin polar coordinatesby its magnitude and its vectorial anglewith an arbitrary reference axis. In Figure 25
772 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>zbzZ=a+jbTAN A=*JZI= Va'+b'Figure 26.Vectors can be transformed from Cartesianinto polar notation as shown in this figure.Figure 25.IN THIS FIGURE A VECTOR HASBEEN REPRESENTED IN POLARINSTEAD OF CARTESIAN CO-ORDINATES.In polar coordinates a vector Is defined bya magnitude and an angle, called the vectorialangle, instead ol by two magnitudesas In Cartesian coordinates.the vector Z has a magnitude 50 and a vector·ial angle of 60 degrees. This will then bewrittenz = 50L60°A vector a + jb can be transformed intopolar notation very simply (see Figure 26)Z = a + jb = Y a 2 + b 2 Ltan-' ~In this connection tan·• means the angle ofwhich the tangent is. Sometimes the notationarc tan b I a is used. Both have the same meaning.A polar notation of a vector can be transformedinto a Cartesian coordinate notation inthe following manner (Figure 27)Z = pLA = p cas A + jp sin A.1\. sinusoidally alternating voltage or currentis symbolically represented by a rotatingvector, having a magnitude equal to the peakvoltage or current and rotating with an angularvelocity of 277"f radians per second or as manyrevolutions per second as there are cycles persecond.The instantaneous voltage, e, is always equalto the sine of the vectorial angle of this rotatingvector, multiplied by its magnitude.Xe = E sin 27TftThe alternating voltage therefore varies withtime as the sine varies with the angle. If weplot time horizontally and instantaneous voltagevertically we will get a curve like thosein Figure 18.In alternating current circuits, the currentwhich flows due to the alternating voltage isnot necessarily in step with it. The rotatingcurrent vector may be ahead or behind thevoltage vector, having a phase difference withit. For convenience we draw these vectors asif they were standing still, so that we can indicatethe difference in phase or the phase angle.In Figure 28 the current lags behind the voltageby the angle 8, or we might say that thevoltage leads the current by the angle 8.Vector diagram.r show the phase relationsbetween two or more vectors (voltages andcurrents) in a circuit. They rna y be added andsubtracted as described; one may add a voltagevector to another voltage vector or a currentvector to a current vector but not a currentvector to a voltage vector (for the same reasonthat one cannot add a force to a speed). Figure28 illustrates the relations in the simple seriescircuit of a coil and resistor. We know that thecurrent passing through coil and resistor mustbe the same and in the same phase, so we drawthis current I along the X-axis. We know alsothat the voltage drop IR across the resistor isin phase with the current, so the vector IR representingthe voltage drop is also along theX-axis.The voltage across the coil is 90 degreesahead of the current through it; IX musttherefore be drawn along the Y-axis. E theapplied voltage must be equal to the vectorialsum of the two voltage drops, IR and /X, andwe have so constructed it in the drawing. Nowexpressing the same in algebraic notation, wehaveDividing by IE = IR + jiXIZ = IR + jiXZ=R+iXDue to the fact that a reactance rotates thevoltage vector ahead or behind the currentvector by 90 degrees, we must mark it with aj in vector notation. Inductive reactance willhave a plus sign because it shifts the voltagevector forwards; a capacitive reactance is neg-
<strong>HANDBOOK</strong>·--------------I: pSIN AFigure 27.Vectors can be transformed from polar intoCartesian notation as shown in this figure.ative because the voltage will lag behind thecurrent. Therefore:XL= + j 2
774 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>y87I FI~ST I1---1-- otE..f~~~T- 1--- b 1-1-t-QUADRANT--5 ,_ '--Q4328-7 -b-5-4-3 -2 I 01 2 3 4 5II-2R-3-4s1---5THIRD- -bFOURTH- 1--QUA~RA\"f-QUADRANT 1-- 1---7-8 yFigure 30.CARTESIAN COORDINATES.The location ol any point can be delined byits distance #rom the X and Y axes.pb-X7 8P 2, 3 or we might say x = 2 and y = 3. Themeasurement x is called the abscissa of thepoint and the distance y is called its ordinate.It is arbitrarily agreed that distances measuredfrom 0 to the right along the X-axis shall bereckoned positive and to the left negative. Distancesmeasured along the Y-axis are positivewhen measured upwards from 0 and negativewhen measured downwards from 0. This isillustrated in Figure 30. The two axes dividethe plane area into four parts called quadrants.These four quadrants are numbered as shownin the figure.It follows from the foregoing statements,that points lying within the first quadrant haveboth x and y positive, as is the case with thepoint P. A point in the second quadrant has anegative abscissa, x, and a positive ordinate, y.This is illustrated by the point Q, which hasthe coordinates x = -4 and y = + 1. Pointsin the third quadrant have both x and y negative.x = - 5 and y = -2 illustrates such apoint, R. The point S, in the fourth quadranthas a negative ordinate, y and a positive abscissaor x.In practical applications we might drawonly as much of this plane as needed to illustrateour equation and therefore, the scalesalong the X-axis and Y-axis might not startwith zero and may show only that part of thescale which interests us.Representation ofFunctionsIn the equation:300,000f=--A-£c200018001600140012001000800600 0 100 200 300 400A-· ..Figure 31.REPRESENTATION OF A SIMPLEFUNCTION IN CARTESIAN COORDINATES.300,000In this chart of the #unction fk, = Am.,.,.,distances along the X axis represent wavelengthin meters, while those along the Yaxis represent frequency in kilocycles. A curvesuch as this helps to lind values betweenthose calculated with su,icient accuracy formost purposes.f is said to be a function of A. For every valueof f there is a definite value of A. A variable issaid to be a function of another variable whenfor every possible value of the latter, or independentvariable, there is a definite value ofthe first or dependent variable. For instance,if y = 5x', y is a function of x and x is calledthe independent variable. When a = 3b' + 5b'- 25b + 6 then a is a function of b.A function can be illustrated in our coordinatesystem as follows. Let us take the equationfor frequency versus wavelength as anexample. Given different values to the independentvariable find the corresponding valuesof the dependent variable. Then plot the pointsrepresented by the different sets of two values.f.,.600800100012001400160018002000/..lnt!ter.o;;500375300250214187167150Plotting these points in Figure 31 and drawinga smooth curve through them gives us thecurve or graph of the equation. This curvewill help us find values of f for other valuesof A (those in between the points calculated)and so a curve of an often-used equation mayserve better than a table which always hasgaps.When using the coordinate system describedso far and when measuring linearly along bothaxes, there are some definite rules regarding500
<strong>HANDBOOK</strong>Representation of Functions775Figure 32.Only two points are needed to define functionswhich result in a straight line as shownin this diagram representing Ohm's Law.the kind of curve we get for any type ofequation. In fact, an expert can draw the curvewith but a very few plotted points since theequation has told him what kind of curve toexpect.First, when the equation can be reduced tothe form y = mx + b, where x and y are thevariables, it is known as a linear or first degreefunction and the curve becomes a straight line.(Mathematicians still speak of a "curve" whenit has become a straight line.)When the equation is of the second degree,that is, when it contains terms like ,?- or y'or xy, the graph belongs to a group of curves,called conic sections. These include the circle,the ellipse, the parabola and the hyperbola. Inthe example given above, our equation is ofthe formxy = c, c being equal to 300,000which is a second degree equation and in thiscase, the graph is a hyperbola.This type of curve does not lend itself read·ily for the purpose of calculation except nearthe middle, because at the ends a very largechange in \ represents a small change in f andvice versa. Before discussing what can be doneabout this let us look at some other types ofcurves.Suppose we have a resistance of 2 ohmsand we plot the function represented by Ohm'sLaw: E = 21. Measuring E along the X-axisand amperes along the Y-axis, we plot thenecessary points. Since this is a first degreeequation, of the form y = mx + b (for E =y, m = 2 and I= x and b = 0) it will be astraight line so we need only two points toplot it.(line passes through origin)E0 05 10The line is shown in Figure 32. It is seen tobe a straight line passing through the origin.Figure 33.A TYPICAL GRID- VOLTAGEPLATE-CURRENT CHARACTER-ISTIC CURVE.The equation represented by such a curve Isso complicated that we do not use it. Datafor such a curve Is obtained experimentally,and intermediate values can be found withsufficient accuracy from the curve.If the resistance were 4 ohms, we should getthe equation E = 41 and this also representsa line which we can plot in the same figure.As we see, this line also passes through theorigin but has a different slope. In this illustrationthe slope defines the resistance and wecould make a protractor which would convertthe angle into ohms. This fact may seem inconsequentialnow, but use of this is made in thedrawing of loadlines on tube curves.Figure 33 shows a typical, grid-voltage,plate-current static characteristic of a triode.The equation represented by this curve israther complicated so that we prefer to dealwith the curve. Note that this curve extendsthrough the first and second quadrant.Families of curves. It has been explainedthat curves in a plane can be made to illustratethe relation between two variables when oneof them varies independently. However, whatare we going to do when there are three variablesand two of them vary independently. Itis possible to use three dimensions and threeaxes but this is not conveniently done. Insteadof this we may use a family of curves. Wehave already illustrated this partly with Ohm'sLaw. If we wish to make a chart which willshow the current through any resistance withany voltage applied across it, we must take the:equation E = IR, having three variables.We can now draw one line representing aresistance of 1 ohm, another line representing2 ohms, another representing 3 ohms, etc., oras many as we wish and the size of our paperwill allow. The whole set of lines is thenapplicable to any case of Ohm's Law fallingwithin the range of the chart. If any two ofthe three quantities are given, the third can befound.
776 Radio Mathematics and Calculations <strong>THE</strong> <strong>RADIO</strong>or.~-r.-ro~-r.-ro-o-ro-ro-o-r~/~.I 1 1/ 1 1/1/AVERAGE PLATE CHARACTERISTICSEt=6.3 v.~ ~t-~~~~-+-+-+-r~~,_,_+-t-t-~~- 1~1~\~1/+r~~~/~v~~+-~~~~\. / / IFigure 34.A FAMILY OF CURVES.An equation such as Ohm's Law has threevariables, but can be represented in Cartesiancoordinates by a family of curves such asshown here. If any two quantities are given,the third can be found. Any point In thechart represents a definite .value each of E,I, and R, which will sat/sly the equation ofOhm's Law. Values of R not situated oa an Rline can be found by Interpolation.Figure 34 shows such a family of curves tosolve Ohm's Law. Any point in the chart representsa definite value each of E, I, and Rwhich will satisfy the equation. The value ofR represented by a point that is not situatedon an R line can be found by interpolation.It is even possible to draw on the same charta second family of curves, representing afourth variable. But this is not always possible,for among the four variables-there should beno more than two independent variables. Inour example such a set of lines could representpower in watts; we have drawn only two ofthese but there could of course be as many asdesired. A single point in the plane now indicatesthe four values of E, I, R, and P whichbelong together and the knowledge of anytwo of them will give us the other two byreference to the chart.Another example of a family of curves isthe dynamic transfer characteristic or platefamily of a tube. Such a chart consists of severalcurves showing the relation between platevoltage, plate current, and grid bias of a tube.Since we have again three variables, we mustshow several curves, each curve for a fixedvalue of one of the variables. It is customaryto plot plate voltage along the X-axis, platecurrent along the Y-axis, and to make differentcurves for various values of .~rrid bias. Such aFigure 35."PLATE" CURVES FOR ATYPICAL VACUUM TUBE.In such curves we have three variables, platevoltage, plate current, and grid bias. Eachpoint on a grid bias line corresponds to theplate voltage and plate current representedby its position with respect to the X and Yaxes. Those lor other values of grid bias maybe found by interpolation. The /oadline shownIn the lower left portion of the chart is explainedIn the text.set of curves is illustrated in Figure 35. Eachpoint in the plane is defined by three values,which belong together, plate voltage, platecurrent, and grid voltage.Now consider the diagram of a resistancecoupledamplifier in Figure 36. Starting withthe B-supply voltage, we know that whateverplate current flows must pass through theresistor and will conform to Ohm's Law. Thevoltage drop across the resistor is subtractedfrom the plate supply voltage and the remainderis the actual voltage at the plate, the kindthat is plotted along the X-axis in Figure 35.We can now plot on the plate family of theFigure 36.PARTIAL DIAGRAM OF A RESISTANCE COUPLED AMPLIFIER.The portion of the supply voltage wastedacross the 50,000-ohm resistor Is representedIn figure 35 as the /oad/lne.
<strong>HANDBOOK</strong>tube the loadline, that is the line showingwhich part of the plate supply voltage is acrossthe resistor and which part across the tube forany value of plate current. In our example, letus suppose the plate resistor is 50,000 ohms.Then, if the plate current were zero, the voltagedrop across the resistor would be zero andthe full plate supply voltage is across the tube.Our first point of the loadline is E = 250,I = 0. Next, suppose, the plate current were1 rna., then the voltage drop across the resistorwould be SO volts, which would leave for thetube 200 volts. The second point of the loadlineis then E = 200, I = 1. We can continuelike this but it is unnecessary for we shall findthat it is a straight line and two points aresufficient to determine it.This loadline shows at a glance what happenswhen the grid-bias is changed. Althoughthere are many possible combinations of platevoltage, plate current, and grid bias, we arenow restricted to points along this line as longas the 50,000 ohm plate resistor is in use. Thisline therefore shows the voltage drop acrossthe tube as well as the voltage drop across theload for every value of grid bias. Therefore, ifwe know how much the grid bias varies, wecan calculate the amount of variation in theplate voltage and plate current, the amplification,the power output, and the distortion.Logarithmic Scales Sometimes it is convenientto measure alongthe axes the logarithms of our variable quantities.Instead of actually calculating the logarithm,special paper is available with logarithmicscales, that is, the distances measuredalong the axes are proportional to the logarithmsof the numbers marked on them ratherthan to the numbers themselves.There is semi-logarithmic paper, havinglogarithmic scales along one axis only, theother scale being linear. We also have fulllogarithmic paper where both axes carry logarithmicscales. Many curves are greatly simplifiedand some become straight lines whenplotted on this paper.As an example let us take the wavelengthfrequencyrelation, charted before on straightcross-section paper.f = 300,000ATaking logarithms:log f = log 300,000 -log AIf we plot log f along the Y-axis and log Aalong the X-axis, the curve becomes a straightline. Figure 3 7 illustrates this graph on fulllogarithmic paper. The graph may be readwith the same accuracy at any point in con-3000lSOO2000 ""'"' 1500""" ""1000"900800700bOO500-400300200Logarithmic Scales 711I 100r ~........ WAVELENGTH IN METERSFigure 37.A LOGARITHMIC CURVE •0 0 0 0.Many functions become greatly simplified andsome become straight lines when plotted tologarithmic scales such as shown in thisdiagram. Here the frequency versus wavelengthcurve of Figure 3 I has been replotted to conformwith logarithmic axes. Note that it isonly necessary to calculate two points inorder to determine the 11 t:urve" since this typeof function results in a straight line.trast to the graph made with linear coordinates.This last fact is a great advantage of logarithmicscales in general. It should be clear thatif we have a linear scale with 100 small divi·sions numbered from 1 to 100, and if we areable to read to one tenth of a division, thepossible error we can make near 100, way upthe scale, is only 1/10th of a percent. But nearthe beginning of the scale, near 1, one tenth ofa division amounts to 10 percent of 1 and weare making a 10 percent error.In any logarithmic scale, our possible errorin measurement or reading might be, say 1/32of an inch which represents a fixed amount ofthe log depending on the scale used. The netresult of adding to the logarithm a fixed quantity,as 0.01, is that the anti-logarithm is multipliedby 1.025, or the error is 21( 2 %. No matterat what part of the scale the 0.01 is added,the error is always 2V 2 %.An example of the advantage due to the use
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Figure 40.REACTANCE-FREQUENCY CHART FOR AUDIO FREQUENCIES~~ Y\V)V )\~ ~ !I\0(X ~ "'- )
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<strong>HANDBOOK</strong>Polar Coordinates781bFigure 42.<strong>THE</strong> SIMPLEST FORM OF NOMOGRAM.plest form, it is somewhat like the lines in Figure42. If the lines a, b, and c are parallel andequidistant, we know from ordinary geometry,that b = % (a + c). Therefore, if we draw ascale of the same units on all three lines, startingwith zero at the bottom, we know that bylaying a straight-edge across the chart at anyplace, it will connect values of a, b, and c,which satisfy the above equation. When anytwo quantities are known, the third can befound.If, in the same configuration we used logarithmicscales instead of lirrear scales, the relationof the quantities would becomelog b = lj2 (log a + log c) orb = vacBy using different kinds of scales, differentunits, and different spacings between the scales,charts can be made to solve many kinds ofequations.If there are more than three variables it isgenerally necessary to make a double chart,that is, to make the result from the first chartserve as the given quantity of the second one.Such an example is the chart for the design ofcoils illustrated in Figure 45. This nomogramis used to convert the inductance in microhenriesto physical dimensions of the coil and viceversa. A pin and a straight edge are required.The method is shown under "R. F. Tank Cir·cuit Calculations" later in this chapter.Polar Coordinates Instead of the Cartesiancoordinate system thereis also another system for defining algebraical·ly the location of a point or line in a plane. Inthis, the polar coordinate system, a point is de·termined by its distance from the origin, 0,and by the angle it makes with the axis 0-X.In Figure 43 the point P is defined by thelength of OP, known as the radius vector andc0///RADIUS /VECTOR////////\"'O ORIGINAXISFigure 43.<strong>THE</strong> LOCATION OF A POINT BYPOLAR COORDINATES.In the polar eoordinote system any point isdetermined by its distonee from the originand the angle formed by a line drawn fromit to the origin and the 0-X axis.by the angle A the vectorial angle. We givethese data in the following formp = 3 /j0°Polar coordinates are used in radio chieflyfor the plotting of directional properties of microphonesand antennas. A typical example ofsuch a directional characteristic is shown inFigure 44. The radiation of the antenna representedhere is proportional to the distance ofthe characteristic from the origin for everypossible direction.sFigure 44.<strong>THE</strong> RADIATION CURVE OF ANANTENNA.Polar eoordlnates are used prlneipal/y In radiowork for plotting the directional characteris·tics of an antenna where the radiation Isrepresented by the distance of the eurve fromthe origin for every possible direction./pX
782 Radio Mathematics and CalculationsReactance CalculationsIn audio frequency calculations, an accuracyto better than a few per cent is seldom required,and when dealing with calculations involvinginductance, capacitance, resonant frequency,etc., it is much simpler to make use ofreactance-frequency charts such as those infigures 40 and 41 rather than to wrestle with acombination of unwieldy formulas. From thesecharts it is possible to determine the reactance ofa condenser or coil if the capacitance or inductanceis known, and vice versa. It follows fromthis that resonance calculations can be madedirect! y irom the chart, because resonancesimply means that the inductive and capacitivereactances are equal. The capacity required toresonate with a given inductance, or the inductancerequired to resonate with a given capacity,can be taken directly from the chart.While the chart may look somewhat formidableto one not familiar with charts of thistype, its application is really quite simple, andcan be learned in a short while. The followingexample should clarify its interpretation.For instance, following the lines to their intersection,we see that 0.1 hy. and 0.1 .ufd. intersectat approximately 1, 500 cycles and 1,000ohms. Thus, the reactance of either the coil orcondenser taken alone is about 1000 ohms, andthe resonan~ frequency about 1, 500 cycles.To find the reactance of 0.1 hy. at, say,10,000 cycles, simply follow the inductanceline diagonally up towards the upper left till itintersects the horizontal 10,000 kc. line. Followingvertically downward from the point ofintersection, we see that the reactance at thisfrequency is about 6000 ohms.To facilitat~ use of the chart and to avoiderrors, simply keep the following in mind: Thevertical lines indicate reactance in ohms, thehorizontal lines always indicate the frequency,the diagonal lines sloping to the lower rightrepresent inductance, and the diagonal linessloping toward the lower left indicate capacitance.Also remember that the scale is logarithmic.For instance, the next horizontal lineabove 1000 cycles is 2000 cycles. Note thatthere are 9, not 10, divisions between the heavylines. This also should be kept in mind wheninterpolating between lines when best possibleaccuracy is desired; halfway between the linerepresenting 200 cycles and the line representing300 cycles is not 250 cycles, but approximately230 cycles. The 250 cycle point is approximately0. 7 of the way between the 200cycle line and the 300 cycle line, rather thanhalfway between.Use of the chart need not be limited by thephysical boundaries of the chart. For instance,the 10-.u.ufd. line can be extended to find whereit intersects the 100-hy. line, the resonant frequencybeing determined by projecting the intersectionhorizontally back on to the chart.To determine the reactance, the logarithmicohms scale must be extended.R.F.TankCircuitCalculationsWhen winding coils for use inradio receivers and transmitters,it is desirable to be able todetermine in advance the fullcoil specifications for a given frequency. Likewise,it often is desired to determine how muchcapacity is required to resonate a given coil sothat a suitable condenser can be used.Fortunately, extreme accuracy is not required,except where fixed capacitors are usedacross the tank coil with no provision for trimmingthe tank to resonance. Thus, even thoughit may be necessary to estimate the stray circuitcapacity present in shunt with the tankcapacity, and to take for granted the likelihoodof a small error when using a chart instead ofthe formula upon which the chart was based,the results will be sufficiently accurate in mostcases, and in any case give a reasonably closepoint from which to start "pruning."The inductance required to resonate with acertain capacitance is given in the chart infigure 41. By means of the r.f. chart , theinductance of the coil can be determined,or the capacitance determined if the inductanceis known. When making calculations, besure to allow for stray circuit capacity, such astube interelectrode capacity, wiring, sockets,etc. This will normally run from 5 to 25 micromicrofarads,depending upon the componentsand circuit.To convert the inductance in microhenriesto physical dimensions of the coil, or viceversa, the nomograph chart in figure 41 isused. A pin and a straightedge are required.The inductance of a coil is found as follows:The straightedge is placed from the correctpoint on the turns column to the correct pointon the diameter-to-length ratio column, thelatter simply being the diameter divided by thelength. Place the pin at the point on the plotaxis column where the straightedge crosses it.From this point lay the straightedge to the correctpoint on the diameter column. The pointwhere the straightedge intersects the inductancecolumn will give the inductance of thecoil.From the chart, we see that a 30 turn coilhaving a diameter-to-length ratio of 0.7 and adiameter of 1 inch has an inductance of approximately12 microhenries. Likewise any oneof the four factors may be determined if theother three are known. For instance, to determinethe number of turns whPn the desired in-
N• orTURNS400300Figure 45. COIL CALCULATOR NOMOGRAPHFor single layer solenoid coils, any wire size. See text for instructions.PLOTAXISINDUCTANCE INMICROHENRIESDIAMETERRATIO LENGTH8&s43DIAMETERINCHESs42001502000a3tOO90807020028050403o------1008040---30 ------20----.atO" ----8 .58--------20 ---152.a.41.5'10.3,755 .2043
784 Radio Mathematics and Calculationsductance, the D/L ratio, and the diameter areknown, simply work backwards from the examplegiven. In all cases, remember that thestraightedge reads either turns and D /L ratio,or it reads inductance and diameter. It canread no other combination.The actual wire size has negligible effectupon the calculations for commonly used wiresizes (no. 10 to no. 30). The number of turnsof insula ted wire that can be wound per inch(solid) will be found in a copper wire table.Significant FiguresIn most radio calculations, numbers representquantities which were obtained by measurement.Since no measurement gives absoluteaccuracy, such quantities are only approximateand their value is given only to a few significantfigures. In calculations, these limitationsmust be kept in mind and one should not finishfor instance with a result expressed in moresignificant figures than the given quantities atthe beginning. This would imply a greater accuracythan actually was obtained and is thereforemisleading, if not ridiculous.An example may make this clear. Many ammetersand voltmeters do not give results tocloser than 1/4 ampere or % volt. Thus if wehave 2% amperes flowing in a d.c. circuit at6% volts, we can obtain a theoretical answerby multiplying 2.25 by 6.75 to get 15.1875watts. But it is misleading to express the answerdown to a ten-thousandth of a watt whenthe original measurements were only good to% ampere or volt. The answer should be expressedas 15 watts, not even 15.0 watts. Ifwe assume J. possible error of lj 8 volt or ampere(that is, that our original data are onlycorrect to the nearest % volt or ampere) thetrue power lies between 14.078 (product of21/ 8 and 6%) and 16.328 (product of 2% and6;;' 8 ). Therefore, any third significant figurewould be misleading as implying an accuracywhich we do not have.Conversely, there is also no point to calculatingthe value of a part down to 5 or 6 significantfigures when the actual part to be usedcannot be measured to better than 1 part inone hundred. For instance, if we are going touse 1% resistors in some circuit, such as anohmmeter, there is no need to calculate thevalue of such a resistor to 5 places, such as1262.5 ohm. Obviously, 1% of this quantityis over 12 ohms and the value should simplybe written as 1260 ohms.There is a definite technique in handlingthese approximate figures. When giving valuesobtained by measurement, no more figures aregiven than the accuracy of the measurementpermits. Thus, if the measurement is good totwo places, we wo1.1ld write, for instance, 6.9which would mean that the true value issomewhere between 6.85 and 6.95. If the measurementis known to three significant figures,we might write 6.90 which means that the truevalue is somewhere between 6.895 and 6.905.In dealing with approximate quantities, theadded cipher at the right of the decimal pointhas a meaning.There is unfortunately no standardized systemof writing approximate figures with manyciphers to the left of the decimal point. 69000does not necessarily mean that the quantity isknown to 5 significant figures. Some indicatethe accuracy by writing 69 x 10 3 or 690 x 10'etc., but this system is not universally employed.The reader can use his own system, butwhatever notation is used, the number of significantfigures should be kept in mind.Working with approximate figures, one mayobtain an idea of the influence of the doubtfulfigures by marking all of them, and productsor sums derived from them. In the followingexample, the doubtful figures have been underlined.60334.6O.l20Multiplication:637.720 answer: 6386540.342130826161962-223.668 answer: 2246540.342196\226\161\308~It is recommended that the system at theright be used and that the figures to the rightof the vertical line be omitted or guessed so asto save labor. Here the partial products arewritten in the reverse order, the most importantones first.In division, labor can be saved wh€n aftereach digit of the quotient is obtained, one figureof the divisor be dropped. Example:1.28527) 67352753 f"i46106sno~
LOW-PASS "SECTIONr-1TI---,~ 2C L2C t·L.---- -----JHIGH-PAS! T S£CTIONVALUESL c.JLOAD RESISTANCE100010000iO.. i.$.4.$...7..i.O20 2.030 3.040 4.000 o.o60 6.070 7.000 a.o90 9.0100 ~o.oi$0 tS.O200 20.0HOOt200'3001400tSOO1000i100•ooo"' ~ i9002000u.,"' uzo(...u::>0~0:.....z...20003000u .4000"' .J:t...i.,000011.1 fOOO::>~ 700000009000eooo70006000"' .Jo(u5000 "'..uz.4000 ~u..o(o(u...3000 ~zu...:t" iii2000 :I:1500...i30:0..300 30.0L c9000Courte•y, Pacific Radio Publi•hing Co.FILTER DESIGN CHARTFor both Pi-type and T -type SectionsTo lind L, connect cut-ofl frequency on left-hand scale (using lelt-slde scale for low-pass and rightsidescale for high-pass) with load on left-hand side of right-hand scale by means of a straight-edge.Then read the value of L from the point where the edge intersects the lelt side of the center scale. ReadIngs ore in henries for frequencies in cycles per second.To lind C, connect cut-off frequency on left-hand scale (using lelt-side scale for low-pass and right·side scale for high pass) with the load on the right-hand side of the right-hand scale. Then read .theya/ue of C from the point where the straightedge cuts the right side of the center scale. Readings arein microfarads for frequencies in cycles per second.for frequencies in kilocycles, C is expressed in thousands of micromicrofarads, L is expressed inmillihenries. For frequencies in megacycles, L is expressed in microhenries and C Is expressed in micromlcrofarads.For eoclo tenfold Increase In floe value of load reslstan
INDEXAcceptor, del. of .................................................. 92A.C.-D.C. power supply ...................... 561, 588, 694''Acorn'' tube ...................................................... 232A-C V- T voltmeter ............................................. .724Adapter, F-M ...................................................... 325Adapter, mechanical filter .................................. 535Admittance, del. of .............................................. 52''A''-frame antenna mast.. ................................. .448Air capacitor ...................................................... 359Air dielectric. ....................................................... 32Air-gap, capacitor .............................................. 264Alignment- B.f.o- Filter ................................ 236" Receiver- 1-F- R-F .............. 148, 235, 236All-driven antenna .............................................. 504Alpha, del. of ...................................................... 93Alternating current - Amplitude - Average ..... .45" Del. of - Effective value .................... 41, 45Generation ............................................... .42Impedance- "J" operator ....................... .47Ohm's law- Phase angle ................... .45, 46Pulsating - R.M.S .................................... .45Transformer .............................................. 61Transient .................... , ............................... 59Voltage divider .......................................... 52Alternator ........................................................... .42Amateur band receiver ........................................ 540Amateur frequency bands - licenses .................. ! 2Amateur radio service, The .................................... llAmmeter ........................................................... .721Ampere, del. of.. ............................................ 22, 23Ampere-turns ................................................ 36, 62Amperite filament regulator ................................ 552Amplification factor ....................................... .73, 74Amplifier- Audio- Cascode .................. 227, 214Cathode - Cathode driven ............ 127, 626Cathode follower ...................................... 134Class A- A2.. .......................... 99, 108, 118Class AB- AB1 .............................. 108, 137Class B .................... 99, 108, 123, 129, 251Class C .................................... l 08, 124, 251Coupling - inductive .............. 113, 11 8, 1 23D.C ......................................................... 117Doherty .................................................. 298Driver ...................................................... 126Equivalent circuit.. .................................... ! 09Feedback ................................................ ] 29Grid circuit.. .............................................. 121Grounded cathode-Grounded screen 132,629Grounded grid ........ 132, 135, 214, 258, 626Hi-li - High mu (JL) ...................... 138, 112Horizontal- I-F .............................. 139, 210Kilowatt, general purpose ........................ 635linear .................................... 138, 286, 620laflin-White ............................................ 118Multi-band linear ...................................... 624Neutralization ........................ 252, 616, 639Noise factor .............................................. ! 2 2Operational ............................................ 199Pentode .................................................. 120Pi-network .............................................. 617Plate efficiency .......................................... 1 29Power .............................................. 11 8, 61 0Printed circuit ............................................ 575A786Amplifier - Push-pull ............................ 121, 612" Push-pull tetrode ...................................... 644R-F- RC.. ...................................... 121, 109Response ................................................!! 2R-F - 4CX-1 OOOA .......................... 21 3, 640Summing .................................................. 201Termon-Woodyard .................................. 298Tetrode .................................................... 614Transistor ................................................] 04Ultra-linear .............................................. 142Vertical .................................................... 138V.h.f. cascade .......................................... 578Video ...................................... 112, 113,128Voltage ....................................................]] 0Williamson .............................................. 140Amplitude - A.C.. ...............................................45" Distortion - Modulation ........ ! 09, 282, 647Analog computers - problems.. ................ 195, 200Angle of radiation - V.h.f.. ............ A 11, 460, 478Anode, del. of - dissipation ........................ 67, 72Antenna - Adjustment ...................................... 509'' All-driven ................................................ 504Bandwidth ....................................405, 413Beam -design chart - 2-element 498, 495Bidirectional - Bi-square .............. 505, 470Bobtail ....................................................472Broadside ................................................468Bruce ......................................................470Center-fed ..............................................438Colinear ..................................................466Combinations of ....................................475Construction .................................... 448, 506Corner reflector ......................................485Couplers - mobile ..............454, 520, 524Coupling systems ....................................451Cubical quad ..........................................471Curtain ..................................................468Delta match ............................................ 501Dipole array - broad-band ........465, 435Directivity ......................................404, 410Discone ..........................................441, 481Doublet ..................................................427Dummy .................................................... 726Efficiency ................................................409Element spacing ......................................468End-effect ................................................406End-fed - end-fire ................426, 437, 473Feed systems ..................................428, 498Franklin array ........................................466Fuchs ......................................................426Gain, beam ............................................495Gamma match ........................................ 503Ground loss ............................................409Ground plane .......................................... 431Hertz ......................................................426High frequency type ................................459Impedance ..............................................408Intercept, v.h.f .......................................477lazy-H .................................................... 468length - length-to-diameter ratio 405, 406long wire ..............................................461Maconi ............................................ 408, 432Matching systems ..................................442
Antenna - Measurements ................................ 730Multee ...... ---------------------------------------------.440Multi-band ................................... .436, 514Mutual coupling ................................... .475Parasitic beam ....................................... .494Patterns ......................................... .404, 460Phasing ................................................. .466Polarization ........................................... .404Polarization, v.h.f .................................. .479Power gain ........................................... .405Q -----------------------------------------------------------.409Reactance - Resonance ............... .408, 404Rhombic ................................................. .464Rotary - Rotary match ................. .494, 502Rotator .................................................... 51 2Single wire feed ................................... .441Six-shooter ............................................. .472Sleeve ................................................... .480Space conserving ................................... .434Stacked ......................................... .468, 498Sterba ................................................... .469Stub adjustment- stub match ....... .445, 503T-match .................................................. 5013-element beam-3-element match 603, 488Triplex .................................................... 475Tuner ..................................................... .456Turnstile ................................................. .480Two-band Marconi .................................. 438Y-type ................................................... .462Vertical ................................................. .430Y.h.f.-8-element-ground plane ... .488, 481Y.h.f. - helical - horn ............... .483, 486Y.h.f. nondirectional ............................... .482Y.h.f. rhombic ....................................... .486Y.h.f. -screen array- Yogi ....... .491, 488W8JK ................................................... .473X-arroy ................................................... .469Yoke match -------------------------------------------.498"Zepplin" ..................................... .427, 467Antennascope ................................................... .738Anti-resonance .... ------------------------------------------------54Aquadag ------------------------------------------------------------85Area, capacitor plate ------------------------------------------33Arrays, antenna ....................................... .465, 412"Assumed'' voltage --------------------------------------------52Atom, def. of - atomic number ........................ 21Attenuation, Bass/Treble .................................. 139Audio - Amplifier .......................................... 227" Distortion ................................................ 135Equalizer ------------------------------------------------139Feedback loop ........................................ 141Filter, SSB .............................................. 333Frequency, def. of ................................... .42Hum --------------------------------------------------------142limiters .................................. 153, 185, 228Phase Inverter ·--------------------------------------- 115Phasing network - SSB ................ 570, 338Wiring technique .................................... 142Autodyne detector .............................................. 208Automatic load control, SSB .............................. 345Automatic modulation control ............................ 648Automatic volume control ----------------------------------225Autotransformer ------------------------------------------63, 390Avalanche voltage ............................................ 697BBalanced modulator - SSB ............ 331, 338, 343Balanced SWR bridge -------------------------------------.736Balanced transmission line measurements ....... .734Bands, amateur --------------------------------------------------1 2Bandspread tuning --------------------------------------------217Bandwidth, antenna -------------------------------.405, 413Bandwidth, 1-F, graph ........................................ 534787Bandwidth, modulation ...................................... 283Base electrode ----------------------------------------------------92Bass suppression, audio .................................... 306Battery bias ------------------------------------------------------269Beam - Design chart - Dimensions ..... .498, 496Element spacing -------------------------------------.497Front-back ratio -------------------------------------.496H-F - 5-element - 2-element ... .496, 494Power tube ............................................. .78Radiation resistance ............................... .495Three element HF- 220 Me. ....... .496, 603Beat note ........................................................ 208Beat oscillator .................. -------------------------------224B-H curve ------------------------------------------------------------36Bias - Battery --------------------------------------------------269Cathode --------------------------------------------------268Def. of - cut-off -----------------------------------.73Grid .............................................. 1 08, 267Grid leak ........................................ 112, 268R-F amplifier ----------------------------------------61 2Safety ----------------------------------------------------268Shift modulator ----------------------------------------307Supply, modulator ----------------------------------653Transistor .................................................. 96Bidirectional antenna ........................................ 505Bifilar coil ------------------------·-·-------------------------------572Billboard antenna -------------------------------------------.477Binary notation ----------------------------------------195, 196Bishop noise limiter ----------------------------------------228Bi-square antenna -------------------------------------------.470Bistable multivibrator ----------------------------------------102Bit --------------------------------------------------------------------196Blanketing interference --------------------------------------380Bleeder resistor - safety .................. 27, 709, 395Blocked grid keying ------------------------------------------398Blocking diodes -----------------------------------------------.401Blocking oscillator .................... 102, 155, 1 89, 190Bob-tail antenna -----------------------------------------------.472Bombardment, cathode ...................................... 70Boost, Bass/Treble --------------------------------------------139Break voltage ----------------------------------------------------203Bridge measurements ........................................ 728Bridge rectifier --------------------------------------------------689Bridge, slide-wire ............................................. .728Bridge, SWR ---------------------------------------------.458, 736Bridge-T oscillator --------------------------------------------192Bridge-type vacuum tube voltmeter .................... 131Bridge, Wheatstone ------------------------------------------728Broad band dipole -------------------------------------------.435Broadcast interference ........................................ 379Broadside antennas - arrays ................. .468, 41 2Bruce antenna -------------------------------------------------.470Butterfly circuit ................................................ 232cCalorimeter -------------------------------------------------------.726Capacitance - Calculation - definition .... 32, 30" lnterelectrode ---------------------------------.76, 1 06Neutralization ----------------------------------------107Tank- Stray --------------------------------261, 218Capacitive coupling --------------------------------------------270Capacitive reactance ......................................... .46Capacitor - A.C. circuit .................................... 34" Air- Characteristics ...................... 359, 358Breakdown .............................................. 264Charge ------------------------------------------------------31Color code .............................................. 531Electrolytic ........................................ 34, 708Equalizing ------------------------------------------------34~:~:~ ·:·_-_-_:::·_·_-_-_::·_-_-_-_-_-_-_-_-_-_::·_-_-_::::·_·_-_-_:·_·_-_:·.:··_~~-~: .. ~~~Leakage ----------------------------------------------------34Low inductance ........................................ 619
Capacitor - Parallel .......................................... 33 Collins feed system ............................................447•• Q ·······-------------------------------------------------------54 Collins filter ------------------------------------------------------222Series ........................................................::03 Color code, standard ................................ 531, 532Temperature coefficient ............................ 32 Colpitts oscillator .............................................. 241Time constant ............................................ 39 Complementary symmetry .................................. 97Vacuum - variable vacuum .......... 360, 620 Complex quantity ................................................49Voltage rating .......................................... 34 Component nomenclature graph ...................... 530Carrier - Distortion ........................................ 31 3 Compound, def. of .............................................. 21Shift ........................................................ 284 Compression, volume ........................................ 648S.S.B ............................. 327, 331, 332, 334 Compressor, A.L.C. ............................................ 345Cascade amplifier- V.H.F ........... 111, 214, 578 Computers, electronic ........................................ 194Cathode- Bias ........................................! 08, 268 Conductance, del. of ............................................ 52" Coupling - coupled inverter .......... 115, 116 Conduction .................................................. 67, 90Current- Definition ..........................79, 67 Conductivity - Conductor .................................. 23Driven amplifier .............................. 258, 626 Constant amplitude recording ............................ 137Follower .................................................. 274 Constant current curve ..............................74, 127Follower amplifier ............................! 27, 134 Constant-K filter ................................................ 64Follower driver ................................ 659, 662 Constant velocity recording ................................ 136Follower modulator .................................. 290 Constants, vacuum tube ......................................! 07Keying .................................................... 397 Construction, transmitter - VFO ............ 663, 676Modulation .............................................. 297 Control circuit, mobile ...................................... 525Cathode ray oscilloscope - tube ............ 138, 84 Control circuit, transmitter ................................ 391Cavity, resonant ................................................ 23 2 Conversion conductance ...................................... 80Cell, Weston ...................................................... 24 Conversion, frequency, SSB ................................ 340Center-fed antenna ............................................438 Converter stage ................................................ 211Ceramic dielectric ................................................ 32 Converter, transistor ........................................ ! 00Charge .................................................. 22, 23, 30 Core loss ............................................................ 55Chassis layout ..................................................726 Corner reflector antenna ..................................485Choke, filter .................................................... 709 Corona static .................................................... 529Choke, R-F ................................................ 272, 361 Coulomb ........................................................ 22, 30Circuit constants, measurement of ....................727 Counter e.m.l ....................................................... 37Circuits - Coupled ..........................................! 23 Counting circuits ........................................ 156, 190" D.C.- Equivalent .............................. 21, 51 Coupler, directional .......................................... 735Input loading ........................................ 122 Coupling - Capactive ...................................... 270Limiting .................................................. 151 Cathode .................................................. 115Magnetic .................................................. 35 Choke ...................................................... 11 4Parallel ...................................................... 25 Critical .............................................. 57, 218Pa~asitic .................................................. 365 Direct ...................................................... 115Q ----------------------------------------·--············-······54 Effect of .................................................. 56Resonant- Series- Tank .... 53, 24, 55, 57 Impedance - Inductive ................ 113, 271Circulating current .............................................. 56 lnterstage - r-f ............................ 270, 619Clamping circuit ................................ 153, 154, 187 Link ................................................ 272, 611Clapp oscillator ................................................ 241 Magnetic .................................................. 38Class A amplifier, del. of ................................ ! 08 ·Parasi~ic ................................................ 364Class A2 amplifier ............................................ 118 R-F feedback .......................................... 275Class AB amplifier, def. of ................................ ! 08 Systems, antenna ....................................451Class ABl amplifiers .............................. 137, 140 Transformer ............................................ 113Class B amplifiers ............ 108, 123, 125, 129, 139 Unity ...................................................... 271Class B modulator ....................................! 26, 294 Critical coupling ................................................ 218Class B, R-F amplifier ...................................... 251 Critical inductance ............................................ 686Class C amplifiers ................! 08, 124, 125, 251 Crosby modulator .............................................. 590Class C bias ...................................................... 267 Cross modulation ............................................ 381Clipper-Amplifier design .................................. 657 Crossover point ..................................................! 36Clipper filter, speech ........................................ 304 Crystal - Current ............................................ 246Clipping circuits ...................................... 151, 185 " Filter - Lattice filter .......... 220, 336, 579Clipping, negative peak .................................... 648 Harmonic ................................................ 246Clipping, phase shift ........................................ 301 Oscillator - Quartz ...................... 244, 245Clipping, speech .............................. 124, 300, 648 Pickup .................................................... 137Closed loop feedback ........................................ 192 Cubical quad antenna ......................................471Coaxial delay line ............................................ 204 Current - Alternating ......................................41Coaxial rellectometer ........................................73 2 Amplification (alpha) .............................. 93Coaxial transmission line ..................................423 Cathode .................................................... 79Coaxial tuned circuit ........................................ 232 Circulating -Closed path .................. 56, 28Code .............................................................. 14-20 Del. of .................................................... 22Code, practice set .............................................. 19 Effective - Effective (a.c.) ............ 723, 45Coefficient of coupling ...................................... 38 Electrode ................................................! 06Coefficient, temperature .................................... 32 Induced- ln~tantaneous ....................42, 44Coercive force .................................................... 37 Inverse .................................................... 693Coil, bifilar ........................................................ 572 Measurement ..................................721, 723Coil, core loss .................................................... 55 Negative screen ...................................... 641Coil, Q -------------·------------------------------------------54, 360 Peak amplifier ........................................ 125Colinear antenna ..............................................466 Rating, power supply .............................. 685Collector electrode .............................................. 92 Saturation - Skin effect ..................72, 54788
Curtain antenna ..............................................468 Double sideband, DSB .............................. 333, 353Cutoff (alpha) .................................................... 93 Doubler, frequency . ....................................... 258Cutoff bias ..........................................................73 Doubler, push-push .......................................... 260Cutoff, extended .............................................. 287 Doublet, antenna - multi-wire ................427, 429Cutoff frequency .................................................. 64 Drift transistor .................................................. 93Cycle ..................................................................41 Driver, amplifier ................................................ ! 26DDriver, cathode follower .......................... 659, 662Duct propagation ........................................... .4 15d'Arsonval meter ..............................................721 Dummy antenna - dummy loads ................... .726D.C. amplifier .................................................. 11 7 Duplex transmission .......................................... 597D.C. clamping circuit .......................................... 187 Dynamic resistance .............................................. 95D.C. power supplies .......................................... 684 Dynamotor, PE-l 03 ............................... 526D.C. restorer - circuit ..............................! 88, 154 Dynatran oscillator ................ 242Defector - autodyne - crystal ...................... 208Deflection plate ................................................ 139Degeneration, transistor ...................................... 96 Eddy current ...................................................... 38Degree, electrical ................................................43 Efficiency, antenna ..........................................409Delay line .......................................................... 204 Electric filters .................................................... 63Delta match - antenna -system .... 501, 430, 442 Electrical energy ................................................ 29Demodulator ...................................................... 207 Electrical potential .............................................. 22Detection, slope ................................................ 21 0 Electromagnetic deflection .................................. 86Detection, synchronous, DSB .............................. 353 Electrolytic capacitor ................................ 34, 708Detector - Diode ............................................ 224 Electrolytic conductor .................................... 2 2, 67Envelope ................................................ 151 Electromagnetism .................................................. 35Fremodyne .............................................. 21 0 Electromotive force (e.m.f.) ........................ 22, 37Gride leak - Impedance - Plate ........ 224 Electron coupled oscillator ................................ 241Product .......................................... 237, 352 Electron, def. of- drift- orbit .................. 67, 21Ratio ........................................................ 323 Electronic computers ......................................... 194Super-regenerative .................................. 209 Electronic conduction ......................................... 67Deviation, FM - Measurement ................ 314, 320 Electronic differentiation - integration ............ 198Dielectric, ceramic- Constant (K) ........ 30, 31, 32 Electronic multiplication ................................... 198Differential keying ............................................400 Electrostatics ..................................................... 30Differentiation, electronic .................................. 198 Electrostatic deflection - energy ............... 83, 30Differentiator ( RC) ............................................ 59 Element, def. of - metal!ic, non-metallic .... 21, 90Digital circuits - computers .............................. 195 Element, reactive, non-reactive ----··············----------21Digital package ................................................ 204 Emmission equation ............................................70Diode - A.v.c ................................................. 225 Emission, photoelectric ...................................... 67Blocking ..................................................401 Emission, secondary .................... .. ..71, 77Crystal .................................................... 234 Emission, spurious ........................................... 383Del. of ....................................................71 Emission, thermionic ............................................ 67Detector ··································-··········-···· 224 Emitter electrode .................................................. 92Gate ........................................................ 204 Enclosures ................................................ 724, 725limiter .................................................... 185 End effect, antenna ............................................406Mixer ...................................................... 212 End-fed antenna ........................................426, 437Modulator .............................................. 332 End-fire antennas - arrays ....................473, 412Semi-conductor .......................................... 90 Energy- Electrical- Electrostatic .............. 29, 30Storage time ............................................! 03 Potential .................................................... 30Voltmeter ................................................724 Storage, capacitor .................................... 31Dipoles ....................................403, 435, 465, 498 Transistor .................................................. 93Direct current circuits .......................................... 21 ENIAC ................................................................ 195Directional coupler ............................................735 Envelope, carrier .............................................. 2 8 2Directive antennas, H-F ....................................459 Envelope detector .............................................. 151Directivity, antenna ..................................404, 41 0 Equalizer, audio ................................................ 139Discharge of capacitor ........................................ 30 Equal-tempered scale ........................................ 135Discane antenna ......................................441, 481 Equivalent circuit ................................................ 51Discontinuities, transmission line ......................425 Equivalent circuit, transistor .................................. 95Discriminator, F-M .............................................. 322 Equivalent noise resistance ................................! 23Dissipation, anode ..............................................72 Error cancellation, feedback .............................. 159Distortion - Amplitude - Frequency .............. 109 Error signal .............................................. 159, 193" Audio ...................................................... 135 Excitation, grid ................................................ 252Carrier .................................................... 31 3 Exciters, SSB .............................................. 346, 349Harmonic ................................................ 119 Extended-Zepp antenna ................................... 467lntermodulation ...................................... 136Modulation .............................................. 307Nonlinear - Phase ......................! 09, 135 Foetor of merit ................................................... 54Products, SSB .......................................... 344 Fading ..............................................................4 18Transient ................................................ 135 Farad, del. of ...................................................... 31Transistor .................................................. 98 Feedback amplifier ............................................ ! 29Divider, frequency ............................................ 190 Feedback, audio ................................................ 141Divider, voltage .................................................. 26 Feedback circuits ................................................ 192Doherty amplifier .............................................. 298 Feedback, control .............................................. 158Dome audio phasing network ............................ 338 Feedback, error cancellation .............................. 159Double conversion circuit .................................... 215 Feedback, Miller .............................................. 199789EF
Feedback, R-F ............................................ 274, 615 Fuchs antenna ..................................................426Feedback, transistor .......................................... 101 Full-wave limiter ................................................ 230Feed systems, antenna ..............................428, 498 Full-wave rectifier .............................................. 689Feed-through power .......................................... 627 function generator- non-linear- ramp 202, 203Field, magnetic .................................................. 35Filament, del. of ................................................ 67GFilament reactivation ............................................ 68 G, conductance, del. of ...................................... 52filament regulator .............................................. 552 Gain, antenna - beam ............................461, 495filter - Capacitor ..................................707, 690 Gain, power - resistance - voltage ................ 94Carrier, SSB ............................................ 334 Gamma match, antenna - system .......... 503, 444Choke ......................................................709 Gas tube - generator .............................. 87, 140Circuits, mobile ...................................... 518 Gate, diode - limiter ............................ 204, 1 85Crystal .................................................... 220 Gauss, del. of ...................................................... 36Crystal lattice - SSB .................... 579, 336 Generator noise ...................................... 528, 740Generator, SSB ...................................... 333 Generator, sawtooth .......................................... 140High pass ...................................... 64, 372 Generator, time base ........................................ 139Inductor input .......................................... 687 Gilbert, del. of .................................................... 36Insertion loss ............................................ 64 Grid bias .................................................. 108, 267Low pass .......................................... 64, 376 Grid, del. of ......................................................72M-derived ................................................ 64 Grid excitation .................................................. 252Mechanical ...................................... 222, 336 Grid leak bias .................................. 109, 112, 268Noise ...................................................... 227 Grid leak detector ............................................ 224Passband, SSB ........................................ 336 Grid limiter, limiting ................................ 153, 187Q-multiplier ............................................ 236 Grid modulation ...................................... 252, 288Resistance-capacitance-Resonance 687, 688 Grid neutralization .......................................... 253Ripple factor .......................................... 68 8 Grid-screen mu factor ........................................78Sections - Series - shunt .................... 64 Ground bus .............................................. 142, 147TVI, receiver - TVI-type ................ 372, 64 Ground currents ................................................ 362Wave ...................................................... 63 Ground lass, antenna ......................................409filter-type exciter, SSB ...................................... 349 Ground plane antenna - VHF ..............431, 481Fixed bias .......................................................... 108 Ground resistance ............................................41 0Flat-top beam antenna ......................................473 Ground, R-F ...................................................... 362Fletcher-Munson curve ........................................ 140 Ground termination ..........................................433Floating Paraphase inverter .............................. 116 Ground, transmitter .......................................... 394Flux, del. of - density .............................. 35, 36 Ground wave ....................................................4 14Flywheel effect .......................................... 57, 259 Grounded-cathode amplifier .............................. 132Folded dipole ............................................429, 498 Grounded-grid amplifier .... 132, 135,140,214, 258Forcing function ................................................ 200 Grounded-grid, Class B amplifier .................... 139Fosler-Seely discriminator ·····························+···322 Grounded-grid r-f amplifier .............................. 626franklin antenna ..............................................466 Grounded neutral wiring .................................. 565Franklin oscillator .............................................. 243 Grounded-screen r-f amplifier ............................ 629Free electrons ...................................................... ~ 1 Guy wires, antenna ..........................................449Free-running multivibrator .................................. 189Fremodyne detector ............................................ 21 0Frequency ....................................................41, 58 Hairpin coupling ................................................ 232" Conversion, SSB ...................................... 340 Half-wavE' rectifier ............................................ 689Cutoff ........................................................ 64 Handie-Talkie, 144 Me ..................................... 547Distortion ................................................ 109 Harmonic - B.f.a ............................................. 225Divider .................................................... 190 Crystal .................................................... 246Interruption ............................................ 209 Del. of ...................................................... 58Maximum useable ..................................417 Distortion ................................................ 119Measurements ........................................729 Music ...................................................... 135Multiplier ................................................ 258 Oscillator ................................................ 247Pulse repetition ...................................... 190 Radiation ................................ 262, 373, 725Range, hi-fidelity .................................... 136 Harmoniker TVI filter ........................................ 379Ratio, Lissajous ........................................ 144 Hartley oscillator .............................................. 240Resonant .................................................. 53 Hash rectifier ............................................ 679, 694Shift keying .......................................... 326 Heat cycle, resistor .......................................... 357Sound .................................................... 134 Heat sink .................................................. 99, 147Spectrum ....................................................41 Heat sink, transistor ................................ 576, 707Spotter ....................................................730 Heater cathode .................................................. 70Standard ...............................................] 29 Heising modulation ............................................ 294Sweep ...................................................... 140 Helical antenna ................................................4R3Frequency Modulation ...................................... 31 2 Henry, del. of .................................................. 37" Adapter .................................................. 325 Hertz antenna ..................................................426Devotion - ratio - index .................. 314 Heterodyne ........................................................ 208Discriminator .......................................... 322 H-F antennas ....................................................459Limiter .................................................... 324 High fidelity .............................................. 134-138Linearity .................................................. 31 8 Interference ............................................ 386Narrow band .......................................... 315 High-pass TVI filter .......................................... 372Pre-emphasis ............................................ 325 Holes in semi-conductor ...................................... 91Reception ................................................ 3 21 Horizontal directivity ........................................4 10Front-back ratio, beom ......................................496Horn antenna, VHF ..........................................4&6790H
Hot cathode oscillator .............................. 580, 676 lnterstuge coupling ............................ 270, 619, 665Hot cathode phase inverter .............................. 116 Intrinsic semi-conductor ........................................ 91Hum, audio ...................................................... 142 Inverse current- Voltage .................................. 693Hysteresis loop - loss .................................. 36, 38 Inverter, phase - voltage divider .... 115, 116, 117ion, del, of- positive .................................. 22, 87Ionosphere, absorption ...................................... 131-f alignment ·······································'···-148, 235 Ionospheric layers - propagation ..........417, 416" Amplifier ................................................ 21 0 IR drop .............................................................. 24Band-width graph .................................. 534 Iron vane meter ................................................723Filter, lattice ............................................ 579 Isotropic radiator .............................................41 0Noise limiter .......................................... 227Pass-band ................................................ 219JRejection notch ....................................... 222 Jitter ........................................................ 154, 188Shape factor .......................................... 219 Johnson Q-feed system ....................................446Tuned circuit .......................................... 2 i 8 ''J'' operator ......................................................47Ignition noise ......'.............................................. 528 Joule, del. of ................................................ 30, 37Images-image intereference-ratio 211, 213, 385 Junction, transistor .............................................. 93Impedance ....................................................46, 47'' Antenna ..................................................408 KComplex .................................................... 51 K, dielectric constant .......................................... 32Coupling .......................................... 56, 113 Key clicks .......................................................... 396match, audio .......................................... 127 Keying - blocked grid .................................. 398reflected ............................................ 56, 63 Cathode circuit ...................................... 397Resonant .................................................... 54 Differential ..............................................400Screen circuit .......................................... 289 Frequency shift ...................................... 326Surge ......................................................424 screen grid .............................................. 399Tank circuit .............................................. 262 Sequence ................................................ 677Transformation .......................................... 63 Transmitter .............................................. 395transistor .................................................. 97 Kilocycle .........•..•.................................................42Transmission line ....................................421 Kinescope tube .................................................. 84Triangle ....................................................48 Kirchhoff's Laws .................................................. 28Incident voltage ................................................ 732 Klystron - reflex .......................................... 81, 82Indicator, Antennascope ....................................738Indicator, selsyn ................................................ 514Indicator - standing wave - twin-lamp 731, 735 Lamb noise limiter ............................................ 227Induced current ....................................................42 Layout, chassis ..................................................726Inductance .......................................................... 37 Lazy-H antenna ................................................ 468Capacitor ................................................ 359 L/C ratio ............................................................!;6Cathode lead ..........................................! 22 Lead inductance .................................................. 81Critical .................................................... 686 Leakage, capacitor ............................................ 34Lead - Magnetic - Mutual ............ 81, 37 Leakage reactance ............................................ 63Parallel - series ...................................... 38 Left-hand rule .................................................... 35Resistor .................................................... 357 Length, antenna ................................................405Screen lead ............................................ 257 L-section filter .................................................... 64Induction, del. of ................................................42 Licenses, amateur .............................................. 12Inductive reactance ............................................ .46 Limiter, audio .................................................. 228Inductive coupling ............................................ 271 Limiter, diode .................................................... 151Inductive tuning ................................................ 617 Limiter, F-M ...................................................... 324Inductor input filter .......................................... 687 Limiter, full-wave .............................................. 230Inductor, iron core .............................................. J8 Limiter, grid ...................................................... 153Inductor, time constant ......................................40 Limiter, noise .................................................... 227Infinite impedance detector .............................. 224 Limiter, series diode .......................................... 185Initial condition voltage ................................. 203 Limiter, TNS ...................................................... 230Injection voltage .............................................. 213 Limiting circuit .................................... 151, 152, 185Input loading .................................................. 122 Limiting, grid ....................................................! 87Input resistance ................................! 07, 216, 276 Line, delay ...................................................... 204Insertion loss, filter .......................................... 64 Line filter ........................................................ 227Instability, R-F amplifier .................................. 615 Line regulation .................................................. 388Instability, static .............................................. 117 Line, Slotted ...................................................... 730Insulation, VHF ..................................................479 Linear amplifier ................................................ 286Insulator, del. of .......................................... 22, 23 " 4CX- 1 OOOA ............................................ 64 1Integration, electronic ...................................... 198 Class B .................................................... 129Integrator ( RC) .................................................. 59 SSB .................................................. 620, 624lnterelectrode capacitance ..................................76 Tuning .................................................... 287Interference - Broadcast ................................ 379 Linear matching transformer ..............................446Harmonic ................................................ 373 Linearity tracer ................................................ 151Hi-li - image- TV .............. 386, 385, 371 Link circuit ........................................................ 611Interlock, power .............................................. 395 Link coupling- antenna .......................... 272, 452lntermodulation distortion .................................. 136 Lissajous figures .............................................. 144lntermodulation test .......................................... 145 Litz, wire ............................................................ 54Internal resistance ..............•..•..........•.........•....... 25 L-network design .............................................. 265International Morse Code .................................... 15 Load line ............................................................74Interruption frequency ...................................... 209 Load line, transistor .......................................... 99791L
Load resistance amplifier .................................. 119Loaded-Q ............................................................ 57Laflin-White amplifier ...................................... 118Long-wire antenna -------------------------------------------.461Loop feedback --------------------------------------------------158Loran, band -----------------------------------·········-··---------1 2Loudness control, audio --------------------------------------140Loudspeaker - del. of - response ........ 139, 140Low-frequency porosities ----------------------------------366Low-pass TVI filter ------··--·---------------------------------376MMagic eye tube --------------------------------------------88, 226Magnetic - Air gap ------------------------------------------38Circuit - field - flux --------------------------35eddy current ······-······--··-----------------------------38hysteresis loss ····-·····---------------------···--·-----38induction --------------------------------------------------37left hand rule ------------------------------------------35Permeability - reluctance - saturation .. 36Magnetism - residual ----------------------------------35, 37Magnetomotive force ( M. M. F.) ----------------------36Magnetron ----------------------------------------------------------83Magnitude, scalar ---------------------------------------------.48Majority carrier --------------------------------------------------93Marconi antenna ---------------------.408, 432, 434, 438Most, A-frame -------------------------------------------------.448Matching stub, antenna ---------------------------.445, 503Matching systems antenna -------------------------------.442Matching transformer -------------------------------------.429Mathematics, radio ------------------------------------------742Maximum usable frequency ------------------------13, 417Maxwell, del. of ----------------------------------------------36M-derived filter --------------------------------------------------64Measurements - Antenna -----------------------------.730Bridge -------------------------------------------······---728Circuit constants -----------------------------------.727Current - voltage -----------------------.721, 723Frequency ----------------------------------------------729Parallel wire line -----------------------------------.735Power ---------------------------------------------.721, 725Transmission line ---------------------------.730, 734Mechanical filter ----------------------------------------------222Mechanical filter adapter - S.S.B . .......... 535, 590Megacycle -----------------------------------------------------------.42Megohm --------------------------------------------------------------24Memory circuit --------------------------------------·---------204Mercury vapor rectifier ------------------------------------694Meteor bursts ---------------------------------------------------.420Meter, d'Arsonval ----------------------------------------------721Meter, iron vane ---------------------------------------------.723Meter, multi-range ------------------------------------------722Meter, rectifier -------------------------------------------------.724Meter, thermocouple -------------------------------.724, 726Mho, del. of -----------------------------------------------------.74Mica dielectric ----------------------------------------------------32Microfarad --------·--········-------------·----·--····------------30Microhenry ------------···--······--····-------·----------·----------37Micro-ohm -------·--------------------·--·---------····--······-----23Middle C ------------------------------------------------------------134Miller effect --------------------------------------107, 112, 220Miller feedback - oscillator ................ 199, 247Milliammeter -----------···--···--····--·-------------······----.721Millihenry, del. of ----------------------------------------------37Mixer - circuits ----------------------------------------------212Diode ----------------·-------------------------------·····--·80Noise ------------------------------------------------------212Products, SSB - spurious .............. 341, 343SSB ------··--------------·--····--------··-----------------340Stage ------------------------------------------------------211Transistor ------------------------------------------------100Triode ----------------------------------------------------213Tube ----------------------------------------------------·---79792Mobile - Amplifier, SSB ----------------------------------620Antenna coupling ----------------------------520, 524Control circuit ------------------------------------------525Dynamotor ----------------------------------------------526Equipment construction -design .... 524, 515Filament regulator --------------------·-------------552Filter circuits ------------------------------------------518Noise limiter ------------------------------------------516Noise sources ------------------------------------------529Power supply --------------------------521, 697, 519Supply, three-phase --------------------------------620Mode of resonance ---------·----------------------------------232Modulation --------------------------··----·--··--···--·······----·239Amplitude --------------------------------------282, 647Automatic control ------------------------------------648Bandwidth ----------------------------------------------283Cathode --------·-------·------------------·--------------297Class B ----------------------------------------------------294Constant efficiency --------------------------------286Distortion ···----------------------------------·-······---307Frequency ------------------------------------------------31 2Grid ------------------------------------------------252, 288Heising ····-------------···--···-----·---·---·-------------294Index, F-M ----------------------------------------------314Pattern, oscilloscope --------------------------------146Percentage ------·----···--··----------------·····--····-284Phase ----------------------------------------------315, 319Plate --------------------------------------251, 293, 295Screen ------------------------------------------------------2 8 8Suppressor ----------------------------------------------292Transformer ·····--------------------------------------·295Variable efficiency ----------------------------------285Velocity ----------------------------------------------------81Modulator - Adjustment --------------------------------654" Balanced - S.S.B. ----------------331, 338, 343Bias-shift -------------------------------···--···-----------307Cathode follower ------------------------------------290Class B ----------------------------------------------------1 25Construction ·----------····--········--··--···----------650Crosby ·····--······----·---------················---······590Diode ·----------···--··---····--··--····-------------------332Matching - impedance match ...... 125, 127Reactance tube ----------------------------------------315SSB ------·-·----·--···--···--------·······--···--------------340Tetrode ···--·····----------------------------·-----------647Zero bias ------------------------------------------------662Molecule --------------------------------------------------------------21Monitor oscilloscope .......................................... 147Morse Code -----------------------------------------------·--------15Mu factor (JL) ---------------------------------------------------.78Multee antenna -----------------------------------------------.440Multi-band antenna -----------------------.......... .436, 514Multiplication, electronic ----------------------------------198Multiplier, frequency ----------------------------------------258Multiplier resistor ---------------------------------------------.72 2Multi-range meters --------------------------------------------722Multivibrator- circuits .................... 1 02, 154, 188Multivibrotor, free running ................................ 155Multi-wire doublet ................................. .429, 442Music, del. of ----------------------------------------------------134Music systems - scale ----------------------- ....... 138, 1 35Mutual conductance -------------------------------------------.73Mutual coupling ------------------------------------------------21 8Mutual coupling, antennas -------------------------------.475Mutual inductance .............................................. 37Mycalex, dielectric ----------------------------------------------32NNarrow-band F-M ----------------------------------------------315NBS bridge-T oscillator --------------------------------------1 92Negative feedback loop ------------------------------------140Negative peak clipping ------------------------------------648Negative screen current ----------------------------------641
Networks, L ond Pi --------------------------------------------265Network, phasing ----------------------------------------------570Neutralization - Amplifier ------------------------------252" Capacitance --------------------------------------------107Capacitor, VHF ----------------------------------------672Grid --------------------------------------------------------253Praced u re ................................................ 2 79R- F .. ------ ............. ---------------------- ..... ----- ..... 2 7 6R-F amplifier ----------------------------------616, 639Shun I ...................................................... 254Tes I .............................. ---------- ................ 2 78Transistor ------------------------------------------------100Neu Ira I izi ng procedure . __ ----------- ...................... 255Nl (ampere turns) ----------------------------------------------62Noise - Factor ------------------------------------------------1 22Check, mobile ----------------------------------------52 8Generator, silicon crystal -------------------------.7 40limiters - mobile ------------------------227, 516Mixer ------------------------------------------------------21 2Regulator - sources ------------------------------528Suppression ----------------------------------2 27, 5 27Thermal agitation ------------------------------------188Voltage --------------------------------------------------1 21Wow and flutter ------------------------------------135Nomenclature, component --------------------------------530Nonconductor ...................................................... 2 2Nondirectional VHF antenna ------------------------------"82Non-linear distortion ------------------------------------------135Non-linear function ------------------------------------------202Non-resonant transmission line --------------------------"21Nonsin usoidal wave --------------------------------------------58N-P-N Ira nsistor .. ----------- ............... ----------- ........... 9 20Octave, del. of --------------------------------------------------135Oersted, del. of --------------------------------------------------36Ohm - Ohm's law ------------------------------------23, 24Ohmmeter, low range ---------------------------------------.723Ohm's Law ( a.c.) ----------------------------------------------"5Ohm's law (complex quantities) ------------------------"9Ohm's Law for magnetic circuit ----------------------------36Ohm's Law (resonant circuit) ·---------------------------540 h ms- per-volt ............ ---------------------------------- ..... .72 2Omega, del. of ----------------------------------------------------"4One-shot multivibrator --------------------------------------1 89On-off circuits --------------------------------------------------195Open wire line --------------------------------------------------"21Operating desk, construction of -----------------------.724Operational amplifier ----------------------------------------199Orbital electron --------------------------------------------------21Oscillation, parasitic ----------------------279, 365, 616Oscillator - Beat --------------------------------------------224Blocking ----------------------102, 155, 156, 189Bridge-T --------------------------------------------------192Circuits ........ --------------- ............................. 250Clapp - Colpitts ------------------------------------241Code practice --------------------------------------------19Crystal ----------------------------------------------------244Dynatron ------------------------------------------------242e. c. o. ----------------------------------------------------241Franklin __ ---------- ........... ----------------- .......... 2 43Free running ------------------------------------------141Harmonic ------------------------------------------------247Hartley .......... ---------- ........................ ___ ..... 240High stability ------------------------------------------604Hot cathode ----------------------------------580, 676Keying ----------------------------------------------------249Miller ------------------------------------------------------247NBS bridge-T ------------------------------------------192Overtone ------------------------------------------------249Phase shift ------------------------------------157, 191Pierce ------------------------------------------------------247RC ------------------------------------------------------------157793Oscillator - Relaxation ------------------------------------102t.p.t.g. ----------------------------------------------------241Transistor ------------------------------------------------101Trans i stron ..................... -------- ............. ---- 24 3V.F.O. ------------------------------------------------------244Wien-bridge ----------------------------------157, 191Oscilloscope ------------------------------------------------------138Circuit ------------------------------------------------------14 2Linearity tracer ----------------------------------------151Lissajous figures --------------------------------------144Modulation pattern-phase patterns 146, 145Monitor ------------------------------------------147, 149Sideband measurements --------------------------150Trapezoidal pattern-wave pattern 146, 148Output, peak power ------------------------------------------124Overloading, TV receiver ----------------------------------371Overtone crysta Is ------ .......................... ___ ........... 249Overtone, music ------------------------------------------------135Oxide filament --------------------------------------------67, 69pPaper dielectric --------------------------------------------------32Parallel circuit - resistance --------------------------------25Diode limiter ------------------------------------------1 86Feed --------------------------------------------------------273Resona nee ............ ----------- ............. _____ ....... 54Wire line measurements -------------------------.735Parasitic antenna, design chart ------------------------"98" Antenna, VHF -----------------------------------------.488Beam design -----------------------------------------.494Check for .. ----------- ................ __ .... ______ ....... 3 6 8Coupling ........................... ------- ................ 3 64Element, antenna ------------------------------------"97Oscillation ------------------------------279, 365, 616Resonance ----------------------------------------------364Pass band, 1-F --------------------------------------------------219Pass band, mechanical filter ------------------------------223Pass band, SSB filters ----------------------------------------336Patterns, antenna ------------------------------------"04, 460PE-l 03 dynamotor --------------------------------------------5 26Peak amplifier current ----------------------------------------1 25Peak current ( a.c.) ----------------------------------------------"5Peak envelope power, SSB --------------------------------327Peak limiter ------------------------------------------------------1 85Peak noise limiter ----------------------------------------------227Peak power output --------------------------------------------1 24Peaked wave ·-----------------------------------------------------59Pentode amplifier ----------------------------------------------1 20Pentode tube -----------------------------------------------------.77Period, sound ----------------------------------------------------1 34Permeability --------------------------------------------------------36Phantom signal --------------------------------------------------3 8 2Phase - angle ( a.c.) ----------------------------------------"6" Angle - difference ----------------------146, 145Distortion ----------------------------------------1 09, 135Inverter ----------------------------------------------------115Modulation ------------------------------------315, 319Shift, clipping ------------------------------------------301Shift, feedback ----------------------------------------193Shift, oscillator ------------------------------157, 191Phasing, antenna ----------------------------------------------"66Phasing generator, SSB --------------------------------------337Phasing networks, audio ------------------------------------338Phasing network, SSB ----------------------------------------570Phonograph reproduction ----------------------------------136Photoelectric emission ------------------------------------------67Pickups --------------------------------------------------------------1 37Pickup, spurious ------------------------------------------------384Pierce oscillator ------------------------------------------------247Pi-network amplifier ------------------------------------------617Pi-network chart - design --------------------267, 265Pi-network coupling ·-----------------------------------------"53Pilot carrier, SSB ------------------------------------------------327
Pitch, sound ...................................................... 134 Radiation, del. of ..............................................403Plate current flow ............................................ 259 Radiation, harmonic ................................ 373, 725Plate detector .................................................... 224 Radiation pattern, distortion ..............................478Plate efficiency amplifier .................................. 129 Radiation resistance ................................404, 495Plate modulation ...................................... 251, 293 Radiator, isotropic ..............................................41 0Plote resistonce ............................................73, 74 Radiator crass-section, VHF ................................ 479P-N-P transistor .................................................. 91 Radio frequency, del. of ....................................42Point contact transistor ........................................ 92 Radio mathematics ............................................742Polar notation ....................................................48 Radio propagation ............................................413Polarization, antenna -VHF ....................404, 479 Radio teletype .................................................. 326Polyphase rectifier ............................................ 693 Ramp function .................................................... 203Potential difference .............................................. 2 2 Ratio detector .................................................... 323Potential, electrode ............................................ 106 Ratio, image ...................................................... 21 3Potential energy .................................................. 30 RC amplifier ...................................................... 109Power amplifier design ...................................... 61 0 RC audio network .............................................. 139Power amplifier triode ...................................... 11 8 RC differentiatar - integrator ............................ 59Power, gain ........................................................ 94 RC oscillator - circuits .......................... 157, 191Power gain antenna ..........................................405 RC time constant ................................................ 60Power gain SSB .................................................. 328 RC transient ........................................................ 38Power measurement ..................................721, 725 Reactance, antenna ............................................408Power interlock ................................................ 395 Reactance, capacitive - inductive - net ........46Power-line filter ................................................ 227 Reactance, leakage .............................................. 63Power loss, VHF ................................................ 663 Reactance, resonance ..........................................47Power, resistive .................................................. 29 Reactance tube modulator .................................. 315Power supplies .................................................. 684 Read-out ............................................................ 20 1'' A.C.-D.C. .................................................. 694 Receiver, alignment .................................. 148, 235Mobile .................................................... 697 Receiver, amateur band ...................................... 540Oscilloscope ............................................ 141 Receiver, superheterodyne .................................. 21 0Regulated ................................................ 711 Receiver, transistor .................................... 103, 533Screen ...................................................... 269 Receivers, UHF .................................................. 231Selenium .................................................. 561 Reception, frequency modulation ........................ 321Three-phase .................................. 521, 620 Reception, mobile ............................................ 515Transistorized ..........................................703 Reception, SSB .................................................. 351V.F.O ....................................................... 604 Recording, constant amplitude, velocity .... 136, 137Voltage multiplier .................................. 695 Recording, crossover point ................................ 136Power system, primary ...................................... 387 Recording, high fidelity ...................................... 136Power transformer ..............................................709 Recording, pre-emphasis - RIAA curve ............ 137Power transistor .................................................. 99 Rectification, stray .............................................. 383Powerstat auto-transformer ................................ 390 Rectified A.C. ......................................................45Preamplifier, hi-li .............................................. 138 Rectifier - bridge - full wave - half wave .... 689Pre-emphasis, recording .................................... 137 " Hash .............................................. 679, 694Preselector ........................................................ 214 Mercury vapor - pa:yphase ........ 693, 694Primary transformer ............................................ 61 Selenium .................................................. 695Printed circuit amplifier ...................................... 575 Si Iicon ............................................ 592, 696Probe, R.F. (v.t.v.m.) ........................................ 133 Type meter ..............................................724Product detector ........................................ 352, 237 Vacuum .................................................... 693Propagation ..............................................413-420 v.t.v.m ..................................................... 133Pulsating alternating current ..............................45 Reflected impedance .................................... 57, 63Pulse-repetition frequency ...................... 103, 190 Reflected voltage ................................................732Push-pull amplifier .................................. 61 2, 121 Reflectometer, coaxial ........................................732Push-pull tetrode amplifier ................................ 644 Regeneration ...................................................... 208Push-pull transformer ........................................ 113 Regulated power supply ....................................711Push-pull tripler - doubler .............................. 260 Regulation, power line ...................................... 388Push-to-talk circuit .............................................. 526 Regulation, voltage ............................................ 686QRegulator, filament ............................................ 552Regulator noise .................................................. 528Q amplifier tank ................................................ 128 Regulator tube (VR) ........................................709Q antenna ........................................................409 Regulator, voltage .............................................. 687Q coils .............................................................. 360 Rejection notch, 1-F ............................................ 222Q del. of ............................................................ 54 Rei, del. of .......................................................... 36Q loaded - unloaded ...................................... 57 Relaxation oscillator .................................. 102, 18 8Q multiplier ...................................................... 236 Reluctance, del. of ............................................ 36Q tank circuit .................................................... 261 Remote cut-off tube ............................................78Q transformer ....................................................429 Residual magnetism ............................................ 37Q tuned circuit .................................................. 216 Resistance: .......................................................... 23Quad antenna ..................................................471 capacitance filter ...................................... 687Quadrant, sine ....................................................43 Dynamic .................................................... 95Quantity, complex ................................................49 gain .......................................................... 94Quench oscillator ............................................. 209 Ground ..................................................41 0RInput .............................................. 1 07, 216Internal - parallel - series .................. 25Radian, del. of ....................................................43 load ........................................................ 119Radiation, angle of ..........................411, 460, 478 plate ..................................................73, 74794
Resistance - Radiation -----------------------------------.404Resistive power ----------------------------------------------------29Resistivity, table of ----------------------------------------------23Resistor, bleeder --------------------------------------27, 709Resistor, characteristics of ----------------------------------356Resistor color code --------------------------------------------531Res is tor, eq ua I izi ng _------------------------------------- ______ 34Resistor, inductance of ----------------------------------------357Res is to• m u Jti pI ier _______ ----- _-------------------- _____ -----.72 2Resistor, typical ----------------------------------------------------24Resonance: __ --------------------------------------------------- ___ .A 7Antenna -----------------------------------------------.404Current -------------------------·----------------------------56Curve - parallel - Series ________________ 53, 54Fi I fer ----------------------- ___ ------------------------------68 8Impedance- Q ----------------------------------·-----54Mode ------------------------------------------------------232Oscilloscope pattern --------------------------------149Parasitic --------------------------------------------------364Speaker --------------------------------------------------140Resonant cavity --------------------------------------------------232Resonant circuit ·-------------------------------------------53, 54Resonant transmission line -------------------------------.424Response, audio --------------------------------------11 2, 136Return trace ------------------------------------------------------140R -F a I ig n men t _____ ------ _---------------------------------------- 236R-F amplifier: --------------------------------------------213, 251" 4CX- 1 OOOA ----------------------------------------------640Cathode driven ----------------------------------------626Construction --------------------------------------------616General purpose --------------------------------------635Grounded grid - grounded screen 626, 629Inductive tuning --------------------------------------61 7Instability- neutralization ____ 615, 616, 639Oscillation ----------------------------------------------61 6Pi-network ----------------------------------------------617Push -pull ------------------------------------------------61 2Self-neutralization ------------------------------------616Tetrode ----------------------------------------------------614R-F chokes ------------------------------------------------272, 361R-F feedback ------------------------------------------------------274R-F ground --------------------------------------------------------362R-F interstage coupling --------------------------------------619R-F shielding ------------------------------------------------------362Rhombic antenna -------------------------------------.464, 486Rh um batron cavity --------------------------------------------23 2RIAA equalizer curve ----------------------------------------1 37Ribbon TV line -------------------------------------------------.478Ring diode modulator ----------------------------------------333Ripple factor, filter --------------------------------------------688Ripple voltage --------------------------------------------------686RL circuit -------------------------------------------------------------.40RL transient --------------------------------------------------------38RLC circuits ---------------------------------------------------------.48Root mean square (a.c.) -----------------------------------.45Rotary beam antenna ---------------------------------------.494Rotator, antenna ----------------------------------------------51 2Ruggedized tube --------------------------------------------------88sSafety bleeder ----------------------------------------------------395Safety precautions --------------------------------------------393Saturation current ------------------------------------------------72Saturation, magnetic ------------------------------------------36Sawtooth wave ------------------------------------------59, 140Scalar notation ---------------------------------------------------.48Scale, musical ----------------------------------------------------135Scatter signals -------------------------------------------------.419Screen, CRT --------------------------------------------------------87Screen current, negative ----------------------------------641Screen grid keying --------------------------------------------399Screen grid tube -------------------------------------------------.77Screen lead inductance --------------------------------------257Screen modulation --------------------------------------------288Screen supply ----------------------------------------------------269Secondary emission _____________________________________ ]], 77Secondary transformer ----------------------------------------61Selective fading, SSB ----------------------------------------331Selectivity, arithmetical --------------------------------------211Selectivity chart --------------------------------------------------342Selectivity control, 1-F ----------------------------------------221Selectivity, mobile reception ------------------------------517Selectivity, resonant circuit ----------------------------------54Selectivity, tuned circuits ----------------------------------341Selenium power supply ----------------------------561, 669Selenium rectifier ----------------------------------------------695Self-inductance ----------------------------------------------------37Self-neutralization amplifier ------------------------------616Self-neutralization, VHF ------------------------------------664Selsyn indicator --------------------------------------------------514Semi-conductor ----------------------------------------------90, 21Semi-conductor rectifier ------------------------------------695Sequence keying ------------------------------------------------677Series-cathode modulator ----------------------------------297Series circuit ------------------------------------------------------24Series-derived filter --------------------------------------------64Series-diode limiter ----------------------------------151, 1 85Series feed --------------------------------------------------------273Series-feed amplifier ------------------------------------------61 2Series-parallel circuit ----------------------------------------25Series resistance - resonance ____________________ 25, 53Shape factor, 1-F ----------------------------------------------219Shielding, R-F ----------------------------------------------------362Shot effect ------------------------------------------------------1 23Shunt derived filter --------------------------------------------64Shunt loading, a.v.c. ------------------------------------------226Shunt neutralization ------------------------------------------254Sidebands, deL of --------------------------------------------28 2Signal, error --------------------------------------------159, 193Signal, phantom ----------------------------------------------382Signal-to-distortion ratio - SSB __________ 151, 344Signal-to-noise ratio ----------------------------------------122Silicon crystal noise generator _________________________ ] 40Silicon rectifier ----------------------------------------592, 696Sine wave -------------------------------------------------.42, 43Single-ended amplifier --------------------------------------611Single sideband (SSB): __________________ 285, 329, 327" Amplifier, 4CX-1 OOOA ----------------------------640Envelopes _____________ -------- _ ---------- ________________ 3 3 0Envelope detector ------------------------------------151Equipment --------- ___ ---------- ______________________ ____ 5 67Filter --------------------------------------------------------3 27Filter system --------------------------------------------590Grounded grid amplifier ------------------------626Grounded screen amplifier ----------------------629Jr. exciter ------------------------------------------------346Linear amplifier --------------------------------------620Linear amplifier, kilowatt ------------------------635Linear operation --------------------------------------138Measurements ------------------------------------------150Reception ------------------------------------------------351Transmission --------------------------------------------3 27Transmitter adjustment --------------------------573Single signal reception --------------------------------------222Single swing oscillator --------------------------------------1 89Single-wire antenna tuner -------------------------------.456Single-wire feed, antenna ---------------------------------.441Single-wire feeder -------------------------------------------.430Six-shooter antenna -----------------------------------------.472Skeleton VHF antenna ---------------------------------------.481Skin effect ------------------------------------------------·---------54Skip distance ---------------------------------------------------.41 8Sky wave ---------------------------------------------------------.414Sleeve antenna -------------------------------------------------.480Slide-wire bridge ---------------------------------------------.728Slope detection --------------------------------------------------210795
Slotted line ........................................................ 730 Tetrode neutralization ...................................... 256S-meter circuits .................................................. 226 Tetrode r-f amplifier ........................................ 614Soldering techniques ........................................727 Tetrode, screen safety circuit ............................ 679Sound in air ...................................................... 134 Tetrode, self neutralization ................................ 664Space charge ..............................................71, 78 Tetrode tube ........................................................77Space wave ........................................................41-4 Thermal agitation noise .................................... 188Speaker response ............................................ 140 Thermionic emission .......................................... 67Specker tweeter ................................................ I 39 Thermocouple meter ..................................724, 726Specific resistance .............................................. 23 Three-phase power supply ...................... 521, 620Speech amplifier construction ............................ 655 Threshold voltage .............................................. 696Speech clipper filter .......................................... 304 Thyratron tube .................................................... 87Speech clipping ...................... 124, 291, 300, 648 Time-base generator .......................................... 139Speech filter, high level .................................... 305 Time constant .............................................. 38, 60Speech waveforms .................... 285, 294, 301, 648 Time sequence keying ........................................400Splatter suppressor .................................. 302, 647 T-match antenna ................................................ 501Sporadic- E propagation ....................................41 8 T-match system ..................................................444Spotter, frequency ..............................................730 TNS limiter ........................................................ 230Spurious emission - spurious pickup ........ 383, 384 Tools, radio ......................................................721Spurious products, mixer .................................. 343 T.P.T.G. oscillator .............................................. 241Square wave - square wave test ........ 58, 61, 62 Trace - cathode ray ................................ 86, 140Stacked antenna - stacked dipole ........498, 468 Tracking capacitor ............................................ 216Standard frequency .......................................... 729 Transceiver, 28 me ........................................... 559Standard pitch .................................................. 135 Transceiver, 50 me ........................................... 552Standing wave ........................................403, 423 Transceiver, VHF .............................................. 578Standing-wave indicator ..................................731 Transconductance ..........................................74, 79Static, corona .................................................... 529 Transducer - pickup .............................. 138, 137Static, wheel ...................................................... 528 Transformation, impedance ....................,............. 63Steering diode .................................................... 103 Transformation ratio .......................................... 62Step-by-step counter ................................ 156, 190 Transformer - ampere-turns ........................ 61, 62Sterba antenna ..................................................469 " Antenna match ........................................ 502Storage time, diode ............................................ 103 Auto .......................................................... 63Stray capacitance .............................................. 21 8 Coupling .................................................. 113Stub match, antenna ........................................ 503 1-F ---------------------------------·····--------------------211Summation voltage ............................................ 197 Linear matching ......................................446Summing amplifier ............................................ 201 Matching- impedance match ..........429, 63Sunspot cycle ....................................................419 Modulator .............................................. 295Superheterodyne receiver .................................. 21 0 Power ...................................................... 709Super-imposition tuning .................................... 553 Vibrator .................................................. 588Super-regenerative detector .............................. 209 Transformerless power supply ............................ 694Supply, a.c.-d.c ................................................. 588 Transient circuit (a.c.) ........................................ 59Suppression, audio ............................................ 306 Transient distortion ------------------------------------------1 35Suppression circuits, porosities ............................ 366 Transient, RC, RL - transient wave ............ 38, 58Suppressor .......................................................... 77 Transistor ...................................................... 90, 92Suppressor modulation ...................................... 292 '' Bias .......................................................... 96Suppressor, splatter .......................................... 302 Code oscillator .......................................... 19Surface wave ....................................................414 Complementary circuit ................................ 97Surge impedance ................................................424 Drift .......................................................... 93Susceptance, del. of ............................................ 52 Equivalent circuit ........................................ 95Sweep frequency ------------------------------------------------140 Heat sink ........................................ 576, 707Switching, transistor action ..............................704 Impedance ................................................ 97SWR bridge ..............................................458, 736 Junction .................................................... 91Symmetry, amplifier .......................................... 612 Mixer ...................................................... 100Synchronization, generator ................................ 173 Multivibrator ............................................ I 02Synchronizing voltage ........................................ 172Oscillator ------------------------------------------------1 0 ISynchronous detection, DSB .............................. 353 Point-contact ............................................ 95Power ........................................................ 99Receiver .................................................. 533TTank capacitance ............................................. 261 SSB equipment ........................................ 574Tank circuit- efficiency ........................ 55, 57, 58 Switching action ......................................704Tank circuit impedance - loading ........ 262, 264 Transistorized mobile supply ..............................703Tank circuit Q .................................................... 128 Transit time ........................................................ 81Technician class amateur license .......................... 12 Transitron oscillator ............................................ 243Teletype, radio .................................................. 326 Transmission, duplex .......................................... 597Television interference ...................................... 371 Transmission line chart ......................................422Temperature coefficient ...................................... 32 Circuits .................................................... 231Ten-A, SSB exciter ............................................ 347 Coaxial ..................................................423Ten-meter transceiver ........................................ 559 Discontinuities ..........................................425Termon-Woodyard amplifier .............................. 298 Impedance ..............................................421Termination, transmission line ............................425 Measurements ................................730, 734Termination, VHF rhombic ..................................487 Resonant - non-resonant ..............424, 421Test equipment ..................................................721 Termination ............................................425Tetrode amplifier, push-pull .............................. 644 VHF ..........................................................478Tetrode modulator ............................................ 647 Transmitter - Construction ...................•.......... 663796
Transmitter - Control circuits .......................... 391 Vacuum - Shot effect ...................................... 123Design .................................................... 356 Space charge ............................................ 89Ground .................................................... 394 Tetrode .................................................... 77Keying .................................................... 395 Thyratron .................................................. 87Oscilloscope, monitor of .......................... 147 Travelling wave ........................................ 84Receiver, VHF .......................................... 597 Triode ...................................................... 72Trapezoidal pattern, oscilloscope ........................ 146 Upper frequency ...................................... 232Trap-type antenna ............................................ 514 Variable mu (~) .................................... 124Travelling wave tube .......................................... 84 VHF .......................................................... 80Travis discriminator ............................................ 322 Voltage regulator .................................... 87Triode amplifier ................................................ 11 0 Voltmeter ........................................ 130, 724Triode mixer ...................................................... 213 V-Antenna ........................................................462Triode power amplifier ...................................... 118 Variable-mu (~) tube ...................................... 124Triode tube ........................................................72 Variable reluctance pickup .............................. 137Tripier, push-pull ................................................ 260 Variac autotransformer ...................................... 390Triplex antenna ................................................475 Vector, sine-wave ................................................43Tropospheric propagation ..................................4 t5 Vehicular noise suppression ................................ 527T-section filter .................................................... 64 Velocity modulation ............................................ 81Tube, vacuum .................................................... 67 Vertical antenna ................................................430Tube, VHF design ............................................ 232 Vertical directivity ..............................................411Tube, water cooled (4W-300B) ........................ 620 VFO, construction .............................................. 676Tubular transmission line ....................................422 VFO, high-stability ............................................ 604Tuned circuit, coaxial ................·........................ 232 VHF Amplifier .................................................... 214Tuned circuit, 1-F ................................................ 218 Antennas ................................................477Tuned circuit, r.f amplifier ................................ 121 Antenna polarization ..............................479Tuned circuit Q .................................................. 216 Antenna relay ........................................478Tuned circuits, selectivity .................................. 341 Bands, characteristics ................................ 14Tuner, antenna ..................................................456 Corner reflector antenna ..........................485Tuning, bandspead .......................................... 217 Coupling, interstage ................................ 665Tuning indicators ................................................ 226 Definition of ............................................477Tuning, inductive ................................................ 617 Discone antenna-ground-plane antenna 481Tuning, super-imposition .................................. 553 Handie-talkie .......................................... 547Turnstile antenna ............................................480 Helical antenna - Horn antenna ....483, 486TVJ filters .................................................. 373, 378 Multi-element antenna ............................488TVI-proof enclosures - openings .......... 724, 725 Neutralization capacitor .......................... 672TVJ suppression ................................................ 375 Nondirectional antenna ..........................48 2TVJ-type filter .................................................... 64 Porosities ................................................ 366Tweeter speaker ................................................ 139 Power Joss ................................................ 663Twin lamp ..........................................................735 Radiation angle - radiation pattern ....478Two-band Marconi antenna ..............................438 Rhombic antenna ......................................486uScreen antenna ....................................... .491Self-neutralization .................................. 664Ultra-linear amplifier ............................. 142, 146 Sleeve antenna ........................................480Unidirectional antenna ...................................... 504 Transceiver .................................... 552, 578Uni-potential cathode ........................................70 Transmission line ....................................478Unity coupling .................................................. 271 Transmitter-receiver .................................. 597Unloaded Q ........................................................ 57 Turnstile antenna ..................................480Wavelength table ....................................479v Vibrator power supply ...................................... 588Vacuum capacitor .............................................. 360 Vibrator, split-reed ............................................ 698Vacuum tube: ...................................................... 67 Video amplifier ........................................ 112, 128Amplification ......................................73, 74 Volt, def. of ........................................................ 22Beam power ..............................................78 Voltage - Amplifier ........................................ 11 0Cathode ray .............................................. 84 Avalanche ................................................ 697Classes- classification .................... 107, 87 Break ...................................................... 203Conductance .............................................. 73 Breakdown ................................................ 3 2Constant-current curve ............................ 127 Decay ........................................................ 40Diode ........................................................?! Divider ................................................ 26, 52Equivalent noise resistance ...................... 123 Divider, phase inverter ............................ 11 7Foreign .................................................... 89 Drop, summation ........................................ 28Gas .......................................................... 87 Gradient .................................................. 39Input loading .......................................... 122 Incident .................................................. 732Klystron .................................................... 81 Injection .................................................. 213Load line .................................................. 74 Initial condition ...................................... 203Magic eye ................................................ 88 Instantaneous ............................................44Magnetron ................................................ 83 Inverse .................................................... 693Mixer ........................................................ 79 Measurement ..................................721, 723Operation .................................................. 75 Multiplier power supply ............................ 695Parameters .............................................. 106 Noise ...................................................... 121Pentode .................................................... 77 Output .................................................... 685Plate resistance ..................................73, 74 '' Reflected ................................................ 732Polarity reversal ........................................76 Regulation ...................................... 686, 687Remote cutoff ............................................78 Regulator tube .................................. 87, 709797
Voltage - Resonant .......................................... 54 Wave-shaping circuits ...................................... 185'' Ripple ...................................................... 686 Weston cell ........................................................ 24Summation .............................................. 197 Wheatstone bridge ............................................ 728Synchronizing .......................................... 141 Wheel static ...................................................... 528Threshold ................................................ 696 Wien-bridge oscillator ............................ 157, 191Voltmeter .......................................................... 722 Williamson amplifier ........................................ 141Voltmeter, A-C V-T ............................................724 Wire, litz ............................................................ 54Voltmeter, diode ..............................................724 Wiring hints ...................................................... 531Voltmeter, vacuum tube .............................. 130, 724 Wiring technique, audio .................................... 142Volt-ohmmeters ..................................................722 Workshop practice - layout ..................720, 729Volume compression .......................................... 648 W8JK antenna ..................................................473WWV transmissions ............................................ 729wWagner ground ................................................729Water cooled tube (4W-300B) ........................ 620X-array antenna ..............................................469Watt, del, of ...................................................... 29Wave - Carrier ................................................ 207yGround ....................................................4 14Harmonic- nonsinusoidal ........................ 58 Yogi, adjustment ............................................... 509Pattern, oscilloscope ................................ 148 Y, (admittance, del. of) .................................... 52Peaked - sawtooth ................................ 59 Yogi antenna - feed systems ................494, 498Sky - space ..........................................4 14 Yogi antenna, VHF ............................................488Square - transient .................................. 58 Yogi, construction .............................................. 506Surface ....................................................4 14 Yoke match ......................................................498Waveform .................................................. 59, 143Waveform, speech ............................ 285, 301, 648zWavelength, del. of ..........................................405 Zeppelin antenna ............................................427Wavelength table, VHF ......................................479 Zero-bias modulator ......................................... 662X$5.00hardboundPREPARES YOU FOR <strong>THE</strong> F.C.C.<strong>RADIO</strong>TELEPHONE LICENSEEXAMINATIONNow, in one convenient volume, complete study-guide questionswith clear, concise answers for preparation for oil U.S.A.commercial radiotelephone operator's license examinations.CONTAINS FOUR ELEMENTS:I. QUESTIONS ON BASIC LAWII. BASIC OPERATING PRACTICEIll. BASIC <strong>RADIO</strong>TELEPHONEI(IIV. ADVANCED <strong>RADIO</strong>TELEPHONEBUY FROM YOUR FAVORITE DISTRIBUTORot above pr~cc or odd 10 °o on d1rcct mod orders toEDITORS and ENGINEERS. ltdSUMMERLAND CALIFORNIABookstores:Order from Boker & Taylor Co., Hillside, N.J.
SURPLUS<strong>RADIO</strong> CONVERSION MANUALSI'IP~ts 1.111111llllSIII llltlLStlrtiS IIIII1111IISIII II!I'ILIN TWO VOLUMESThis set of conversion data has become standard for mostcommonly used items of surplus electronic equipment. Allconversions shown are practical and yield a useful item ofequipment; all have been proven by testing an several units.VOLUME IBC-221 Frequency MeterBC-342 ReceiverBC-348 ReceiverBC-312 ReceiverBC-412 Oscilloscope as atest scope or as a televisionreceiver.BC-645 420 . Me. Transmit·ter I ReceiverBC-453A Series ReceiversBC-457A Series Tran>mittersSCR-522 144-Mc. Transmit·ter /ReceiverTBY Transceiver with XtalControlPE-103A DynamotorBC-1068A V-h·f ReceiverElectronics Surplus IndexCross Index of VT-NumbertubesOrder BookNo. E&E-SMlVOLUME IIARC-5 and BC-454 Receiversfor 28 Me.ARC-5 and BC-457 Tx for22-Mc. MobileART-13 and A TC X millerSurplus Beam RotatingMechanismsSelenium~Rect. Power UnitsHi-Fi Tuner from BC-946BReceiverARC-5 v.h.f TransmittersG0-9 and TBW Xmitters9-W Amplifier from AM-26TA-12B & TA-12C XmittersAVT-112A Aircraft XmitterBC-375 & BC-191 XmittersModel LM Freq. MeterPrimary Power RequirementsChartARB Recr. Diagram OnlyOrder Book No. E&E-SM2$2.50 for either volumeBUY FROM YOUR FAVORITE DISTRIBUTORat above pr1.:e or udd 10 °o on d1rcct mod order~ toEDITORS and ENGINEERS LtdSummerland. CalofornooOrder from Baker & Taylor Co., Hillside, N.J.
WORLD'S <strong>RADIO</strong> TUBES(<strong>RADIO</strong> TUBES VADE MECUM)Characteristics of all existing radio tubesmade in all countries. The world'.- mostauthoritative tub~ book. All types classifiednumerically and alphabetically. To find agiven tube is only a matter of seconds! Alsob~~t~~~en~=~:~~~s. 0 n $5.00WORLD'S EQUIVALENT TUBES(EQUIVALENT TUBES VADE MECUM)All replacement tubes for a given type, bothexact and near-equivalents (with points ofdifference detailed). From normal tubes tothe most advanced comparisons. Also militarytubes of 7 nations, with commercialequivalents. Over 43,900 comparisons! $5.00WORLD'S TELEVISION TUBES(TELEVISION TUBES VADE MECUM)Characteristics of all TV picture and cathoderay tubes. Also special purpose electronictubes. Invaluable to technicians andall specialists in the electronic field. $5.00BUY FROM YOUR FAVORITE DISTRIBUTORat above pncc or add l 0 °o on d~rcct mod order
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