III. Gm-C Filtering - Epublications - Université de Limoges

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Reference [II.20] is based on the same architecture as the latter filter. However, the two transconductances of the gyrator are tunable. This second degree of tunability allows the enhancement of the tuning range. Frequencies from 48 to 780 MHz are now covered at the cost of reduced linearity and higher noise. II.4.c.iii Performances Comparison of Gm-C Filters Many RF bandpass filters, making use of single MOS transistors as transconductors, were previously reported [II.8, II.21]. Operating in the GHz range, they can reach very high Q-factors for a very low power consumption, but with signal distortion and noise. In parallel, the possibility to use an active selectivity for RF front-ends under certain conditions on the LNA (2dB NF and gain limited to 10dB) was shown in [II.22]. The literature also reports on more complex Gm-C filters, used at lower frequencies (40-300MHz), which achieve both interesting RF performances and large tuning range [II.5]. However such filter, with a lowpass structure, does not have an influence on adjacent channels rejection. Finally, as previously described, it has been reported a Gm-C bandpass filter for TV tuners that reaches a good dynamic range while being very low power [II.19]. The performances of these RF filters are summarized in Table 5. Table 5. Gm-C & Gyrator-C RF Filters Ref. [II.8] [II.21] [II.22] [II.5] [II.19] Units Type BPF BPF BPF LPF BPF - Mode single single differential differential differential - Freq. 400-1050 500-1300 1900-3800 50-300 50-300 MHz Q 2 to 80 60 40 6 - NF 8.5 18 14 20 dB IIP3 -15 -26 16 4 dBm Power 51 274 10.8 72 7.6 mW Supply 5 1.8 1.8 1.2 V Techno 350 350 180 180 130 nm II.4.d Rm-C Filtering Rm based filters are very similar to the Gm-cells based ones. As explained previously, a Gm-cell converts a voltage into a current with a certain gain gm whereas a trans-resistance amplifier converts a current into a voltage with a certain resistive gain Rm, called “transresistance”. A capacitors associated to this trans-resistance amplifier creates a derivator, as depicted in Figure 89, which is essential to synthesize a filter transfer function. Indeed, this leads to a derivative behaviour since: Vout = jRmCω . (III.69) ΔV in - 72 -
Vin Rm - 73 - C Vout Figure 89. Derivator using an Rm amplifier Rm-C filters are rarely used above 10MHz because of their frequency limitation. Only two publications were classified in APPENDIX C. In [II.23] Y. Cheng et al present an interesting structure, which consists in a CMOS inverted-based Rm-C amplifier that is connected to a capacitor in order to form an Rm-C biquad. A Q-enhancement circuit is added which also helps to adjust the filter operating frequency and gain. The entire circuit is illustrated in Figure 90. The Rm amplifier is realized with a CMOS inverter with feedback and current-source load. The Q-enhancement is realized thanks to a negative resistance which generates a higher gain, resulting in a higher Q-factor: the higher the gain, the higher the extra-gain given by the positive feedback. Figure 90. Reported Rm-C filter As far as the performances are concerned, this structure is tunable between 41 and 178MHz with 20 dB gain. It also achieves a noise figure of 15 dB and an IIP3 of -5 dBm. Further details are given in APPENDIX C.
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UNIVERSITE DE LIMOGES ECOLE DOCTORA
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Acknowledgments En préambule à ce
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Summary ACKNOWLEDGMENTS ...........
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V. A PERSPECTIVE FOR FUTURE DEVELOP
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French Introduction Les signaux TV
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I. Introduction to TV tuners and to
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I.1.b.ii Digital modulations In dig
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makes the covering of a large terri
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I.1.c.iii Terrestrial spectrum Firs
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I.1.e Broadband Reception Systems F
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I.2.b TV Tuner High Performance Spe
Page 25 and 26: I.2.b.ii High linearity Although co
Page 27 and 28: I.3 RF Filter Specifications I.3.a
Page 29 and 30: 27. This leads to the downconversio
Page 31 and 32: These adjacent channel rejection re
Page 33 and 34: Above 870MHz, there are no more ter
Page 35 and 36: Newest silicon tuners generations,
Page 37 and 38: I.6 References [I.1] B. Razavi, RF
Page 39 and 40: II. Challenges of RF Selectivity Fo
Page 41 and 42: Figure 39. H3 and N+5 rejections of
Page 43 and 44: Figure 43. Second order Low-pass Fi
Page 45 and 46: Figure 46 illustrates the transfer
Page 47 and 48: H HPF s ( s) = s 1 + ω This corres
Page 49 and 50: Topologies comparison for a same Q-
Page 51 and 52: Figure 56. Low-pass to Notch Transf
Page 53 and 54: II.2 Passive LC Second Order Bandpa
Page 55 and 56: A possible solution is to split the
Page 57 and 58: II.2.c Second Order Bandpass Filter
Page 59 and 60: Figure 69. Band Switching Figure 70
Page 61 and 62: Another paper [II.7] provides a FOM
Page 63 and 64: A parallel self resonating circuit
Page 65 and 66: This gives the following values for
Page 67 and 68: Vb - 63 - M1 M2 Zin Figure 76. Anot
Page 69 and 70: when given, reported noise figures
Page 71 and 72: In all publications dedicated to Gm
Page 73 and 74: II.4.c Second Order Bandpass Filter
Page 75: Figure 87. Biquad Schematic The Gm-
Page 79 and 80: Figure 93 depicts the RF performanc
Page 81 and 82: A second advantage of advanced BiCM
Page 83 and 84: II.7 Conclusion As a conclusion, th
Page 85 and 86: [II.13] C. Andriesei, L. Goras, and
Page 87 and 88: III.I Theoretical Study III. Gm-C F
Page 89 and 90: g V m3 in III.1.b Linearity Conside
Page 91 and 92: Assuming an ideal input transconduc
Page 93 and 94: Figure 102. Linearity of the filter
Page 95 and 96: Furthermore, a Gm-C circuit is not
Page 97 and 98: In terms of noise, the Flicker nois
Page 99 and 100: Table 7 summarizes the specificatio
Page 101 and 102: Idiff (A) levels reached with this
Page 103 and 104: III.2.e Unbalanced Differential Pai
Page 105 and 106: Hence, the combination of the expon
Page 107 and 108: Technique Current Increase Source D
Page 109 and 110: Using this structure and considerin
Page 111 and 112: The dimensioning of the transistors
Page 113 and 114: III.3.c Gm-cells with MGTR III.3.c.
Page 115 and 116: Furthermore, the high sensitivity t
Page 117 and 118: Figure 136. Filter in-band IIP3 ver
Page 119 and 120: III.4 Comparison of the filters III
Page 121 and 122: III.5 Conclusion Once the Gm-C seco
Page 123 and 124: IV. Rauch Filtering IV.1 Sallen-Key
Page 125 and 126: Thus, an equation linking Q and Gai
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IV.1.c.ii Proposed positive feedbac
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For both Rauch and Sallen-Key filte
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The high sensitivity to passive com
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Furthermore, it should also be take
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IV.2.b Innovative Implementation of
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esulting gain K, constituted of the
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Table 25. Sum-up of the OA specific
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Figure 153. OA voltage gain versus
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Filtering in the mirror, by means o
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However, this feedback makes the fi
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Figure 160. OA Gain and phase versu
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IV.3.c Implemented Filter and Test
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Figure 165 depicts the layout of th
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Figure 169. PVT variations of the f
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IV.4 Rauch Filter Performances: Mea
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Figure 178. IIP3 versus input power
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IV.4.c Performances Sum-up The filt
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Table 30. Frequency Limitations of
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IV.7 References [IV.1] Y. Tsividis
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V. A Perspective for Future Develop
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V.2 4-path Filter Simulations V.2.a
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The frequency tunability is ensured
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V.3 State-of-the-Art V.3.a 4-path F
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V.4 Conclusion In this chapter, it
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IEEE International, 2009, pp. 222-2
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RF selectivity challenges In this t
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The positive feedback Rauch filter
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It is worth highlighting that FOM1
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étudiée. Il s’avère que la lin
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It is worth noticing that S0 only d
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A.3 Signal-to-Noise Ratio These dif
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A.5 Friis’ Formula In case of mul
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A.7 References [A.1] Friis, H.T., N
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To estimate the influence of the di
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Now, let’s have a look to the cub
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Figure 200. Third order intercept p
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B.2 RF Filter Linearity Measurement
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B.2.c IIP2 measurement To measure t
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- 199 -
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C.1 Gyrator-C filtering year Ref. f
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C.2 Gm-C Filtering year Ref. f0 (MH
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year C.3 Rm-C filtering Ref. f0 (MH
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C.5 References C.5.a Gyrator-C filt
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[C.23] J. De Lima and C. Dualibe,
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- 211 -
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this gives: I I I md d1 d 2 = I I =
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D.2 MOS Degenerated Common-Source C
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⎛ I s ⎞ Vout = Vin + U T ln ⎜
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Computing, this leads to: V out + v
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- 221 -
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- 223 -
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FIGURE 54. H3 AND N+5 REJECTIONS AC
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FIGURE 168. PVT VARIATIONS OF THE F
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