III. Gm-C Filtering - Epublications - Université de Limoges

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Figure 58. Double Notch Filtering II.1.e Comparison of the Topologies The previously studied topologies are compared in Table 4. The table summarizes the different advantages and drawbacks of each solution. Table 4. Comparison of the Topologies Topology Advantages Drawbacks Low-pass Harmonics rejection for order>3 Adjacent channels rejection above fc Bandpass Double notch Selectivity depends on Q Harmonics rejection for order>2 Adjacent channel rejection for order>2 Adjacent channel rejection Frequency tuning - 48 - Harmonics rejection From this, the bandpass filtering can be considered as the most appropriate solution since it is able to attenuate both harmonics and adjacent channels. The Q-factor also appears as a strong advantage in order to reject interferers in a controlled way. Moreover, a second order bandpass filter is sufficient to reach the specifications given in introduction, while higher order filters are required for the low-pass. This enables the minimization of reactive components. That is why the second order bandpass topology has been chosen to handle the RF selectivity.
II.2 Passive LC Second Order Bandpass Filtering II.2.a Frequency Tunability As explained before, cable and terrestrial TV bands cover a range from 45MHz up to 1002MHz. Given f0 formula, some relations and orders of magnitude for L and C can be deduced. It has to be kept in mind that usual integrated capacitances in microelectronics vary between 10fF and 100pF. According to the technology, inductors may be integrated on silicon as well. However, above some tens of nano-Henrys, a spiral inductor becomes too large and so very costly in terms of silicon on-chip space. Hence, it is not efficient to integrate 50nH or more, which are values needed to reach the bottom of the TV bands in VHF low. To achieve a frequency sweep of the central frequency by steps smaller than a channel width of 6MHz as explained in introduction, the value either of the inductance or of the capacitance, or even both of them, has to be changed gradually. Technically, it is not optimum to switch inductors. Indeed, parallel switching of inductors, as described in Figure 59.a, requires even higher inductance values since N parallel L inductances L are equivalent to a value of inductance of . The addition of inductors in N series, depicted in Figure 59.b, is efficient in terms of inductance values. However, a large difference of quality factor happens between Lmax and Lmin. Lmin is made of a small inductance value in series with several switches Ron in series, while Lmax is constituted of a large inductance value with no switch Ron in series. Figure 59. a) parallel inductors switching, b) series inductors switching Capacitance values can be tuned by different means. The two most popular techniques are the varactor (Figure 60.a) and the capacitor bank (Figure 61.a). The varactor, which actually is a reverse-biased varicap diode with a voltage Vb, has a capacitance Cd which depends on the voltage bias, as it may be seen in Figure 60.b. Figure 60. a) Varactor symbol, b) C(V) curve of a varactor - 49 -
Page 1: UNIVERSITE DE LIMOGES ECOLE DOCTORA
Page 5 and 6: Acknowledgments En préambule à ce
Page 7 and 8: Summary ACKNOWLEDGMENTS ...........
Page 9 and 10: V. A PERSPECTIVE FOR FUTURE DEVELOP
Page 11 and 12: French Introduction Les signaux TV
Page 13 and 14: I. Introduction to TV tuners and to
Page 15 and 16: I.1.b.ii Digital modulations In dig
Page 17 and 18: makes the covering of a large terri
Page 19 and 20: I.1.c.iii Terrestrial spectrum Firs
Page 21 and 22: I.1.e Broadband Reception Systems F
Page 23 and 24: I.2.b TV Tuner High Performance Spe
Page 25 and 26: I.2.b.ii High linearity Although co
Page 27 and 28: I.3 RF Filter Specifications I.3.a
Page 29 and 30: 27. This leads to the downconversio
Page 31 and 32: These adjacent channel rejection re
Page 33 and 34: Above 870MHz, there are no more ter
Page 35 and 36: Newest silicon tuners generations,
Page 37 and 38: I.6 References [I.1] B. Razavi, RF
Page 39 and 40: II. Challenges of RF Selectivity Fo
Page 41 and 42: Figure 39. H3 and N+5 rejections of
Page 43 and 44: Figure 43. Second order Low-pass Fi
Page 45 and 46: Figure 46 illustrates the transfer
Page 47 and 48: H HPF s ( s) = s 1 + ω This corres
Page 49 and 50: Topologies comparison for a same Q-
Page 51: Figure 56. Low-pass to Notch Transf
Page 55 and 56: A possible solution is to split the
Page 57 and 58: II.2.c Second Order Bandpass Filter
Page 59 and 60: Figure 69. Band Switching Figure 70
Page 61 and 62: Another paper [II.7] provides a FOM
Page 63 and 64: A parallel self resonating circuit
Page 65 and 66: This gives the following values for
Page 67 and 68: Vb - 63 - M1 M2 Zin Figure 76. Anot
Page 69 and 70: when given, reported noise figures
Page 71 and 72: In all publications dedicated to Gm
Page 73 and 74: II.4.c Second Order Bandpass Filter
Page 75 and 76: Figure 87. Biquad Schematic The Gm-
Page 77 and 78: Vin Rm - 73 - C Vout Figure 89. Der
Page 79 and 80: Figure 93 depicts the RF performanc
Page 81 and 82: A second advantage of advanced BiCM
Page 83 and 84: II.7 Conclusion As a conclusion, th
Page 85 and 86: [II.13] C. Andriesei, L. Goras, and
Page 87 and 88: III.I Theoretical Study III. Gm-C F
Page 89 and 90: g V m3 in III.1.b Linearity Conside
Page 91 and 92: Assuming an ideal input transconduc
Page 93 and 94: Figure 102. Linearity of the filter
Page 95 and 96: Furthermore, a Gm-C circuit is not
Page 97 and 98: In terms of noise, the Flicker nois
Page 99 and 100: Table 7 summarizes the specificatio
Page 101 and 102: Idiff (A) levels reached with this
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III.2.e Unbalanced Differential Pai
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Hence, the combination of the expon
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Technique Current Increase Source D
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Using this structure and considerin
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The dimensioning of the transistors
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III.3.c Gm-cells with MGTR III.3.c.
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Furthermore, the high sensitivity t
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Figure 136. Filter in-band IIP3 ver
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III.4 Comparison of the filters III
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III.5 Conclusion Once the Gm-C seco
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IV. Rauch Filtering IV.1 Sallen-Key
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Thus, an equation linking Q and Gai
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IV.1.c.ii Proposed positive feedbac
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For both Rauch and Sallen-Key filte
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The high sensitivity to passive com
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Furthermore, it should also be take
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IV.2.b Innovative Implementation of
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esulting gain K, constituted of the
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Table 25. Sum-up of the OA specific
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Figure 153. OA voltage gain versus
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Filtering in the mirror, by means o
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However, this feedback makes the fi
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Figure 160. OA Gain and phase versu
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IV.3.c Implemented Filter and Test
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Figure 165 depicts the layout of th
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Figure 169. PVT variations of the f
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IV.4 Rauch Filter Performances: Mea
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Figure 178. IIP3 versus input power
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IV.4.c Performances Sum-up The filt
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Table 30. Frequency Limitations of
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IV.7 References [IV.1] Y. Tsividis
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V. A Perspective for Future Develop
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V.2 4-path Filter Simulations V.2.a
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The frequency tunability is ensured
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V.3 State-of-the-Art V.3.a 4-path F
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V.4 Conclusion In this chapter, it
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IEEE International, 2009, pp. 222-2
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RF selectivity challenges In this t
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The positive feedback Rauch filter
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It is worth highlighting that FOM1
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étudiée. Il s’avère que la lin
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It is worth noticing that S0 only d
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A.3 Signal-to-Noise Ratio These dif
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A.5 Friis’ Formula In case of mul
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A.7 References [A.1] Friis, H.T., N
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To estimate the influence of the di
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Now, let’s have a look to the cub
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Figure 200. Third order intercept p
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B.2 RF Filter Linearity Measurement
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B.2.c IIP2 measurement To measure t
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C.1 Gyrator-C filtering year Ref. f
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C.2 Gm-C Filtering year Ref. f0 (MH
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year C.3 Rm-C filtering Ref. f0 (MH
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C.5 References C.5.a Gyrator-C filt
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[C.23] J. De Lima and C. Dualibe,
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- 211 -
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this gives: I I I md d1 d 2 = I I =
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D.2 MOS Degenerated Common-Source C
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⎛ I s ⎞ Vout = Vin + U T ln ⎜
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Computing, this leads to: V out + v
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- 223 -
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FIGURE 54. H3 AND N+5 REJECTIONS AC
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FIGURE 168. PVT VARIATIONS OF THE F
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