III. Gm-C Filtering - Epublications - Université de Limoges

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Figure 21 illustrates the uncoded BER, which stands for Bit Error Rate, versus the Es/N0, which is the energy of a symbol Es normalized to the noise power N0, in the case of a 16 QAM modulation. Assuming that the symbol and the sampling periods are equal, Es/N0 then corresponds to the SNR. The required BER depends on the communication system. However, this figure shows that BER strongly depends on the SNR. To reach bit error rates of 10 -4 or 10 -5 , SNR of 18 to 20dB are needed. Besides, it is worth pointing out that a 1dB variation at these SNRs leads to a factor 10 variation on the BER. Hence, noise is a very critical issue in our applications. Figure 21. BER versus Es/N0 in case of a 16QAM modulation As explained in APPENDIX A, the noise factor (or the noise figure) is a way to measure the degradation of the SNR by a system between its input and its output. Applying Friis’ formula to a receiver system made of a LNA of gain GLNA and of noise factor FLNA, and considering the noise factor of the rest of the receiver chain Frest, it can be written that: F F rest receiver = FLNA + . (I.5) G LNA Thus, the receiver noise factor is strongly dependent on the noise and the gain of the first stage LNA. To get a very low Freceiver, it is required to have a high gain with a very low noise factor, and this stage is performed by an LNA. When FLNA is low while GLNA is high, the noise constraints over the rest of the receiver chain, described by the noise factor Frest, can be relaxed. - 20 - −1
I.2.b.ii High linearity Although components such as amplifiers are considered as linear elements for small signals, they actually have non-linear characteristics. When dealing with large signals, distortion phenomena like compression or intermodulation occur and may strongly degrade the wanted signal. The theory about distortions and its effects are discussed more extensively in APPENDIX B. Figure 22 shows the degradation of a since wave due to second order distortion (a) and to third and fifth order distortion (b). Figure 22. Second order (a) and Third and Fifth order (b) distortion of a sine wave References [I.5 and I.6], written by Charles Rhodes, highlight how crucial the linearity of the TV receivers is. It demonstrates that nearly all interferences are due to third order nonlinearity of the receiver when overloaded. Indeed, the receiver is then subject to multiple distortion products which may fall in the desired channel, as described in Figure 23. Thus, like noise, intermodulation products falling can mask and degrade the wanted signal. To describe the degradation of the signal by noise as well as non-linearities, a factor similar to the SNR is often used. It is called SNIR, standing for Signal to Noise plus Interference Ratio. Figure 23. Intermodulation products interfering with a desired signal - 21 -
Page 1: UNIVERSITE DE LIMOGES ECOLE DOCTORA
Page 5 and 6: Acknowledgments En préambule à ce
Page 7 and 8: Summary ACKNOWLEDGMENTS ...........
Page 9 and 10: V. A PERSPECTIVE FOR FUTURE DEVELOP
Page 11 and 12: French Introduction Les signaux TV
Page 13 and 14: I. Introduction to TV tuners and to
Page 15 and 16: I.1.b.ii Digital modulations In dig
Page 17 and 18: makes the covering of a large terri
Page 19 and 20: I.1.c.iii Terrestrial spectrum Firs
Page 21 and 22: I.1.e Broadband Reception Systems F
Page 23: I.2.b TV Tuner High Performance Spe
Page 27 and 28: I.3 RF Filter Specifications I.3.a
Page 29 and 30: 27. This leads to the downconversio
Page 31 and 32: These adjacent channel rejection re
Page 33 and 34: Above 870MHz, there are no more ter
Page 35 and 36: Newest silicon tuners generations,
Page 37 and 38: I.6 References [I.1] B. Razavi, RF
Page 39 and 40: II. Challenges of RF Selectivity Fo
Page 41 and 42: Figure 39. H3 and N+5 rejections of
Page 43 and 44: Figure 43. Second order Low-pass Fi
Page 45 and 46: Figure 46 illustrates the transfer
Page 47 and 48: H HPF s ( s) = s 1 + ω This corres
Page 49 and 50: Topologies comparison for a same Q-
Page 51 and 52: Figure 56. Low-pass to Notch Transf
Page 53 and 54: II.2 Passive LC Second Order Bandpa
Page 55 and 56: A possible solution is to split the
Page 57 and 58: II.2.c Second Order Bandpass Filter
Page 59 and 60: Figure 69. Band Switching Figure 70
Page 61 and 62: Another paper [II.7] provides a FOM
Page 63 and 64: A parallel self resonating circuit
Page 65 and 66: This gives the following values for
Page 67 and 68: Vb - 63 - M1 M2 Zin Figure 76. Anot
Page 69 and 70: when given, reported noise figures
Page 71 and 72: In all publications dedicated to Gm
Page 73 and 74: II.4.c Second Order Bandpass Filter
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Figure 87. Biquad Schematic The Gm-
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Vin Rm - 73 - C Vout Figure 89. Der
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Figure 93 depicts the RF performanc
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A second advantage of advanced BiCM
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II.7 Conclusion As a conclusion, th
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[II.13] C. Andriesei, L. Goras, and
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III.I Theoretical Study III. Gm-C F
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g V m3 in III.1.b Linearity Conside
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Assuming an ideal input transconduc
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Figure 102. Linearity of the filter
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Furthermore, a Gm-C circuit is not
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In terms of noise, the Flicker nois
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Table 7 summarizes the specificatio
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Idiff (A) levels reached with this
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III.2.e Unbalanced Differential Pai
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Hence, the combination of the expon
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Technique Current Increase Source D
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Using this structure and considerin
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The dimensioning of the transistors
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III.3.c Gm-cells with MGTR III.3.c.
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Furthermore, the high sensitivity t
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Figure 136. Filter in-band IIP3 ver
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III.4 Comparison of the filters III
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III.5 Conclusion Once the Gm-C seco
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IV. Rauch Filtering IV.1 Sallen-Key
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Thus, an equation linking Q and Gai
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IV.1.c.ii Proposed positive feedbac
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For both Rauch and Sallen-Key filte
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The high sensitivity to passive com
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Furthermore, it should also be take
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IV.2.b Innovative Implementation of
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esulting gain K, constituted of the
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Table 25. Sum-up of the OA specific
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Figure 153. OA voltage gain versus
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Filtering in the mirror, by means o
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However, this feedback makes the fi
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Figure 160. OA Gain and phase versu
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IV.3.c Implemented Filter and Test
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Figure 165 depicts the layout of th
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Figure 169. PVT variations of the f
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IV.4 Rauch Filter Performances: Mea
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Figure 178. IIP3 versus input power
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IV.4.c Performances Sum-up The filt
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Table 30. Frequency Limitations of
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IV.7 References [IV.1] Y. Tsividis
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V. A Perspective for Future Develop
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V.2 4-path Filter Simulations V.2.a
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The frequency tunability is ensured
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V.3 State-of-the-Art V.3.a 4-path F
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V.4 Conclusion In this chapter, it
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IEEE International, 2009, pp. 222-2
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RF selectivity challenges In this t
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The positive feedback Rauch filter
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It is worth highlighting that FOM1
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étudiée. Il s’avère que la lin
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It is worth noticing that S0 only d
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A.3 Signal-to-Noise Ratio These dif
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A.5 Friis’ Formula In case of mul
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A.7 References [A.1] Friis, H.T., N
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To estimate the influence of the di
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Now, let’s have a look to the cub
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Figure 200. Third order intercept p
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B.2 RF Filter Linearity Measurement
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B.2.c IIP2 measurement To measure t
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C.1 Gyrator-C filtering year Ref. f
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C.2 Gm-C Filtering year Ref. f0 (MH
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year C.3 Rm-C filtering Ref. f0 (MH
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C.5 References C.5.a Gyrator-C filt
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[C.23] J. De Lima and C. Dualibe,
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this gives: I I I md d1 d 2 = I I =
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D.2 MOS Degenerated Common-Source C
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⎛ I s ⎞ Vout = Vin + U T ln ⎜
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Computing, this leads to: V out + v
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FIGURE 54. H3 AND N+5 REJECTIONS AC
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FIGURE 168. PVT VARIATIONS OF THE F
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