III. Gm-C Filtering - Epublications - Université de Limoges

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I.2 TV Receivers Architecture & Specifications I.2.a TV Tuner Architecture As said before and as in most receivers, the signal is first amplified by a LNA before being mixed, as shown in Figure 17. The mixing stage actually consists in multiplying two signals. Let’s consider the RF signal and the LO (which stands for Local Oscillator) one, assuming that both are sine wave signals, mixed by a linear system: 1 cos ( ω RFt ) cos( ωLOt ) = ( cos( ( ωRF − ωLO ) t) + cos( ( ωRF + ωLO ) t) ) . (I.4) 2 Hence, the RF signal is actually translated to lower frequency by the LO signal. Two new frequencies then come out: - ωRF − ωLO , called IF frequency, which is the useful frequency; - ω RF + ωLO , called image frequency which has to be rejected. Depending on the value of the intermediate frequency (IF) fIF =fRF-fLO, receivers are called IF, low-IF or even ZIF (which stands for Zero-IF) when the signal is directly converted to baseband. Figure 17. RF receiver front-end simplified architecture As it may be seen on Figure 18, once the signal is converted to IF, the wanted channel is filtered. It is a very selective and sharp filter so that only this channel remains among the entire received spectrum. Then, an analog-to-digital conversion of the channel is performed by an ADC in order to provide the signal to be demodulated. Figure 18. TV receiver architecture One might wonder why this tuner architecture is necessary. Indeed, a simpler concept would be designed with an analog-to-digital converter (ADC) much closer to the antenna. However, such an ADC would require very high sampling rate and resolution to be designed, since two billion samples per second are needed [I.2] to convert up to 1GHz. Though this type of architecture is emerging to handle cable TV signals [I.4], the presence of very high power differences between channels still prevents from using it for terrestrial applications. - 18 -
I.2.b TV Tuner High Performance Specifications I.2.b.i Low noise As explained in APPENDIX A, noise is a random fluctuation of energy which can be found in all electronic circuits. In the case of an RF receiver, a high noise level results in an undesirable signal that masks or degrades the useful signal, as illustrated on Figure 19 by a noisy sine wave. Figure 19. Noise added to a sine signal A noisy reception chain makes the wanted signal be degraded. Hence, it becomes more difficult to finally demodulate in good conditions and the number of errors may increase. To decrease demodulation errors, noise has to be as low as possible compared to the wanted signal. This is expressed by a signal-to-noise ratio (SNR) that should be high enough at the input of the demodulator. This is one of the specifications for the TV tuner. On Figure 20 are shown two 16QAM constellations drawn for two different SNR. It clearly shows that noise random fluctuations on the desired signal can lead to decision errors on bit values. Figure 20. 16QAM constellations for two different SNR - 19 -
Page 1: UNIVERSITE DE LIMOGES ECOLE DOCTORA
Page 5 and 6: Acknowledgments En préambule à ce
Page 7 and 8: Summary ACKNOWLEDGMENTS ...........
Page 9 and 10: V. A PERSPECTIVE FOR FUTURE DEVELOP
Page 11 and 12: French Introduction Les signaux TV
Page 13 and 14: I. Introduction to TV tuners and to
Page 15 and 16: I.1.b.ii Digital modulations In dig
Page 17 and 18: makes the covering of a large terri
Page 19 and 20: I.1.c.iii Terrestrial spectrum Firs
Page 21: I.1.e Broadband Reception Systems F
Page 25 and 26: I.2.b.ii High linearity Although co
Page 27 and 28: I.3 RF Filter Specifications I.3.a
Page 29 and 30: 27. This leads to the downconversio
Page 31 and 32: These adjacent channel rejection re
Page 33 and 34: Above 870MHz, there are no more ter
Page 35 and 36: Newest silicon tuners generations,
Page 37 and 38: I.6 References [I.1] B. Razavi, RF
Page 39 and 40: II. Challenges of RF Selectivity Fo
Page 41 and 42: Figure 39. H3 and N+5 rejections of
Page 43 and 44: Figure 43. Second order Low-pass Fi
Page 45 and 46: Figure 46 illustrates the transfer
Page 47 and 48: H HPF s ( s) = s 1 + ω This corres
Page 49 and 50: Topologies comparison for a same Q-
Page 51 and 52: Figure 56. Low-pass to Notch Transf
Page 53 and 54: II.2 Passive LC Second Order Bandpa
Page 55 and 56: A possible solution is to split the
Page 57 and 58: II.2.c Second Order Bandpass Filter
Page 59 and 60: Figure 69. Band Switching Figure 70
Page 61 and 62: Another paper [II.7] provides a FOM
Page 63 and 64: A parallel self resonating circuit
Page 65 and 66: This gives the following values for
Page 67 and 68: Vb - 63 - M1 M2 Zin Figure 76. Anot
Page 69 and 70: when given, reported noise figures
Page 71 and 72: In all publications dedicated to Gm
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II.4.c Second Order Bandpass Filter
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Figure 87. Biquad Schematic The Gm-
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Vin Rm - 73 - C Vout Figure 89. Der
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Figure 93 depicts the RF performanc
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A second advantage of advanced BiCM
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II.7 Conclusion As a conclusion, th
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[II.13] C. Andriesei, L. Goras, and
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III.I Theoretical Study III. Gm-C F
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g V m3 in III.1.b Linearity Conside
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Assuming an ideal input transconduc
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Figure 102. Linearity of the filter
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Furthermore, a Gm-C circuit is not
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In terms of noise, the Flicker nois
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Table 7 summarizes the specificatio
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Idiff (A) levels reached with this
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III.2.e Unbalanced Differential Pai
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Hence, the combination of the expon
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Technique Current Increase Source D
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Using this structure and considerin
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The dimensioning of the transistors
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III.3.c Gm-cells with MGTR III.3.c.
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Furthermore, the high sensitivity t
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Figure 136. Filter in-band IIP3 ver
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III.4 Comparison of the filters III
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III.5 Conclusion Once the Gm-C seco
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IV. Rauch Filtering IV.1 Sallen-Key
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Thus, an equation linking Q and Gai
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IV.1.c.ii Proposed positive feedbac
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For both Rauch and Sallen-Key filte
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The high sensitivity to passive com
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Furthermore, it should also be take
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IV.2.b Innovative Implementation of
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esulting gain K, constituted of the
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Table 25. Sum-up of the OA specific
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Figure 153. OA voltage gain versus
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Filtering in the mirror, by means o
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However, this feedback makes the fi
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Figure 160. OA Gain and phase versu
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IV.3.c Implemented Filter and Test
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Figure 165 depicts the layout of th
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Figure 169. PVT variations of the f
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IV.4 Rauch Filter Performances: Mea
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Figure 178. IIP3 versus input power
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IV.4.c Performances Sum-up The filt
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Table 30. Frequency Limitations of
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IV.7 References [IV.1] Y. Tsividis
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V. A Perspective for Future Develop
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V.2 4-path Filter Simulations V.2.a
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The frequency tunability is ensured
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V.3 State-of-the-Art V.3.a 4-path F
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V.4 Conclusion In this chapter, it
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IEEE International, 2009, pp. 222-2
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RF selectivity challenges In this t
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The positive feedback Rauch filter
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It is worth highlighting that FOM1
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étudiée. Il s’avère que la lin
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It is worth noticing that S0 only d
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A.3 Signal-to-Noise Ratio These dif
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A.5 Friis’ Formula In case of mul
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A.7 References [A.1] Friis, H.T., N
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To estimate the influence of the di
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Now, let’s have a look to the cub
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Figure 200. Third order intercept p
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B.2 RF Filter Linearity Measurement
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B.2.c IIP2 measurement To measure t
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C.1 Gyrator-C filtering year Ref. f
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C.2 Gm-C Filtering year Ref. f0 (MH
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year C.3 Rm-C filtering Ref. f0 (MH
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C.5 References C.5.a Gyrator-C filt
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[C.23] J. De Lima and C. Dualibe,
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this gives: I I I md d1 d 2 = I I =
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D.2 MOS Degenerated Common-Source C
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⎛ I s ⎞ Vout = Vin + U T ln ⎜
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Computing, this leads to: V out + v
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FIGURE 54. H3 AND N+5 REJECTIONS AC
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FIGURE 168. PVT VARIATIONS OF THE F
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