Power Switch/Amplifier - Closed

Introduction
The CA3094 unique monolithic programmable power
switch/amplifier IC consists of a high-gain preamplifier driving
a power-output amplifier stage. It can deliver average power of
3 watts of peak power of 10W to an external load, and can be
operated from either a single or dual power supply. This Note
briefly describes the characteristics of the CA3094, and
illustrates its use in the following circuit applications:
• Class A Instrumentations and Power Amplifiers
• Class A Driver-Amplifier for Complementary Power
Transistors
• Wide Frequency Range Power Multivibrators
• Current- or Voltage Controlled Oscillators
• Comparators (Threshold Detectors)
• Voltage Regulators
• Analog Timers (Long Time Delays)
• Alarm Systems
• Motor-Speed Controllers
• Thyristor Firing Circuits
• Battery Charger Regulator Circuit
• Ground Fault Interrupter Circuits
DIFF.
VOLTAGE
INPUT
2
DIFF.
VOLTAGE
INPUT
AMPLIFIER
BIAS INPUT
I ABC
Q4
3
Q1
D2
Q2
Q5
4-165
D3
5 Q3
D1
Some Applications of a Programmable
Power Switch/Amplifier
Application Note April 1994
AN6048.1
Authors: L.R. Campbell and H.A. Wittlinger
EXTERNAL FREQUENCY
COMPENSATION OR INHIBIT INPUT
Q7
Q6
D4
Q11
Q9
D5
Q8
Q10
1
R1 47kΩ
D6
4
Circuit Description
FIGURE 1. CA3094 CIRCUIT SCHEMATIC DIAGRAM
7
V-
Q12
The CA3094 series of devices offers a unique combination
of circuit flexibility and power-handling capability. Although
these monolithic ICs dissipate only a few microwatts when
quiescent, they have a high current-output capability (100mA
average, 300mA peak) in the active state, and the premiumgrade
devices can operate at supply voltages up to 44V.
Figure 1 shows a schematic diagram of the CA3094. The
portion of the circuit preceding transistors Q12 and Q13 is
the preamplifier section and is generically similar to that of
the CA3080 Operational Transconductance Amplifier (OTA)
[1, 2]. The CA3094 circuits can be gain-programmed by
either digital and/or analog signals applied to a separate
Amplifier-Bias-Current (I ABC ) terminal (No. 5 in Figure 1) to
control circuit sensitivity. Response of the amplifier is
essentially liner as a function of the current at terminal 5.
This additional signal input “port” provides added flexibility in
may applications. Thus, the output of the amplifier is a
function of input signals applied differentially at terminals 2
and 3 and/or in a singly-ended configuration at terminal 5.
The output portion of the monolithic circuit in the CA3094
consists of a Darling connected transistor pair with access
provided to both the collector and emitter terminals to
provide capability to “sink” and/or “source” current.
The CA3094 series of circuits consists of six types that differ
only in voltage-handling capability and package options, as
shown below; other electrical characteristics are identical.
V+
R 1
2kΩ
8
“SINK”
OUTPUT
Q13
6
“SOURCE”
(DRIVE)
OUTPUT
OUTPUT
MODE
OUTPUT
TERM
INPUTS
INV NON-INV
“Source” 6 2 3
“Sink” 8 3 2
1-800-4-HARRIS or 407-727-9207 | Copyright © Harris Corporation 1994

PACKAGE OPTIONS MAXIMUM VOLTAGE RATING
CA3094S, CA3094T 24V
CA3094AS, CA3094AT 36V
CA3094BS, CA3094BT 44V
The suffix “S” indicates circuits packaged in TO-5 enclosures
with leads formed to an 8 lead dual in line configuration (0.1
inch pin spacing). The suffix “T” indicates circuits packaged
in 8 lead TO-5 enclosures with straight leads. The generic
CA3094 type designation is used throughout the Note.
Class A Instrumentation Amplifiers
One of the more difficult instrumentation problems frequently
encountered is the conversion of a differential input signal to
a single-ended output signal. Although this conversion can
be accomplished in a straightforward design through the use
of classical op amps, the stringent matching requirements of
resistor ratios in feedback networks make the conversion
particularly difficult from a practical standpoint. Because the
gain of the preamplifier section in the CA3094 can be
defined as the product of the transconductance and the load
resistance (g m R L ), feedback is not needed to obtain
predictable open-loop gain performance. Figure 2 shows the
CA3094 in this basic type of circuit.
DIFFERENTIAL
THERMOCOUPLE
10kΩ
100Ω
IABC 260μA
110kΩ
7
5
2kΩ
2kΩ
8
1W
3 +
RO 2kΩ
EDIFF CA3094A
≤13kΩ
2 -
6
IO IO 10kΩ RL 36kΩ
1
4
RE 560Ω
NOTES:
-30V
1. Preamp gain (AV ) = gm RL = (5) (10 -3 ) (36) (10 3 ) = 180
(Output at terminal 1).
2. For linear operation: Differential Input ≤| ±26mV |
(With approx. 1% deviation from linearity).
3. Output voltage (EO ) = AV (±eDIFF ) = (180) (±26mV) = ±4.7V
4.7V
( g
4.
m
R
L
) ( e
DIFF
)
IO ≈ -------------- = 8.35mA
560Ω
IO ≈ -----------------------------------------
RE FIGURE 2. OPEN LOOP INSTRUMENTATION AMPLIFIER
WITH DIFFERENTIAL INPUT AND SINGLE
ENDED OUTPUT
The gain of the preamplifier section (to terminal No. 1) is g m
R L = (5 x 10 -3 ) (36 x 103) = 180. The transconductande g m
is a function of the current into terminal No. 5, I ABC , the
4-166
Application Note 6048
amplifier-bias-current. In this circuit an I ABC of 260μA results
in a g m of 5mS. The operating point of the output stage is
controlled by the 2kΩ potentiometer. With no differential input
signal (e DIFF = 0), this potentiometer is adjusted to obtain a
quiescent output current I O of 12mA. This output current is
established by the 560Ω emitter resistor, R E , as follows:
( gm R
L
) ( e
DIFF
)
I
O
≈ -------------------------------------------
R
E
Under the conditions described, an input swing e DIFF of
±26mV produces a variation in the output current I O of
±8.35mA. The nominal quiescent output voltage is 12mA
times 560Ω or 6.7V. This output level drifts approximately
-4mV, or -0.0595%, for each o C change in temperature. Output
drift is caused by temperature induced variations in the base
emitter voltage of the two output transistors, Q12 and Q13.
R
(NOTE 5)
5kΩ
50kΩ/ARM
TRANSDUCER
BRIDGE
97.6kΩ
100kΩ
±1%
10μA
750kΩ
3
2
+
-
5
7
CA3094
4
82Ω
1000pF
8
1
6
0-1mA
8.4V
820Ω
200Ω
CAL
M
NOTE:
5. Set to optimize CMRR.
FIGURE 3. SINGLE SUPPLY DIFFERENTIAL BRIDGE AMPLIFIER
1kΩ
100Ω
±1%
THERMO-
COUPLE
240kΩ
1kΩ
100kΩ
ZERO ADJ
120K
100Ω
±1%
110kΩ
3
2
+
5
7
CA3094
-
4
1000pF
8
1
100kΩ
±1%
6
82Ω
+12V
M
910Ω
±1%
200Ω
V OUT
OUTPUT 1V FULL SCALE
0-1mA
1mV FROM
THERMO-
COUPLE
FULL SCALE
OUTPUT
CURRENT
IN914
FIGURE 4. SINGLE SUPPLY AMPLIFIER FOR THERMO-
COUPLE SIGNALS

Figure 3 shows the CA3094 used in conjunction with a
resistive-bridge input network; and Figure 4 shows a singlesupply
amplifier for thermocouple signals. The RC networks
connected between terminals 1 and 4 in Figure 3 and Figure
4 provide compensation to assure stable operation.
The components of the RC network are chosen so that
1
--------------- ≈ 2MHz
2πRC
Class A Power Amplifiers
The CA3094 is attractive for power-amplifier service
because the output transistor can control current up to
100mA (300mA peak), the premium devices.
CA3094B can operate at supply voltages up to 44V, and the
TO-5 package can dissipate power up to 1.6W when
equipped with a suitable heat sink that limits the case
temperature to 55 o C.
Figure 5 shows a Class A amplifier circuit using the CA3094A
that is capable of delivering 280mW to a 350Ω resistive load.
This circuit has a voltage gain of 60dB and a 3dB band width
of about 50kHz. Operation is stable without the use of a
phase-compensation network. Potentiometer R is used to
establish the quiescent operating point for class A operation.
E IN
-15V
10kΩ
R
10kΩ
+15V
3
2
100Ω 100Ω
The circuit of Figure 6 illustrates the use of the CA3094 in a
class a power-amplifier circuit driving a transformer-coupled
load. With dual power supplies of +7.5V and -7.5V, a base
resistor R B of 30kΩ, and an emitter resistor R E of 50Ω,
CA3094 dissipation is typically 625mW. With supplies of
+10V and -10V, R B of 40kΩ, and R E of 45Ω, the dissipation
is 1.5W. Total harmonic distortion is 0.4% at a power output
level of 220mW with a reflected load resistance R P of 310Ω,
and is 1.4% for an output of 600mW with an R P of 128Ω. The
setting of potentiometer R establishes the quiescent
operating point for class A operation. The 1kΩ resistor
connected between terminals 6 and 2 provides DC feedback
to stabilize the collector current of the output transistor. The
AC gain is established bye the ratio of the 1MΩ resistor
4-167
56kΩ
-
5
+
7
CA3094A
1
4
RL 100kΩ 350Ω
8
6
-15V
+15V
V OUT
FIGURE 5. CLASS A AMPLIFIER -280mW CAPABILITY INTO
A RESISTIVE LOAD
Application Note 6048
connected between terminals 8 and 3 the 1kΩ resistor
connected to terminal 3. Phase compensation is provided by
the 680pF capacitor connected to terminal 1.
-V
10kΩ
1kΩ
51Ω
R
10kΩ
1kΩ
3
2
+V
R B
680pF
-
5
+
CA3094A
1
Class A Driver Amplifier for
Complementary Power Transistors
7
4
1MΩ
0.01μF
The CA3094 configuration and characteristics are ideal for
driving complementary power-output transistors [3]; a typical
circuit is shown in Figure 7. This circuit can provide 12W of
audio power output into an 8Ω load with intermodulation
distortion (IMD) of 0.2% when 60Hz and 2kHz signals are
mixed in a 4:1 ratio. Intermodulation distortion is shown as a
function of power output in Figure 8.
The large amount of loop gain and the flexibility of feedback
arrangements with the CA3094 make it possible to
incorporate the tone controls into a feedback network that is
closed around the entire amplifier system. The tone controls
in the circuit of Figure 7 are part of the feedback network
connected from the amplifier output (junction of the 330Ω
and 47Ω resistors driven by the emitters of Q2 and Q3) to
terminal 3 of the CA3094. Figure 9 shows voltage gain as a
function of frequency with tone controls adjusted for “flat”
response and for responses at the extremes of tone-control
rotation. The use of tone controls incorporated in the feedback
network results in excellent signal to noise ratio. Hum and
noise are typically 700μV (83dB down) at the output.
8
6
R P
R E
+V
-V
R L
GEN. RADIO
TYPE 1840-A
OUTPUT POWER
METER OR
EQUIVALENT
+
1000μF
-
TYPICAL DATA
DEVICE
DISSIPATION 625MW 1.5W
RB 30kΩ 40kΩ
V+ +7.5V +10V
V- -7.5V -10V
RE 50Ω 45Ω
PO 220mW 600mW
THD 0.4% 1.4%
RP 310Ω 128Ω
FIGURE 6. CLASS A AMPLIFIER WITH TRANSFORMER
COUPLED LOAD

In addition to the savings resulting from reduced parts count
circuit size, the use of the CA3094 leads to further savings in the
power supply system. Typical values of power supply rejection
and common mode rejection are 90dB and 100dB, respectively.
An amplifier with 40dB gain and 90dB power supply rejection
would require a 31mV power-supply ripple to produce 1mV of
hum at the output. Therefore, no filtering is required other than
that provided by the energy storage capacitors at the output of
the rectifier system shown in Figure 7.
For application in which the operating temperature range is
limited (e.g., consumer service the thermal compensation
network (shaded area) can be replaced by a more
economical configuration consisting of a resistor diode
combination (8.2Ω and 1N5391) as shown in Figure 7.
“BOOST”
(CW) 15kΩ
0.12μF
1800Ω
68Ω
INPUT
25μF
+
0.001μF
C 1
TREBLE
R 1
0.2μF
1kΩ
“CUT”
(CCW)
0.001μF
VOLUME
2
3
0.01μF
5600Ω
5μF
+
680
kΩ
4-168
+
5
Power Multivibrators (Astable and
Monostable)
The CA3094 is suitable for use in power multivibrators
because its high current output transistor can drive low
impedance circuits while the input circuitry and the frequency
determining elements are operating at micropower levels. A
typical example of an astable multivibrator using the CA9034
with a dual power supply is shown in Figure 10. The output
frequency fOUT is determined as follows:
1
fOUT = --------------------------------------------------------------
2RC
IN
[ ( 2R1 ⁄ R2)
+ 1]
If R2 is equal to 3.08 R1, then f OUT is simply the reciprocal
of RC.
TYPICAL PERFORMANCE DATA FOR 12W AUDIO AMPLIFIER CIRCUIT
Power Output (8Ω load, Tone Control set at “Flat”)
Music (at 5% THD, regulated supply). . . . . . . . . . . . . . . . . . . 15W
Continuous (at 0.2% IMD, 60Hz and 2kHz
mixed in a 4:1 ratio, unregulated supply)
See Figure 8 in AN6048. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12W
Total Harmonic Distortion
At 1W, unregulated supply . . . . . . . . . . . . . . . . . . . . . . . . . .0.05%
At 12W, unregulated supply . . . . . . . . . . . . . . . . . . . . . . . . .0.57%
NOTES:
6. For standard input: Short C2 ;R1 = 250kΩ,C1 = 0.047μF; remove R2 .
7
CA3094B
“BOOST” 100kΩ “CUT”
(CW)
(CCW)
BASS
-
4
820Ω
0.47
μF
0.02μF
1
10kΩ
6
6.8pF
1Ω
8
8 LEAD
TO-5
Application Note 6048
220Ω
1W
220Ω
1W
30Ω†
27Ω
15μF
+
Q2 2N6292
Voltage Gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .40dB
Hum and Noise (below continuous power output) . . . . . . . . . . .83dB
Input Resistance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .250kΩ
Tone Control Range. . . . . . . . . . . . . . . . . . .See Figure 9 in AN6048
7. For ceramic cartridge input: C 1 = 0.0047μF, R 1 = 2.5MΩ, remove jumper from C 2 ; leave R 2.
FIGURE 7. 12W AUDIO AMPLIFIER CIRCUIT FEATURING TRUE COMPLEMENTARY SYMMETRY OUTPUT STAGE WITH CA3094 IN
DRIVER STAGE
Q 1
Q 3
THERMAL
COMPENSATION
NETWORK†
C 2
0.47μF
JUMPER
+
2N6292
0.47Ω
0.47Ω
2N6107
4700μF
+
D1 - D4 1N5391
4700
μF
330Ω
47Ω
V+
V-
D 1
D 2
D 3
D 4
3μH
22Ω
R 2
1.8MΩ
R L
8Ω
120V
60Hz
STANCOR
NO. P-8609
OR EQUIVALENT
(120VAC TO
26.8VCT AT 1A)
†OPTIONAL THERMAL
COMPENSATION
NETWORK
8.2Ω
1N5391

INTERMODULATION DISTORTION (%)
2.0
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
UNREGULATED SUPPLY LOAD 8Ω
60Hz AND 12kHz
60Hz AND 2kHz
60Hz AND 7kHz
0 2 4 6 8 10 12 14
POWER OUTPUT (W)
FIGURE 8. INTERMODULATION DISTORTION vs POWER OUTPUT
VOLTAGE GAIN (dB)
60
55
50
45
40
35
30
25
20
15
10
51kΩ
43kΩ
10
R2
R1
BASS CUT
2 4 68
100
BASS BOOST
2 4 68 2 4 6 8 2 4 68
1000 10K 100K
FREQUENCY (Hz)
Figure 11 is a single supply astable multivibrator circuit which
illustrates the use of the CA3094 for flashing an incandescent
lamp. With the component values shown, this circuit produces
one flash per second with a 25% “on” time while delivering
output current in excess of 100mA. During the 75% “off” time
it idles with micropower consumption. The flashing rate can
be maintained with in ±2% of the nominal value over a battery
voltage range from 6V to 15V and a temperature excursion
from 0 o C to 70 o C. The CA3094 series of circuits can supply
peak-power output in excess of 10W when used in this type of
4-169
TREBLE BOOST
TREBLE CUT
FLAT
FIGURE 9. VOLTAGE GAIN vs FREQUENCY
1000pF
R
100kΩ
2
3
C
300kΩ
+
-
5
7
CA3094A
4
8
6
-15V
1kΩ
+15V
OUTPUT
f OUT ≈ 5kHz
FIGURE 10. ASTABLE MULTIVIBRATOR USING DUAL SUPPLY
Application Note 6048
circuit. The frequency of oscillation f OSC is determined by the
resistor rations, as follows:
I
fOSC = ------------------------------------------------------------------
( 2RCln)
[ ( 2R1 ⁄ R2)
+ 1]
where:
R
A
R
B
R1 = ---------------------
R
A
+ R
B
3MΩ
12MΩ
where:
R A
R B
R2
18MΩ
C
0.47μF
R
4.3MΩ
2
3
1.2MΩ
NOTES:
1. Flash/s.
I
f
OSC
= ------------------------------------------------------------------
( 2RCln)
[ ( 2R1 ⁄ R2)
+ 1]
R
A
R
B
R1 = ----------------------
R
A
+ R
B
CA3094
Provisions can easily be made in the circuit of Figure 11 to
vary the multivibrator pulse length while maintaining an
essentially constant pulse repetition rate. The circuit shown
in Figure 12 incorporates a potentiometer R P for varying the
width of pulses generated by the astable multivibrator driving
light emitting diode (LED).
+
-
5
4
7
6
8
+V
NO. 57
2. 25% Duty Cycle
3. Frequency Independent of V+ from 6V - 15VDC FIGURE 11. ASTABLE MULTIVIBRATOR USING SINGLE SUPPLY
27kΩ
50kΩ
27kΩ
R P
100kΩ
R
2
3
300kΩ
100kΩ
C
560pF
+
-
5
CA3094A
4
7
8
6
+30V
510Ω
LED
FIGURE 12. ASTABLE POWER MULTIVIBRATOR WITH
PROVISIONS FOR VARYING DUTY CYCLE

Figure 13 shows a circuit incorporating independent controls
(r ON and r OFF ) to establish the “on” and “off” periods of the
current supplied to the LED. The network between points “A”
and “B” is analogous in function to that of the 100kΩ resistor R
in Figure 12.
47kΩ
47kΩ
“A” IN914
r OFF
“B”
r ON
100kΩ
C
560pF
The CA3094 is also suitable for use in monostable
multivibrators as shown in Figure 14. In essence, this circuit
is a pulse counter in which the duration of the output pulses
is independent of trigger pulse duration. The meter reading
is a function of the pulse repetition rate which can be
monitored with the speaker.
Current or Voltage Controlled Oscillators
Because the transconductance of the CA9034 varies linearly
as a function of the amplifier bias current (I ABC ) supplied to
terminal 5, the design of a current or voltage controlled
oscillator is straightforward, a shown in Figure 15. Figure 16
and Figure 17 show oscillator frequency as a function of I ABC
for a current controlled oscillator for two different values of
2
3
4-170
300kΩ
+
5
7
CA3094A
-
4
8
6
+30V
510Ω
LED
FIGURE 13. ASTABLE POWER MULTIVIBRATOR WITH
PROVISIONS FOR INDEPENDENT CONTROL OF
LED “ON-OFF” PERIODS
100pF
1V
0V
1MΩ
1N914
33kΩ
0.1μF
TRIGGER
INPUT
220kΩ
1N914
1MΩ
100μF
+15V
0.005μF
300kΩ
2
3
+
5
CA3094A
-
4
+
-
7
+5V
270Ω
8
6
†
M
200Ω
200μF
-3V
0-1mA
5V
0V
+
-
OUTPUT
1KΩ
100Ω
100Ω
SPEAKER
1N914
1ms
† Full Scale Deflection = 83P/s
FIGURE 14. POWER MONOSTABLE MULTIVIBRATOR
Application Note 6048
capacitor C in Figure 15. The addition of an appropriate resistor
(R) in series with terminal 5 in Figure 15 converts the circuit into
a voltage controlled oscillator. Linearity with respect to either
current or voltage control is within 1% over the middle half of
the characteristics. However, variation in the symmetry of the
output pulses was a function of frequency is an inherent
characteristic of the circuit in Figure 15, and leads to distortion
when this circuit is used to drive the phase detector in phaselocked-loop
applications. This type distortion can be eliminated
by interposing an appropriate flip-flop between the output of the
oscillator and the phase locked discriminator circuits.
47kΩ
47kΩ
20kΩ
1N914
100Ω
C
2
CURRENT INPUT
OR
VOLTAGE INPUT
+
7
CA3094A
3 -
6
5
R
4
8
1kΩ
+15V
OUTPUT
FIGURE 15. CURRENT OR VOLTAGE CONTROLLED OSCILLATOR
FREQUENCY (Hz)
SUPPLY VOLTAGE V+ = 15V
500
C = 1000pF TERMINAL NO. 3 TO GND IN FIGURE 15
AMBIENT TEMPERATURE TA = 25
400
o C
300
200
100
0 100 200 300 400 500
AMPLIFIER BIAS CURRENT (μA)
600
FIGURE 16. FREQUENCY vs I ABC FOR C = 1000pF FOR
CIRCUIT IN FIGURE 15
FREQUENCY (kHz)
500
400
300
200
100
SUPPLY VOLTAGE V+ = 15V
C = 1000pF TERMINAL NO. 3 TO GND IN FIGURE 15
AMBIENT TEMPERATURE T A = 25 o C
0 100 200 300 400 500
AMPLIFIER BIAS CURRENT (μA)
600
FIGURE 17. FREQUENCY vs I ABC FOR C = 100pF FOR
CIRCUIT IN FIGURE 15

Comparators (Threshold Detectors)
Comparator circuits are easily implemented with the CA3094,
as shown by the circuits in Figure 18. The circuit of Figure 18A
is arranged for dual supply operation; the input voltage
exceeds the positive threshold, the output voltage swings
essentially to the negative supply voltage rail (it is assumed
that there is negligible resistive loading on the output
terminal). An input voltage that exceeds the negative
threshold value results in a positive voltage output essentially
equal to the positive supply voltage. The circuit in Figure 18B,
connected for single supply operation, functions similarly.
± THRESHOLD =
± SUPPLY
R1
R1 + R2
300kΩ
2kΩ
R =
INPUT
R1 R2
R1 + R2
3
5
+
7
8
51kΩ
2
CA3094A
-
6
R1
100kΩ
R2
4 -15V
51kΩ
RA
200kΩ
150kΩ
3
2
100kΩ
FIGURE 18A. DUAL SUPPLY
+
200kΩ RB
R1
100kΩ
5
7
CA3094
-
4
8
6
Figure 19 shows a dual limit threshold detector circuit in
which the high level limit is established by potentiometer R1
and the low level limit is set by potentiometer R2 to actuate
the CA3080 low limit detector [1, 2]. A positive output signal
exceeds either the high limit or the low limit values
established by the appropriate potentiometer settings. This
output voltage is approximately 12V with the circuit shown.
The high current handling capability of the CA3094 makes it
useful in Schmitt power trigger circuits such as that shown in
Figure 20. In this circuit, a relay coil is switched whenever
the input signal traverses a prescribed upper or lower trip
point, as defined by the following expressions:
4-171
+15V
2kΩ
V+
OUTPUT
V-
+15V
OUTPUT
RB
R1 RA
R1 + RA +RB
R1 RB
R1 + RB
R1RB
R1 + RB +RA
FIGURE 18B. SINGLE SUPPLY
FIGURE 18. COMPARATORS (THRESHOLD DETECTORS)
DUAL AND SINGLE SUPPLY TYPES
R3
Upper Trip Point = 30⎛------------------------------------
⎞
⎝R1 + R2 + R3⎠
Lower Trip Point ( 30 – 0.026R1 ) R3
≅
---------------------
R2 + R3
Application Note 6048
The circuit is applicable, for example, to automatic ranging.
With the values shown in Figure 20, the relay coil is energized
when the input exceeds approximately 5.9V and remains
energized until the input signal drops below approximately 5.5V.
-15V
INPUT
1.5kΩ
HIGH
DEAD
ZONE
LOW
INPUT
5kΩ
10kΩ
3.9kΩ
5kΩ
10kΩ
33kΩ
33kΩ
+15V
Power Supply Regulators
2
3
3
2
150kΩ
-
CA3094
The CA3094 is an ideal companion device to the CA3085
series regulator circuits [4] in dual voltage tracking regulators
that handle currents up to 100mA. In the circuit of Figure 21,
the magnitude of the regulated positive voltage provided by
the CA3085A is adjusted voltage supplies the power for the
CA3094A negative regulator and also supplies a reference
voltage to its terminal 3 to provide tracking. This circuit
provides a maximum line regulation equal to 0.075% per volt
of input voltage change and a maximum load regulation of
0.075% of the output voltage.
7
+
+
-
5
CA3080
30kΩ
4
5
7
4
8
6
100Ω
6
1N914
FIGURE 19. DUAL LIMIT THRESHOLD DETECTOR
100kΩ
3
2.7MΩ 100Ω
5
2 -
1
100kΩ
+
CA3094A
4
6
7
8
0.01μF
12V
450Ω
COIL
620Ω
R1
82Ω
R2
24kΩ
R3
5.9kΩ
+15V
OUTPUT
1kΩ
+30V
FIGURE 20. PRECISION SCHMITT POWER TRIGGER CIRCUIT

Figure 22 shows a regulated high voltage supply similar to the
type used to supply power for Geiger-Mueller tubes. The
CA3094, used as an oscillator, drives a step-up transformer
which develops suitable high voltages for rectification in the
IN4007 diode network. A sample of the regulated output
voltage is fed to the CA3080A operational transconductance
amplifier through the 198MΩ and 910kΩ divider to control the
pulse repetition rate of the CA3094. Adjustment of
potentiometer R determines the magnitude of the regulated
output voltage. Regulation of the desired output voltage is
maintained within 1% despite load current variations of 5μAto
26μA. The DC to DC conversion efficiency is about 48%.
Timers
The programmability feature inherent in the CA3094 (and
operational transconductance amplifiers in general) simplifies
the design of presettable timers such as the one shown in
Figure 23. Long timing intervals (e.g., up to 4hrs) are achieved
by discharging a timing capacitor C1 into the signal input
terminal (e.g., No. 3) of the CA3094. This discharge current is
controlled precisely by the magnitude of the amplifier bias
current I ABC programmed into terminal 5 through a resistor
selected by switch S2. Operation of the circuit is initiated by
charging capacitor C1 through the momentary closing of
switch S1. Capacitor C1 starts discharging and continues
discharging until voltage E1 is less than voltage E2. The
differential input transistors in the CA3094 then change state,
and terminal 2 draws sufficient current to reverse the polarity
of the output voltage (terminal 6). Thus, the CA3094 not only
has provision for readily presetting the time delay, but also
provides significant output current to drive control devices
such as thyristors. Resistor R5 limits the initial charging
current for C1. Resistor R5 limits the initial charging current for
C1. Resistor R7 establishes a minimum voltage of at least 1V
at terminal 2 to insure operation within the common mode
input range of the device. The diode limits the maximum
910kΩ
R
2MΩ 1.8MΩ
2
3
3.6MΩ
1MΩ
-
7
1.8MΩ
CA3080A
+
4
4-172
5
+6V
1.5μA
6
560pF
0.47μF
820
kΩ
Application Note 6048
430
kΩ
820
kΩ
2
differential input voltage to 5V. Gross changes in time range
selection are made with switch S2, and vernier trimming
adjustments are made with potentiometer R6.
V+ INPUT
(NOTE 1)
V- INPUT
(NOTE 2)
REF 1.6V
2 CA3085A
VOLTAGE
REG
3
0.0056μF
5.1kΩ
4
6
7
0.01μF
100Ω
1
200kΩ
5
2
+
CA3094A
7
1
8
R
5kΩ
10kΩ
1.5kΩ
3
-
8
6
4 10kΩ
±1%
10kΩ ±1%
5.6Ω
NOTES:
1. V+ input range = 19V to 30V for 15V output
2. V- input range = -16 to -30V for -15V output
3. REGULATION:
47μF 150kΩ
+
7
3 -
6
470kΩ
1N914
4
MAX I OUT = ±100mA
+15V
REG OUTPUT
COMMON
RETURN
0.03μF
MAX LINE =
ΔVOUT --------------------------------------------------------------- × 100
VOUT( INITIAL)
ΔVIN = 0.075%/V
-15V
REG OUTPUT
MAX LINE =
ΔVOUT ------------------------------------------- × 100 =
VOUT( INITIAL)
0.075% V
OUT
(IL FROM 1mA to 50mA)
CA3094
FIGURE 21. DUAL VOLTAGE TRACKING REGULATOR
5
25μA
FIGURE 22. REGULATED HIGH VOLTAGE SUPPLY
8
198MΩ
(NINE 22MΩ RESISTORS IN SERIES)
1N914
0.01μF
IN4007
MINIATURE
MICROPHONE
INPUT TRANSFORMER
600Ω CT TO 50kΩ
R L
IN4007
IN4007
0.01μF
+900V
0.01μF

120V
AC
+
30V
DC
COMMON
R5
START
SWITCH
R6
R7
S1
E1
100Ω
3
2
C1 E2
2μF
+
7
CA3094A
-
4
In some timer applications, such as that shown in Figure 24, a
meter readout of the elapsed time is desirable. This circuit uses
the CA3094 and CA3083 transistor array [5] to control the
meter and a load switching triac. The timing cycle starts with
the momentary closing of the start switch to charge capacitor
C1 to an initial voltage determined by the 50kΩ vernier timing
adjustment. During the timing cycle the output of the CA3094,
4-173
TIME RANGE
SECTOR
S2
8
5
6
R8
R1
R2
R3
R4
R LOAD
G
40529 TURNS “OFF” AFTER
EXPIRATION OF TIME DELAY
TIME RESISTOR VALUE
3Min R1 = 0.5MΩ R5 = 2.7kΩ
30Min R2 = 5.1MΩ R6 = 50kΩ
2Hrs R3 = 22MΩ R7 = 2.7kΩ
4Hrs R4 = 44MΩ R8 = 1.5kΩ
FIGURE 23. PRESETTABLE ANALOG TIMER
+30V
5.1kΩ
50kΩ
5.1kΩ
START
SWITCH
VERNIER
TIME
ADJUST
+30V
3
68kΩ
68kΩ
C1
0.5μF
0.01μF
270kΩ
2
3
2
7
+
-
7
MT2
MT1
8
CA3094
4
5
which is also the collector voltage of Q1, is “high”. The base
drive for Q1 is supplied from the positive supply through a 91kΩ
resistor. The emitter of Q1, through the 75Ω resistor, supplies
gate-trigger current to the triac. Diode connected transistors Q4
and Q5 are connected so that transistor Q1 acts as a constant
current source to drive the triac. As capacitor C1 discharges,
the CA3094 output voltage at terminal 6 decreases until it
becomes less than the V CESAT of Q1. At this point the flow of
drive current to the triac ceases and the timing cycle is ended.
The 20kΩ resistor between terminals 2 and 6 of the CA3094 is
a feedback resistor. Diode connected transistor Q2 and Q3 and
their associated networks serve to compensate for
nonlinearities in the discharge circuit network by bleeding
corrective current into the 20kΩ feedback resistor. Thus, current
flow in the meter is essentially linear with respect to the timing
period. The time periods as a function of R1 and indicated on
the Time-Range Selection Switch in Figure 24.
Alarm Circuit
Q2 Q3 6
Q1 16 Q4
4
Application Note 6048
8
20kΩ
68kΩ
R1
METER
READOUT
(0-200μA)
M
Figure 25 shows an alarm circuit utilizing two “sensor” lines. In
the “no-alarm” state, the potential at terminal 5 (I ABC ) is driven
with sufficient current though resistor R5 to keep the output
voltage “high”. If either “sensor” line is opened, shorted to
ground, or shorted to the other sensor line, the output goes
“low” and activates some type of alarm system. The back-toback
diodes connected between terminals 2 and 3 protect the
CA3094 input terminals against excessive differential voltages.
22MΩ
1.5MΩ 4MIN
1HR
500kΩ 1.5MIN
100MΩ 20s
6
1
15
TIME RANGE
SELECTION SWITCH
75Ω
9
11
14
12
1kΩ
FIGURE 24. PRESETTABLE TIMER WITH LINEAR READOUT
Q5
10
13
CA3083
91kΩ
G
120V
60Hz
R L
MT2
MT1

R5
100kΩ
R4
100kΩ
SENSOR
LINES
Motor-Speed Controller System
Figure 26 illustrates the use of the CA3094 in a motor-speed
controller system. Circuity associated with rectifiers D1 and D2
comprises a full wave rectifier which develops a train of
half-sinusoid voltage pulse to power the DC motor. The motor
speed depends on the peak value of the half-sinusoids and the
period of time (during each half cycle) the SCR is conductive.
The SCR conduction, in turn, is controlled by the time
duration of the positive signal supplied to the SCR by the
phase comparator. The magnitude of the positive DC voltage
supplied to terminal 3 of the phase comparator depends on
motor-speed error as detected by a circuit such as that
shown in Figure 27. This DC voltage is compared to that of a
MOTOR SPEED
ERROR
SIGNAL
R3
150kΩ
INPUT TO
TERMINAL 2
INPUT TO
TERMINAL 3
(NOTE 1)
R1
510kΩ
1N914
D2
R2
510kΩ
4-174
3
D1
2
+
7
5
100Ω
CA3094
-
I ABC
FIGURE 25. ALARM SYSTEM
SEE FIGURE 27
MOTOR SPEED
ERROR
DETECTOR
SEE FIGURE 28
SYNCHRONOUS
RAMP
GENERATOR
+
MOTOR
CURRENT
1N914
510kΩ
510kΩ
3
1N914
2
8
4
Application Note 6048
+12V
OUTPUT
FOR
TRIGGERING
ALARM
5
6
Ø COMPARATOR
330kΩ
+
7
CA3094A
-
8
4
-15V
+15V
6
OUTPUT
FOR
TRIGGERING
ALARM
FIGURE 26A.
fixed amplitude ramp wave generated synchronously with
the AC line voltage frequency. The comparator output at
terminal 6 is “high” (to trigger the SCR into conduction)
during the period when the ramp potential is less than that of
the error voltage on terminal 3. The motor current
conduction period is increased as the error voltage at
terminal 3 is increased in the positive direction. Motor-speed
accuracy of ±1% is easily obtained with this system.
Motor-Speed Error Detector
Figure 27A shows a motor-speed error detector suitable for use
with the circuit of Figure 26. A CA3080 operational
transconductance amplifier is used as a voltage comparator.
The reference for the comparator is established by setting the
potentiometer R so that the voltage at terminal 3 is more
positive than that at terminal 2 when the motor speed is too low.
An error voltage E1 is derived from a tachometer driven by the
motor. When the motor speed is too low, the voltage at terminal
2 of the voltage comparator is less positive than that at terminal
3, and the output voltage at terminal 6 goes “high”. When the
motor speed is too high, the opposite input conditions exist, and
the output voltage at terminal 6 goes “low”. Figure 27B also
shows these conditions graphically, with a linear transition
region between the “high” and “low” output levels. This linear
transition region is known as “proportional bandwidth”. The
slope of this region is determined by the proportional bandwidth
control to establish the error correction response time.
1.5kΩ
SCR
TIME
FIGURE 26B.
FIGURE 26. MOTOR SPEED CONTROLLER SYSTEM
DC MOTOR
+
0
LINE
D2
D1
NOTE:
1. This level will vary depending on
motor speed. (See Text).

+
E1
-
10kΩ
100kΩ
R
82kΩ
1N914
1N914
1N914
2
1N914
Synchronous Ramp Generator
Figure 28 shows a schematic diagram and signal waveforms
for a synchronous ramp generator suitable for use with the
motor controller circuit of Figure 26. Terminal 3 is biased at
approximately +2.7V (above the negative supply voltage). The
input signal E IN at terminal 2 is a sample of the half-sinusoids
(at line frequency) used to power the motor in Figure 26. A
synchronous ramp signal is produced by using the CA3094 to
charge and discharge capacitor C1 in response to the
synchronous toggling of E IN . The charging current for C1 is
supplied by terminal 6. When terminal 2 swings more positive
than terminal 3, transistors Q12 and Q13 in the CA3094
(Figure 1) lose their base drive and become non-conductive.
Under these conditions, C1 discharges linearly through the
external diode D3 and Q10, D6 path in the CA3094 to
produce the ramp wave. The E OUT signal is supplied to the
phase comparator in Figure 26.
Thyristor Firing Circuits
Temperature Controller
In the temperature control system shown in Figure 29, the
differential input of the CA3094 is connected across a bridge
circuit comprised of a Positive Temperature Coefficient
(PTC) temperature sensor, two 75kΩ resistors, and an arm
containing the temperature set control. When the
temperature is “low”, the resistance of the PTC type sensor
is also low; therefore, terminal 3 is more positive than
terminal 2 and an output current from terminal 6 of the
CA3094 drives the triac into conduction. When the
temperature is “high”, the input conditions are reversed and
the triac is cut off. Feedback from terminal 8 provides
hysteresis to the control point to prevent rapid cycling of the
system. The 1.5kΩ resistor between terminal 8 and the
3
4-175
PROPORTIONAL
BANDWIDTH CONTROL
-
7
CA3080A
+
4
330kΩ
5
-15V
RECTIFIED AND FILTERED SIGNAL DERIVED FROM
TACHOMETER DRIVEN BY MOTOR BEING CONTROLLED
FIGURE 26A. VOLTAGE COMPARATOR
EOUT HIGH
MOTOR SPEED
MOTOR SPEED FAST
SLOW
EOUT LOW
1MΩ
PROPORTIONAL
BANDWIDTH
+15V
6
EOUT TO PHASE
COMPARATOR
FIGURE 26B.
FIGURE 26. MOTOR SPEED ERROR DETECTOR
Application Note 6048
positive supply limits the triac gate current and develops the
voltage for the hysteresis feedback. The excellent power
supply rejection and common mode rejection ratios of the
CA3094 permit accurate repeatability of control despite
appreciable power supply ripple. The circuit of Figure 29 is
equally suitable for use with Negative Temperature
Coefficient (NTC) sensors provided the positions of the
sensor and the associated resistor R are interchanged in the
circuit. The diodes connected back to back across the input
terminals of the CA3094 protect the device against
excessive differential input signals.
E IN
330kΩ
33kΩ
D1
1N914
2
1N914
D2
3
Thyristor Control from AC Bridge Sensor
Figure 30 shows a line operated thyristor firing circuit
controlled by a CA3094 that operates from an AC Bridge
sensor. This circuit is particularly suited to certain classes of
sensors that cannot be operated from DC. The CA3094 is
inoperative when the high side of the AC line is negative
because there is no IABC supply to terminal 5. When the
sensor bridge is unbalanced so that terminal 2 is more
positive than terminal 3, the output stage of the CA3094 is
cut off when the AC line swings positive, and the output level
at terminal 8 of the CA3094 goes “high”. Current from the
line flows through the IN3193 diode to charge the 100μF
reservoir capacitor, and also provides current to drive the
triac into conduction. During the succeeding negative swing
of the AC line, there is sufficient remanent energy in the
reservoir capacitor to maintain conduction in the triac.
When the bridge is unbalanced in the opposite direction so that
terminal 3 is more positive than terminal 2, the output of the
CA3094 at terminal 8 is driven sufficiently “low” to “sink” the
current supplied through the 1N4003 diode so that the triac
gate cannot be triggered. Resistor R1 supplies the hysteresis
feedback to prevent rapid cycling between turn on and turn off.
+
7
5
330kΩ
8
CA3080A
- 1
4
FIGURE 27A.
1.5kΩ
6
1N914
+15V
E OUT
TO PHASE
COMPARATOR
D3
C1
0.033
μF
-15V
+
BIAS LEVEL
E AT TERMINAL NO. 3
IN
0
C1 DISCHARGING (RAMP)
+
C1 CHARGING
EOUT 0
FIGURE 27B.
FIGURE 27. SYNCHRONOUS RAMP GENERATOR WITH INPUT
AND OUTPUT WAVEFORMS

T
117V
60Hz
IN4001
10Ω
26V
60Hz
+
-
47μF
50V
Battery Charger Regulator Circuit
The circuit for battery charger regulator circuit using the
CA3094 is shown in Figure 31. This circuit accurately limits
the peak output voltage to 14V, as established by the zener
diode connected across terminals 3 and 4. When the output
voltage rises slightly above 14V, signal feedback through a
100kΩ resistor to terminal 2 reduces the current drive
supplied to the 2N3054 pass transistor from terminal 6 of the
CA3094. An incandescent lamp serves as the indicator of
charging current flow. Adequate limiting provisions protect
the circuit against damage under load short conditions. The
advantage of the circuit over certain the types of regulator
circuits is that the reference voltage supply doesn’t drain the
battery when the power supply is disconnected. This feature
is important in portable service applications, such as in a
trailer where a battery is kept “on-charge” when the trailer is
parked and power is provided from an AC line.
AC IN
12V TO 24V
RMS
D2
D1
2.2kΩ
D3
D4
D5
4-176
PTC TEMP.
SENSOR
75kΩ
100Ω
100kΩ
100kΩ
5
R
7
75kΩ
68kΩ
10kΩ
TEMP.
SET
75kΩ
1N914
FIGURE 28. TEMPERATURE CONTROLLER
MURALITES
LAMP L10/20
680Ω
1/2W
16Ω
10W
2 +
8
2N3054
100Ω
1N914 CA3094A 6
100kΩ
3 - 1
14V
4
0.001μF
1kΩ
EOUT TO
BATTERY
14V PEAK
FIGURE 29. BATTERY CHARGER REGULATOR CIRCUIT
Application Note 6048
2
3
1.5MΩ 1N914
330kΩ
7
1.5kΩ
5
8
CA3094B 6
4
HEATER
MT 2
MT 1
Ground Fault Interrupters (GFI)
1kΩ
0.01μF
Ground fault interrupter systems are used to continuously
monitor the balance of current between the high and neutral
lines of power distribution networks. Power is interrupted
whenever the unbalance exceeds a preset value (e.g., 5mA).
An unbalance of current can occur then, for example,
defective insulation in the high side of the line permits
leakage of current to an earth ground. GFI systems can be
used to reduce the danger of electrocution from accidental
contact with “high” line because the unbalance caused by
the leakage of current from the “high” line through a human
body to ground results in an interruption of current flow.
The CA3094 is ideally suited for GFI applications because it
can be operated from a single supply, has adequate
sensitivity, and can drive a relay or thyristor directly to effect
power interruption. Figure 32 shows a typical GFI circuit.
Vernier adjustment of the trip point is made by the R TRIP
potentiometer. When the differential current sensor supplies
a signal that exceeds the selected trip-point voltage level
(e.g., 60mV), the CA3094 is toggled “on” and terminal 8
goes “low” to energize the circuit breaker trip coil. Under
quiescent conditions, the entire circuit consumes
approximately 1mA. The resistor R, connected to one leg of
the current sensor, provides current limiting to protect the
CA3094 against voltage spikes as large as 100V. Figure 32
also shows the pertinent waveform for the GFI circuit.
Because hazards of severe electrical shock are a potential
danger to the individual user in the event of malfunctions in
GFI apparatus, it is mandatory that the highest standard of
good engineering practice be employed in designing
equipment for this service. Every consideration in design
and application must be given to the potentially serious
consequences of component malfunction in such
equipment. Use of “reliability through redundancy” concepts
and so called “fail-safe” features is encouraged.
G
NOTE: All Resistors
are 1/2W.

200mV
RANGE
R
47K
L C
0.02μF
0.1μF C2
120V
AC LINE
4-177
HIGH
27kΩ
1kΩ
SENSOR
LOW
1MΩ
27kΩ
27kΩ
R1 180kΩ
330kΩ
0.01μF
2
3
1kΩ
+
-
7
CA3094B
5
4
6
8
2kΩ
5W
1N4003
10kΩ
References
510Ω
+
- 100μF
G
15V
LOAD E OUT
FIGURE 30. LINE OPERATED THYRISTOR FIRING CIRCUIT CONTROLLED BY AC BRIDGE SENSOR
33K
200Ω
3.3K
47K
100K
3
2
-
5
3.3MΩ
7
CA3094B
+
6
4
100Ω
DIFFERENTIAL CURRENT
SENSOR PROVIDES
1K 0.001μF
60mV SIGNAL
≈ 5mA
OF UNBALANCE
(TRIP) CURRENT
NOTES:
1. All resistors in Ω, 1/2W, 10%.
2. RC selected for 3dB point at 200Hz.
3. C2 = AC bypass.
4. Offset ADJ included in RTRIP. CIRCUIT
BREAKER
CONTROL
SOLENOID
5. Input impedance from 2 to 3 equals 800K.
6. With no input signal terminal 8 (Output) at +36V.
3
VOLTS
R TRIP
1mA
3V
CIRCUIT TRIPS ON POSITIVE
PEAKS - WILL SWITCH
WITHIN 1.5 CYCLES
60mV
TYPICAL
t
GROUND FAULT
SIGNAL 60Hz
10μA
IABC 8
+36V
20μA
IA VOLTAGE BETWEEN
TERMINALS 2 AND 4
I LOAD
VOLTAGE BETWEEN
TERMINALS 3 AND 4
(ADJUSTABLE WITH R TRIP )
FIGURE 31. GROUND FAULT INTERRUPTER (GFI) AND
WAVEFORM PERTINENT TO GROUND FAULT
DETECTOR
Application Note 6048
MT 2
MT 1
For Harris documents available on the internet, see web site
http://www.semi.harris.com/
Harris AnswerFAX (407) 724-7800.
[1] CA3080 and CA3080A Data Sheet, Harris Corporation,
Semiconductor Communications Division, FN475.
[2] AN6668 Application Note, Harris Corporation,
Semiconductor Communications Division, “Applications
of the CA3080 and CA3080A High Performance
Operational Transconductance Amplifiers”.
[3] AN6077 Application Note, L. Kaplan and H. A.
Wittlinger, Harris Corporation, Semiconductor
Communications Division, “An IC Operational
Transconductance Amplifier (OTA) with Power
Capability”. Paper originally presented at the IEEE
Chicago Springs Conference on Broadcast and TV
Receivers, June 1972. Reprinted as AN6077.
[4] CA3085 Data Sheet, Harris Corporation, Semiconductor
Communications Division, FN491.
[5] CA3083 Data Sheet, Harris Corporation, Semiconductor
Communications Division, FN481.